CN109120290B - RFID receiving circuit, storage medium, and electronic device - Google Patents

RFID receiving circuit, storage medium, and electronic device Download PDF

Info

Publication number
CN109120290B
CN109120290B CN201810878331.3A CN201810878331A CN109120290B CN 109120290 B CN109120290 B CN 109120290B CN 201810878331 A CN201810878331 A CN 201810878331A CN 109120290 B CN109120290 B CN 109120290B
Authority
CN
China
Prior art keywords
signal
carrier
port
baseband
circuit
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
CN201810878331.3A
Other languages
Chinese (zh)
Other versions
CN109120290A (en
Inventor
余云忠
谢理
吴雷
胡攀攀
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Wuhan Wanji Information Technology Co Ltd
Original Assignee
Wuhan Wanji Information Technology Co Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Wuhan Wanji Information Technology Co Ltd filed Critical Wuhan Wanji Information Technology Co Ltd
Priority to CN201810878331.3A priority Critical patent/CN109120290B/en
Publication of CN109120290A publication Critical patent/CN109120290A/en
Application granted granted Critical
Publication of CN109120290B publication Critical patent/CN109120290B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06K—GRAPHICAL DATA READING; PRESENTATION OF DATA; RECORD CARRIERS; HANDLING RECORD CARRIERS
    • G06K7/00—Methods or arrangements for sensing record carriers, e.g. for reading patterns
    • G06K7/10—Methods or arrangements for sensing record carriers, e.g. for reading patterns by electromagnetic radiation, e.g. optical sensing; by corpuscular radiation
    • G06K7/10009—Methods or arrangements for sensing record carriers, e.g. for reading patterns by electromagnetic radiation, e.g. optical sensing; by corpuscular radiation sensing by radiation using wavelengths larger than 0.1 mm, e.g. radio-waves or microwaves
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/20—Modulator circuits; Transmitter circuits

Landscapes

  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Health & Medical Sciences (AREA)
  • Toxicology (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • General Health & Medical Sciences (AREA)
  • Artificial Intelligence (AREA)
  • Computer Vision & Pattern Recognition (AREA)
  • General Physics & Mathematics (AREA)
  • Theoretical Computer Science (AREA)
  • Transceivers (AREA)

Abstract

The invention provides an RFID receiving circuit, a storage medium and an electronic device, wherein the circuit comprises: the passive carrier cancellation module is connected with the directional coupler and used for carrying out target operation on a coupling signal received from a first port of the directional coupler to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting; and the second port of the directional coupler is used for performing vector superposition on the modulation signal and a carrier leakage signal generated by the transmitting carrier to obtain a receiving signal, wherein the amplitude of the carrier leakage signal carried in the receiving signal is smaller than the amplitude of the carrier leakage signal generated by the transmitting carrier. Therefore, the problem that the carrier leakage signal cannot be effectively processed in the related art can be solved, and the effects of reducing the amplitude of the carrier leakage signal and reducing the influence of the carrier leakage signal on the circuit can be achieved.

Description

RFID receiving circuit, storage medium, and electronic device
Technical Field
The present invention relates to the field of communications, and in particular, to an RFID receiving circuit, a storage medium, and an electronic device.
Background
In recent years, the problem of carrier leakage in a Radio Frequency Identification (RFID) reader-writer has been a popular research subject, and the problem of carrier leakage is a key problem that must be solved in the design of an ultrahigh Frequency RFID reader-writer because a carrier leakage signal is much larger than an echo signal of a tag, which seriously interferes with a receiver system and causes the sensitivity of the receiver to be seriously deteriorated.
Aiming at the problem that carrier leakage signals cannot be effectively processed in the prior art, an effective solution is not provided in the related art.
Disclosure of Invention
The embodiment of the invention provides an RFID receiving circuit, a storage medium and an electronic device, which at least solve the problem that carrier leakage signals cannot be effectively processed in the related art.
According to an embodiment of the present invention, there is provided an RFID receiving circuit including: the passive carrier cancellation module is connected with the directional coupler and used for carrying out target operation on a coupling signal received from a first port of the directional coupler to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting; and the second port of the directional coupler is used for carrying out vector superposition on the modulation signal and a carrier leakage signal generated by the transmitting carrier to obtain a receiving signal.
According to another embodiment of the present invention, there is also provided a signal processing method including: performing target operation on the received coupling signal to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting; and carrying out vector superposition on the modulation signal and a carrier leakage signal generated by the transmitting carrier to obtain a receiving signal.
There is also provided, in accordance with another embodiment of the present invention, an RFID receiving circuit including: the low-noise amplifier is connected with the second port and used for amplifying a received signal, wherein the second port is positioned in the directional coupler and used for carrying out vector superposition on a modulation signal and a carrier leakage signal generated by a transmission carrier to obtain the received signal; and the analog correlation demodulation circuit is connected with the low noise amplifier and is used for demodulating the received signal to generate a baseband signal.
According to another embodiment of the present invention, there is also provided an electronic device, including a memory and a processor, the memory having a computer program stored therein, the processor being configured to execute the computer program to perform the method of the above.
According to another embodiment of the present invention, there is also provided a storage medium having a computer program stored therein, wherein the computer program is configured to execute the method of the above when executed
According to the invention, a passive carrier cancellation module is adopted to perform target operation on a coupling signal received from a first port of a directional coupler to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting; and the second port of the directional coupler carries out vector superposition on the modulation signal and a carrier leakage signal generated by the emission carrier to obtain a receiving signal. I.e., the carrier leakage signal can be cancelled. Therefore, the problem that the carrier leakage signal cannot be effectively processed in the related art can be solved, and the effects of reducing the amplitude of the carrier leakage signal and reducing the influence of the carrier leakage signal on the circuit can be achieved.
Drawings
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this application, illustrate embodiment(s) of the invention and together with the description serve to explain the invention without limiting the invention. In the drawings:
FIG. 1 is a block diagram of an RFID receiving circuit according to an embodiment of the present invention;
FIG. 2 is a flow chart of a signal processing method according to an embodiment of the present invention;
fig. 3 is a schematic structural diagram of an analog demodulation-related circuit in the present embodiment;
fig. 4 is a functional block diagram of an electronic apparatus in the present embodiment.
Detailed Description
The invention will be described in detail hereinafter with reference to the accompanying drawings in conjunction with embodiments. It should be noted that the embodiments and features of the embodiments in the present application may be combined with each other without conflict.
It should be noted that the terms "first," "second," and the like in the description and claims of the present invention and in the drawings described above are used for distinguishing between similar elements and not necessarily for describing a particular sequential or chronological order.
In this embodiment, an RFID receiving circuit is provided, and fig. 1 is a structural diagram of an RFID receiving circuit according to an embodiment of the present invention, and as shown in fig. 1, the RFID receiving circuit includes:
a passive carrier cancellation module 106, connected to the directional coupler 104, configured to perform a target operation on a coupled signal received from the first port 1 of the directional coupler 104 to obtain a modulated signal 107, where the target operation includes reflection and phase shifting;
and a second port 2 of the directional coupler 104, configured to perform vector superposition on the modulated signal 107 and a carrier leakage signal generated by the transmit carrier to obtain a receive signal 108, where an amplitude of the carrier leakage signal carried in the receive signal 108 is smaller than an amplitude of the carrier leakage signal generated by the transmit carrier 101.
In this embodiment, the passive carrier cancellation module 106 phase shifts to invert the coupled signal.
The transmitting carrier 101 generates a carrier leakage signal when transmitting a signal, and the transmitting circuit includes a power divider 102 for transmitting a local oscillator signal; the power amplifier 103, for the method signal, sends the signal to the directional coupler 104.
The directional coupler 104 includes four ports, wherein port 3 is used for receiving signals transmitted by the transmitting circuit, and port 4 is used for transmitting signals to the antenna 105.
Optionally, the passive carrier cancellation module 106 includes: the microstrip line TL is used for receiving the coupling signal and carrying out target operation on the coupling signal by adjusting the length of the TL so as to adjust the phase of the modulation signal; for example, the characteristic impedance of the microstrip line TL is 50 ohms, the length l is related to the phase of the modulation signal, and changing the length l can change the phase of the modulation signal; and the resistor R is connected with the microstrip line TL and is used for carrying out target operation on the coupling signal by adjusting the resistance value of R so as to adjust the amplitude of the modulation signal. That is, the resistance of the resistor R is related to the magnitude of the modulation signal, and changing the resistance of the resistor R can modulate the amplitude of the modulation signal; the length l of the microstrip line TL and the resistance value of the resistor R are selected to enable the modulation signal and the carrier leakage signal to be close to equal-amplitude phase inversion, and therefore a good cancellation effect is achieved.
The first port is also used for receiving the modulation signal reflected by the passive carrier cancellation module and sending the modulation signal to the second port, and the second port carries out vector superposition on the modulation signal and the carrier leakage signal. It is assumed that the leakage signal can be expressed as
Figure BDA0001753868360000047
The first port 1 of the directional coupler outputs a signal of Aej(ωt)Wherein A > α; the generation process of the modulation signal is as follows; the first port 1 of the directional coupler outputs a signal of Aej(ωt)And the modulated signal is reflected back to the first port of the directional coupler through the passive carrier cancellation module and then is output from the second port to form a modulated signal.
Assuming that the first port output signal of the directional coupler is: aej(ωt)(ii) a The reflection coefficient of the passive carrier cancellation module is
Figure BDA0001753868360000041
Wherein the propagation constant
Figure BDA0001753868360000042
λgThe transmission wavelength of the signal on the microstrip line; the transmission coefficient from the first port to the second port of the directional coupler is B; the expression of the modulated signal is therefore
Figure BDA0001753868360000043
The modulation signal and the carrier leakage signal are vector-superposed at the second port of the directional coupler, and when the modulation signal and the carrier leakage signal are equal in amplitude and opposite in phase (namely, equal in amplitude and opposite in phase), depth cancellation can be achieved (cancellation degree is greater than 30dB), so that if and only if
Figure BDA0001753868360000044
Or
Figure BDA0001753868360000045
When the depth compensation is achieved, the corresponding microstrip line length 1 and the resistance R are solved in two ways, namely
Figure BDA0001753868360000046
Due to the limited precision of the existing resistor or the factor of environment change, the carrier cancellation cannot reach complete cancellation. The carrier leakage signal after cancellation has smaller amplitude and is mixed with the echo signal of the tag to form a receiving signal;
in an optional embodiment, the circuit further comprises: a low noise amplifier 109 connected to the second port for amplifying the received signal to improve the signal-to-noise ratio of the received signal; and the analog correlation demodulation circuit 120 is connected with the low-noise amplifier and is used for demodulating the received signal with low power consumption, optimizing the signal-to-noise ratio of the generated baseband signal and randomly adjusting the phase within (0 degrees and 180 degrees).
Furthermore, the received signal also contains the echo signal of the tag
Figure BDA0001753868360000051
And carrier leakage signal after cancellation
Figure BDA0001753868360000052
Where n (t) is noise; the phase deviation of the electrically-adjusted phase shifter in the initial state is 0 DEG, and the phase deviation is expressed as when a received signal reaches a 3dB coupler
Figure BDA0001753868360000053
The electrically-adjusted phase shifter is used for adjusting the phase shift of the received signal; the 3dB coupler is connected with the electrically-adjusted phase shifter and is used for performing sum and difference operation on the adjusted received signal and the local oscillation signal 122 to obtain a sum and difference signal, wherein the local oscillation signal is sent by a circuit for transmitting a carrier; the first power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum-difference signal to obtain a first baseband signal; and the second power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum-difference signal to obtain a second baseband signal. Suppose the local oscillator signal is μ ejωtAfter passing through the 3dB coupler, 2 paths of sum and difference signals are output as follows:
Figure BDA0001753868360000054
Figure BDA0001753868360000055
the sum and difference signals respectively pass through a power detector to generate two paths of baseband signals:
BB + output signal (corresponding to the first baseband signal in the above):
Figure BDA0001753868360000056
BB-output signal (corresponding to the second baseband signal in the above):
Figure BDA0001753868360000057
and (3) subtracting the two baseband signals to obtain a BB signal (corresponding to the third baseband signal in the above step): 2 μ A (t) sin θ1+2βn(t)μsinθ2；
Optionally, if the phase shift of the electrically-adjusted phase shifter in the initial state is θ0From the above inference process, the baseband signal BB is derived: 2 μ A (t) sin (θ)1+θ0)+2βn(t)μsin(θ2+θ0) (ii) a Wherein 2 μ A (t) sin (θ)1+θ0) As a signal component, 2 β n (t) μ sin (θ)2+θ0) For noise components, there is one θ0So that the signal-to-noise ratio of the BB signal is maximized.
In an alternative embodiment the circuit further comprises: the baseband filtering and amplifying circuit 121 is configured to receive the first baseband signal and the second baseband signal, and perform difference operation on the first baseband signal and the second baseband signal to obtain a third baseband signal, where a signal-to-noise ratio of the third baseband signal is greater than signal-to-noise ratios of the first baseband signal and the second baseband signal. The first baseband signal is denoted by BB +; the second baseband signal is denoted by BB-; the third baseband signal is represented by the BB signal. The baseband filtering and amplifying circuit is also used for amplifying and filtering the baseband signals.
According to the method, the passive carrier cancellation scheme is adopted, so that the cost is extremely low, the structure is simple, and no extra noise is introduced; the related analog related demodulation circuit is adopted to replace the traditional IQ mixing demodulation, the requirement on the local oscillation power is reduced, the system power consumption is reduced, and the receiving dynamic range of the receiver is improved.
In this embodiment, the passive carrier cancellation module and the analog related demodulation circuit can be implemented separately.
In this embodiment, a signal processing method is further provided, and fig. 2 is a flowchart of the signal processing method according to the embodiment of the present invention, as shown in fig. 2, the flowchart includes the following steps:
step S202, carrying out target operation on the received coupling signal to obtain a modulation signal, wherein the target operation comprises reflection and phase shift;
and step S204, carrying out vector superposition on the modulation signal and a carrier leakage signal generated by the emission carrier to obtain a receiving signal.
Through the steps, a passive carrier cancellation module is adopted to perform target operation on a coupling signal received from a first port of a directional coupler to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting; and the second port of the directional coupler carries out vector superposition on the modulation signal and a carrier leakage signal generated by the emission carrier to obtain a receiving signal. I.e., the carrier leakage signal can be cancelled. Therefore, the problem that the carrier leakage signal cannot be effectively processed in the related art can be solved, and the effects of reducing the amplitude of the carrier leakage signal and reducing the influence of the carrier leakage signal on the circuit can be achieved.
Optionally, the main body of the above steps may be a terminal, but is not limited thereto.
In an optional embodiment, the microstrip line TL in the passive carrier cancellation module 106 is used to receive the coupling signal, and the length of TL is adjusted to perform target operation on the coupling signal, so as to adjust the phase of the modulation signal; for example, the characteristic impedance of the microstrip line TL is 50 ohms, the length 1 is related to the phase of the modulation signal, and changing the length 1 can change the phase of the modulation signal; and the resistor R is connected with the microstrip line TL and is used for carrying out target operation on the coupling signal by adjusting the resistance value of R so as to adjust the amplitude of the modulation signal. That is, the resistance of the resistor R is related to the magnitude of the modulation signal, and changing the resistance of the resistor R can modulate the amplitude of the modulation signal; the length 1 of the microstrip line TL and the resistance value of the resistor R are selected to be appropriate, so that the modulation signal and the carrier leakage signal are close to equal amplitude and opposite phase, and a good cancellation effect is achieved.
The first port is also used for receiving the modulation signal reflected by the passive carrier cancellation module and sending the modulation signal to the second port, and the second port carries out vector superposition on the modulation signal and the carrier leakage signal. It is assumed that the leakage signal can be expressed as
Figure BDA0001753868360000071
The first port 1 of the directional coupler outputs a signal of Aej(ωt)Wherein A > α; the generation process of the modulation signal is as follows; the first port 1 of the directional coupler outputs a signal of Aej(ωt)And the modulated signal is reflected back to the first port of the directional coupler through the passive carrier cancellation module and then is output from the second port to form a modulated signal.
Assuming that the first port output signal of the directional coupler is: aej(ωt)(ii) a The reflection coefficient of the passive carrier cancellation module is
Figure BDA0001753868360000072
Wherein the propagation constant
Figure BDA0001753868360000073
λgThe transmission wavelength of the signal on the microstrip line; the transmission coefficient from the first port to the second port of the directional coupler is B; the expression of the modulated signal is therefore
Figure BDA0001753868360000074
The modulation signal and the carrier leakage signal are subjected to vector superposition at the second port of the directional coupler, and when the modulation signal and the carrier leakage signal have equal amplitudes and are opposite, depth cancellation can be achieved, so that if and only if
Figure BDA0001753868360000075
Or
Figure BDA0001753868360000076
When the depth compensation is achieved, the corresponding microstrip line length 1 and the resistance R are solved in two ways, namely
Figure BDA0001753868360000077
Due to the limited precision of the existing resistor or the factor of environment change, the carrier cancellation cannot reach complete cancellation. The carrier leakage signal after cancellation has smaller amplitude and is mixed with the echo signal of the tag to form a receiving signal;
in an alternative embodiment, the low noise amplifier 109 amplifies the received signal to improve the signal-to-noise ratio of the received signal; and the analog correlation demodulation circuit 120 is used for demodulating the received signal with low power consumption, optimizing the signal-to-noise ratio of the generated baseband signal, and enabling the phase to be randomly adjustable within (0 degrees and 180 degrees).
Furthermore, the received signal also contains the echo signal of the tag
Figure BDA0001753868360000081
And carrier leakage signal after cancellation
Figure BDA0001753868360000082
Where n (t) is noise; the phase deviation of the electrically-adjusted phase shifter in the initial state is 0 DEG, and the phase deviation is expressed as when a received signal reaches a 3dB coupler
Figure BDA0001753868360000083
Adjusting the phase shift of the received signal by using an electrically-adjusted phase shifter; performing sum and difference operation on the adjusted received signal and a local oscillator signal 122 by using a 3dB coupler to obtain a sum and difference signal, wherein the local oscillator signal is sent by a circuit for transmitting a carrier; the first power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum-difference signal to obtain a first baseband signal; and the second power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum-difference signal to obtain a second baseband signal. Suppose a local oscillator signalIs μ ejωtAfter passing through the 3dB coupler, 2 paths of sum and difference signals are output as follows:
Figure BDA0001753868360000084
Figure BDA0001753868360000085
the sum and difference signals respectively pass through a power detector to generate two paths of baseband signals:
BB + output signal (corresponding to the first baseband signal in the above):
Figure BDA0001753868360000086
BB-output signal (corresponding to the second baseband signal in the above):
Figure BDA0001753868360000087
Figure BDA0001753868360000091
and (3) subtracting the two baseband signals to obtain a BB signal (corresponding to the third baseband signal in the above step): 2 μ A (t) sin θ1+2βn(t)μsinθ2；
Optionally, if the phase shift of the electrically-adjusted phase shifter in the initial state is θ0From the above inference process, the baseband signal BB is derived: 2 μ A (t) sin (θ)1+θ0)+2βn(t)μsin(θ2+θ0) (ii) a Wherein 2 μ A (t) sin (θ)1+θ0) As a signal component, 2 β n (t) μ sin (θ)2+θ0) For noise components, there is one θ0So that the signal-to-noise ratio of the BB signal is maximized.
In an optional embodiment, a baseband filtering and amplifying circuit is used to receive the first baseband signal and the second baseband signal, and perform difference operation on the first baseband signal and the second baseband signal to obtain a third baseband signal, where a signal-to-noise ratio of the third baseband signal is greater than a signal-to-noise ratio of the first baseband signal and the second baseband signal. The first baseband signal is denoted by BB +; the second baseband signal is denoted by BB-; the third baseband signal is represented by the BB signal. The baseband filtering and amplifying circuit is also used for amplifying and filtering the baseband signals.
According to the method, the passive carrier cancellation scheme is adopted, so that the cost is extremely low, the structure is simple, and no extra noise is introduced; the related analog related demodulation circuit is adopted to replace the traditional IQ mixing demodulation, the requirement on the local oscillation power is reduced, the system power consumption is reduced, and the receiving dynamic range of the receiver is improved.
The present invention will be described in detail with reference to the following specific examples:
in order to solve the problem of carrier leakage in the related art, a vector modulation circuit is adopted to generate a signal with the same amplitude and opposite phase with a leakage signal, and vector superposition is carried out on the leakage signal, but the vector modulation circuit is realized through an adjustable attenuator, a variable capacitance diode or a PIN (positive-intrinsic-negative) transistor, power needs to be supplied to a chip, a control circuit is complex, and the hardware cost is high. In addition, the demodulation circuit of the ultrahigh frequency RFID reader-writer is demodulated through the traditional IQ quadrature mixing, and the method is mature in application and good in stability. However, the power requirement for the local oscillator is high, and amplification of the local oscillator signal is usually required, which increases the power consumption of the circuit, introduces corresponding noise, and deteriorates the sensitivity of the receiver. In addition, the dynamic range of the conventional IQ quadrature mixing demodulation circuit is limited, and the linearity of the mixer is limited, so that the mixer is saturated when the received power is high, and the performance of a receiver is affected.
The embodiment aims to provide the RFID receiving and demodulating circuit and the method which are low in cost, small in power consumption, large in dynamic range, high in sensitivity and simple in structure.
The main scheme is as follows:
a low-power consumption RFID receiving and demodulating circuit and method includes a directional coupler, a passive carrier cancellation module, a low noise amplifier, an analog correlation demodulating circuit, a baseband filtering amplifying circuit; the passive carrier cancellation module comprises a microstrip line TL and a resistor R and is used for reflecting and phase-shifting a coupling signal of a port 3 of the directional coupler, outputting the coupling signal from a port 4 of the directional coupler to generate a modulation signal, enabling the modulation signal to have the same amplitude and opposite phase with a carrier leakage signal, and carrying out vector superposition to eliminate the carrier leakage signal;
further, the characteristic impedance of the microstrip line TL is 50 ohms, the length 1 is related to the phase of the modulation signal, and changing the length 1 can change the phase of the modulation signal; the resistance value of the resistor R is related to the magnitude of the modulation signal, and the amplitude of the modulation signal can be modulated by changing the resistance value of the resistor R; selecting the proper length 1 of the microstrip line TL and the resistance value of the resistor R, the modulation signal and the carrier leakage signal can be close to equal amplitude and opposite phase, and therefore a good cancellation effect is achieved;
it is assumed that the leakage signal can be expressed as
Figure BDA0001753868360000101
The output signal of the 3 rd port of the directional coupler is Aej(ωt)Wherein A > α;
the generation process of the modulation signal is as follows; the output signal of the 3 rd port of the directional coupler is Aej(ωt)And the signal is reflected to the 3 rd port of the directional coupler through the passive carrier cancellation module and then is output from the 4 th port to form a modulation signal.
Suppose that the output signal of the 3 rd port of the directional coupler is: aej(ωt)(ii) a The reflection coefficient of the passive carrier cancellation module is
Figure BDA0001753868360000102
Wherein the propagation constant
Figure BDA0001753868360000103
λgThe transmission wavelength of the signal on the microstrip line; the transmission coefficient from the 3 rd port to the 4 th port of the directional coupler is B;
the expression of the modulated signal is therefore
Figure BDA0001753868360000104
The modulation signal and the carrier leakage signal are subjected to vector superposition at the 4 th port of the directional coupler, and when the modulation signal and the carrier leakage signal have equal amplitudes and are opposite, depth cancellation can be achieved, so that if and only if
Figure BDA0001753868360000111
Or
Figure BDA0001753868360000112
When the depth compensation is achieved, the corresponding microstrip line length 1 and the resistance R are solved in two ways, namely
Figure BDA0001753868360000113
Due to the limited precision of the existing resistor or the factor of environment change, the carrier cancellation cannot reach complete cancellation. The carrier leakage signal after cancellation has smaller amplitude and is mixed with the echo signal of the tag to form a receiving signal;
the low-noise amplifier amplifies the received signal and improves the signal-to-noise ratio; the analog correlation demodulation circuit comprises an electric tuning phase shifter, a 3dB coupler, a first power detector and a second power detector, wherein the electric tuning phase shifter is connected with the 3dB coupler and used for changing the phase of a received signal, and the phase is randomly adjustable within (0 degree and 180 degrees); the 3dB coupler is connected with the first power detector and the second power detector and is used for performing sum and difference operation on a received signal and a local oscillator signal to generate sum and difference signals V1 and V2; the first power detector and the second power detector respectively carry out square rate detection on the sum and difference signals V1 and V2 to generate baseband signals BB & lt + & gt and BB & lt- & gt; echo signal with received signal containing tag
Figure BDA0001753868360000114
And carrier leakage signal after cancellation
Figure BDA0001753868360000115
Where n (t) is noise.
Furthermore, the phase deviation of the electrically-adjusted phase shifter in the initial state is 0 DEG, and the time table when the received signal reaches the 3dB couplerShown as
Figure BDA0001753868360000116
Suppose the local oscillator signal is μ ejωtAfter passing through the 3dB coupler, 2 paths of sum and difference signals are output as follows:
Figure BDA0001753868360000117
Figure BDA0001753868360000118
the sum and difference signals respectively pass through a power detector to generate two paths of baseband signals:
BB + output signal:
Figure BDA0001753868360000121
BB-output signal:
Figure BDA0001753868360000122
and (3) subtracting the two paths of baseband signals to obtain BB signals: 2 μ A (t) sin θ1+2βn(t)μsinθ2
Further, if the phase shift of the initial state of the electrically-adjusted phase shifter is theta0From the above inference process, the baseband signal BB is derived: 2 μ A (t) sin (θ)1+θ0)+2βn(t)μsin(θ2+θ0) (ii) a Wherein 2 μ A (t) sin (θ)1+θ0) As a signal component, 2 β n (t) μ sin (θ)2+θ0) For noise components, there is one θ0Maximizing the signal-to-noise ratio of the BB signal;
and the baseband filtering and amplifying circuit is used for amplifying and filtering the baseband signal.
As shown in fig. 1, an RFID receiving and demodulating circuit with low power consumption includes: directional couplerThe device comprises a combiner, a passive carrier cancellation module, a low-noise amplifier, an analog related demodulation circuit and a baseband filtering amplification circuit. Wherein, the directional coupler is used for the isolation of the receiving and transmitting path and simultaneously used as a cancellation branch of the carrier cancellation, and the 6dB coupler 1D1304-6 can be selected and used [ S ]]The parameter matrix is
Figure BDA0001753868360000123
Where A, B, C is a real number, θ1∈(-π，π)；
The passive carrier cancellation module comprises a microstrip line TL and a resistor R, and is used for reflecting and phase-shifting a coupling signal at a port 3 of the directional coupler, outputting the coupling signal from a port 4 of the directional coupler, generating a modulation signal, enabling the modulation signal to have the same amplitude and opposite phase with a carrier leakage signal, and carrying out vector superposition to eliminate the carrier leakage signal; it is assumed that the leakage signal can be expressed as
Figure BDA0001753868360000124
The output signal of the 3 rd port of the directional coupler is Aej(ωt)Wherein A > α;
the generation process of the modulation signal is as follows; the output signal of the 3 rd port of the directional coupler is Aej(ωt)And the signal is reflected to the 3 rd port of the directional coupler through the passive carrier cancellation module and then is output from the 4 th port to form a modulation signal.
Suppose that the output signal of the 3 rd port of the directional coupler is: aej(ωt)(ii) a The reflection coefficient of the passive carrier cancellation module is
Figure BDA0001753868360000131
Wherein the propagation constant
Figure BDA0001753868360000132
λgThe transmission wavelength of the signal on the microstrip line; the transmission coefficient from the 3 rd port to the 4 th port of the directional coupler is B; the expression of the modulated signal is therefore
Figure BDA0001753868360000133
Modulating a signal with a carrierThe wave leakage signal is vector-superposed at the 4 th port of the directional coupler, and when the amplitude of the modulation signal is equal to that of the carrier leakage signal, the depth cancellation can be achieved, so that if and only if the amplitude of the modulation signal is equal to that of the carrier leakage signal, the carrier leakage signal is inverted, the depth cancellation can be achieved
Figure BDA0001753868360000134
Or
Figure BDA0001753868360000135
When the depth compensation is achieved, the corresponding microstrip line length 1 and the resistance R are solved in two ways, namely
Figure BDA0001753868360000136
The length l of the microstrip line is changed by cutting the microstrip line by a knife or other methods, and the value of the resistor R is changed to meet the requirement
Figure BDA0001753868360000137
Or
Figure BDA0001753868360000138
The carrier cancellation effect is very good;
due to factors of limited resistance accuracy or environmental changes, carrier cancellation cannot achieve complete cancellation. The carrier leakage signal after cancellation has smaller amplitude and is mixed with the echo signal of the tag to form a receiving signal; the noise amplifier amplifies the received signal to improve the signal-to-noise ratio, and sky67101-396LF of sky works company can be selected; the analog correlation demodulation circuit demodulates the received signal into a baseband signal; the baseband filtering and amplifying circuit is used for filtering and amplifying the baseband signal; the analog correlation demodulation circuit and the baseband filtering amplification circuit do not need special limitation, and can be common analog correlation demodulation circuits and baseband filtering amplification circuits.
Fig. 3 is a schematic structural diagram of an analog correlation demodulation circuit in this embodiment, and as shown in fig. 3, the analog correlation demodulation circuit (corresponding to the analog correlation demodulation circuit 120 in the above description) includes an electrically-tunable phase shifter 123, a 3dB coupler 124, a first power detector 125, and a second power detector 126, where the electrically-tunable phase shifter is connected to the 3dB coupler and is used for changing the phase of a received signal, and the phase is arbitrarily tunable within (0 ° and 180 °); the 3dB coupler is connected with the first power detector and the second power detector and is used for performing sum and difference operation on a received signal and a local oscillator signal to generate sum and difference signals V1 and V2; the first power detector and the second power detector respectively carry out square rate detection on the sum and difference signals V1 and V2 to generate baseband signals BB & lt + & gt and BB & lt- & gt;
because carrier cancellation can not reach complete cancellation, the amplitude of carrier leakage signals after cancellation is small, and the carrier leakage signals and echo signals of the tags are mixed together to form receiving signals; echo signal with received signal containing tag
Figure BDA0001753868360000141
And carrier leakage signal after cancellation
Figure BDA0001753868360000142
Where n (t) is noise; the phase shift of the electrically-tunable phase shifter in the initial state is theta0The received signal is represented as passing through the electrically-adjusted phase shifter
Figure BDA0001753868360000143
The 3dB coupler can be selected from an ultra-wideband 3dB coupler C0810J5003AHF of Anaren corporation.
Further, suppose that the local oscillator signal is μ ejωtAfter passing through the 3dB coupler, 2 paths of sum and difference signals are output as follows:
Figure BDA0001753868360000144
Figure BDA0001753868360000145
the sum and difference signals respectively pass through a power detector to generate two paths of baseband signals:
BB + output signal:
Figure BDA0001753868360000146
BB-output signal:
Figure BDA0001753868360000151
and (3) subtracting the two paths of baseband signals to obtain BB signals: 2 μ A (t) sin (θ)1+θ0) +2 β nt μ sin (θ 2+ θ 0), where 2 μ Atsin (θ 1+ θ 0) is the signal component and 2 β nt μ sin θ 2+ θ 0 is the noise component, there is one θ0Maximizing the signal-to-noise ratio of the BB signal;
the electrically-tunable phase shifter does not need special limitation, and can meet the requirements of frequency and phase shifting range by adopting a common electrically-tunable phase shifter.
Through the above description of the embodiments, those skilled in the art can clearly understand that the method according to the above embodiments can be implemented by software plus a necessary general hardware platform, and certainly can also be implemented by hardware, but the former is a better implementation mode in many cases. Based on such understanding, the technical solutions of the present invention may be embodied in the form of a software product, which is stored in a storage medium (e.g., ROM/RAM, magnetic disk, optical disk) and includes instructions for enabling a terminal device (e.g., a mobile phone, a computer, a server, or a network device) to execute the method according to the embodiments of the present invention.
Embodiments of the present invention also provide a storage medium having a computer program stored therein, wherein the computer program is arranged to perform the steps of any of the above method embodiments when executed.
According to another aspect of the embodiments of the present invention, there is also provided an electronic device for implementing the method for adjusting the power output signal, as shown in fig. 4, the electronic device includes a memory 402 and a processor 404, the memory 402 stores a computer program, and the processor 402 is configured to execute the steps in any one of the method embodiments through the computer program.
Alternatively, in the present embodiment, the above-described storage medium may be configured to store a computer program for executing the following steps.
S1, carrying out target operation on the received coupling signal to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting;
and S2, carrying out vector superposition on the modulation signal and the carrier leakage signal generated by the transmitting carrier to obtain a receiving signal.
Optionally, the storage medium is further configured to store a computer program for executing the steps included in the method in the foregoing embodiment, which is not described in detail in this embodiment.
Alternatively, in this embodiment, a person skilled in the art may understand that all or part of the steps in the methods of the foregoing embodiments may be implemented by a program instructing hardware associated with the terminal device, where the program may be stored in a computer-readable storage medium, and the storage medium may include: flash disks, Read-Only memories (ROMs), Random Access Memories (RAMs), magnetic or optical disks, and the like.
Optionally, in this embodiment, the electronic apparatus may be located in at least one network device of a plurality of network devices of a computer network.
Optionally, in this embodiment, the processor may be configured to execute the following steps by a computer program:
s1, carrying out target operation on the received coupling signal to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting;
and S2, carrying out vector superposition on the modulation signal and the carrier leakage signal generated by the transmitting carrier to obtain a receiving signal.
Alternatively, it will be understood by those skilled in the art that the structure shown in fig. 4 is merely illustrative, and the charger component is included in the electronic device. The structure of the electronic device is not limited. For example, the electronic device may also include more or fewer components (e.g., network interfaces, etc.) than shown in FIG. 4, or have a different configuration than shown in FIG. 4.
The storage 402 may be, but not limited to, specifically used for storing information such as sample characteristics of the item and the target virtual resource account number. As an example, as shown in fig. 4, the memory 402 may include, but is not limited to, the processing circuit element 408.
Optionally, the transmission device 406 is configured to receive or transmit power via a power line.
Optionally, in this embodiment, the storage medium may include, but is not limited to: various media capable of storing computer programs, such as a usb disk, a Read-Only Memory (ROM), a Random Access Memory (RAM), a removable hard disk, a magnetic disk, or an optical disk.
Embodiments of the present invention also provide an electronic device comprising a memory having a computer program stored therein and a processor arranged to run the computer program to perform the steps of any of the above method embodiments.
Optionally, the specific examples in this embodiment may refer to the examples described in the above embodiments and optional implementation manners, and this embodiment is not described herein again.
It will be apparent to those skilled in the art that the modules or steps of the present invention described above may be implemented by a general purpose computing device, they may be centralized on a single computing device or distributed across a network of multiple computing devices, and alternatively, they may be implemented by program code executable by a computing device, such that they may be stored in a storage device and executed by a computing device, and in some cases, the steps shown or described may be performed in an order different than that described herein, or they may be separately fabricated into individual integrated circuit modules, or multiple ones of them may be fabricated into a single integrated circuit module. Thus, the present invention is not limited to any specific combination of hardware and software.
The above description is only a preferred embodiment of the present invention and is not intended to limit the present invention, and various modifications and changes may be made by those skilled in the art. Any modification, equivalent replacement, or improvement made within the principle of the present invention should be included in the protection scope of the present invention.

Claims (15)

1. A radio frequency identification, RFID, receiving circuit, comprising:
the passive carrier cancellation module is connected with the directional coupler and used for carrying out target operation on a coupling signal received from a first port of the directional coupler to obtain a modulation signal, wherein the target operation comprises reflection and phase shifting;
the second port of the directional coupler is used for performing vector superposition on the modulation signal and a carrier leakage signal generated by a transmission carrier to obtain a receiving signal;
wherein the circuit further comprises:
the electrically-adjusted phase shifter is used for adjusting the phase shift of the received signal;
the 3dB coupler is connected with the electrically-adjusted phase shifter and is used for performing sum and difference operation on the adjusted received signal and a local oscillator signal to obtain a sum and difference signal, wherein the local oscillator signal is sent by a circuit for transmitting a carrier;
the first power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum and difference signal to obtain a first baseband signal;
and the second power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum and difference signal to obtain a second baseband signal.
2. The circuit of claim 1, wherein the passive carrier cancellation module comprises:
a microstrip line TL for receiving the coupled signal, wherein a length variation of the TL is associated with a phase variation of the modulated signal;
and the resistor R is connected with the microstrip line TL, wherein the resistance value change of R is related to the amplitude change of the modulation signal.
3. The circuit of claim 1,
the first port is further configured to receive the modulation signal reflected by the passive carrier cancellation module, and send the modulation signal to the second port, where the second port performs vector superposition on the modulation signal and a carrier leakage signal.
4. The circuit of claim 1, further comprising:
a low noise amplifier connected to the second port for amplifying the received signal;
and the analog correlation demodulation circuit is connected with the low noise amplifier and is used for demodulating the received signal to generate a baseband signal.
5. The circuit of claim 1, further comprising:
and the baseband filtering and amplifying circuit is used for receiving the first baseband signal and the second baseband signal and carrying out difference operation on the first baseband signal and the second baseband signal to obtain a third baseband signal.
6. The circuit of claim 1,
the directional coupler also comprises a third port for receiving a carrier leakage signal generated by a transmitting carrier and transmitting the carrier leakage signal to a fourth port.
7. A radio frequency identification, RFID, receiving circuit, comprising:
the low-noise amplifier is connected with the second port and used for amplifying a received signal, wherein the second port is positioned in the directional coupler and used for carrying out vector superposition on a modulation signal and a carrier leakage signal generated by a transmission carrier to obtain the received signal;
the analog correlation demodulation circuit is connected with the low noise amplifier and is used for demodulating the received signal to generate a baseband signal;
the analog correlation demodulation circuit includes:
the electrically-adjusted phase shifter is used for adjusting the phase shift of the received signal;
the 3dB coupler is connected with the electrically-adjusted phase shifter and is used for performing sum and difference operation on the adjusted received signal and a local oscillator signal to obtain a sum and difference signal, wherein the local oscillator signal is sent by a circuit for transmitting a carrier;
the first power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum and difference signal to obtain a first baseband signal;
and the second power detector is connected with the 3dB coupler and used for carrying out square rate detection on the sum and difference signal to obtain a second baseband signal.
8. The circuit of claim 7, further comprising:
and the baseband filtering and amplifying circuit is used for receiving the first baseband signal and the second baseband signal and carrying out difference operation on the first baseband signal and the second baseband signal to obtain a third baseband signal.
9. A signal processing method, comprising:
performing target operation on the received coupled signal to obtain a modulated signal, wherein the target operation comprises reflection and phase shifting;
vector superposition is carried out on the modulation signal and a carrier leakage signal generated by a transmitting carrier to obtain a receiving signal, wherein the amplitude of the carrier leakage signal carried in the receiving signal is smaller than that of the carrier leakage signal generated by the transmitting carrier;
wherein the method further comprises: the method further comprises the following steps:
adjusting a phase offset of the received signal;
performing sum and difference operation on the adjusted received signal and a local oscillator signal to obtain a sum and difference signal, wherein the local oscillator signal is sent by a circuit for transmitting a carrier;
carrying out square rate detection on the sum and difference signal to obtain a first baseband signal;
and carrying out square rate detection on the sum and difference signal to obtain a second baseband signal.
10. The method of claim 9, wherein performing a target operation on the received coupled signal to obtain the modulated signal comprises:
performing the target operation on the coupling signal by adjusting the length of the microstrip line TL to adjust the phase of the modulation signal;
and performing the target operation on the coupled signal by adjusting the resistance value of R so as to adjust the amplitude of the modulation signal.
11. The method of claim 9,
reflecting the modulation signal to a first port of a directional coupler, instructing the first port to transmit the modulation signal to a second port of the directional coupler, instructing the second port to vector-add the modulation signal and a carrier leakage signal, wherein the directional coupler is used for generating the coupling signal.
12. The method of claim 9, wherein after performing square-rate detection on the sum-difference signal to obtain a second baseband signal, the method further comprises:
and performing difference operation on the first baseband signal and the second baseband signal to obtain a third baseband signal.
13. The method of claim 12,
and adjusting the signal-to-noise ratio of the third baseband signal by using the phase offset value of the electrically-adjusted phase shifter, so that the signal-to-noise ratio of the third baseband signal reaches the highest value.
14. A computer-readable storage medium, in which a computer program is stored, wherein the computer program is configured to carry out the method of any one of claims 9 to 13 when executed.
15. An electronic device comprising a memory and a processor, wherein the memory has stored therein a computer program, and wherein the processor is arranged to execute the computer program to perform the method of any of claims 9 to 13.
CN201810878331.3A 2018-08-03 2018-08-03 RFID receiving circuit, storage medium, and electronic device Active CN109120290B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN201810878331.3A CN109120290B (en) 2018-08-03 2018-08-03 RFID receiving circuit, storage medium, and electronic device

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN201810878331.3A CN109120290B (en) 2018-08-03 2018-08-03 RFID receiving circuit, storage medium, and electronic device

Publications (2)

Publication Number Publication Date
CN109120290A CN109120290A (en) 2019-01-01
CN109120290B true CN109120290B (en) 2021-03-16

Family

ID=64853028

Family Applications (1)

Application Number Title Priority Date Filing Date
CN201810878331.3A Active CN109120290B (en) 2018-08-03 2018-08-03 RFID receiving circuit, storage medium, and electronic device

Country Status (1)

Country Link
CN (1) CN109120290B (en)

Families Citing this family (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110739982A (en) * 2018-07-20 2020-01-31 四川华大恒芯科技有限公司 devices and methods for realizing carrier wave elimination
CN114650201A (en) * 2022-03-14 2022-06-21 深圳市国芯物联科技有限公司 UHF RFID carrier suppression method

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN203786753U (en) * 2014-04-16 2014-08-20 成都信息工程学院 Ultra high frequency RFID reader system based on RFID middleware technology
CN205540822U (en) * 2016-01-20 2016-08-31 成都信息工程大学 RFID read -write system is handed to hyperfrequency

Family Cites Families (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR100961260B1 (en) * 2007-12-04 2010-06-03 한국전자통신연구원 Apparatus and method for suppressing carrier feedthrough in quadrature modulator system
JP5158052B2 (en) * 2009-09-24 2013-03-06 株式会社デンソーウェーブ Wireless communication device
CN102480308A (en) * 2010-11-23 2012-05-30 天津中兴软件有限责任公司 Method for restraining carrier leak
CN106100690B (en) * 2016-08-03 2018-06-05 江苏本能科技有限公司 Radio frequency offsets signal generation apparatus
CN107770106A (en) * 2016-08-15 2018-03-06 天津科畅慧通信息技术有限公司 The system and method that a kind of self-interference offsets
CN107832818B (en) * 2017-12-08 2024-03-29 江苏本能科技有限公司 Radio frequency identification reader-writer based on multi-region identification and road side parking management system

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN203786753U (en) * 2014-04-16 2014-08-20 成都信息工程学院 Ultra high frequency RFID reader system based on RFID middleware technology
CN205540822U (en) * 2016-01-20 2016-08-31 成都信息工程大学 RFID read -write system is handed to hyperfrequency

Also Published As

Publication number Publication date
CN109120290A (en) 2019-01-01

Similar Documents

Publication Publication Date Title
US6194980B1 (en) Quadrature hybrid RF combining system
CN105703719B (en) The method and apparatus for correcting envelope tracking system intermediate power amplifier load characteristic
US9467118B2 (en) RFID positioning and tracking apparatus and methods
CN102006102B (en) Coupling apparatus for dividing receiving and transmitting signals and control method thereof
US20060098765A1 (en) Interference cancellation in RFID systems
US11265055B2 (en) Flexible beamforming using frequency-division multiplexing
US8712356B2 (en) Apparatus and method for phase synchronization in radio frequency transmitters
US20100216413A1 (en) Leakage suppressing circuit
US9998171B2 (en) IBFD transceiver with non-reciprocal frequency transposition module
CN106301458A (en) Self-interference cancellation system and method
CN208938118U (en) A kind of passive self-interference cancellation circuit of RFID reader
CN109120290B (en) RFID receiving circuit, storage medium, and electronic device
US20200259520A1 (en) Rfid systems
Ensworth et al. Full-duplex bluetooth low energy (BLE) compatible backscatter communication system for mobile devices
Mirzavand et al. A ZERO-power sensor using multi-port direct-conversion sensing
WO2012050982A1 (en) Systems and methods of transmitter protection for wireless communications
US6894657B2 (en) Bi-directional vector modulator
CN108282171B (en) Self-interference offset device and electronic license plate reader-writer
US10521619B2 (en) Method and RFID reader for obtaining a backscatter modulated receive signal
GB2428940A (en) Carrier leakage cancellation in an RFID reader
Heino et al. Design of RF self-interference cancellation circuit for 100-W full-duplex radio at 225–400 MHz
CN101208869B (en) Transmitter device with quasi-constant LO leakage, for wireless communication equipment with a direct conversion architecture.
US20080076360A1 (en) Apparatus for combining two radios on a single antenna
Kruse et al. Distributed System Architecture for Software-Defined Radio/Radar with Optical Signal Distribution
You et al. Self‐interference cancellations for full‐duplex RF front‐end using a broadband phase‐slope adjuster

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant