CN101124794A - Pilot transmission and channel estimation for multiple transmitters - Google Patents
Pilot transmission and channel estimation for multiple transmitters Download PDFInfo
- Publication number
- CN101124794A CN101124794A CNA2005800318935A CN200580031893A CN101124794A CN 101124794 A CN101124794 A CN 101124794A CN A2005800318935 A CNA2005800318935 A CN A2005800318935A CN 200580031893 A CN200580031893 A CN 200580031893A CN 101124794 A CN101124794 A CN 101124794A
- Authority
- CN
- China
- Prior art keywords
- impulse response
- code
- pilot
- transmitter
- symbols
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
Images
Landscapes
- Mobile Radio Communication Systems (AREA)
Abstract
Each transmitter is assigned a time-only pilot code, a frequency-only pilot code, or a time-frequency pilot code to use for pilot transmission. The pilot codes may be pseudo-random, orthogonal, and/or cyclic-shift codes. To obtain a channel estimate for a transmitter using a time-frequency pilot code composed of a time-only code and a frequency-only code, a receiver multiplies a set of received symbols for each symbol period with a set of code values for the frequency-only code to obtain a set of detected symbols and performs an IDFT on the set of detected symbols to obtain an initial impulse response estimate. The receiver performs code matching on multiple initial impulse response estimates derived for multiple symbol periods with the time-only code to obtain a final impulse response estimate for the desired transmitter. The receiver retains the first L channel taps and zeroes out remaining channel taps, where L is the expected channel length.
Description
Cross Reference to Related Applications
[0001] This patent application claims priority to provisional application No.60/590,864, entitled "Pilot Transmission Schemes for a Multi-Antenna System," filed on 7, 22/2004, assigned to the assignee of the present application and hereby expressly incorporated by reference.
Technical Field
[0002] The present disclosure relates generally to communication, and more specifically to pilot symbol transmission and channel estimation in a wireless communication system.
Technical Field
[0003] In wireless communication systems, it is often desirable to estimate the response of a wireless channel from a transmitter to a receiver. Channel estimation may be used for various purposes, such as data detection, time synchronization, frequency correction, spatial processing, rate selection, and so on. Channel estimation is typically performed by transmitting a pilot signal that includes pilot symbols (pilot symbols) known a priori by both the transmitter and the receiver. The receiver can then estimate the channel gain based on the ratio of the received pilot symbols to the known pilot symbols.
[0004] Noise and interference typically affect the pilot signal. These effects degrade the quality of the channel estimates obtained by the receiver based on the received pilot signal. The sources of noise are many, such as wireless channels, receiver electronics, etc. In general, the noise impact problem can be addressed by transmitting the pilot signal in an appropriate manner and/or for a sufficient period of time so that the receiver can obtain the desired channel estimate quality. Interference may be caused by multiple transmitters transmitting pilot signals simultaneously. These transmitters may correspond to different base stations in the system, different antennas of the same base station, etc. In any case, the pilot signal from each transmitter is interference to the pilot signals from other transmitters. This pilot interference degrades the quality of the channel estimate.
[0005] Therefore, in this field, there is a need for a technique to obtain high quality channel estimates when there are multiple transmitters.
Disclosure of Invention
[0006] Techniques for transmitting pilot signals from multiple transmitters and channel estimation for the transmitters are described. These techniques can suppress or eliminate pilot interference and provide high quality channel estimates. Each transmitter is assigned a time-pilot code, frequency-pilot code, or time-frequency pilot code for pilot symbol transmission. The pilot codes may be pseudo-random codes, orthogonal codes (e.g., walsh codes), cyclic shift codes, etc., or a combination thereof. Each transmitter transmits its pilot signal on a fixed set of frequency subbands in a different symbol period or on a different set of frequency subbands and using the pilot code assigned to it.
[0007] The processing performed by the receiver to obtain the channel estimates for each associated transmitter, and the quality of the channel estimates, is dependent on the pilot code used by the transmitter and the manner in which the pilot signal is transmitted, as will be described below. For example, each transmitter may use a time-frequency pilot code comprised of a time code and a frequency code for pilot symbol transmission. The receiver may then obtain a channel estimate for the intended transmitter as follows. For each symbol period in which a pilot symbol is transmitted, the receiver obtains a set of received symbols for a set of frequency subbands used for pilot symbol transmission in the symbol period. The receiver multiplies the set of received symbols with a set of code values of a frequency code assigned to a desired transmitter and obtains a set of detected symbols. The receiver performs an inverse transform (e.g., an Inverse Discrete Fourier Transform (IDFT)) on each set of detected symbols to obtain an initial impulse response estimate. The receiver then Code matches (Code Matching) multiple initial impulse response estimates obtained over multiple symbol periods using the time Code assigned to the intended transmitter to obtain a final impulse response estimate for the intended transmitter, as described further below. The receiver may perform a truncation process to preserve the first L channel taps of the final impulse response estimate and zero the remaining channel taps. L may be the expected length of the channel impulse response of the expected transmitter. The receiver may also perform channel estimation in other manners, as will be described below.
[0008] Various aspects and embodiments of the invention are described in further detail below.
Brief Description of Drawings
[0009] The features and nature of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout.
[0010] Fig. 1 shows a block diagram of a transmitter and a receiver;
[0011] FIG. 2 illustrates an interlaced subband structure;
[0012] fig. 3 shows a fixed pilot symbol transmission scheme;
[0013] fig. 4 illustrates an interleaved pilot symbol transmission scheme;
[0014] fig. 5 illustrates a process for transmitting pilot signals;
[0015] FIG. 6 shows a channel estimation process for pilot code matching of time pilot codes in the time domain;
[0016] FIG. 7 illustrates a channel estimation process for pilot code matching of time-pilot codes in the frequency domain;
[0017] FIG. 8 illustrates a channel estimation process for frequency-directed codes;
[0018] FIG. 9 illustrates a channel estimation process for interlace steering using cyclic shift codes;
[0019] FIG. 10 shows a phase diagram of cyclic shift codes for four transmitters;
[0020] FIG. 11 shows channel impulse estimation for multiple transmitters using a cyclic shift code;
[0021] fig. 12 shows the channel estimation process for time-domain pilot codes.
Detailed Description
[0022] The term "exemplary" as used in this application means "serving as an example, instance, or illustration. Any embodiment or design described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments or designs.
[0023] The pilot symbol transmission and channel estimation techniques described herein may be used for various wireless communication systems. These techniques may be used for the downlink as well as the uplink. The downlink or forward link refers to the communication link from the base stations to the wireless devices, and the uplink or reverse link refers to the communication link from the wireless devices to the base stations. A base station is generally a fixed station that communicates with the wireless devices and may also be referred to as a Base Transceiver System (BTS), access point, or other terminology. A wireless device may be fixed or mobile and may also be referred to as a user terminal, mobile station, handset, or other terminology.
[0024] Communication systems are widely deployed to provide various communication services such as voice, packet data, and so on. These systems may be time, frequency, and/or code division multiple access systems capable of supporting simultaneous communication with multiple users by sharing the available system resources. Examples of such multiple-access systems include Code Division Multiple Access (CDMA) systems, multi-carrier CDMA (MC-CDMA), wideband CDMA (W-CDMA), high Speed Downlink Packet Access (HSDPA), time Division Multiple Access (TDMA) systems, frequency Division Multiple Access (FDMA) systems, and Orthogonal Frequency Division Multiple Access (OFDMA) systems.
[0025] The techniques described herein may also be used with various multicarrier modulation techniques, such as Orthogonal Frequency Division Multiplexing (OFDM). OFDM effectively partitions the overall system bandwidth into multiple (K) orthogonal frequency subbands. These sub-bands are also referred to as audio bands (tones), sub-carriers, frequency bins (bins), and frequency channels. With OFDM, each subband is associated with a respective subcarrier that may be modulated with data.
[0026] Fig. 1 shows a block diagram of a transmitter 110 and a receiver 150 in a wireless communication system 100. Transmitter 110 may be part of a base station and receiver 150 may be part of a wireless device. Alternatively, transmitter 110 may be part of a wireless device and receiver 150 may be part of a base station. In general, a base station may include any number of transmitters and any number of receivers. A wireless device may also include any number of transmitters and any number of receivers.
[0027] At the transmitter 110, a Transmit (TX) data processor 120 receives different types of data, e.g., traffic/packet data and overhead/control data. TX data processor 1 processes (e.g., encodes, interleaves, and symbol maps) the data to generate data symbols and provides the data symbols to an OFDM modulator (Mod) 130. As used herein, a data symbol is a modulation symbol of data, a pilot symbol is a modulation symbol of a pilot signal, a code symbol is a pilot symbol multiplied by a code value, and a modulation symbol is a complex value of one point in a signal group of a certain modulation scheme (e.g., M-PSK, M-QAM, etc.). The pilot symbols and the code symbols are known a priori by both the transmitting entity and the receiving entity. A pilot processor 122 receives the pilot codes assigned to transmitter 110, generates coded symbols based on the pilot codes, as described further below, and provides the coded symbols to an OFDM modulator 130.
[0028] OFDM modulator 130 multiplexes the data and code symbols onto the appropriate subbands and OFDM symbol periods and OFDM modulates the multiplexed symbols to generate OFDM symbols. In each OFDM symbol period, OFDM modulator 130 performs K-point Inverse Fast Fourier Transform (IFFT) on K multiplexed symbols corresponding to K total subbands and generates a transformed symbol including K time-domain samples. Each sample is a complex value to be transmitted in one sample period. OFDM modulator 130 then repeats a portion of each transformed symbol to form an OFDM symbol that includes K + C samples, where C is the number of samples repeated. The repeated portion, commonly referred to as a cyclic prefix, is used to overcome inter-symbol interference (ISI) caused by frequency selective fading. An OFDM symbol period (or symbol period, in short) is the duration of one OFDM symbol, which is equal to K + C sample periods. OFDM modulator 130 provides a stream of OFDM symbols to a transmitter unit (TMTR) 132. Transmitter unit 132 processes (e.g., converts to analog, amplifies, filters, and frequency upconverts) the OFDM symbol stream and generates a modulated signal, which is then transmitted from an antenna 134.
[0029] At receiver 150, an antenna 152 receives the modulated signal transmitted by transmitter 110 and other transmitters and provides a received signal to a receiver unit (RCVR) 154. Receiver unit 154 processes (e.g., filters, amplifies, frequency upconverts, and digitizes) the received signal and provides a stream of received samples. An OFDM demodulator (Demod) 160 OFDM demodulates the received samples. In each OFDM symbol period, OFDM demodulator 160 removes the cyclic prefix appended to the received OFDM symbol, performs K-point Fast Fourier Transform (FFT) on the K received samples in the OFDM symbol period, and generates K received symbols for all K subbands. OFDM demodulator 160 provides received data symbols to a detector 170 and received pilot symbols to a channel estimator 172.
[0030] Channel estimator 172 derives a channel estimate for the wireless channel between transmitter 110 and receiver 150 based on the received pilot symbols, as will be described further below. Detector 170 performs data detection (e.g., equalization or matched filtering) on the received data symbols using the channel estimates from channel estimator 172 and provides data symbol estimates, which are estimates of the data symbols transmitted by transmitter 110. A Receive (RX) data processor 180 processes (e.g., symbol demaps, deinterleaves, and decodes) the data symbol estimates and provides decoded data. In general, the processing by OFDM demodulator 160, RX data processor 180, and channel estimator 172 is complementary to the processing by OFDM modulator 130, TX data processor 120, and pilot processor 122, respectively, at transmitter 110.
[0031] Controllers 140 and 190 control the operation of transmitter 110 and receiver 150, respectively. Memory units 142 and 192 store program codes and data used by controllers 140 and 190, respectively.
[0032] Fig. 2 illustrates an interlace subband structure 200 that may be used for pilot symbol transmission in system 100. The overall system bandwidth is divided into K frequency subbands using OFDM. For simplicity, the following description assumes that all K subbands are available for transmission and are assigned an index of K =1. The K total subbands are arranged into S "interlaces," which may also be referred to as groups or sets of subbands. Each interlace includes P subbands with indices P =1. The P subbands in each interlace may be evenly distributed across K total subbands such that consecutive subbands in the interlace are spaced S subbands. The pilot signal may be transmitted on one interlace in each symbol period in which pilot symbols are transmitted. The pilot subbands are subbands used for pilot symbol transmission.
[0033] In general, system 100 may utilize an OFDM structure with any total number of pilot subbands. For clarity, the following description is for the OFDM structure shown in fig. 2 with P pilot subbands evenly distributed over K total subbands. The steering structure enables the receiver to estimate the channel gain over the entire system bandwidth and further enables the receiver to perform an inverse P-point transform rather than an inverse K-point transform of the channel estimation.
[0034] In the following description, the intended transmitter is the transmitter whose wireless channel is to be estimated. An interfering transmitter is a transmitter that interferes with the intended transmitter and affects the channel estimation. The receiver may receive pilot signals from any number of transmitters, which may correspond to the same or different transmitting entities (e.g., base stations). The receiver may derive channel estimates for multiple transmitters based on pilot signals received from the transmitters. Each transmitter is an intended transmitter for its channel estimate, but an interfering transmitter for the channel estimates of the other transmitters.
[0035] The transmitters may be synchronized and may transmit their pilot signals simultaneously over the same pilot subband in the same symbol period. The pilot signals will then interfere with each other. To facilitate channel estimation when multiple transmitters are present, each transmitter may multiply its pilot symbols with the pilot code assigned to that transmitter. The pilot codes of different transmitters are designed to obtain a better channel estimate for any associated transmitter.
[0036]Table 1 lists some pilot code types and schemes that may be used for pilot symbol transmission. A time-only code is a code that depends only on time or symbol period. A frequency-only code is a code that depends only on frequency or sub-band. Different hairTime code T of shooting machine i (n) may be mutually pseudo-random or orthogonal. Frequency codes F of different transmitters i (k) May be pseudo-random with respect to each other or may be different cyclic shift codes. A pseudo-random code (PN) generator may be used to generate the pseudo-random code. The orthogonal codes may be Walsh codes, orthogonal Variable Spreading Factor (OVSF) codes, and the like. The pilot code assigned to each transmitter may be a time code, a frequency code, or a combination of time and frequency codes. Various types of pilot codes are described below.
TABLE 1
| Guiding code square Table (A table) | Time code T i (n) | Frequency code F i (k) | Guide code S i (k，n) | Guide codes Model (III) |
| 1 | Pseudo-random | - | S i (k，n)＝T i (n) | |
| 2 | Orthogonal | - | S i (k，n)＝T i (n) | |
| 3 | - | Pseudo-random | S i (k，n)＝F i (n) | Frequency guideGuiding device Code |
| 4 | - | Cyclically shifted | S i (k，n)＝F i (n) | |
| 5 | Pseudo-random | Pseudo-random | S i (k，n)＝T i (n)·F i (k) | Time-frequency guidance Code |
| 6 | Orthogonal | Pseudo-random | S i (k，n)＝T i (n)·F i (k) | |
| 7 | Pseudo-random | Cyclically shifted | S i (k，n)＝T i (n)·F i (k) | |
| 8 | Orthogonal | Cyclically shifted | S i (k，n)＝T i (n)·F i (k) |
[0037] As shown in table 1, the time domain pilot code may be formed by the product of a time domain code and a frequency code as follows:
S i (k，n)＝T i (n)·F i (k) Wherein i = 0.., M, formula (1)
Wherein, T i (n) is the time code assigned to the ith transmitter;
F i (k) Is the frequency code assigned to the ith transmitter; and
S i (k, n) are the time-frequency pilot codes assigned to the ith transmitter.
For clarity, in the following description, the index i =1 corresponds to the intended transmitter and the index i =1. Each of the pilot values is a complex value having a unit measure, and thus
Wherein S is * (k, n) is the complex conjugate of S (k, n).
[0038] The transmitter may transmit its pilot signals using various pilot symbol transmission schemes. The manner in which the pilot signal is transmitted affects the processing of the channel estimate by the receiver. Two exemplary pilot symbol transmission schemes are described below.
[0039]Fig. 3 shows a fixed pilot symbol transmission scheme 300 that employs a single interlace for pilot symbol transmission in all symbol periods. The code symbols for each transmitter may be based on the time code T assigned to that transmitter i (n) and/or frequency code F i (k) To generate the signal. Fig. 3 shows the code symbols of the intended transmitter. The pilot signal may be transmitted in each symbol period, as shown in fig. 3, or only in certain symbol periods.
[0040]Fig. 4 shows an interleaved pilot employing two interlaces for pilot symbol transmission in alternating symbol periodsReference is made to symbol transmission scheme 400. In general, any number of interlaces may be used for pilot symbol transmission. The particular interlace used for pilot symbol transmission in each symbol period may be selected based on a predetermined pattern or in a pseudo-random manner. The code symbols for each transmitter may be based on the time code T assigned to that transmitter i (n) and/or frequency code F i (k) To be generated. The code symbols of the intended transmitter are shown in fig. 4.
[0041]For a fixed or interleaved pilot symbol transmission scheme, the P subbands used for pilot symbol transmission in each symbol period n are referred to as pilot subbands and are denoted K pilot (n) of (a). The pilot subbands are known to the transmitter and receiver.
[0042] Fig. 5 shows a process 500 for transmitting a pilot signal by a transmitter. Initially, the transmitter determines a time code and/or frequency code assigned to the transmitter for pilot symbol transmission (block 512). The transmitter determines the pilot subbands for each symbol period in which pilot symbols are transmitted (block 514). Different interlaces may be used in different symbol periods, and the frequency codes may depend on the particular subbands used for pilot transmission. The transmitter generates coded symbols for a symbol period for which pilot symbols are transmitted based on (1) the assigned time code and/or frequency code and (2) the pilot subbands for the symbol period (block 516). The transmitter then transmits the coded symbols on the pilot subbands (block 518).
1.Time-oriented code
[0043] For time-pilot codes, the pilot code is the same for all pilot subbands and only varies with symbol period. The code symbols transmitted by the intended transmitter may be represented as:
C(k，n)＝S 0 (n). P (K), where k.epsilon.K pilot (n, formula (2)
Where P (k) is a pilot symbol for subband k; and
c (k, n) is the coded symbol sent on subband k in symbol period n. Equation (2) assumes that the same pilot symbol P (k) is used for subband k in all symbol periods.
[0044] The code symbols transmitted by each interfering transmitter may be represented as:
X i (k，n)＝S i (n). P (k), where k.epsilon. Kpilot (n) and i =1,.. M, (equation 3)
Wherein, X i (k, n) are the coded symbols sent by the ith interfering transmitter on subband k in symbol period n.
[0045] At the receiver, the received symbols for the P pilot subbands (or simply, the received pilot symbols) may be expressed as:
Wherein Y (k, n) is the received pilot symbol corresponding to subband k in symbol period n;
h (k, n) is the channel gain from the intended transmitter to the receiver for subband k in symbol period n;
E d is the transmit power used by the intended transmitter for steering;
G i (k, n) is the channel gain from the ith interfering transmitter to the receiver corresponding to subband k in symbol period n;
E i is the transmit power used by the ith interfering transmitter for steering; and
w (k, n) is the noise at the receiver on subband k in symbol period n. It can be assumed that the noise W (k, N) is Additive White Gaussian Noise (AWGN) with zero mean and variance N 0 。
[0046] Each transmitter transmits its pilot signal at a power level determined by system specifications, regulatory constraints, etc. The prospective transmitter and the interfering transmitter may transmit their pilot signals at the same or different power levels. The pilot power level affects the quality of the channel estimate but does not generally change the receiver's processing of the channel estimate.
[0047]For simplicity, the following description assumes that all transmitters transmit their pilot signals at unity power, so that E for all i d ＝E i And =1. Further, assume that the value of the pilot symbol is P (k) =1+ j0, so that, for the time pilot code, C (k, n) = C (n) = T 0 (n) and X i (k，n)＝X i (n)＝T i (n) of (a). Then, the received pilot symbols mayExpressed as:
[0048] The receiver processes the received pilot symbols to obtain an estimate of the response of the wireless channel from the intended transmitter to the receiver. The wireless channel may be characterized by a channel impulse response or a channel frequency response. As used herein, a "channel impulse response" or "impulse response" is a time-domain response of a wireless channel, and "channel frequency response" and "frequency response" are frequency-domain responses of the wireless channel, consistent with conventional terminology. In a sampled data system, the channel frequency response is the Discrete Fourier Transform (DFT) of the channel impulse response.
[0049] The receiver may obtain a channel estimate for the intended transmitter by performing pilot code matching (or simply, code matching) in the time or frequency domain. Code matching refers to processing using the pilot code of the intended transmitter to obtain a channel estimate for the intended transmitter and to suppress or eliminate pilot interference from other transmitters.
[0050]Fig. 6 shows a flow 600 of code matching time-pilot codes in the time domain to derive channel estimates. For this embodiment, the receiver obtains an initial impulse response estimate (which is a sequence of time-domain samples) for each symbol period, and then code matches the channel taps to obtain a final impulse response estimate for the intended transmitter. Initially, the receiver pairs each symbolP received pilot symbols Y (K, n) with a number of periods n (where K ∈ K) pilot (n)) is inverse transformed to obtain an initial impulse response estimate y (l, n) for the symbol period (where l =1. The inverse transform may be an IDFT or other linear transform.
[0051] The initial impulse response estimate includes a phase ramp over P channel taps. The slope of the phase ramp depends on the index of the first steering subband. The phase ramp may be rejected by multiplying each channel tap (block 614), as follows:
Wherein k is n Is the index of the first pilot subband of symbol period n; and
Term "k" in the index of equation (6) n -1 "(instead of" k) n ") and" l-1 "(rather than the generation" l ") are due to the index starting from 1 rather than 0. Index k n Is fixed for all symbol periods of a fixed pilot symbol transmission scheme and staggered pilot symbol transmissionThe scheme is different for different symbol periods. It is also possible to perform both phase ramp culling and inverse transform in one step using a suitable inverse transform matrix.
[0052] The receiver code matches the initial impulse response estimate for multiple symbol periods (after the phase ramp is removed) using a time code (block 616). Code matching for each tap index (or tap position) l may be performed as follows:
Wherein the content of the first and second substances,
is the l-th channel tap of the intended transmitter; and
r is the length of the time code.
As shown in equation (7), the R channel taps in the initial impulse response estimate at each tap index/position/are first multiplied by the conjugate code value of the pilot code assigned to the intended transmitter. The results are then accumulated to obtain the channel taps for that tap index/position/.
[0053] It can be assumed that the channel impulse response of the intended transmitter has a length L, where L < P. In this case, the receiver may truncate the impulse response estimate obtained from equation (7) and set each of the channel taps L +1 through P to 0 (block 618). This truncation (truncation) process can be expressed as follows:
Wherein the content of the first and second substances,
is the l-th channel tap of the final impulse response estimate of the intended transmitter. This truncation process removes noise in extra channel taps with tap indices greater than L. If the channel taps L +1 through P are truncated, the receiver may code match only the first L channel taps in block 616, omitting block 618.
[0054] The time domain code matching shown in fig. 6 may be used for both fixed and interleaved pilot symbol transmission schemes. Any interlace may be used for pilot symbol transmission. The receiver rejects the phase ramp associated with the interlace used for pilot symbol transmission prior to code matching, as shown in fig. 6.
[0055]FIG. 7 shows in the frequency domainProcess 700 for code matching time-pilot codes to derive channel estimates. For this embodiment, the receiver code matches the received pilot symbols to obtain detected symbols and then derives an impulse response estimate for the intended transmitter. Most preferablyInitially, the receiver will receive P pilot symbols Y (K, n) for each symbol period n (where K ∈ K) pilot (n)) is multiplied by the pilot code of the intended transmitter to obtain the detected symbols Z (K, n) for that symbol period (where K e K) pilot (n)) (block 712). The detection symbol may be expressed as:
Wherein, C (n). C * (n)＝1，R i (n)＝X i (n)·C * (n)，Equation (9) shows that the P detected symbols for each symbol period are P channel gain estimates for P pilot subbands. These channel gain estimates are degraded by pilot interference and noise from other transmitters.
[0056] The receiver may then inverse transform (e.g., IDFT) the P detected symbols for each symbol period to obtain a P-tap initial channel impulse response estimate z (l, n) for that symbol period, where l =1. The initial impulse response estimate may be expressed as:
Where h (l, n) is the l-th channel tap of the intended transmitter;
g i (l, n) is the l channel tap of the i-th interfering transmitter;
z (k, n) is an estimate of the l-th channel tap of the intended transmitter;
In the formula (10), the term R i (n)·g i (l, n) for the l channel tap of the wireless channel from the l interfering transmitter to the receiver. Equation (10) indicates that each channel tap h (l, n) of the intended transmitter observes M corresponding channel taps of M transmitters.
[0057] The receiver strips out the phase ramp on the P channel taps for each symbol period (block 716), as follows:
Wherein, the first and the second end of the pipe are connected with each other,
is the l-th channel tap of symbol period n after phase ramp culling. The receiver will then take R channel taps over the pilot code length for each tap index/
(where n = 1.., P) are accumulated (block 718). The receiver can carry out truncationThe first L channel taps are retained and the remaining channel taps are zeroed out as shown in equation (8) (block 720).
[0058]For pilot code scheme 1 in table 1, the time pilot codes assigned to different transmitters are pseudo-random in time with each other. Thus, each interfering transmitter R i (n) randomize pilot interference observed by each channel tap. For pilot code scheme 2 in table 1, the time pilot codes assigned to different transmitters are orthogonal in time to each other. Thus, by encodingMatching cancels the channel estimate of the interfering transmitter unless time variations of the radio channels of the intended transmitter and the interfering transmitter affect orthogonality.
[0059] The length (R) of the time-pilot code determines the number of transmitters that can be identified by the receiver using code matching. The pilot code length is typically limited by (and chosen to be shorter than) the duration over which the wireless channel can be assumed to be static.
2.Frequency pilot code
[0060] For frequency pilot codes, the pilot code is the same for all symbol periods and varies only across subbands. The code symbols transmitted by the intended transmitter may be represented as:
C(k)＝S 0 (k) P (K), where K ∈ K pilot (n, formula (12)
Where C (k) is the code symbol transmitted by the intended transmitter on subband k.
[0061] The code symbols transmitted by each interfering transmitter may be represented as:
X i (k)＝S i (k) P (K), where K ∈ K pilot (n) and i =1,.., M, formula (13)
Wherein, X i (k) Is the code symbol sent by the ith interfering transmitter on subband k.
[0062] Fig. 8 shows a flow 800 for deriving a channel estimate for an intended transmitter for a frequency-directed code. Initially, the receiver multiplies the P received pilot symbols for a symbol period with the pilot code of the intended transmitter to obtain P detected symbols for the symbol period (block 812), as follows:
Wherein, C (k). C * (k) =1 and R i (k)＝X i (k)·C * (k)。
[0063] The receiver then inverse transforms (e.g., IDFT) the P detected symbols to obtain P tap initial impulse response estimates (block 814). The initial impulse response estimate may be expressed as:
Wherein v is i (l) (where l = 1.... P) is the P-tap "effective" channel impulse response of the ith interfering transmitter. The effective channel impulse response may be expressed as:
v i ＝g i r i equation (16)
Wherein the content of the first and second substances,g i is G for P pilot subbands i (k) P-point IDFT of (1);
r i is R for P pilot subbands i (k) P-point IDFT of (1);
denotes circular convolution; and
v i is V for P pilot subbands i (k)＝G i (k)·R i (k) P-point IDFT. In the following description, underlinedfDenotes the sequence of f (l) at the index l of the appropriate range. For example,v i is composed of v i (l) Wherein l =1.
[0064]The impulse response estimate is shown in equation (15)zEqual to the actual channel impulse response of the intended transmitterhPlus effective channel impulse response of each interfering transmitterv i Plus noise. The effective channel impulse response of each interfering transmitter is shown in equation (16)v i Equaling the actual channel impulse responseg i And the interference emissionTime domain code sequence of machiner i Is performed by the cyclic convolution of (c). The time domain code sequencer i Is a pilot code X of an interfering transmitter i (k) And the complex conjugate pilot code C of the intended transmitter * (k) IDFT of the product of (a).
[0065]For the pilot code scheme 3 in table 1, the frequency pilot codes assigned to different transmitters are pseudo-random with respect to each other in frequency. In this case, the time domain code sequence of each interfering transmitterr i Is to interfere with the L-tap channel impulse response of the transmitterg i The pseudo-random sequence over P taps is spread (or smeared). If L < P and via truncation process, the impulse response is estimatedzThe remaining channel taps L +1 through P are set to zero, the pilot interference in these remaining channel taps is effectively cancelled. Due to the spreading nature of the pseudo-random frequency pilot code,thus, pilot interference in channel taps 1 through L is suppressed.
[0066]For pilot code scheme 4 in table 1, frequency pilot codes assigned to different transmitters are defined to achieve pilot interference cancellation (rather than suppression) under certain conditions. Frequency-directed codes may be defined such that each interfering transmitter's time-domain code sequencer i Has the following form:
For the delta function shown in equation (17), the effective channel impulse response of each interfering transmitterv i Is only equal to being cyclically delayed (i.e., cyclic delay or time rotation) by T i Undesired channel impulse response of individual tapsg i . If T is for each interfering transmitter i > L and T i + L is less than or equal to P, theng i Is cyclically delayed by T i And the channel taps for these L cyclic delays fall between tap indices L +1 and P. Can be estimated by responding to the impulsezIs cut off and will zSet to zero and completely eliminate the L cyclic delayed channel taps (block 816). It is then expected that the final impulse response estimate for the transmitter will comprise onlyzAll remaining channel taps are zero. For example, if there is only one interfering transmitter (or M = 1) and if T 1 = L +1 and P > 2L, theng i Is cyclically delayed by L +1 taps and can be passedzThe channel taps L +1 to P of (a) are set to zero and are rejected.
[0067]The time domain code sequence shown in equation (17) may be providedr i The cyclic shift code of (a) can be expressed as:
Wherein, for example, T can be i Is selected as T i And (= i · L). The function given by equation (18) corresponds to a progressive phase shift over K total subbands. The phase shift changes at different rates (or with different phase slopes) for different transmitters. For each transmitter i, the P phase values in the P pilot subbands are calculated according to equation (18) and form the frequency code value for that transmitter. Will be the same function F i (k) For each transmitter, but the P phase values depend on the particular subband (or interlace) used for pilot symbol transmission.
[0068]FIG. 10 shows K total of four cyclic shift codes at four transmittersA plot of phase values across subbands. Function F in equation (18) i (k) A linear phase shift is generated over the K total subbands. Each transmitter i (where i = 0.. Multidot.m) has a phase slope 2 pi · T as in fig. 10 i is/K correlated, wherein T i = i · T and M =3. For the transmitter i, the phase value of each subband k is expressed as 2 π (k-1). T i and/K. The phase shift is such that the central subband at K = K/2+1 has a phase value of zero.
[0069] The cyclic shift code shown in equation (18) is applied over the P pilot subbands at the transmitter and results in a periodic or cyclic delay in the channel response estimate for the transmitter. The cyclic delay is relatively long and may be chosen to be longer than the channel length L. This is in contrast to the usual use of cyclic delays to achieve diversity, which typically requires a cyclic delay much less than the channel length L.
[0070] FIG. 11 shows channel impulse estimation for multiple transmitters using a cyclic shift code; the channel impulse responses of transmitters 0, 1 and M are shown in the upper part of fig. 11. The channel impulse response of each transmitter starts with a different tap index depending on its cyclic shift code. The received pilot symbols are multiplied by the cyclic shift code of the intended transmitter and the channel impulse responses of all transmitters are then cyclically shifted such that the channel impulse response of the intended transmitter begins at the tap index 1. The truncation of the channel taps with indices greater than L removes the channel impulse responses of all interfering transmitters.
[0071]Pilot interference cancellation using cyclic shift codes may be used for any number of interfering transmitters. The pilot code may be defined such that the time domain code sequence ri of the i-th interfering transmitter has a value of 1 for tap index L = i · L +1 and a value of 0 for all other tap indices. Then, the channel impulse response of the ith interfering transmitterg i Is cyclically shifted by i · L + 1. If P ≧ M · L, the impulse response can be estimated byzSet to zero and completely cancel the channel impulse responses of all M interfering transmitters.
[0072]If P < M.L, then interfere with a portion of the channel impulse response of the transmitterg i Can fall into the cycle to delayr i Subsequent impulse response estimationzWithin the first L taps of (a). This portion cannot be eliminated, so that the expected channel impulse response is reducedhIs estimated. P can be effectively increased by using interleaved pilot signals, as shown in fig. 4. The interleaved pilot signal increases the total number of pilot subbands observed by the receiver without increasing the number of pilot subbands used for any symbol periodThe number of pilot subbands in which pilot symbols are transmitted.
[0073] Fig. 9 shows a flow 900 for deriving a channel estimate for a desired transmitter that transmits a pilot signal using a cyclic shift code. The receiver multiplies the P received pilot symbols for each symbol period n by the cyclic shift code of the intended transmitter and obtains P detected symbols for that symbol period (block 912). The detection symbol may be expressed as:
wherein K ∈ K pilot (n, formula (19)
Wherein, F 0 * (k, n) is the cyclic shift code that the transmitter is expected to use for symbol period n. In each symbol period n, F 0 * (K, n) (where K ∈ K) pilot (n)) provides P phase values generated based on equation (18), where K pilot (n)＝{k n ，k n +S，2k n +S，...，k n +(P-1)·S}。F 0 (K, n) is a function of the symbol period, since K pilot (n) is a function of the symbol period.
[0074] Each interlace corresponds to a different set of pilot subbands and, therefore, to a different set of phase values for the cyclic shift code. The S sets of phase values for the S interlaces are all generated using the same cyclic shift code assigned to the transmitter, as shown in equation (18). The P received pilot symbols for each symbol period are multiplied by P phase values of the interlaces used for pilot transmission in that symbol period. Because different interlaces are used for pilot transmission in different symbol periods of the interleaved pilot signal, different sets of phase values are also used for different symbol periods.
[0075]If S is p A different interlace is at S p The receiver acquires the S for pilot symbol transmission in one symbol period p S of one symbol period p And P detection symbols are formed in a group. The receiver uses the S p The group of P detection symbols forms P T A sequence of detected symbols, whereinP T ＝S p P (block 914). This P T Each detected symbol corresponds to P T A different pilot subband. Then, the receiver pairs the P T The sequence of detected symbols is inverse transformed to obtain a symbol having P T Initial impulse response estimates for the channel taps (block 916). The receiver then performs truncation processing to tap the channel L +1 through P T Set to zero (block 918).
[0076]If P < M.L for a fixed pilot symbol transmission scheme or P for an interleaved pilot symbol transmission scheme T < M.L, the cyclic shift code can be designed to interfere with the time domain code sequence of the transmitterr i At e.g. T i = i.P/M or T i ＝i·P T There is a channel impulse response at/M separated by less than L taps. The semi-orthogonal pilot codes may be aligned withThe loop removes most of the interfering energy, and even if P < M.L or P T A larger amount of pilot interference cancellation can be achieved below M.L.
3.Time-frequency guide code
[0077] For time-frequency pilot codes, the pilot code varies across subbands and symbol periods. The code symbols transmitted by the intended transmitter may be represented as:
C(k，n)＝S 0 (K, n) P (K), where K ∈ K pilot (n), formula (20)
Where C (k, n) is the coded symbol transmitted by the intended transmitter on subband k in symbol period n.
[0078] The code symbols transmitted by each interfering transmitter may be represented as:
X i (k，n)＝S i (K, n) P (K), where K ∈ K pilot (n) and i =1,.., M, formula (21)
Wherein, X i (k, n) are the coded symbols sent by the ith interfering transmitter on subband k in symbol period n.
[0079] The received pilot symbols acquired by the receiver in the case of the transmitter using time-domain pilot codes are shown in equation (4). The receiver may obtain the channel estimate for the intended transmitter in various ways.
[0080]FIG. 12 shows a flow 1200 for deriving a channel estimate for a desired transmitter using a channel estimation derived from a frequency code F 0 (k) And time code T 0 (n) constituent time domain pilot codes S 0 (k, n) as shown in equation (1). Frequency code F 0 (k) May be a pseudo-random code, cyclic shift code, etc. The time code may be a pseudo-random code, an orthogonal code, etc. The process 1200 may be used for the pilot code schemes 5 through 8 in table 1.
[0081]The receiver combines the P received pilot symbols Y (K, n) for each symbol period n (where K ∈ K) pilot (n)) is multiplied by the frequency code to obtain P detected symbols Z (K, n) for the symbol period (where K ∈ K) pilot (n)) (block 1210). The detection symbol may be expressed as:
wherein K ∈ K pilot (n, formula (22)
Wherein, the first and the second end of the pipe are connected with each other,
for the interleaved pilot case, the frequency code F 0 (k) Can beA function of the symbol period of the cyclic shift code is used, as will be described below.
[0082]The receiver then inverse transforms (e.g., IDFT) the P detected symbols for each symbol period to obtain a P-tap initial impulse response estimate for the symbol period (block 1212). The receiver then strips out the phase ramp over the P channel taps for each symbol period (block 1214), as shown in equation (11). The receiver then uses the time code T 0 (n) time code matching (after phase ramping) the initial impulse response estimate for a plurality of symbol periods (block)1216). Each tap index/position/may be time code matched (e.g., similar to that shown in equation (7)). For each tap index/, the receiver multiplies the channel taps at that tap index for different symbol periods by the conjugate time code T 0 * (k) And accumulated over the length of the time code to obtain the channel taps for the tap index. The receiver then performs a truncation process to retain the first L channel taps and to zero out the remaining taps (block 1218).
[0083] In an alternative channel estimation scheme for time-domain pilot codes, the receiver multiplies the received pilot symbols with a frequency code and a time code to obtain detected symbols with all pilot codes removed. The receiver then accumulates the detected symbols over multiple symbol periods for each subband k to obtain a combined symbol for that subband. The receiver then performs an IDFT on the combined symbols to obtain an initial channel response estimate and further performs a truncation process to preserve the first L channel taps.
[0084]The pilot code schemes 7 and 8 in table 1 may be used in various ways. In one embodiment, the different delays T i Different cyclic shift codes F i (k) Are assigned to each transmitter. For this embodiment, the channel impulse response of each transmitter may be obtained using a cyclic delay (implemented by frequency code matching) and a truncation process. Time code matching over multiple symbol periods improves the quality of the channel estimate. In another embodiment, multiple groups of transmitters are formed and the same cyclic shift code is assigned to each group of transmitters. By combining different (e.g. orthogonal) time codes T i (n) further to the transmitters in each group. The receiver can be tuned by using cyclic shift codes F assigned to each group of transmitters i (k) Frequency code matching is performed to obtain channel estimates for the group. Frequency code matching cyclically shifts the channel impulse responses of all other transmitters to the higher taps so that these can be cancelled by the truncation process. The receiver may then perform the time-slicing by using the different time codes assigned to the individual transmitters in the selected groupCode matching to obtain channel estimates for these transmitters.
[0085] For all pilot code schemes, the receiver can obtain the final impulse response estimate for the intended transmitter as described above. The receiver may perform filtering to further improve the quality of the final impulse response estimate. The receiver may also perform a thresholding process to zero out channel taps with energy below a predetermined threshold. The receiver can extend the final impulse response estimate to length K by adding zeros. The receiver may then perform a K-point forward transform (e.g., DFT) on the spread impulse response estimate to obtain a frequency response estimate with channel gain for all K full subbands.
[0086] The pilot symbol transmission and channel estimation techniques described herein may be implemented in various ways. For example, these techniques may be implemented in hardware, software, or a combination of hardware and software. For a hardware implementation, the processing units used at the transmitter for pilot symbol transmission may be implemented within one or more Application Specific Integrated Circuits (ASICs), digital Signal Processors (DSPs), digital Signal Processing Devices (DSPDs), programmable Logic Devices (PLDs), field Programmable Gate Arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, other electronic units designed to perform the functions described herein, or a combination thereof. The processing units used for channel estimation at the receiver may also be implemented in one or more ASICs, DSPs, processors, and so on.
[0087] For a software implementation, the techniques described herein may be implemented with modules (e.g., procedures, functions, and so on). These software codes may be stored in memory units (e.g., memory units 142 and 192 in fig. 1) and executed by processors (e.g., controllers 140 and 190). The memory unit may be implemented within the processor or external to the processor.
[0088] Headings are included herein for ease of reference, which are intended to locate particular chapters. These headings are not intended to limit the scope of the concepts described therein under, which concepts may be applied to other sections throughout the specification.
[0089] The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.
Claims (57)
1. A method for channel estimation in a wireless communication system, comprising:
deriving a plurality of initial impulse response estimates corresponding to a plurality of symbol periods; and
code matching the plurality of initial impulse response estimates using a first code of a transmitter to obtain a final impulse response estimate for the transmitter.
2. The method of claim 1, wherein deriving the plurality of initial impulse response estimates comprises:
obtaining a set of received symbols corresponding to a set of frequency subbands in each of the plurality of symbol periods; and
inverse transforming the set of received symbols for each symbol period to derive an initial impulse response estimate corresponding to the symbol period.
3. The method of claim 1, wherein deriving the plurality of initial impulse response estimates comprises:
obtaining a set of received symbols corresponding to a set of frequency subbands in each of the plurality of symbol periods, wherein at least two different sets of frequency subbands are used for pilot symbol transmission; and
inverse transforming the set of received symbols for each symbol period to derive an initial impulse response estimate corresponding to the symbol period.
4. The method of claim 1, further comprising:
and eliminating the phase slope in each initial impulse response estimation of the plurality of initial impulse response estimations.
5. The method of claim 1, further comprising:
determining a set of frequency subbands used for pilot symbol transmission in each symbol period, wherein at least two different sets of frequency subbands are used for pilot symbol transmission in the plurality of symbol periods; and
based on the set of frequency subbands used for pilot symbol transmission in each symbol period, the phase ramp in the initial impulse response estimate corresponding to the symbol period is removed.
6. The method of claim 1, wherein code matching comprises: for each of a plurality of tap positions,
identifying a plurality of channel taps of the plurality of initial impulse response estimates located at the tap positions;
multiplying the plurality of channel taps by a plurality of coded values of the first code to obtain a plurality of intermediate values; and
accumulating the plurality of intermediate values to obtain channel taps at the tap positions in the final impulse response estimate.
7. The method of claim 1, further comprising:
deriving a plurality of groups of detected symbols corresponding to the plurality of symbol periods based on a second encoding of the transmitter.
8. The method of claim 7, wherein deriving the plurality of initial impulse response estimates comprises:
and carrying out inverse transformation on a group of detection symbols corresponding to each symbol period so as to derive an initial impulse response estimation corresponding to the symbol period.
9. The method of claim 7, wherein deriving the plurality of groups of detected symbols comprises: for each of the plurality of symbol periods,
determining a set of code values based on the second code and a set of frequency subbands used for pilot transmission in the symbol period; and
and multiplying a group of received symbols corresponding to the symbol period by the group of coded values to obtain a group of detected symbols corresponding to the symbol period.
10. The method of claim 1, further comprising:
reserving first L channel taps in the final impulse response estimate, wherein L is an integer greater than 1; and
setting the remaining channel taps in the final impulse response estimate to zero.
11. The method of claim 1, further comprising:
setting channel taps in the final impulse response estimate having a magnitude below a predetermined threshold to zero.
12. An apparatus in a wireless communication system, comprising:
a channel estimator for deriving a plurality of initial impulse response estimates corresponding to a plurality of symbol periods, and performing code matching on the plurality of initial impulse response estimates using a first code of a transmitter to obtain a final impulse response estimate of the transmitter.
13. The apparatus of claim 12, further comprising:
a demodulator to provide a set of received symbols for a set of frequency subbands in each of the plurality of symbol periods; and
wherein the channel estimator is configured to perform inverse transformation on the set of received symbols corresponding to each symbol period to derive an initial impulse response estimate corresponding to the symbol period.
14. The apparatus of claim 12, wherein the channel estimator is configured to: for each of a plurality of tap positions,
identifying a plurality of channel taps of the plurality of initial impulse response estimates located at the tap positions;
multiplying the plurality of channel taps by a plurality of coded values of the first code to obtain a plurality of intermediate values; and
accumulating the plurality of intermediate values to obtain channel taps at the tap positions in the final impulse response estimate.
15. The apparatus of claim 12, wherein the channel estimator is configured to reserve the first L channel taps in the final impulse response estimate and to set remaining channel taps in the final impulse response estimate to zero, wherein L is an integer greater than 1.
16. The apparatus of claim 12, wherein the channel estimator derives multiple sets of detected symbols for the symbol periods based on a second encoding of the transmitter, and wherein a set of detected symbols for each symbol period is inverse transformed to derive an initial impulse response estimate for the symbol period.
17. The apparatus of claim 12, wherein the first code of the transmitter is orthogonal to at least one other first code of at least one other transmitter.
18. The apparatus of claim 12, wherein the first code of the transmitter is pseudo-random with respect to at least one other first code of at least one other transmitter.
19. The apparatus of claim 16, wherein the second encoding of the transmitter is pseudo-random with respect to at least one other second encoding of at least one other transmitter.
20. The apparatus of claim 16, wherein the second encoding of the transmitter is a cyclic shift code.
21. An apparatus in a wireless communication system, comprising:
an initial impulse response estimate derivation module, configured to derive a plurality of initial impulse response estimates corresponding to a plurality of symbol periods; and
and the code matching module is used for performing code matching on the plurality of initial impulse response estimates by using a first code of a transmitter so as to obtain a final impulse response estimate of the transmitter.
22. The apparatus of claim 21, wherein means for deriving the plurality of initial impulse response estimates comprises:
an obtaining module, configured to obtain a set of received symbols corresponding to a set of frequency subbands in each of the plurality of symbol periods; and
and the inverse transformation module is used for carrying out inverse transformation on the group of received symbols corresponding to each symbol period so as to derive the final impulse response estimation of the symbol period.
23. The apparatus of claim 21, wherein the code matching module comprises: for each of a plurality of tap positions,
an identifying module that identifies a plurality of channel taps of the plurality of initial impulse response estimates that are located at the tap positions;
a multiplying module that multiplies the plurality of channel taps by a plurality of coded values of the first code to obtain a plurality of intermediate values; and
and the accumulation module accumulates the plurality of intermediate values to obtain the channel tap positioned at the tap position in the final impulse response estimation.
24. The apparatus of claim 21, further comprising:
a reserving module, configured to reserve first L channel taps in the final impulse response estimate, where L is an integer greater than 1; and
a setting module that sets remaining channel taps in the final impulse response estimate to zero.
25. The apparatus of claim 21, further comprising:
a detection symbol derivation module for deriving a plurality of groups of detection symbols corresponding to the plurality of symbol periods based on a second encoding of the transmitter; and
wherein the means for deriving the plurality of initial impulse response estimates comprises an inverse transform means for inverse transforming a set of detected symbols corresponding to each symbol period to derive an initial impulse response estimate for the symbol period.
26. A method for channel estimation in a wireless communication system, comprising:
deriving a plurality of detection symbols based on a plurality of received symbols corresponding to a plurality of pilot subbands and a cyclic shift code of a transmitter, wherein the plurality of pilot subbands is a subset of a plurality of frequency subbands of the system;
deriving an initial impulse response estimate based on the plurality of detected symbols; and
deriving a final impulse response estimate for the transmitter based on the initial impulse response estimate.
27. The method of claim 26, wherein deriving the plurality of detection symbols comprises:
determining a plurality of phase values based on the cyclic shift code and the plurality of pilot subbands; and
multiplying the plurality of received symbols by the plurality of phase values to obtain the plurality of detected symbols.
28. The method of claim 26, wherein deriving the final impulse response estimate comprises:
setting first L channel taps in the final impulse response estimate to first L channel taps in the initial impulse response estimate, wherein L is an integer greater than 1; and
setting the remaining channel taps in the final impulse response estimate to zero.
29. A method for channel estimation in a wireless communication system, comprising:
deriving a plurality of sets of detected symbols based on a plurality of sets of received symbols corresponding to a plurality of symbol periods and a cyclic shift code of a transmitter, wherein each set of received symbols corresponds to a set of pilot subbands, the pilot subbands being a subset of a plurality of frequency subbands of the system;
deriving at least one initial impulse response estimate based on the plurality of groups of detected symbols; and
a final impulse response estimate is derived based on the at least one initial impulse response estimate.
30. The method of claim 29, wherein deriving the plurality of sets of detection symbols comprises: for each of a plurality of symbol periods,
deriving a set of phase values based on the cyclic shift code and the set of pilot subbands used for pilot transmission in the symbol period;
multiplying a set of received symbols of the symbol period by the set of phase values to obtain a set of detected symbols of the symbol period.
31. The method of claim 29, wherein deriving the at least one initial impulse response estimate comprises:
averaging the multiple groups of detection symbols to obtain a detection symbol sequence; and
inverse transforming the sequence of detected symbols to derive a single initial impulse response estimate.
32. The method of claim 29, wherein deriving the at least one initial impulse response estimate comprises:
forming a sequence of detected symbols using the plurality of groups of detected symbols; and
inverse transforming the sequence of detected symbols to derive a single initial impulse response estimate.
33. The method of claim 32, wherein deriving the final impulse response estimate comprises:
setting first L channel taps in the final impulse response estimate to first L channel taps in the single initial impulse response estimate, wherein L is an integer greater than 1; and
setting the remaining channel taps in the final impulse response estimate to zero.
34. The method of claim 29, wherein deriving the at least one initial impulse response estimate comprises: for each of a plurality of encoded values of a second encoding for the transmitter,
forming a sequence of detected symbols using at least one group of detected symbols to which the coded values are applied; and
and performing inverse transformation on the detection symbol sequence so as to derive an initial impulse response estimation corresponding to the coding value.
35. The method of claim 34, wherein deriving the final impulse response estimate comprises:
code matching a plurality of initial impulse response estimates derived for the plurality of coded values using the second code.
36. The method of claim 29, further comprising:
reserving first L channel taps in the final impulse response estimate, wherein L is an integer greater than 1; and
setting the remaining channel taps in the final impulse response estimate to zero.
37. An apparatus in a wireless communication system, comprising:
a demodulator for providing a plurality of sets of received symbols corresponding to a plurality of symbol periods; and
a channel estimator for:
deriving a plurality of sets of detection symbols based on the plurality of sets of received symbols and a cyclic shift code of a transmitter;
deriving at least one initial impulse response estimate based on the plurality of groups of detected symbols; and
deriving a final impulse response estimate for the transmitter based on the at least one initial impulse response estimate.
38. The apparatus of claim 37, wherein the channel estimator is configured to form a sequence of detected symbols using the plurality of groups of detected symbols, and to inverse transform the sequence of detected symbols to derive a single initial impulse response estimate.
39. The apparatus of claim 37, wherein the channel estimator is configured to derive a plurality of initial impulse response estimates for a plurality of code values of a second code for the transmitter, and to code match the plurality of initial impulse response estimates using the second code.
40. An apparatus in a wireless communication system, comprising:
the detection symbol derivation module is used for deriving a plurality of groups of detection symbols based on a plurality of groups of receiving symbols corresponding to a plurality of symbol periods and the cyclic shift code of the transmitter;
an initial impulse response estimate derivation module that derives at least one initial impulse response estimate based on the plurality of groups of detected symbols;
a final impulse response estimate derivation module that derives a final impulse response estimate for the transmitter based on the at least one initial impulse response estimate.
41. The apparatus of claim 40, wherein means for deriving at least one initial impulse response estimate comprises:
a forming module for forming a detection symbol sequence by using the plurality of groups of detection symbols; and
and the inverse transformation module is used for performing inverse transformation on the detection symbol sequence to derive a single initial impulse response estimation.
42. The apparatus of claim 40, wherein means for deriving the at least one initial impulse response estimate comprises: for each of a plurality of encoded values of a second encoding for the transmitter,
a forming module for forming a detection symbol sequence using at least one group of detection symbols to which the coded values are applied; and
an inverse transform module inverse transforms the sequence of detected symbols to derive an initial impulse response estimate for the encoded values.
43. The apparatus of claim 42, wherein means for deriving the final impulse response estimate comprises:
a code matching module that code matches a plurality of initial impulse response estimates derived for the plurality of code values using the second code.
44. A method for channel estimation in a wireless communication system, comprising:
obtaining a plurality of receiving symbols corresponding to a plurality of frequency sub-bands;
deriving a plurality of detected symbols based on the plurality of received symbols and a frequency code of a transmitter;
deriving an initial impulse response estimate based on the plurality of detected symbols; and
performing a truncation process on the initial impulse response estimate to obtain a final impulse response estimate for the transmitter.
45. The method of claim 44, further comprising:
a plurality of initial impulse response estimates obtained for a plurality of symbol periods are averaged.
46. The method of claim 44, further comprising:
setting channel taps in the final impulse response estimate having a magnitude below a predetermined threshold to zero.
47. A method for transmitting pilot symbols in a wireless communication system, comprising:
generating a set of coded symbols for a set of pilot subbands based on a cyclic shift code assigned to a transmitter, wherein the set of pilot subbands is a subset of a plurality of frequency subbands in the system; and
the set of coded symbols is transmitted on the set of pilot subbands.
48. The method of claim 47, wherein generating the set of encoded symbols comprises: for each symbol period in which pilot symbol transmission is performed,
identifying the set of pilot subbands used for pilot transmission in the symbol period, wherein at least two different sets of pilot subbands are used for pilot transmission in different symbol periods;
determining a set of phase values based on the cyclic shift code and the set of pilot subbands for the symbol period; and
the set of encoded symbols is generated based on the set of phase values.
49. The method of claim 47, wherein generating the set of encoded symbols comprises: for each symbol period in which pilot symbol transmission is performed,
the set of code symbols is generated based on the cyclic shift code and a code value of a second code assigned to the transmitter.
50. An apparatus in a wireless communication system, comprising:
a pilot processor that generates a set of coded symbols for a set of pilot subbands based on a cyclic shift code assigned to a transmitter, wherein the set of pilot subbands is a subset of a plurality of frequency subbands in the system; and
a transmitter unit to transmit the set of coded symbols on the set of pilot subbands.
51. The apparatus of claim 50, wherein the pilot processor is configured to, for each symbol period in which pilot symbol transmission occurs:
identifying the set of pilot subbands used for pilot transmission in the symbol period;
determining a set of phase values based on the cyclic shift code and the set of pilot subbands for the symbol period; and
the set of encoded symbols is generated based on the set of phase values, wherein at least two different sets of pilot subbands are used for pilot transmission in different symbol periods.
52. The apparatus of claim 50, wherein the pilot processor is configured to generate the set of code symbols for each symbol period in which pilot symbol transmission is performed based on the cyclic shift code and a second coded code value assigned to the transmitter.
53. The apparatus of claim 52, wherein the second code is orthogonal to at least one other second code assigned to at least one other transmitter.
54. The apparatus of claim 52, wherein the second code is pseudo-random with respect to at least one other second code assigned to at least one other transmitter.
55. An apparatus in a wireless communication system, comprising:
a generating module that generates a set of coded symbols for a set of pilot subbands based on a cyclic shift code assigned to a transmitter, wherein the set of pilot subbands is a subset of a plurality of frequency subbands of the system; and
a transmit module transmits the set of coded symbols over the set of pilot subbands.
56. The apparatus of claim 55, wherein the means for generating the set of coded symbols comprises: for each symbol period in which pilot symbol transmission is performed,
an identifying module that identifies the set of pilot subbands used for pilot transmission in the symbol periods, wherein at least two different sets of pilot subbands are used for pilot transmission in different symbol periods;
a determining module that determines a set of phase values based on the cyclic shift code and the set of pilot subbands for the symbol period; and
a module that generates the set of encoded symbols based on the set of phase values.
57. The apparatus of claim 55, wherein the means for generating the set of coded symbols comprises: for each symbol period in which pilot symbol transmission is performed,
and means for generating the set of code symbols based on the cyclic shift code and a code value of a second code assigned to the transmitter.
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US59086404P | 2004-07-22 | 2004-07-22 | |
| US60/590,864 | 2004-07-22 | ||
| US11/022,146 | 2004-12-22 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN101124794A true CN101124794A (en) | 2008-02-13 |
Family
ID=39086116
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNA2005800318935A Pending CN101124794A (en) | 2004-07-22 | 2005-07-11 | Pilot transmission and channel estimation for multiple transmitters |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN101124794A (en) |
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102780660A (en) * | 2012-06-19 | 2012-11-14 | 深圳数字电视国家工程实验室股份有限公司 | Channel estimation method and device |
| CN102067539B (en) * | 2008-07-01 | 2014-10-29 | 高通股份有限公司 | Adaptive thresholding for OFDM channel estimation |
| CN106855620A (en) * | 2015-12-08 | 2017-06-16 | 德尔福技术有限公司 | Remnants for automated vehicle MIMO radar offset |
| CN111431827A (en) * | 2020-05-07 | 2020-07-17 | 中国人民解放军63921部队 | FFT-based step-by-step progressive high-precision frequency estimation method |
-
2005
- 2005-07-11 CN CNA2005800318935A patent/CN101124794A/en active Pending
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102067539B (en) * | 2008-07-01 | 2014-10-29 | 高通股份有限公司 | Adaptive thresholding for OFDM channel estimation |
| CN102780660A (en) * | 2012-06-19 | 2012-11-14 | 深圳数字电视国家工程实验室股份有限公司 | Channel estimation method and device |
| CN102780660B (en) * | 2012-06-19 | 2015-09-09 | 深圳数字电视国家工程实验室股份有限公司 | A kind of channel estimation methods and device |
| CN106855620A (en) * | 2015-12-08 | 2017-06-16 | 德尔福技术有限公司 | Remnants for automated vehicle MIMO radar offset |
| CN106855620B (en) * | 2015-12-08 | 2021-09-21 | 安波福技术有限公司 | Residual cancellation for automated vehicle MIMO radar |
| CN111431827A (en) * | 2020-05-07 | 2020-07-17 | 中国人民解放军63921部队 | FFT-based step-by-step progressive high-precision frequency estimation method |
| CN111431827B (en) * | 2020-05-07 | 2023-03-07 | 中国人民解放军63921部队 | FFT-based step-by-step progressive high-precision frequency estimation method |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| US7418046B2 (en) | Pilot transmission and channel estimation for multiple transmitters | |
| US7339999B2 (en) | Pilot transmission and channel estimation for an OFDM system with excess delay spread | |
| EP2961075B1 (en) | Pilot transmission and channel estimation for a communication system utilizing frequency division multiplexing | |
| JP5048481B2 (en) | Staggered pilot transmission for channel estimation and time tracking | |
| KR100630196B1 (en) | Apparatus and method for acquiring synchronization in a mobile communication system using an orthogonal frequency division multiplexing scheme | |
| US7630465B2 (en) | Wireless communications device providing time and frequency-domain channel estimates interpolation and related methods | |
| US20090154625A1 (en) | Joint estimation apparatus of channel and frequency offset based on multiband-orthogonal frequency division multiplexing and thereof | |
| WO2005081486A1 (en) | Method and apparatus providing time domain interpolated channel estimation with advanced noise suppression for multicarrier transmissions | |
| JP2011502453A (en) | Synchronization in broadcast OFDM systems using time division multiplexed pilots | |
| Filippi et al. | OFDM symbol synchronization using frequency domain pilots in time domain | |
| CN101124794A (en) | Pilot transmission and channel estimation for multiple transmitters | |
| JP2011502454A (en) | Synchronization in broadcast OFDM systems using time division multiplexed pilots | |
| KR101203861B1 (en) | method for channel estimation, channel estimator, mobile station and base station | |
| Rangaraj et al. | Low-complexity channel estimation for transmitter diversity OFDM systems | |
| Fan et al. | Modified UWB channel estimation | |
| KR20050119592A (en) | Apparatus and method for channel estimation in a mobile communication system using an frequency hopping - orthogonal frequency division multipl access scheme | |
| Inderjeet et al. | Orthogonal Frequency Division Multiplexing: An Overview |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| C06 | Publication | ||
| PB01 | Publication | ||
| C10 | Entry into substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| REG | Reference to a national code |
Ref country code: HK Ref legal event code: DE Ref document number: 1112349 Country of ref document: HK |
|
| C02 | Deemed withdrawal of patent application after publication (patent law 2001) | ||
| WD01 | Invention patent application deemed withdrawn after publication |
Open date: 20080213 |
|
| REG | Reference to a national code |
Ref country code: HK Ref legal event code: WD Ref document number: 1112349 Country of ref document: HK |