CA1141437A - Large dynamic range multiplier for a maximal-ratio predetection diversity combiner - Google Patents
Large dynamic range multiplier for a maximal-ratio predetection diversity combinerInfo
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- CA1141437A CA1141437A CA000360720A CA360720A CA1141437A CA 1141437 A CA1141437 A CA 1141437A CA 000360720 A CA000360720 A CA 000360720A CA 360720 A CA360720 A CA 360720A CA 1141437 A CA1141437 A CA 1141437A
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Abstract
Abstract An improved large dynamic range multiplier is dis-closed that provides substantially ideal multiplication for signals within a maximal-ratio predetection diversity combiner. Linear product multiplication is necessary in the second mixing stage of the maximal-ratio predetection diversity combiner, where it is essential that the mixer product signal be proportional to the product of its input signals, which each have a dynamic range in excess of 40 db. The multiplier, which in the preferred embodiment of the present invention is a field-effect-transistor device, provides a product signal that has a linear dynamic range approaching 130 db, far in excess of the 80 db necessary to accommodate typical input signals having a 40 db dynamic range.
Description
3~
LARGE DYNAMIC RANGE MULTIPLIER FOR A
MAXIMAL-RATIO PREDETECTION DIVERSITY COMBINER
Background of the Invention This invelltion relates to diversity combining systems and, more particularly, to a large dynamic range mult.iplier for maximal-ratio divers.ity comb.iners.
The need for space d.iversity combining arises in moblle radio systems because the rad.io-frequency (RF) s.ignal pa-th between a mobile transmitter and a base receiver is general-ly not l.ine of sight, but instead consists of many reflected and scattered RF signal paths having varying amplitudes and phases. Furthermore, in mobile radio systems operating at relatively high frequencies, for example at 800 MHz, deep, rapid fading, commonly referred to as Rayleigh fad.ing, must be contended with. By utilizing an antenna array having space diversity, the foregoing effects may be substantially reduced. According to space diversity, antennas of an an-tenna array are spaced at predetermined distances from one another, for example, at a distance of at least one-quarter wavelength from one another. The probability that deep fades will occur simultaneously at all antennas of a space-diversity antenna array will be extremely low. Thus, a composite signal formed by coherently combining each of the RF signals from a space-diversi-ty antenna array w.ill theo retically have a s.ignal level at least as high as the strongest RF signal received by the antenna array.
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LARGE DYNAMIC RANGE MULTIPLIER FOR A
MAXIMAL-RATIO PREDETECTION DIVERSITY COMBINER
Background of the Invention This invelltion relates to diversity combining systems and, more particularly, to a large dynamic range mult.iplier for maximal-ratio divers.ity comb.iners.
The need for space d.iversity combining arises in moblle radio systems because the rad.io-frequency (RF) s.ignal pa-th between a mobile transmitter and a base receiver is general-ly not l.ine of sight, but instead consists of many reflected and scattered RF signal paths having varying amplitudes and phases. Furthermore, in mobile radio systems operating at relatively high frequencies, for example at 800 MHz, deep, rapid fading, commonly referred to as Rayleigh fad.ing, must be contended with. By utilizing an antenna array having space diversity, the foregoing effects may be substantially reduced. According to space diversity, antennas of an an-tenna array are spaced at predetermined distances from one another, for example, at a distance of at least one-quarter wavelength from one another. The probability that deep fades will occur simultaneously at all antennas of a space-diversity antenna array will be extremely low. Thus, a composite signal formed by coherently combining each of the RF signals from a space-diversi-ty antenna array w.ill theo retically have a s.ignal level at least as high as the strongest RF signal received by the antenna array.
~' . ' ~ -. :
-2- ~ 3'7 One practical prior art technique of coherently combin-ing the antenna RF signals from a space-diversity antenna array is known as "equal-gain predetect.ion diversity combln-ing". Exemplary equal-gain predetection diversity combiners are those described in an art.icle by D. Brennan entitled, "Linear Diversity Comblning Techniques", publ.ished in IRE
Proceedings, June 1959, at pp. 1075 to 1101 and in U.S.
patent no. 3,471,788 to W.S. B.ickford et al. In these prior art comb.iners, the antenna slgnals are converted to inter-mediate frequency (IF) signals which are then cophased wi-th one another and thereafter linearly combined to provide a composite IF signal. For example, the IF signals developed from each antenna RF signal may be phase allgned w.ith a locally generated signal of a reference frequency, or may be phase aligned to a selected one of the IF signals, or may be phase aligned with respect to the compos.ite IF s.ignal. Once the IF signals from each antenna RF signal are cophased with one another, they may then be linearly added by appropriate circuitry to prov.ide a coherent composite IF signal which is the vector sum of the individual IF signals.
In order to cophase each IF signal, prior art equal-gain predetection diversity combiners include circuitry, commonly referred to as a "branch", for dividing the IF sig-nal into first and second portions, mixing the first portion with a reference signal to provide a firs-t product signal that has a phase equal to the difference in phase between the first portion and the reference signal, and mixing the first product signal with the second portion of the IF s.ig-nal to provide a second product signal that will be cophased with the reference signal. S.ince the second product signals of each branch are cophased with one another, they may then be linearly added by appropriate circuitry to provide the composite coherent IF signal. If the second portion of the IF signal and the first product signal were each amplified i ' . ~3~ ~14~37 , l,inearly so that the magnitude of each would be proportional to the .input IF signal, the magnitude of the second product signal would theoretically be proport,ional to the square of the magnitude of the ,input IF signal. However, ,in the con-ceptual design of the prior art equal-gain comb.iner, the f.irst product signal is ampl.itude l.im.ited prior to the input to the second stage of mixing. Consequently, the second product s.ignal will be directly proportional to the magni-tude of the input IF signal rather than to .its square.
In such equal-gain combining systems, it is necessary that all of the antenna ~F signals must have substantially the same mean signal level due to the fact that the first product signal is amplltude l.imited prior to the second mix-.ing. ~eca~se the first product signal is amplitude limited, the second product signal from the second stage of mix,ing will not be proportional to the square of the magnitude of the input IF signal. Thus, if IF signals rece,ived by all branches do not have substantially the same mean signal level, the composite IF signal may be significantly degraded in signal-to-noise ratio s.ince a weak signal received by one branch w,ill be weighted substantially equally with a strong signal received by another branch.
The foregoing inadequacy of prior art equal-gain pre-detection diversity combiners may be improved by utilizing the prior art combining technique known as "maximal-ratio predetection combining". In such maximal-ratio predetection combining systems, signals are not limited prior to the sec-ond mixing in each of the branches. It is desired that all signals are proportionally related to the input IF signal so that the magnitude of the second product signal will be pro-portional to the square of the magnitude of the input IF
signal. As a result, branches receiving strong signals will receive more emphasis than branches receiving weak signals.
~4~ 11~3~
Since Rayleigh fading experienced in 800 MHz systems may c~use instantaneous amplitude variations between the RF
signals received at different antennas of a space-diversity antenn~ array that are in excess of 40 decibels (db~, the linear dynamic range of the branch circuitry must accommo-date IF signals ha~ing at least a 40 db dynamic range ~nd second product signals having amplitude variations in excess of 80 db, which is twice the dynamic range of the input IF
signals due to the squaring by the second mixing operation.
Thus, the particular circuit implementation for providing the second mixing operation must provide substantially idealized multiplication over an extremely large dynamic range.
A prior art product multiplier capable of providing the .desired performance over an output dynamic range in excess of 80 db has not been practically achieved in the past.
Commercially available line~r inte~rated-circuit balanced mixers such as the Motorola~ C15g6 have been utilized as the second mixer in maximal ratio combiners designed for mili-tary applications~ These integrated circuit mixers consist of a quad differential amplifier with cross-coupled outputs to provide full-wave balanced multiplication of the two input signals. Each differential pair is powered by a con-stant current source. Such a mixer will not accommodate input signals having a dynamic range in excess of 30 db, since its linear output dynamic range is only 50 to 60 db.
A doubly~balanced FET mixer, such as that described by Highleyman and Jacob in the article~ "An Analog Multiplier Using Two Field Effect Transistors", IRE Transactions on Communications Systems, Vol. CS-10, pp. 311-317, September, 1962, may also be used for the second mixing operation with some expected impr~vement in dynamic range, but without any appreciable reduction in complexity or cost~ :
Accordingly, it is an object of the present invention to provide an improved low-cost, large dynamic range product ,~
multiplier for a maximal-ratio predetection diversity com-biner that coherently combines a plurality of amplitude and phase varying input signals of substantially the same fre-quency.
It is another object of -the present invention to pro-vide an improved low-cost, large dynamic range product mul-tiplier for a ~aximal-ratlo predetectlon divers.ity combiner su.itable for use in a d.iversity recelver for coherently com-b.ining RF signals havlng a dynamic range in excess of forty decibels (40 dB).
Summary of the Invention According to the .invent.ion, there is provided a maxl-mal-ratio predetection diversity combiner for coherently combining over a predetermined dynamic range a plurality of input signals of substantially the same frequency that have unknown and varying phases and magnitudes with respect to one another. The plurality of input signals are co-phased with each other within the combiner so that these signals are phase-coherent with each other prior to being linearly summed together at the output of the combiner. Furthermore, the amplitude of each input signal is, in effect, squared within the combiner prior to the l.inear summation to give greater emphasis to the siynals having the larger magni-tudes.
For each input signal, the maximal-ratio predetection combiner includes: circuitry for dividing the input signal into first and second portions; a first mixer for multiply-ing the first portion of the lnput slgnal with a reference signal that is produced within the combiner to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference sig-nal; a filter network for prov.iding a variable phase shift , to the first product signal with the phase shift being a function of the frequency of the first product signal; and a second mixer for multiplying the second portion of the input signal with the phase-shifted first product signal to provide a second product signal that is substantially independent of the phase of the origianl input signal. The second product sig~als, which are each cophased with the reference signal, may then be linearly combined by combining circuitry to provide a composite coherent IF signal.
The second mixer of the maximal-ratio diversity combiner preferably is a field-effect transistor devic~ that propor-tionally multiplies, over the predetermined dynamic range of the combiner, the second portion of the input signal and the first product signal to provide the second product signal. Thus, the magni-tude of the second product slgnal is proportional to the square o the magnitude o~ the respective input signal. Therefore, since strong input signals receive more emphasis than weak input signals, the signaL-to-noise degradation from the weak input signals is greatly reduced. Since a single FET device provides essentially ideal multiplication, the second product signal is proportional to the square of the respective input signal over the entire dynamic range of the input signal, which, in some applications, may be in excess of forty decibels (40 db) for the input signal.
More particularly, there is provided:
A maximal-ratio predetection diversity combiner for coherently combining a plurality of input signals each having a linear dynamic range exceeding forty decibels (40 db), having substantially the same predetermined frequency and further having unknown and varying phases and magnitudes with respect to one another, said maximal ratio predetection diversity combiner comprising:
means for generating a reference signal having a pre-determined reference frequency for each input signal;
.~
43~
-6a-means for dividing each input signal into first and second portions;
first means for multiplying the first portion of each input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product signal;
second means for multiplying the second portion of each input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and substantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (F~
said FET being predeterminedly biased for multiplying the second portion of the input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the ~o magnitudes of the second portion of the input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each input signal to provide a phase coherent composite signal.
There is also provided:
A diversity receiving system comprising antenna array means having a plurality of substantially independent antennas for receiving a radio signal of a predetermined frequency, each antenna providing an input signal, the input signals from the antennas having unknown and varying phases and magnitudes with respect to one another; converting means coupled to the input signals from the antennas for converting the frequency of the in-put signals to an intermediate frequency; intermediate frequency amplifying and filtering means coupled to the converted input signals from the converting means for filtering the converted ,' .
~' :
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-6b-input signals; and maximal-ratio predetection combining means coupled to the filtered input signals from the intermediate frequency amplifyling and filtering means for coherently combining the filtered input signals to provide a coherent composite signal, said combining means including:
means for generating a reference signal having a predeter-mined reference frequency for each filtered input signal;
means for dividing each filtered input signal into first and second portions;
first means for multiplying the first portion of each filtered input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the filtered input signal and the reference signal;
lS means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency o~ said first product signal;
second means for multiplying the second portion of each filtered input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and sub-stantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being perdeterminedly biased for multiplying the second portion of the filtered input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the filtered input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each filtered input signal to provide a phase coherent compoiste signal.
~,
Proceedings, June 1959, at pp. 1075 to 1101 and in U.S.
patent no. 3,471,788 to W.S. B.ickford et al. In these prior art comb.iners, the antenna slgnals are converted to inter-mediate frequency (IF) signals which are then cophased wi-th one another and thereafter linearly combined to provide a composite IF signal. For example, the IF signals developed from each antenna RF signal may be phase allgned w.ith a locally generated signal of a reference frequency, or may be phase aligned to a selected one of the IF signals, or may be phase aligned with respect to the compos.ite IF s.ignal. Once the IF signals from each antenna RF signal are cophased with one another, they may then be linearly added by appropriate circuitry to prov.ide a coherent composite IF signal which is the vector sum of the individual IF signals.
In order to cophase each IF signal, prior art equal-gain predetection diversity combiners include circuitry, commonly referred to as a "branch", for dividing the IF sig-nal into first and second portions, mixing the first portion with a reference signal to provide a firs-t product signal that has a phase equal to the difference in phase between the first portion and the reference signal, and mixing the first product signal with the second portion of the IF s.ig-nal to provide a second product signal that will be cophased with the reference signal. S.ince the second product signals of each branch are cophased with one another, they may then be linearly added by appropriate circuitry to provide the composite coherent IF signal. If the second portion of the IF signal and the first product signal were each amplified i ' . ~3~ ~14~37 , l,inearly so that the magnitude of each would be proportional to the .input IF signal, the magnitude of the second product signal would theoretically be proport,ional to the square of the magnitude of the ,input IF signal. However, ,in the con-ceptual design of the prior art equal-gain comb.iner, the f.irst product signal is ampl.itude l.im.ited prior to the input to the second stage of mixing. Consequently, the second product s.ignal will be directly proportional to the magni-tude of the input IF signal rather than to .its square.
In such equal-gain combining systems, it is necessary that all of the antenna ~F signals must have substantially the same mean signal level due to the fact that the first product signal is amplltude l.imited prior to the second mix-.ing. ~eca~se the first product signal is amplitude limited, the second product signal from the second stage of mix,ing will not be proportional to the square of the magnitude of the input IF signal. Thus, if IF signals rece,ived by all branches do not have substantially the same mean signal level, the composite IF signal may be significantly degraded in signal-to-noise ratio s.ince a weak signal received by one branch w,ill be weighted substantially equally with a strong signal received by another branch.
The foregoing inadequacy of prior art equal-gain pre-detection diversity combiners may be improved by utilizing the prior art combining technique known as "maximal-ratio predetection combining". In such maximal-ratio predetection combining systems, signals are not limited prior to the sec-ond mixing in each of the branches. It is desired that all signals are proportionally related to the input IF signal so that the magnitude of the second product signal will be pro-portional to the square of the magnitude of the input IF
signal. As a result, branches receiving strong signals will receive more emphasis than branches receiving weak signals.
~4~ 11~3~
Since Rayleigh fading experienced in 800 MHz systems may c~use instantaneous amplitude variations between the RF
signals received at different antennas of a space-diversity antenn~ array that are in excess of 40 decibels (db~, the linear dynamic range of the branch circuitry must accommo-date IF signals ha~ing at least a 40 db dynamic range ~nd second product signals having amplitude variations in excess of 80 db, which is twice the dynamic range of the input IF
signals due to the squaring by the second mixing operation.
Thus, the particular circuit implementation for providing the second mixing operation must provide substantially idealized multiplication over an extremely large dynamic range.
A prior art product multiplier capable of providing the .desired performance over an output dynamic range in excess of 80 db has not been practically achieved in the past.
Commercially available line~r inte~rated-circuit balanced mixers such as the Motorola~ C15g6 have been utilized as the second mixer in maximal ratio combiners designed for mili-tary applications~ These integrated circuit mixers consist of a quad differential amplifier with cross-coupled outputs to provide full-wave balanced multiplication of the two input signals. Each differential pair is powered by a con-stant current source. Such a mixer will not accommodate input signals having a dynamic range in excess of 30 db, since its linear output dynamic range is only 50 to 60 db.
A doubly~balanced FET mixer, such as that described by Highleyman and Jacob in the article~ "An Analog Multiplier Using Two Field Effect Transistors", IRE Transactions on Communications Systems, Vol. CS-10, pp. 311-317, September, 1962, may also be used for the second mixing operation with some expected impr~vement in dynamic range, but without any appreciable reduction in complexity or cost~ :
Accordingly, it is an object of the present invention to provide an improved low-cost, large dynamic range product ,~
multiplier for a maximal-ratio predetection diversity com-biner that coherently combines a plurality of amplitude and phase varying input signals of substantially the same fre-quency.
It is another object of -the present invention to pro-vide an improved low-cost, large dynamic range product mul-tiplier for a ~aximal-ratlo predetectlon divers.ity combiner su.itable for use in a d.iversity recelver for coherently com-b.ining RF signals havlng a dynamic range in excess of forty decibels (40 dB).
Summary of the Invention According to the .invent.ion, there is provided a maxl-mal-ratio predetection diversity combiner for coherently combining over a predetermined dynamic range a plurality of input signals of substantially the same frequency that have unknown and varying phases and magnitudes with respect to one another. The plurality of input signals are co-phased with each other within the combiner so that these signals are phase-coherent with each other prior to being linearly summed together at the output of the combiner. Furthermore, the amplitude of each input signal is, in effect, squared within the combiner prior to the l.inear summation to give greater emphasis to the siynals having the larger magni-tudes.
For each input signal, the maximal-ratio predetection combiner includes: circuitry for dividing the input signal into first and second portions; a first mixer for multiply-ing the first portion of the lnput slgnal with a reference signal that is produced within the combiner to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference sig-nal; a filter network for prov.iding a variable phase shift , to the first product signal with the phase shift being a function of the frequency of the first product signal; and a second mixer for multiplying the second portion of the input signal with the phase-shifted first product signal to provide a second product signal that is substantially independent of the phase of the origianl input signal. The second product sig~als, which are each cophased with the reference signal, may then be linearly combined by combining circuitry to provide a composite coherent IF signal.
The second mixer of the maximal-ratio diversity combiner preferably is a field-effect transistor devic~ that propor-tionally multiplies, over the predetermined dynamic range of the combiner, the second portion of the input signal and the first product signal to provide the second product signal. Thus, the magni-tude of the second product slgnal is proportional to the square o the magnitude o~ the respective input signal. Therefore, since strong input signals receive more emphasis than weak input signals, the signaL-to-noise degradation from the weak input signals is greatly reduced. Since a single FET device provides essentially ideal multiplication, the second product signal is proportional to the square of the respective input signal over the entire dynamic range of the input signal, which, in some applications, may be in excess of forty decibels (40 db) for the input signal.
More particularly, there is provided:
A maximal-ratio predetection diversity combiner for coherently combining a plurality of input signals each having a linear dynamic range exceeding forty decibels (40 db), having substantially the same predetermined frequency and further having unknown and varying phases and magnitudes with respect to one another, said maximal ratio predetection diversity combiner comprising:
means for generating a reference signal having a pre-determined reference frequency for each input signal;
.~
43~
-6a-means for dividing each input signal into first and second portions;
first means for multiplying the first portion of each input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product signal;
second means for multiplying the second portion of each input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and substantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (F~
said FET being predeterminedly biased for multiplying the second portion of the input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the ~o magnitudes of the second portion of the input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each input signal to provide a phase coherent composite signal.
There is also provided:
A diversity receiving system comprising antenna array means having a plurality of substantially independent antennas for receiving a radio signal of a predetermined frequency, each antenna providing an input signal, the input signals from the antennas having unknown and varying phases and magnitudes with respect to one another; converting means coupled to the input signals from the antennas for converting the frequency of the in-put signals to an intermediate frequency; intermediate frequency amplifying and filtering means coupled to the converted input signals from the converting means for filtering the converted ,' .
~' :
L4~
-6b-input signals; and maximal-ratio predetection combining means coupled to the filtered input signals from the intermediate frequency amplifyling and filtering means for coherently combining the filtered input signals to provide a coherent composite signal, said combining means including:
means for generating a reference signal having a predeter-mined reference frequency for each filtered input signal;
means for dividing each filtered input signal into first and second portions;
first means for multiplying the first portion of each filtered input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the filtered input signal and the reference signal;
lS means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency o~ said first product signal;
second means for multiplying the second portion of each filtered input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and sub-stantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being perdeterminedly biased for multiplying the second portion of the filtered input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the filtered input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each filtered input signal to provide a phase coherent compoiste signal.
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-6c-Brief Description of the Drawings .
Fig. 1 is a block diagram of a diversity receiving system that may advantageously utilize the present invention.
Fig. 2 is a block diagram of a maximal-ratio predetection diversity combiner which may advantageously utilize the present invention.
Fig. 3, including Figs. 3A and 3B taken together, is a detailed circuit diagram of a portion of the circuitry in each branch of Fig. 2, together with a portion of the circuitry of Fig. 2 common to all of the branches.
1L4~3~
Detailed Description of the Preferred Embodiment In Fig. 1, there is illustrated a diversity receiver including six directional sector antennas in an antenna array 100, conversion stages 110-115, IF selectivity stages 120-125 and a maximal-ratio predetection diversity combiner 130. The signal received by each directional sector antenna is tuned to the same radio frequency by the conversion stages 110-115.
The diversity receiver of Fig. 1 may be advantageously utilized at the base station of a mobile radio system, such as the system described in the Federal Communications Commission filing by American Radio Telephone Service, Inc., of Baltimore, Maryland, entitled "An Application For A Developmental Cellular Mobile and Portable Radio Telephone System In The Washington-Baltimore Northern Virginia Area", filed on February 14, 1977, and in copending application, Serial No. 345,723 by Frank J.
Cerny, Jr., and James J. Mikulski, entitled "Instantaneously Acquiring Sector Antenna Combining System", filed on February 15, 1980 and assi~ned to the same assignee as the instant application. Prior to the aforementioned copending application, ~0 radio-telephone systems have typically utilized omnidirectional antennas, as shown in U.S. Patent No. 3,471,788, instead of high-gain directional sector antennas as shown in this copending application, for providing omnidirectional coverage. By utilizing a maximal-ratio predetection diversity combiner 130 with a directional antenna array 100 comprised of a plurality of directional gain antennas whose patterns are spatially distributed to provide an omnidirectional pattern, the effective coverage area of the antenna array may be substantially increased while providing an omnidirectional receiving pattern. However, systems utilizing antenna arrays , comprised of either d,irect.ional or omnidirectional antennas may advantageously utilize the present invention.
The signals received by each antenna of the antenna array 100 are converted to an .intermediate frequency by con-version stages 110-115, filtered by IF stages 120-125, and applied to max.imal ratio predetect.ion combiner 130. The combiner 130 continuously cophases the branch IF signals received from each antenna of the antenna array 100 and thereafter linearly adds these cophased branch IF s.ignals together to provide a compos.ite IF signal. ~n combining the branch IF signals, it is desirable that signal-to-no.ise rat.io degradations which may be .introduced by low-level s.ig-nals or deep nulls received by one or more of the sector antennas of the antenna array 100, be avo.ided. Thus, the entire diversity rece,iver should be capable of linearly accommodating the expected dynamic range of each sector antenna signal from the antenna array 100. The expected dynam,ic range should be linearly accommodated not only by the conversion stages 110-115 and the IE' stages 120-125 but also by the maximal-ratio predetection diversity combiner 130. It has been found that, in order to provide optimum suppression of the noise pops due to Rayle.igh fading, a l.inear dynamic range of at least 40 db should be maintained throughout the diversity receiver of Fig. 1. Thus, in order to prov.ide a linear dynamic range for the maximal-ratio pre-detection diversity combiner 130, it is necessary that the second multiplier (230 of Fig. 2) provlde essentially ideal multiplication so that the multiplier product signal is pro-port.ional to the product of the magnitudes of the multiplier input signals over the entire dynamic range of each multi-plier input signal. Prior art combiners have failed to pro-vide such proportional multiplication for signals having a linear dynamic range in excess of 40 db.
g -In accordance with the present invention, it has been found that a FET device will provide essentially ideal multiplication over the wide dynamic range encountered in maximal ratio predetection combiners. The FET device is preferably biased such that the qulescent gate-to-source voltage is approximately one-half of the gate pinchoff volt-age. The -two signals to be multiplied are applied to the gate and source terminals of the FET device, respectively.
By filtering the FET drain current, id~ at the d~ifference frequency, wl-w2, the ~ollowing equation is obtained:
id = (IDSs/vp2) Vl V2 CS(Wl-W2)t where IDSs = the steady state drain saturation current;
Vp = the gate pinchoff voltage;
Vl = the magnitude of the input signal at radian frequency wl; and V2 = the magnitude of the input signal at radian frequency w2.
The development of the foregoing equation is described in detail in my prior U.S. patent no. 3,716,730.
According to the present invention, a FET device will provide substantially ideal multiplication as predicted by the foregoing equation as long as the FET input signal lev-els are properly ratioed. For example, a Siliconix U-310 FET iS capable of providing a product signal having a dynam-ic range in excess of 130 db. For the U-310 FET, one deci-bel of output compression (indicating the limit of linear product multiplication) occurs when the gate signal level is -3 dbm or the source signal level is +9 dbm. Assuming that a receiver requires an input signal level of -112 dbm to provide a 20 db quieting signal, the threshold levels for the gate signal may be set at -62 dbm and the source signal at -50 dbm (total of -112 dbm). Thus, the signals at the gate and source terminals of the FET may each vary over a : ` ' '' ~ .
-10- ~ 3~
, , dynamic range of 59 db (e.g. for the gate, -3dbm -[-62dbm] =
59db) without producing output compression. Furtherm~re, if the signal-to-noise ratio of the receiver IF circuitry is 7 db for a 20 db quleting signal, the siynals at the gate and source terminals may each vary over a dynamic range of 66 db (i.e. 59db + 7db = 66db). Thus, the FET product signal has a dynamic range that .is 132 db, twice the dynamic range of the s.ignals at the gate and source terminals.
In Fig. 2, there .is .illustrated a more deta.iled block d.iagram of the diversity receiver of Fig. 1, as shown in the aforementioned copending application. The diversity rece.iv-er of Fig. 2 shows only three of the six branches of the diversity receiver of Fig. 1, although any number of branch-es may be utilized in practicing the present invention.
Branches 200, 201 and 202 are comprised of substantially identical c.ircu.itry, each branch provid.ing a product signal that .is both phase coherent with the other branch product signals and proport.ional to the square of the magnitude of the signal received by its respective sector antenna.
In the diversity receiver of Fig. 2, the frequency of local oscillator 208 determines which radio channel the diversity receiver is tuned to. The RF signal received by each branch sector antenna 220 is combined by mixer 221 with the signal from local oscillator 208 to provide an IF signal at 21.4 MHz. The IF signal from m.ixer 221 is then applied to IF bandpass filter 222, which may be a monolithic band-pass f.ilter of conventional design similar to that described in U.S. patent no. 3,716,808. The filtered IF signal from filter 222 is then applied to IF amplif.ier 223. The output from IF amplifier 223 is split and fed forward via two paths to mixer 230. The first portion of the IF signal is linear-ly amplif.ied by IF ampl.ifier 229 and thereafter appl.ied to mixer 230. IF amplifier 229 linearly ampl.ifies the first portion of the IF signal to provide a signal level , ' . ` ` . .
that is within the input dynamic range of mixer 230. The second portion of the IF signal is appl.ied to mixer 225 together with -the 1.72 MHz composite IF signal which is fed back v.ia amplifier 206 and fllter 224. By feed.ing ~ack the S IF slgnal, the IF strlp of the dlverslty recelver forms a closed feedback loop that is regenerative on noise. Thus, the randomly varylng phase of each branch IF slgnal relatlve to the compos.ite IF signal is added into the closed loop via mixer 225/ and then subtracted out at mixer 230. By th.is process, the random phase variations are removed from each branch IF signal in relation to the composite IF signal.
The result is that each branch IF output signal is cophased to the composite IF signal. Alternatively, in other ar-rangements, the branch IF signals need not be cophased with the composite IF signal, but may be cophased to a locally generated reference signal.
Referring back to branch 200, the component of the out-put signal from mlxer 225 at the dlfference frequency of 19.68 MHz has a relatlve phase which ls the dlfference be-tween the phase of the branch IF slgnal at 21.4 MHz and the composlte IF signal at 1.72 MHz. This resultant output slg-nal is llnearly amplified by second IF amplifler 226 and applled to bandpass filter 227 to provide a variable phase shift to the resultant signal. Filter 227 may be a two-pole crystal fllter havlng a center frequency of 19.68 MHz and passband bandwidth of 2 KHz. The phase shift, provlded by fllter 227, is a function of the absolute frequency of the resultant signal. The slgnal from filter 227 ls linearly ampl.ified by thlrd IF ampllfler 228 to provlde a signal level that is within the input dynamlc range of mixer 230;
thls ampllfied signal is applied to the second input of mixer 230. Mixer 230 multiplles the ampllfied 19.68 MHz difference product signal from ampl.ifier 228 wlth the ampll-fied 21.4 MHz IF signal from ampllfler 229 to provlde a ; `
:
resultan-t output product signal having a 1.72 MHz difference frequency that is cophased with the compos.ite IF signal, thus being substantially free of the random phase variations of the input IF s.ignals. The phase d.ifference result.ing from mixer 225 .is subtrac-ted from the phase of the ampl.ified 21.4 MHz IF s.ignal from amplifier 229 to produce ~he 1.72 MHz difference product s.ignal from m.ixer 230, which is co-phased wlth the compos.ite IF signal and free from the random phase variations of the branch IF signal. The resultant output signal from mixer 230 is proportional to the square of the level of the input IF signal to that branch. The resultant output signals from the mixer 230 of each branch are linearly added together to form one composite IF s.ignal at 1.72 MHz. This composite IF signal is the output s.ignal from the maximal ratio predetection diversity combiner; it .is fed to a 1.72 MHz IF bandpass filter 204 and fourth IF
amplifier 205. The composite IF signal from amplif.ier 205 may then be applied to a conventional demodulator that is appropriate for recovering the method of information modula-tion being utilized within the system. The composite IF
signal from amplifier 205 is further amplified and then amplitude limited by fifth IF amplifier 206 to provide a high-level amplitude-limited composite IF signal which is applied to each mixer 225 through each filter 224. Filter 224 may be either a bandpass filter or a low-pass filter having an operating fre~uency of 1.72 M~z. Automatic gain control is applied to all branches of the combiner by con-trolling the gain of each IF amplifier 223 with an AGC con-trol voltage from AGC circuitry 207. This control voltage may be obtained by rectifying, amplifying, and low-pass fil-tering a portion of the composite IF signal from the output of mixer 230.
Figs. 3A and 3B taken together illustrate in detail the corresponding blocks of the circuitry of Fig. 2. Each -13~ 37 branch 200, 201 and 202 ~f Fig. 2 contains essentially the same circuitry that is illustrated in Figs. 3A and 3B. In Fig. 3A, the branch IF signal applied to FET device 310 is the ~ignal provided by IF filter 222 of Fig. 2. Each desig-nated portion of Figs. 3A and 3B corresp~nds to the block of Fig. 2 identified in parentheses after each designation.
Referring to Fig. 3A, the branch IF signal, having a nominal frequency of 21.4 MHz, is amplified by FET device 310, which may be a Motorola~ N~04, before application to the IF amplifier and A~C stage consisting of am~ ier 311, which may be a Motorola~ C1350P. The output of ~his IF
amplifier is then divided into two portions, one portion being applied to FET device 313, and the other portion being applied ~o amplifier 315. FET device 313, which may be a Motorola~N4416, is biased to operate as a mixer~ The FET
device 313 mixes the signal from the this IF amplifier with a feedback signal derived from the composite IF signal. The output of FET device 313 is then transf~rmer-coupled to amplifier 314, which may be a Motorol ~ MC1350P. Amplifier 314 is biased to provide linear amplification to signals from FET device 313. The output ~f amplifier 314 is then filtered by a two-pole monolithic crystal filter 316, which has a center frequency of 19.6B MHz and a passband of 2 KHz.
The monolithic crystal filter 316 provides for both n3rrow-band filtering and phase-shifting that is ~ ~unction of the frequency of the signal from amplifier 314. The phase-shifted output from monolithic crystal filter~316 is then applied to amplifier 317, which may be an RCA~ A3~86.
Amplifier 317 likewise provides linear amplifica~ion ~o sig-3Q nals from the mon~lithic crystal filter 316. The output of amplifier 317 is transfcrmer-coupled to the source terminalS
of ~ET device 31B. The second portion of the branch IF sig-nal from IF a~plifier 311 is coupled to amplifier 315, wh~ch also provides linear amplification. The output fr~m ampli --14- ~ ~
fier 315, which may be a Motorola~MC1350P, is ~ransformer-coupled to the gate terminal ~ of FET device 318.
In accordance with the present inventi~n, FET device 318 proportionally multiplies the signals from amplifiers 317 and 315 to provide the 1.72 MHz difference pr~duct sig-nal at its drain teI~unal D. The FET device 318 is preferably a large ~cwer FET
device, such as the Siliconi ~U-310. According to another feature of the present invention, the product signals ~rom the FET devices 318 of each branch may be combined simply by interconnecting the respective FET drain terminals. The drain terminals of FET devices 318 may be paralleled since the drain of a FET is essentially a constant-current source when the load conductance is at least an order of magnitude larger than the output conductance of the FET device; the paralleling of the drain terminals of the FET devices of each branch does not degrade either the gain or th2 dynamic range of the individual FET devices. According to yet another feature of the present invention, additional ~ET
devices 3181 may be paralleled with the FET device 318 in each branch to provide an increased dynamic range frsm the paral-leled FET devices. Thus, nei~her the intercoupling of the FET devices nor the parallelinq of ~dditional FET devices requires any additional circuitry. In addition, the connec-tions from the amplifiers 315 and 317 to the FET device 318' 2~ may be reversed, such that the amplifier 317 is coupled to the gate terminal and amplifier 315 is coupled to the source terminal indicated on Fiq. 3B by "G" and "S!! in parentheses.
The additional circuitry shown in Fig 3B is primarily circuitry that is common to all of the branches of the diversity combiner. The AGC circuitry includes transistors 319 and 320, which are capacitively coupled to the drain terminals of the branch FET devices 318 for providing an AGC
voltage therefrom. The transistors 319 and 320 may be any suitable silic~n transistors, ~uch as the Motorola MPS6517 r .~ .
~L4~437 and Motorc>l(~MPS6513, respectively. The composite IF signal is coupled to amplifier 322 and ~hereafter to limiter 321 for providing a feedback signal to FET device 313 of each branch of the diversity combiner. The feedback signal from limiter 321 is applied via a low-pass filter 323 to the FET
device 313 of Fig. 3A. Amplifier 322 ~ay be a Motorola(~) MC1350P, and limiter 321 may be an RC 3086.
In summary, a maximal~ratio predetection diversity com-biner has been described hereinabove which provides a com-posite IF signal having a dynamic range in excess of 100 db.
The large dynamic range of the diversity combiner has been achieved by utilizing an active FET device to proportionally multiply linear IF signals over their entire dynamic range.
The circuit configuration for achieving this result is very simple, not requiring a multitude of current sources, dif-ferential pair arrangements or balanced circuit arrange-ments. Thus, the cost of the inventive diversity combiner has been substantially reduced. The diversity combiner of ` the present invention may be expanded to accommodate a large number of branches simply by paralleling additional branches with the drain terminal of the branch FET devices 318 of Fig. 3B.
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-6c-Brief Description of the Drawings .
Fig. 1 is a block diagram of a diversity receiving system that may advantageously utilize the present invention.
Fig. 2 is a block diagram of a maximal-ratio predetection diversity combiner which may advantageously utilize the present invention.
Fig. 3, including Figs. 3A and 3B taken together, is a detailed circuit diagram of a portion of the circuitry in each branch of Fig. 2, together with a portion of the circuitry of Fig. 2 common to all of the branches.
1L4~3~
Detailed Description of the Preferred Embodiment In Fig. 1, there is illustrated a diversity receiver including six directional sector antennas in an antenna array 100, conversion stages 110-115, IF selectivity stages 120-125 and a maximal-ratio predetection diversity combiner 130. The signal received by each directional sector antenna is tuned to the same radio frequency by the conversion stages 110-115.
The diversity receiver of Fig. 1 may be advantageously utilized at the base station of a mobile radio system, such as the system described in the Federal Communications Commission filing by American Radio Telephone Service, Inc., of Baltimore, Maryland, entitled "An Application For A Developmental Cellular Mobile and Portable Radio Telephone System In The Washington-Baltimore Northern Virginia Area", filed on February 14, 1977, and in copending application, Serial No. 345,723 by Frank J.
Cerny, Jr., and James J. Mikulski, entitled "Instantaneously Acquiring Sector Antenna Combining System", filed on February 15, 1980 and assi~ned to the same assignee as the instant application. Prior to the aforementioned copending application, ~0 radio-telephone systems have typically utilized omnidirectional antennas, as shown in U.S. Patent No. 3,471,788, instead of high-gain directional sector antennas as shown in this copending application, for providing omnidirectional coverage. By utilizing a maximal-ratio predetection diversity combiner 130 with a directional antenna array 100 comprised of a plurality of directional gain antennas whose patterns are spatially distributed to provide an omnidirectional pattern, the effective coverage area of the antenna array may be substantially increased while providing an omnidirectional receiving pattern. However, systems utilizing antenna arrays , comprised of either d,irect.ional or omnidirectional antennas may advantageously utilize the present invention.
The signals received by each antenna of the antenna array 100 are converted to an .intermediate frequency by con-version stages 110-115, filtered by IF stages 120-125, and applied to max.imal ratio predetect.ion combiner 130. The combiner 130 continuously cophases the branch IF signals received from each antenna of the antenna array 100 and thereafter linearly adds these cophased branch IF s.ignals together to provide a compos.ite IF signal. ~n combining the branch IF signals, it is desirable that signal-to-no.ise rat.io degradations which may be .introduced by low-level s.ig-nals or deep nulls received by one or more of the sector antennas of the antenna array 100, be avo.ided. Thus, the entire diversity rece,iver should be capable of linearly accommodating the expected dynamic range of each sector antenna signal from the antenna array 100. The expected dynam,ic range should be linearly accommodated not only by the conversion stages 110-115 and the IE' stages 120-125 but also by the maximal-ratio predetection diversity combiner 130. It has been found that, in order to provide optimum suppression of the noise pops due to Rayle.igh fading, a l.inear dynamic range of at least 40 db should be maintained throughout the diversity receiver of Fig. 1. Thus, in order to prov.ide a linear dynamic range for the maximal-ratio pre-detection diversity combiner 130, it is necessary that the second multiplier (230 of Fig. 2) provlde essentially ideal multiplication so that the multiplier product signal is pro-port.ional to the product of the magnitudes of the multiplier input signals over the entire dynamic range of each multi-plier input signal. Prior art combiners have failed to pro-vide such proportional multiplication for signals having a linear dynamic range in excess of 40 db.
g -In accordance with the present invention, it has been found that a FET device will provide essentially ideal multiplication over the wide dynamic range encountered in maximal ratio predetection combiners. The FET device is preferably biased such that the qulescent gate-to-source voltage is approximately one-half of the gate pinchoff volt-age. The -two signals to be multiplied are applied to the gate and source terminals of the FET device, respectively.
By filtering the FET drain current, id~ at the d~ifference frequency, wl-w2, the ~ollowing equation is obtained:
id = (IDSs/vp2) Vl V2 CS(Wl-W2)t where IDSs = the steady state drain saturation current;
Vp = the gate pinchoff voltage;
Vl = the magnitude of the input signal at radian frequency wl; and V2 = the magnitude of the input signal at radian frequency w2.
The development of the foregoing equation is described in detail in my prior U.S. patent no. 3,716,730.
According to the present invention, a FET device will provide substantially ideal multiplication as predicted by the foregoing equation as long as the FET input signal lev-els are properly ratioed. For example, a Siliconix U-310 FET iS capable of providing a product signal having a dynam-ic range in excess of 130 db. For the U-310 FET, one deci-bel of output compression (indicating the limit of linear product multiplication) occurs when the gate signal level is -3 dbm or the source signal level is +9 dbm. Assuming that a receiver requires an input signal level of -112 dbm to provide a 20 db quieting signal, the threshold levels for the gate signal may be set at -62 dbm and the source signal at -50 dbm (total of -112 dbm). Thus, the signals at the gate and source terminals of the FET may each vary over a : ` ' '' ~ .
-10- ~ 3~
, , dynamic range of 59 db (e.g. for the gate, -3dbm -[-62dbm] =
59db) without producing output compression. Furtherm~re, if the signal-to-noise ratio of the receiver IF circuitry is 7 db for a 20 db quleting signal, the siynals at the gate and source terminals may each vary over a dynamic range of 66 db (i.e. 59db + 7db = 66db). Thus, the FET product signal has a dynamic range that .is 132 db, twice the dynamic range of the s.ignals at the gate and source terminals.
In Fig. 2, there .is .illustrated a more deta.iled block d.iagram of the diversity receiver of Fig. 1, as shown in the aforementioned copending application. The diversity rece.iv-er of Fig. 2 shows only three of the six branches of the diversity receiver of Fig. 1, although any number of branch-es may be utilized in practicing the present invention.
Branches 200, 201 and 202 are comprised of substantially identical c.ircu.itry, each branch provid.ing a product signal that .is both phase coherent with the other branch product signals and proport.ional to the square of the magnitude of the signal received by its respective sector antenna.
In the diversity receiver of Fig. 2, the frequency of local oscillator 208 determines which radio channel the diversity receiver is tuned to. The RF signal received by each branch sector antenna 220 is combined by mixer 221 with the signal from local oscillator 208 to provide an IF signal at 21.4 MHz. The IF signal from m.ixer 221 is then applied to IF bandpass filter 222, which may be a monolithic band-pass f.ilter of conventional design similar to that described in U.S. patent no. 3,716,808. The filtered IF signal from filter 222 is then applied to IF amplif.ier 223. The output from IF amplifier 223 is split and fed forward via two paths to mixer 230. The first portion of the IF signal is linear-ly amplif.ied by IF ampl.ifier 229 and thereafter appl.ied to mixer 230. IF amplifier 229 linearly ampl.ifies the first portion of the IF signal to provide a signal level , ' . ` ` . .
that is within the input dynamic range of mixer 230. The second portion of the IF signal is appl.ied to mixer 225 together with -the 1.72 MHz composite IF signal which is fed back v.ia amplifier 206 and fllter 224. By feed.ing ~ack the S IF slgnal, the IF strlp of the dlverslty recelver forms a closed feedback loop that is regenerative on noise. Thus, the randomly varylng phase of each branch IF slgnal relatlve to the compos.ite IF signal is added into the closed loop via mixer 225/ and then subtracted out at mixer 230. By th.is process, the random phase variations are removed from each branch IF signal in relation to the composite IF signal.
The result is that each branch IF output signal is cophased to the composite IF signal. Alternatively, in other ar-rangements, the branch IF signals need not be cophased with the composite IF signal, but may be cophased to a locally generated reference signal.
Referring back to branch 200, the component of the out-put signal from mlxer 225 at the dlfference frequency of 19.68 MHz has a relatlve phase which ls the dlfference be-tween the phase of the branch IF slgnal at 21.4 MHz and the composlte IF signal at 1.72 MHz. This resultant output slg-nal is llnearly amplified by second IF amplifler 226 and applled to bandpass filter 227 to provide a variable phase shift to the resultant signal. Filter 227 may be a two-pole crystal fllter havlng a center frequency of 19.68 MHz and passband bandwidth of 2 KHz. The phase shift, provlded by fllter 227, is a function of the absolute frequency of the resultant signal. The slgnal from filter 227 ls linearly ampl.ified by thlrd IF ampllfler 228 to provlde a signal level that is within the input dynamlc range of mixer 230;
thls ampllfied signal is applied to the second input of mixer 230. Mixer 230 multiplles the ampllfied 19.68 MHz difference product signal from ampl.ifier 228 wlth the ampll-fied 21.4 MHz IF signal from ampllfler 229 to provlde a ; `
:
resultan-t output product signal having a 1.72 MHz difference frequency that is cophased with the compos.ite IF signal, thus being substantially free of the random phase variations of the input IF s.ignals. The phase d.ifference result.ing from mixer 225 .is subtrac-ted from the phase of the ampl.ified 21.4 MHz IF s.ignal from amplifier 229 to produce ~he 1.72 MHz difference product s.ignal from m.ixer 230, which is co-phased wlth the compos.ite IF signal and free from the random phase variations of the branch IF signal. The resultant output signal from mixer 230 is proportional to the square of the level of the input IF signal to that branch. The resultant output signals from the mixer 230 of each branch are linearly added together to form one composite IF s.ignal at 1.72 MHz. This composite IF signal is the output s.ignal from the maximal ratio predetection diversity combiner; it .is fed to a 1.72 MHz IF bandpass filter 204 and fourth IF
amplifier 205. The composite IF signal from amplif.ier 205 may then be applied to a conventional demodulator that is appropriate for recovering the method of information modula-tion being utilized within the system. The composite IF
signal from amplifier 205 is further amplified and then amplitude limited by fifth IF amplifier 206 to provide a high-level amplitude-limited composite IF signal which is applied to each mixer 225 through each filter 224. Filter 224 may be either a bandpass filter or a low-pass filter having an operating fre~uency of 1.72 M~z. Automatic gain control is applied to all branches of the combiner by con-trolling the gain of each IF amplifier 223 with an AGC con-trol voltage from AGC circuitry 207. This control voltage may be obtained by rectifying, amplifying, and low-pass fil-tering a portion of the composite IF signal from the output of mixer 230.
Figs. 3A and 3B taken together illustrate in detail the corresponding blocks of the circuitry of Fig. 2. Each -13~ 37 branch 200, 201 and 202 ~f Fig. 2 contains essentially the same circuitry that is illustrated in Figs. 3A and 3B. In Fig. 3A, the branch IF signal applied to FET device 310 is the ~ignal provided by IF filter 222 of Fig. 2. Each desig-nated portion of Figs. 3A and 3B corresp~nds to the block of Fig. 2 identified in parentheses after each designation.
Referring to Fig. 3A, the branch IF signal, having a nominal frequency of 21.4 MHz, is amplified by FET device 310, which may be a Motorola~ N~04, before application to the IF amplifier and A~C stage consisting of am~ ier 311, which may be a Motorola~ C1350P. The output of ~his IF
amplifier is then divided into two portions, one portion being applied to FET device 313, and the other portion being applied ~o amplifier 315. FET device 313, which may be a Motorola~N4416, is biased to operate as a mixer~ The FET
device 313 mixes the signal from the this IF amplifier with a feedback signal derived from the composite IF signal. The output of FET device 313 is then transf~rmer-coupled to amplifier 314, which may be a Motorol ~ MC1350P. Amplifier 314 is biased to provide linear amplification to signals from FET device 313. The output ~f amplifier 314 is then filtered by a two-pole monolithic crystal filter 316, which has a center frequency of 19.6B MHz and a passband of 2 KHz.
The monolithic crystal filter 316 provides for both n3rrow-band filtering and phase-shifting that is ~ ~unction of the frequency of the signal from amplifier 314. The phase-shifted output from monolithic crystal filter~316 is then applied to amplifier 317, which may be an RCA~ A3~86.
Amplifier 317 likewise provides linear amplifica~ion ~o sig-3Q nals from the mon~lithic crystal filter 316. The output of amplifier 317 is transfcrmer-coupled to the source terminalS
of ~ET device 31B. The second portion of the branch IF sig-nal from IF a~plifier 311 is coupled to amplifier 315, wh~ch also provides linear amplification. The output fr~m ampli --14- ~ ~
fier 315, which may be a Motorola~MC1350P, is ~ransformer-coupled to the gate terminal ~ of FET device 318.
In accordance with the present inventi~n, FET device 318 proportionally multiplies the signals from amplifiers 317 and 315 to provide the 1.72 MHz difference pr~duct sig-nal at its drain teI~unal D. The FET device 318 is preferably a large ~cwer FET
device, such as the Siliconi ~U-310. According to another feature of the present invention, the product signals ~rom the FET devices 318 of each branch may be combined simply by interconnecting the respective FET drain terminals. The drain terminals of FET devices 318 may be paralleled since the drain of a FET is essentially a constant-current source when the load conductance is at least an order of magnitude larger than the output conductance of the FET device; the paralleling of the drain terminals of the FET devices of each branch does not degrade either the gain or th2 dynamic range of the individual FET devices. According to yet another feature of the present invention, additional ~ET
devices 3181 may be paralleled with the FET device 318 in each branch to provide an increased dynamic range frsm the paral-leled FET devices. Thus, nei~her the intercoupling of the FET devices nor the parallelinq of ~dditional FET devices requires any additional circuitry. In addition, the connec-tions from the amplifiers 315 and 317 to the FET device 318' 2~ may be reversed, such that the amplifier 317 is coupled to the gate terminal and amplifier 315 is coupled to the source terminal indicated on Fiq. 3B by "G" and "S!! in parentheses.
The additional circuitry shown in Fig 3B is primarily circuitry that is common to all of the branches of the diversity combiner. The AGC circuitry includes transistors 319 and 320, which are capacitively coupled to the drain terminals of the branch FET devices 318 for providing an AGC
voltage therefrom. The transistors 319 and 320 may be any suitable silic~n transistors, ~uch as the Motorola MPS6517 r .~ .
~L4~437 and Motorc>l(~MPS6513, respectively. The composite IF signal is coupled to amplifier 322 and ~hereafter to limiter 321 for providing a feedback signal to FET device 313 of each branch of the diversity combiner. The feedback signal from limiter 321 is applied via a low-pass filter 323 to the FET
device 313 of Fig. 3A. Amplifier 322 ~ay be a Motorola(~) MC1350P, and limiter 321 may be an RC 3086.
In summary, a maximal~ratio predetection diversity com-biner has been described hereinabove which provides a com-posite IF signal having a dynamic range in excess of 100 db.
The large dynamic range of the diversity combiner has been achieved by utilizing an active FET device to proportionally multiply linear IF signals over their entire dynamic range.
The circuit configuration for achieving this result is very simple, not requiring a multitude of current sources, dif-ferential pair arrangements or balanced circuit arrange-ments. Thus, the cost of the inventive diversity combiner has been substantially reduced. The diversity combiner of ` the present invention may be expanded to accommodate a large number of branches simply by paralleling additional branches with the drain terminal of the branch FET devices 318 of Fig. 3B.
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Claims (12)
1. A maximal-ratio predetection diversity combiner for coherently combining a plurality of input signals each having a linear dynamic range exceeding forty decibels (40db), having substantially the same predetermined frequency and further having unknown and varying phases and magnitudes with respect to one another, said maximal ratio predetection diversity combiner comprising:
means for generating a reference signal having a pre-determined reference frequency for each input signal;
means for dividing each input signal into first and second portions;
first means for multiplying the first portion of each input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product sign 1;
second means for multiplying the second portion of each input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and sub-stantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being pre-determinedly biased for multiplying the second portion of the input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each input signal to provide a phase coherent composite signal.
means for generating a reference signal having a pre-determined reference frequency for each input signal;
means for dividing each input signal into first and second portions;
first means for multiplying the first portion of each input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product sign 1;
second means for multiplying the second portion of each input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and sub-stantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being pre-determinedly biased for multiplying the second portion of the input signal and the first product signal to provide the second product signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each input signal to provide a phase coherent composite signal.
2. The diversity combiner according to claim 1, wherein the FET includes source, gate and drain terminals and is arranged such that the second portion of the input signal is coupled to the source terminal thereof, the first product signal is coupled to the gate terminal thereof, and the second product signal is provided at the drain terminal thereof.
3. The diversity combiner according to claim 1, wherein the FET includes source, gate and drain terminals and is arranged such that the first product signal is coupled to the source terminal thereof, and the second portion of the input signal is coupled to the gate terminal thereof and the second product signal is provided at the drain terminal thereof.
4. The diversity combiner according to claim 1, wherein the second multiplying means is a power FET that is biased such that the quiescent gate-to-source voltage is substantially one half the gate pinch-off voltage of the power FET.
5. The diversity combiner according to claims 1, 2 or 3, wherein the second multiplying means is comprised of a plurality of substantially identical FET's coupled in para-llel with each other for increasing the dynamic range of the second product signal.
6. The diversity combiner according to claim 1, wherein the reference-signal generating means is coupled to the composite IF signal for developing a reference signal therefrom.
7. A diversity receiving system comprising antenna array means having a plurality of substantially independent antennas for receiving a radio signal of a predetermined fre-quency, each antenna providing an input signal, the input signals from the antennas having unknown and varying phases and magnitudes with respect to one another; converting means coupled to the input signals from the antennas for converting the frequency of the input signals to an intermediate frequency;
intermediate frequency amplifying and filtering means coupled to the converted input signals from the converting means for filtering the converted input signals; and maximal-ratio pre-detection combining means coupled to the filtered input signals from the intermediate frequency amplifying and filtering means for coherently combining the filtered input signals to provide a coherent compoiste signal, said combining means including:
means for generating a reference signal having a pre-determined reference frequency for each filtered input signal;
means for dividing each filtered input signal into first and second portions;
first means for multiplying the first portion of each filtered input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the filtered input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product signal;
second means for multiplying the second portion of each filtered input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and substantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being predeterminedly biased for multiplying the second portion of the filtered input signal and the first product signal to provide the second pro-duct signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the filtered input signal and the first product sig-nal over substantially twice the dynamic range of the corres-ponding input signal and of the second portion of the filtered input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each filtered input signal to provide a phase coherent composite signal.
intermediate frequency amplifying and filtering means coupled to the converted input signals from the converting means for filtering the converted input signals; and maximal-ratio pre-detection combining means coupled to the filtered input signals from the intermediate frequency amplifying and filtering means for coherently combining the filtered input signals to provide a coherent compoiste signal, said combining means including:
means for generating a reference signal having a pre-determined reference frequency for each filtered input signal;
means for dividing each filtered input signal into first and second portions;
first means for multiplying the first portion of each filtered input signal with the reference signal to provide a first product signal having a phase that is the difference between the phase of the filtered input signal and the reference signal;
means for providing a variable phase shift to said first product signal, said phase shift being a function of the frequency of said first product signal;
second means for multiplying the second portion of each filtered input signal and the corresponding phase-shifted first product signal to provide a second product signal that is substantially co-phased with the reference signal and substantially independent of the phase of the input signal, said second multiplying means being comprised of only a single field-effect transistor (FET), said FET being predeterminedly biased for multiplying the second portion of the filtered input signal and the first product signal to provide the second pro-duct signal, such that the magnitude of the second product signal is proportional to the product of the magnitudes of the second portion of the filtered input signal and the first product sig-nal over substantially twice the dynamic range of the corres-ponding input signal and of the second portion of the filtered input signal and the first product signal over substantially twice the dynamic range of the corresponding input signal; and means for intercoupling the second product signals developed from each filtered input signal to provide a phase coherent composite signal.
8. The diversity receiving system according to claim 7, wherein the FET includes source, gate and drain terminals and is arranged such that the second portion of the input signal is coupled to the source terminal thereof, the first product signal is coupled to the gate terminal thereof, and the second product signal is provided at the drain terminal thereof.
9. The diversity receiving system according to claim 7, wherein the FET includes source, gate and drain terminals and is arranged such that the first product signal is coupled to the source terminal thereof, and the second portion of the input signal is coupled to the gate terminal thereof, and the second product signal is provided at the drain terminal thereof.
10. The diversity receiving system according to claim 7, wherein the second multiplying means is a power FET
that is biased such that the quiescent gate-to-source voltage is substantially one-half the gate pinch-off voltage of the power FET.
that is biased such that the quiescent gate-to-source voltage is substantially one-half the gate pinch-off voltage of the power FET.
11. The diversity receiving system according to claims 7, 8 or 9, wherein the second multiplying means is com-prised of a plurality of substantially identical FET's coupled in parallel with each other for increasing the dynamic range of the second product signal.
12. The diversity receiving system according to claim 7, wherein the reference signal generating means is coupled to the composite IF signal for developing a reference signal therefrom.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US8498079A | 1979-10-15 | 1979-10-15 | |
| US06/084,980 | 1979-10-15 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1141437A true CA1141437A (en) | 1983-02-15 |
Family
ID=22188423
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000360720A Expired CA1141437A (en) | 1979-10-15 | 1980-09-22 | Large dynamic range multiplier for a maximal-ratio predetection diversity combiner |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1141437A (en) |
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO1984000654A1 (en) | 1982-08-03 | 1984-02-16 | Motorola Inc | Method and apparatus for assigning duplex radio channels and scanning duplex radio channels assigned to mobile and portable radiotelephones in a cellular radiotelephone communications system |
| US5339184A (en) * | 1992-06-15 | 1994-08-16 | Gte Laboratories Incorporated | Fiber optic antenna remoting for multi-sector cell sites |
-
1980
- 1980-09-22 CA CA000360720A patent/CA1141437A/en not_active Expired
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| WO1984000654A1 (en) | 1982-08-03 | 1984-02-16 | Motorola Inc | Method and apparatus for assigning duplex radio channels and scanning duplex radio channels assigned to mobile and portable radiotelephones in a cellular radiotelephone communications system |
| US5339184A (en) * | 1992-06-15 | 1994-08-16 | Gte Laboratories Incorporated | Fiber optic antenna remoting for multi-sector cell sites |
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