CA1110356A - Mesfet device surface-wave-device channel selector - Google Patents

Abstract

MESFET DEVICE SURFACE-WAVE-DEVICE CHANNEL SELECTOR

Abstract of the Disclosure An improved channel selector for a television receiver which includes a metal semiconductor field effect transistor, i.e. MESFET, mixer in combination with an acoustic surface wave device filter. The MESFET mixer simultaneously receives a plurality of channels and mixing signals of a selectable frequency, and in response thereto frequency shifts selectable channels to a predetermined center frequency. The MESFET mixer has almost perfect square law characteristics, and accordingly introduces extremely small distortion into the system. The acoustic surface wave device filter has an input coupled to the output of the MESFET mixer. The surface wave device filter provides essentially all of the channel selectivity of the receiver.

Description

'1,64~6 ~.~$~:335;~

BACKGROUND OF THE INVENTION
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This invention relates to electronic devices for receiving a plurality of radiated electromagnetic signals, filtering a selectable channel of frequencies from the received signals, and demodulating the signals of the selected filtered channel. More particularly, the invention relates to television receivers.
Television receivers of the prior art include a radio frc-quency (RF) section and an intermediate frequency (IF) scction.
The RF section includes RF filters which are tuned to coarsely filter a band of channels centered about a manually selected channel. The output of the RF filter couples to the input of an RF amplifier. Typically, total gain through the RF scction is at least 20 dB to 30 dB.This gain increases the arnplitude of signals within the selected channel and additionally makes the noise figure of the system essentially independent of subsequent elements in the receiver. The output of the RF amplifier couples to onc input of a mixer, while a second input of the mixer receives mixing signals of a selectable frequency. T~e selectable frequency is generated such that the selected channel is frequency shifted to approximately 45 MHZ. The output of the mixer couples to a channel selectioD filter which provides a relatively hiyh impedance path for frequencies outside of the selected channel, and a relatively low impedance path for signals inside the selected channel. Signals at the output of the channel selection ~ilter are therefore primarily comprised of frequencies within the selected channel.
Each television channel contains audio in~ormation, video information, and frame synchronizing information. The output of the channel selection filter couples to an audio demodulator ,: .

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T I - 6 4 o which separates the audio information from the selected channel;
and the output of the audio demodulator couples to a speaker which generates audible sounds. Similarly, ~he output of thc channel selection filter couples to a video processing unit which separates the video and frame synchronizing information from the selected channel; and the output of the video processor couples to a picture tube which converts the video and frame synchronizing information to pictures.
As described a~ove, a basic function which all telcvision receivers perform is to frcquency shift the selected channel from an RF frequency to a predetermined IF frequcncy by a mixing device. This mixing operation has been perfor~ed in the past by a variety of nonlinear devices. These devices include vacuum tu~es, diodes, MOSFET transistors, and bipolar transistors.
However, the prior art mixers also generate o~put terms which are proportional to their inputs cubcd or raised to higher order odd powers. In a tele~ision receiver, these odd power terms may generate interfering signals in the desired channel. ~or example, such interfering signals are generated when signals are present in channels on one side of the scl~ctcd ch~el which are onc and two channels removed from the selected channel. This phenomena is known as intermodulation distortion. Similarly, the cubic terms and higher order odd power terms yencrate interf-erin~ sig-nals in the desired channel when a carrier with amplitude modulation is present in anyone of the undesired channels. This phenomena is known as cross modulation distortion. The frequencies which are generated in a desired channel as a result of intermodulation distortion or cross modulation distortion cannot be separated from the information signals lying therein.

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Thus, as the magnitude of the inte~fering frequencies increases, perceptible picture distortion or sound distortion occurs.
A principal advantage of the television receiver herein disclosed is that it has greatly reduced cross modulation and intermodulation distortion. This is accomplished to a large extent by means of a unique RF m~xer~ The mixer utilizes a MESFET transistor which has almost perfect square law current-voltage characteristics. Since the-~ESFET mixer has almost perfect square law characteristics, the mixer introduces extremely small distortion into the system. ln particular, the MESFET
mixer handles an interfering signal level of greater than ~6dBm on its output with less than 1% cross modulation distortion and -40dB intermodulation distortion products.
The square law current-voltage characteristics of a MESFET transistor and their application to reduce intermodulation and cross modulation distortion in a television receiver are not -taught by the prior art. In the past, MESFET devices were used ;
primarily to achieve high speed operation. For example, they were utilized in pico-second digital switching circuits. See, for example, a paper by Cahen et al entitled "A Subnanosecond Switching Circuit" which was presented at the IEEE International Solid State Circuits Conference on February 14, 1974 at Orsay, France. See also, for example, a paper entitled "X and KU Band Amplifiers with GaAs Schottky Barrier Field Effect Transistors"
by Weiner Baechtold in the IEEE Journal of Solid State Circuits Volume, SC-8, No. 1, February 1973. High speed digital switching is combined with high speed linear amplification in a high speed pulse amplitude modulation device that is described in the paper entitled "Performance of Dual Gate GaAs MESFETS as Gain Contro]led ~.

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Low Noise Amplifiers and High Speed Modulators" by Liechti in the IEEE Transactions on ~licrowave Theory and Techniques Vol~
MIT-23, No. 6, June 1975. However, all of the devices in the above cited references only utilize MESFETS for h;gh speed.
The television receiver herein disclosed is also novel in that it includes a unique combination of surface wave device technology with MESFET device technology in the RF-IF
section. A MESFET mixer provides frequencv shifting while channel selectivity is provided by a single low loss surface acoustic wave device (i.e. SWD) bandpass filter. This SWD
filter has a sharp passband - stop band transition. For example, signals 1~5 MHZ above the sound carrier in the passband are attenuated by greater than 65dB. Conversely, the filter has a low insertion loss for signals in the passband. The preferred , - .
embodiment is a unidirectional SWD filter which has an inband insertion loss of less than 3.5dB, ~ ~:
The SWD filter also has a high center frequency and a passband which is a small percentage of the center frequency.
Thus, the SWD filter is readily implemented with~piezoelectric ., ~
material having a relatively low coefficient of coupling.
~uartz has such a characteristic, and it is a preferred substrate material. By comparison, prior art television receivers have - . ,:
employed SWD filters but only at a much lower IF fre~uency of45 ~'Z Accordingly, their passband was a relatively large percentage of their center fre~uency and thus the devices needed piezoelectric material having a relatively large co- -efficient of coupling such as lithium niobate. Lithium niobate, however, has piezoelectric characteristics which are highly sensitive to temperature change in the 0-70 C temperature range. As a result, they require .~r .

temp rature compensation circuitry. By comparison, the piezoelectric characteristics of quartz are relatively insensitive to t~mperature change in the 0~70 temperature range; and thus no temperature compensation circuitry is required.
Prior art television receivers include RF tuned filters before the mixer to insure that image frequencies of the selected channel are sufficien~ly attenuated at the mixer input so as to not produce interfering mixer output signals. To accomplish this, the bandwidth of the tuned RF filters are only several channels wide, and the center frequency is adjusted to align with the selected channel. RF filters which pass all of the channels of the VHF and UHF band at one time cannot be utilized because image frequencies would destroy reception in the selected channel.
To demonstrate the abovepoint, consider the following. VHF
television channels exist from 55.~5 ~HZ to 71.75 MHZ, and from 177.25 MHZ to 215.75 M~Z; while UHF channels exist from 471.25 MHZ to 889.75 MHZ. Also, as is known in the art, the image frequencies of a selected channel lie at 2XlF above the selected channel. Thus, channel 6, for example, which is .
centered at 87075 MHZ, has an image frequency in a conventional television receiver at (87.75 +90) MHZ. This equals 177~75 MHZ-which is within channel 7. Similarly, the picture carrier for u ch~nnel 14 is 471 MHZ; and thus its image ~requency is 561 MHZ-which is within channel 29. Accordingly, prior art receivers need an RF tuner before the mixer to filter image frequencies.

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35~i In the prior art, RF tuning is accomplished by varactor ilters, mechanically variable capacitors, ctc. Ilowcvcr, thesc are both expensive and difficult to align. By comparison, the television receiver o~ the disclosed invention has no tuners in the RF section. In one preferred embodiment, ~ fixcd bandpass filter is included which pass~s the entire low VHF band, a second filter is included which passes the entire high V~-lF band, and a third filter is included which passes the entire UHF
band. These flxed fiLters are simple in design and eliminate alignment problems. The disclosed in~ention also includes a mixer having an IF output fre~uency of between 300 MHZ and 400 MHZ. As a result, image frequencies of the selected channel are placed at least 600 MHZ above the selected channeL
where they are easily rejected by the fixed bandpass RF filters.
The mixer output of the disclosed receiver couples to a channel selection filter which is implemented by an improved surface wave device. Prior art television receivers also used surface wave device channel selection filters. ~owever, the SWD filter of the disclosed inve~tion is an improvement from the prior art in that it is constructed on a relatively small substrate area. The area used by a SWD filter is proportional to its center frequency. Conventi ~ 1 receivers have an IF frequency of 45 MHZ, and thus, the SWD's used therein require substantially more space.

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Another novel aspect of the disclosed system is that the IF section includes two mixers at the output of the SWD
filter for ~requency shifti~g the selected channel back to baseband. The irst IF mixer frequency shifts the selected channel to approximately 45 MHZ. Most of the gain of the system is then added to the selected channel signals. The second IF mixer is utilized to synchronously detect signals in the selected channel. This architecture permits the described high IF system advantages to be achieved, withou~ introducing high frequency feedback problems.
As previously described, prior art television receivers insert at least 2Q to 30 dB of gain in the RF section to achieve a low system noise figureO System noise~fi~ure~equals NFl N3~2-1 ~ NF2-1 + NF2-1 + ...
Gl Gl G2 -- -wherein NFi and Gi are the nQise figure and~ain of the ith functional block in the system. Thus, inserting a large gain in the first functional blocks (i.e. the RF section) lowers system noise figure by making it independent of noi~e figure of subsequent circuitry.
The disclosed invention has a unique architecture which achieves;low noise figure, low in~ermodulation distortion and low cross-modulation distortion simultaneously. ~ In the preferred embodiment, the disclosed invention has a maximum gain in the channel selecting section which is less than 10 dB;
that utilizes circuit elements therein which individually have low noi~e figures yield a low system noise figure, while the low gain allows RF and IF circuit elements to operate within their dynamic range without generating high odd order output terms.

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The architecture of the disclosed invention is also novel in that it has two IF frequencies about which the system gain is selectively distributed. In one preferred embodiment, a irst mixer shifts the selected channel to about 330 MHz. The output o~ this mixer couples to the channel selection filter;
and the output of the channel selection filter couples to a second mixer which frequency shifts the filtered selected channel to a second IF frequency of approximately 45 MHz. Most of ~he gain of the system (approximately 60 dB) is inserted at the lower IF
frequency after the channel selection ilter. Gain at the high IF is small as pointed out above. As a result, several advantages of a high IF system - such as simplified filtering of image frequencies-are~obtained, while feedback inherent in a high gain-high IF section is avoided.
The disclosed system is also simpler and potentially less expensive than prior art television receivers in that the low gain RF section makes feasible integration of a major portion of the receiver on a single semiconductor chip.
Incorporated in the disclosure are two differently organized ~E-IF sections which are suitable for semiconductor chip integration.

SUMMARY OF_THE INVENTION c~ s~ I Pc to~
In one aspect o the invention, a ~ n r is provided which includes a MESEET mixer in combination with an acoustic surface wa~e device filter. The MESFET mixer simultaneously receives a plurality of channels and mixing signals of a selectable frequency, and in response thereto frequency shifts sele~table channels to a predetermined center frequency.

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The ~SFET mixer has almost pe~fect square law characteristics, and accordingly introduces extremely small distortion into the system. In one embodiment, the MESFET mixer handles interfering signal levels of greater than -~6dBm on its output with less than 1% cross-modulation distortion and -40dB
intermodulation distortion products. The acoustic surface wave device (SWD) filter has an input coupled to the output of the MESFET mixer. The SWD filter provides essentially all of the channel selectivity of the receiver. In a preferred embodiment, the ST~D filter is a three-phase low-loss unidir-ectional filter having a finger withdrawal type impulse response.This SWD has an in-band insertlon loss of < 3.5dB, and a stop-band loss at 1.5MHZ above and below the passband of > 65dB.

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:BRIEF DESCRIPTION OF THE DR~ INGS
I
The novel features believed characteristic of the invention are set forth in the appended claims; the invention itself, however, as well as other features and advantages thereof, may best be understood by referring to the following detailed description of particular embodiments when read in reference to the accompanying drawings, wherein:

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FIGURE ]. is a block diagram illustrating a channel selector constructed according to the invention.
FIGURES 2a-2e are a series of frequency diagrams illustrating signals at selected points on the channel selector of FIGURE 1.
FIGURE 3 is a timing diagram illustrating the operation of a phase locked loop within the channel selector of FIGURE 1.

FIGURES 4a and 4b are graphs, respectively, illustrating the amplitude of in-band and out-of-band signals at various points in the channel selector of FIGURE 1.

~ FIGURES 5a, 5b and 5c are detailed circuit diagrams of ; RF filters within the channel selector of FIGURE 1. -FIGURE 6 is a detailed circuit diagram of a two-by-one switch within the channel selector of FIGURE 1.
FIGURE 7 is a detailed circuit diagram of an RF
amplifier within the channel selector of FIGURE 1~ ~
FIGURE 8 is a detailed circuit diagram of a MESFET ~-mixer within the channel selector of FIGURE 1.
FIGURE 9a is a plan view of a MESFET transistor suitable for application in the MESFET mixer of FIGURE 8.
FIGVRE 9b is an enlarged cross-sectional view taken along the line A-A of FIGURE 9 a.
FIGVRE 9c is a greatly enlarged cross-sectional view of a dual gate MESFET device showing one source-drain pair thereof and which comprises another MESFET transistor suit-able for application in the MESFET amplifier of FIGURE 7.
FIGURES 9d and 9e are graphs illustrating the operational characteristics of the MESFET transistor shown in FIGURE 9a.

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FIGURE 10 is a circuit diagram of a second two-by-one switch within the channel selector of FIGURE 1.
FIGURE lla is a circuit diagram of a surface wave devi.ce acoustic filter within the channel selector of FIGVRE 1.

; FIGURE llb is a graph illustrating the magnitude-frequency characteristic of the surface wave device included in the filter of FIGURE lla.
FIGURE llc is a perspective view illustrating one ~.
embodiment of the surface wave device included in the filter of FIGURE lla.
~FIGURE lld is a partially perspective diagrammatic ;view illustrating the surface wave device included in the filter of FIGURE lla wherein the impulse response of the surface wave device has been shaped by electrode finger withdrawal FIGURE lle is a diagram of a phase shifting circuit .-for use in the input circuitry o the filter of FIGURE lla.
FIGURE llf is a phase diagram illustrating how the 20 phase shif-ting circuit of FIGURE lle can produce a 120 :~
phase shift.
FIGURE llg is a diagram of a circuit for producing a 60 phase lag. ..
FIGURE llh is a circuit diagram of the input circuitry ~ -included in the filter of FIGURE lla.
FIGURES 12a-12b are detailed circuit diagrams of a linear amplifier within the channel selector of FIGURE 1.
FIGURE 13 is a detailed circuit diagram of a mixer within the channel selector of FIGURE 1.
30FIGURE 14a is a detailed circuit diagram of a surface wave device oscillator within the channel selector of ~, 3~
FIGURE 1.

FIGURE 14b is a top plan view illustrating the surface wave device included in the oscillator of FIGURE 14a. :
FIGURE 15 is a detailed eircuit diagram of a high gain linear amplifier within the channel selector of FIGURE 1.
FIGURE 16 is a detailed eircuit diagram of a synchronous detector within the ehannel selector of FIGURE

FIGURE 17 is a detailed circuit diagram of a L-C ~ ~ -oscillator within the channel selector of FIGURE 1.
FIGURE 18 is a detailed circult diagram of a phase ` :
detector, a ramp-generator, and a loop filter within the channel selector of FIGURE 1.
FIGURE 19 is a detailed circuit diagram of a voltage .
controlled oscillator within the channel seleetor of .
FIGURE 1.

FIGURE 20 iq a block diagram of a television reeeiver whieh includes the`ehannel seleetor of FIGURE 1.
FIGURE 21 is a bloek diagram of an alternative embodiment for the channel seleetor of FIGURE 1.

- 12a -DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS
Referring to Figure 1, the channel selecting portion of a television receiver constructed according to the invention is illustrated in block diagram form. The selector includes a VHF antenna 10 having an output coupled via a lead 11 to a fixed bandpass filter 12 and to another fixed bandpass filter 130 Signals on lead 11 are herein designated Sl(f). Filter 12 has a center freauency of 69 MHZ and a 3dB bandwidth of 34.6 MHZ to thereby permit passage of the low VHF television channels. Similarly, filter 13 has a center frequency of 193 MHZ and a 3dB bandwidth of 44.5 ~IZ to provide filtering -of the high VHF television channels. The output of filter 12 is coupled via a lead 14 to~one of the inputs of a 2 x 1 switch 15, while the output of filter 13 is coupled via a lead 16 ~-to a second input of switch 15. Switch 15 operates to select either signals from filter 12 or 13. The filter-switch combination inserts approximately a 1 dB loss on signals in -the passbands.
The output of switch 15 is coupled to an RF amplifier 17 via lead 18. Amplifier 17 has a varia~le gain which is con- ~;
trolled by signals on an AGC line 19. The m~m gain of ampli~
fier 17 is approximately 4 dB. The noise figure of the amplifier is approximately 3dB.
The output of amplifier 17 couples to one input of a MESFET mixer 20 via a lead 21. Mixer 20 has a fixed gain of approximately 4dB, and a noise figure of approximately 8 dB.
A second input of mixer 20 is coupled to a VHF voltage controlled oscillator 22 via a lead 23. Oscillator 22 generates local oscillator (LO) signals on lead 23 of a selectable frequency -in the range 385-541 M~IZ. In response thereto, mixer 20 frequency shifts the RF signals on lead 21 to a new IF frequency ~r . . ~ . .

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range. The frequency of the LO signals on lead 23 is selected such that the channel to be received is frequency shifted to a predetermined high IF of between 300 MHZ to 400 M~IZ. In one embodiment, this predetermined high IF is approximately 330 MHZ. The frequency shifted signals are generated on a lead 24 and are designated S2(f).

-13a-"~ ~, Lead 24 couples to one input of a 2 x 1 switch 25, while a second input of switch 25 is coupled via a lead 26 to receive frequency shifted UHF television channels. Switch 25 is identical in construction to switch 15. The circuitry for frequency shifting the UHF television channels, which is labeled as UHF RF
SECTION, is similar in construction to the VHF-RF section, and is described in detail infra.
The output of switch 25 is coupled to a surface wave device (SWD) filter 28 via a lead 27~ Filter 28 has a passband which is shaped to pass only one of the television channels on lead 27. In particular, filter 28 passes th~ television channel at the predetermined high IF. The fixed RF filters in combination wi~h the SWD filter provide essentially all the filtering in the system. In a preferred embodiment, filter 28 B is a three phase unidirectional filter which has ~ insertion loss. Typically, the 108s through ilte~ 28 is less than 3.5 dB
in the passband. Conversely, out of band signals are greatly attenuated.
The output of filter 28 is coupled to an IF amplifier 29 via a lead 30. ~he signals on lead 30 are designated herein as S3 (f) . Amplifier 29 has a variable gain which is controlled by AGC signals on a lead 31. The maximum gain of amplifier 29 is ~pproximately 30 dB. Thust amplifier 29 is the first high gain element in the system. To this point, the gain of the system is characterized as belng no larger than necessary to obtain the desired system noise figure. Noise figure for amplifier 29 is approximately 4 dB.
: The output of amplifier 29 couples to a mixer 32 via a lead 33. A second input of mixer 3~ is coupled via a lead 34 to the output of an oscillator 35. Osclllator 35 has a surface wave device resonator 36 as its frequency controllins element. In a preferred embodiment, resonator 36 and oscillator 35 operate to generate a mixing signal on lead 34 having a frequency o~ 285 MHZ.

.'3~6 The 285 MHZ signal on lead 34 is mixed with the si~nal on lead 33 to thereby generate signals S4(f) on the output of mixer 32. Signal S4(f) is similar to signal S3(f) except that it is shifted down in fre~uency by 285 MHZ. Thus it has a picture c rrier at approximately 45 MHZ.
Additional gain is add~d to the signal of the selected channel after it has been frequency shifted to the lower IF fre-quency of 45 M~Z. Mixer 32 adds a fixed ~ain of +10 dB~ }
the output of mixer 32 couples via a lead 37 to an IF amplifier 38 which has a maxLmum gain of +50 dB. The gain of amplifier 38 is ~aried by an AGC signal on a lead 39. By inserting most of the gain at the lower IF frequency, system stability is increased since feedback through parasitic capacitances, radiation, etc. is much less at 45 MHZ than 330 MHZ.
The output of amplifler ~ is coupled via lead 40 to a tank circuit 41, a synchronous detector 42 and the phase detector 43. Tank circuit 41 has a center ~requency of approximately 4S MHZ. Synchronous detector 42 has a second input coupled via a.
lead 44 to an oscillator 45~ Oscillator 45 generates clock signals at a fixed frequency of 45 MHZ on lead 44. The signals on lead 44 are in phase with the 45 MHZ picture carrier on lead 40.
Detector 42 mixes the signals on leads 40 and 44 to generate output signals S5(f) on a lead 46. Signal S5(f) contains the selected television channel with the picture carrier at zero Hz and the sound carrier at 4.5 MHæ. ~ead 46 is then coupled to conventional televis~on circuitry for sepaxating the sound signal from the picture signals and for reproducing the sound and the picture from these si~nals respéctively in a conventional manner.
The synchronous clock signal on lead 44 is kept in phase with the picture carrier on lead 40 by means of the phase detector 43 operating in conjunction with oscillator 45. Phase detector 43 generates phase detection signals PD2 on lead 48 which indicate the phase difference between the picture carrier on lead 40 and the oscillator signal on lead 47~ Signals PD2 maintain a 90~ phase difference between the signals on leads 40 and 47. This phase difference is compensated for by oscillator 45 which maintains a compensating 90 phase difference between the oscillator signals on leads 44 and 47.
To complete the phase locked loop, lead 48 couples to a loop filter 49, and he output of filter 49 couples to a summer 50 via leads 51. Summer 50 has a second input coupled to a con~
troller 52 via a lead 53, and a third input coupled to a ramp oscillator 54 via lead 55. ControlLer 52 and oscillator 54 operate to provide a coarse voltage for selecting one channel ~rom another.
Channel selection switches 56 are coupled to controller 52 via leads 57. Switches 56 generate digital signals on leads 57 indicating the selected channel. Controller 52 contains 2 digital to analog converter which generates a coarse channel selection voltage on lead 53 in response to the digital signals. The signals on lead 53 are summed with the coarse signais on leads 51 and 55 to thereby pro~ide a loop which is phase locked to the picture carrier of the selected channel. The output of summer 50 is coupled to VHF voltage controlled oscillator 22 via leads 58. Oscillator 22 generates LO
signals on lead 23 of a ~requency ranging from 385 MHZ to 541 MHZ
i~ response~to the phase detection signals on leads 58 thereby completing the loop.
The UHF-RF section of the FIGURE 1 channel selector begins with a UHF antenna 59. Antenna 59. is coupled to a high pass filter 60 via a-lead 61. Filter 60 has a fixed 3dB cutoff frequency of approximately 380 MHZ. The output of filter 60 couples to the input : of an RF amplifier 6Z via a lead 63. Amplifier 62 also has an AGC
input which is connected to receive AGC signals on lead 19. The maxLmum gain of amplifier 62 is 4dB, and its noise figure is also approximately 4dB. A low pass filter 65 has its input coupled to the output of amplifier 62. Filter ~5 has a fixed 3dB cutoff frequency of approximately 936 MHZ. Thus, filter 65 in combination with filter 60, pass the entire UHF band and reject other frequencies.
A mixer 67 receives the UHF band of signals from filter 65 via a lead 66. Mixer 67 simultaneously receives mixing signals of a selectable frequency from a UHF voltaye controlled oscillator 68.
These selectable frequencies range from approximately 801 MHZ to 1215 MHZ. The particular frequency at any time instant is generated in response to channel selection switches 56 so as to frequency shift the selected channel to-the predetermined high IF.
The overal1 oper.ation of the above described FIGURE 1 structure is as follows. Antennas lO,and 59 receive radiated electromagnetic signals which inc:lude the VHF and UHF frequency spectrum. The signals received by antenna 10 are applied to non-tuned filters 12 and 13 which respectively pass the entire low VHF and high ~F frequency spectrum. Similarly, the signals received by antenna 59 are applied to non-tuned.filbers 60 and 65 5~;t~s which pass the entire UHF frequency spectrum. Sw~*c~ 15 and 25 select one of these three frequency spectrums in response to logic signals from the channel selection switches 56.
~ ESFET mixers 20 or 67 then frequency shift the selec~ed frequency spec~rum ~o a predetermined IF in the 300 MHZ-400 MHZ
range. Mixer 20 frequency shi-fts VHF signals, whereas mixer 67 frequency shifts UHF signals. Both mixers have a large dynamic range o~er which their output is an almost perfect product of their inputs. As a re~ult, the system has improved performance.
For example, mixers 20 and 67 can handle >+6dBm interfering signal levels on their output with less than 1% cross-modulation distortion.
. Switch 25 then couples one of the mixers to the input of SWD filter 28. Filter 28 ~reatly attenuates signals outside the selected channel. In particular, the stopband of filter 28 is notched such that the lower adjacent sound carrier and upper adjacent picture carrier are attenuated by more than 6SdB. All other out of band signals are attenuated by at least 55dB.
Conversely, in band insertion loss of ilter 28 is onIy 3.5dB.
Output signals from filter 28 are sent to amplifier 29-which is the first high gain component of the system. Amplifier 29 has a maximum gain of 30dB. In comparison, the total gain of the system before amplifier ~9 is less than lQdB. As a result of the low RF-IF gain, intermodulation distortion and cross modulation distoxtion is decreased. Further, system stability is improved since high requency feedback due to parasitic capacitance, radiation, etc. is avoided. At the same time,~low system noise igure is achleved ~y circuit elements which indlvidually have a low noise flgure in combination with a sufficiently high gain.
- The low gain RF-IF section aLso makes feasible integration of all of the channel selectors, except the fixed filters, on a single semiconductor chip. An outline of such a chip t S
designated by the dashed line 69 in FIGU~E 1. The process f~r fabricating the chip includes a combination of steps presently well known for constructing MESFET devices and bipolar devices. A
second chip is utilized to construct SWD filter 28 and S~D

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resonator 36. The fixed filters 12, 13, 60 and 65 are construc.ted of discrete components.
Signals in the selected channel at the output of amplifier 29 are frequency shifted to 45 MHZ by mixer 32. Then they are further amplified to -lOdBm by amplifier 38. The selected channel video signals are then reduced to baseband by synchronous detector 42; while the sound carrier of the selected channel is shifted to 4.5 MHZ. A 4.5 MHZ trap 70 removes the sound from the video signals, and a low pass filter 71 removes all signals except the video of the selected channel~ The video signals at the output of filter 71 are sent via lead 72 to video processing circuitry, while the audio ignals at ~.5 MHZ on lead 46 are sent to audio processing circuitry. This audio-video circuitry is described infra in conjunction with FIGURE 20.
Referring now to FIGURES 2a-2e, there is illustrated a set of frequency diagrams of signals Sl(f) - S5(f). In FIGURE
2a, the low VHF band is shown generally at 75a,the high VHF band is shown generally at 75brand the UHF band is shown at 75c. Each of the bands is comprised of a plurality of channels; and each channel has a ~requency spectrum which is assigned as illustrated in detail ~t 76~ The frequency allocation and type of modulation of signals within each channel is a well known standard that is fixed by the FCC. FIGURæ 2b is an exemplary frequency diagram of signal S2(f)- In the example illustrated, signàl S2(f) contains channels in the low VHF spectrum. The selected channel is ne.ar 330 MHZ. The frequency of the LO signal on lead 23 minus the frequency of the picture carxier from the selected channel equals 330 MHZ. Since the mixing frequency is higher than the picture carrierl the frequency spectrum at the output of mixer 20 is inverted from the input frequency spectrum as shown at 77.
FIGURE 2c illustrates the frequency spectrum of signal _ .

s3(f). Signal S3(f) is the output of the channel selecting filter 280 Thus it basically contains only frequencies within the selected channel. Signal S3(f) is then amplified and frequency shifted down by 285 MHZ. The result is signal s4(f) as illustrated in FIGURE 2d.
Signal S4~f) is further amplified, and then synchronously . detected by detector 42. These operations producP signal S5(f) as illustrated in FIGU~E 2e. Note that the mlxing action of synchronous detector 42 again inverts the frequency spectrum of the seIected channel~ Thus, the picture carrier of the selected channel is at 0 HZ, and the sound carrier of the selected channel is at 4.5 M~Z as shown at 78.
FIGURE 3 is a timing diagram illustrating the operation of the channel selecting phase locked loop of FIGUR~ 1. As shown therein, signal PD4 on lead 58 is comprised of components PDl-PD3.
Signal PD3 constitutes a coarse channel selection voltage which is produced by controller 52. Signa:Ls PD2 and PDl provide the fine tuning for the phase locked loop. In the example of FIGURE
3, one particular channel'is selected duri'ng a first time inter-valQTl, while another channel is selected duxing a'time interval T2. Signal PD3 provides the coarse voltage for channel selection.

Signal PDl compensates for any DC offset between signal PD3 and ,a~ .
the desired voltage level.,~ signal PD2 provides a dynamic correction voltage to eompensate for instantaneous phase or frequency differences between the signals on leads 40 and 44.
F IG U~e~s 2,a ~,e!.
The magnitude of,the signals of ~ are illus~
trated in FIGURES.4a and 4b. FIGURE 4a illustrates the magnitude of signals in the desired channel, wherPas FIGURE 4b illustrates ~he magnitude of interfering signals in a channel that is two .channels removed. Curves 81-84 iIlustrate the magnitude of signals in the ~esired channel at various points in the system when the incoming signal strength is 0 dBM, -35 dBM, -55dBM, and -85 dBM respectively. As these curves illustrate, gain is first added by amplifier 38, then by amplifier 29, and finally i35~

by amplifier 17 as the input signal strength in the selected channel decreases. In particular, the RF section has no gain unless the input signal strength in the selected channel is less than -55 dBm. And the RF section reaches its maximum gain of +3 dBm when the input signal strength of the selected channel is between -55 d~m and -85 dBm As previously pointed out, the low gain RF section in combination with the MESFET RF amplifiers and MESFET RF mixer provides a system having superior channel discrimation capability.
This is exemplified in FIGURE 4b by the relative signal strength `, of the selected and unselected channels. For example, curve 85 of FIGURE 4b illustrates the case where the input signal strength of the selected channel is~-55 dE~ and input signal strength two - channels removed is +1 dBm. Similarly, curve 86 illustrates B the case where the input,signal strength in the selected channel is -35'd~m and the input signal strengt:h two channels removed is +2 dBm. The most stringent re~uirement for the receiver is when the ~irst RF amplifier,xequires gain (i.e., when the signal strength of the ~esired channel i5 less than -55 dBm). This is because the added gain increases nonlinearities in the RF section and aggravates the cross modulation and intermodulation dis-tortion~ Thus, curve 85 illustrates the most stringent condition for the system. Under the conditions of curve 85, RF amplifier 17 and mixer 20 must have cross modulation distortion and inter-Th,s modulation distortion of less than 1%. ~ c~i~ requirement is ' met, since the MESFET designs herein described have less than 1%
a~l cross modulation distortion~intermodulation distortion when their output signal level is less than +6 dgm.
After the selected channel has been shifted to 330 , MHZ~signals two, channels removed rom the selected channel are greatly-attenuated. Mixer 20 has a tuned output which attenuates ~21-3~;~

two channel removed signals by -~ DB. SWD filter 28 atten uates all out of band signals by at least -53 DB. At the same time, mixer 20 adds 4 DB of gain to the desired signal, and filter 28 inserts only 3.5 DB of loss to signals in the selected channel.
The details of the various block of FIGURE 1 will now be descri~ed in conjunction with FIGURES 5-19. Referring first to FIGURE 5a, a circuit diagram of bandpass filter 12 is therein illustratedO Basically, filter 12 is comprised of two series resonant LC circuits 91 and 92 and one parallel resonant LC cixcuit 93. Filter 13 is similarly structured.
A primary function of RF filters 12 and 13 is to pass one band of channels while rejecting image frequencies by an amount sufficient to eliminate perceptible picture interferenceD Since this system uses a 330 MHZ IF, all image frequencies are 660 MHZ above the frequency of the desired channel. Thus, the low VHF images are rejected by the three poll bandpass filter of FIGURE 5a by greater than 80 DB. In the system of FIGURE 1, just perceptible picture interference occurs when image frequency signals are less than 36 DB below the level of the desired signal at the filter output. Thus, for example, the filter of FIGURE 5a provides adequate image frequency rejection when the desired picture signal level is ;~
at -55 DBM and the image frequency level is at -11 DBM. Image frequencies for the low VHF and the high VHF spectrum lie within the UHF spectrum, and the level of UHF signal at the input of the V~IF antenna can normally be expected to be less than -11 DBM.
FIGURES 5b and Sc are circuit diagrams of a high pass filter and a low pass filter suitable for use by the UHF
radio frequency section of FIGURE 1. All of the UHF images fall outside of the TV band. The first image falls at approximately .~, .
..

3~

1130 MHZ and the last image falls at approximately 1545 MHZ. Thesefrequencies are allocated for aircraft navigation, with the TACAN
system having the highest power output. But TACAN power output is only 5 KW with a 1.~ x 10 to 1 duty cycle. And thus the TACAN
signal is 50 DB lower than that transmitted by a 1 MW TV trans-mitter. Therefore, the filters of FIGURES 5b and 5c provide adequate rejection of image signals for the UHF band.
FIGURE 6 is a detailed circuit diagram of switch 15.
one portion is basically comprise~ of a diode 101 connecting ieads 14 and 18, and an RLC bi~s network 102 having a control input 103. A DC voltage control signal SELLOVHF is applied to lead 103 for selectively turning diode 101 on or off to thereby select or deselect the low VHF signals on lead 14. Another portion identical to the one descxibed above couples lead 18 to lead 16, and is utilized to select and deselect the high VHF
channels.
Referring next to FIGURES 7 and 8, there is illustrated a detailed circui~ diagram of RF ampli~ier 17 and mixer 20, respectively. Basically, amplifier 17 is comprised of a dual gate MESFET transistor 1~1 having a source coupled to a bias resistor and coupling capacitor 112, and a drain coupled to an L-C bandpass circuit 113. The gain of transistor 111 is varied about the levels indicated in FIGURE 4a by an automatic gain control signal AGCRF.
Signal AG~RF couples to a gate of transistor 111 through a voltage dividing network 114.
SLmilarly, mixer 20 is comprised of a single gate MESFET transistor 121 having a source coupled to a bias resistor and coupling capacitor 122, and a drain coupled to an L-C bandpass circuit 123. Circuit 123 has a center frequency of 330 MHZ. The mixing signals from VHF VCO 22 are coupled to the gate of transistor 121 via an ~C circuit 124.

;hl~,3~

It should b~ emphasized that the Schottky har~ier gate structure of the MESFET transistor of amplifier 17 and mixer 20 yield significant performance improvements over other known devices. Narrow gate depletion mode MOSFET devices have a high frequency response, but they approximate square-law transfer characteristics over only a very narrow range of gate bias. Since departure from square-law opera*ion results in cross modulation and intermodulation distortion, the devices are restricted to a small dynamic range. In comparison, the MESFET devices 111 and 121 have nearly ideal square-law characteristics over the entire operating range indicated in EIGURE 4a. JFET devices also have a good square-law transfer characteristic, but their high frequency performance is greatly reduced because of parasitic capacitance and process difficultites that limit their usable geometries. ~
Referring now to FIGURE 9a, there is illustrated a `
complete MESFET device suitable for use as mixer 20. The MESFET of FIGURE 9a has a closed gate 131. Gate 131 is approximately 80 mils in length. The gate metal is approximately 0.3 mils wide, while the Schottky barrier gate in contact with the gate metal i5 approximately 0.15 mils wide. The source consists of five fingers 132-136 having lengths of 32.7 mils, - 4.1 mils, 4.1 mils, 7.7 mils, and 12.1 mils respectively.
The width of these fingers is approximately 0.3 mils. The drain consists of fingers 137-140.
FIGURE 9b is an enlarged cross-sectional view taken alon~ the line A-A of FIGURE 9a. The MESFET is constructed on a silicon substrate 140~having a P-type impurity. The resistivity of substrate 140 is approximately 50 ohm-centimeters. Each of the source electrodes 132-L36 ,;

, 3~

couples to an N~ doped region 142~146 respectively. These doped regions have a resistivity of approximately 0.005 ohm-centimeters. The doped regions extend beyond their cor=esponding electrodes by ~,' '', .

~, : ,. .

, '` ,:
:`
', -24a~ -35i~

approximately 0.3 mils and are separated from the spaced apart ; gate electrode by approximately 0.15 mils. Similarly, the metal electrodes 137 140 which form the drain electrodes are coupled -~
to underlying N+ doped regions 147-150, respectively. The ~e~-~e 5'~ v~
and geometry of the drain doped regions ~ similar to that of the source doped regions.
FIGURE 9c is a greatly enlarged cross sectional view of one source drain pair within a dual gate MESFET device suitable for use with amplifier 17. The entire device is constructed similar ~o that of FIGURE 9a, with the modification that two gates are interleaved between the source and drain electrodes. In FIGU~E 9c, source electrode 132a and drain electrode 137a correspond to electrodes 132 and 137 of FIGURE 9a. Gate electrodes 131a and 131b occupy the space of electrode 131 in FIGURE 9a.
FIGU~E 9d illustrates the I-V characteristics of the devices of FIGURE 9a, while FIGURE 9e illustrates their trans-conductance as a function of gate voltage. In an ideal square-law device, the drain current is proportional to the square of the gate voltage. And since transcon~uctance equals the partial derivative of drain current with respect to gate voltage, the transconductance is directly proportional to gate voltage for ideal square-law operation. FIGURE 9e demonstrates such a Iinear relation between the transconductance and gate voltage for the MESFET device of FIGURE 9a. !'' Referring now to FIGURE 10, a circuit diagram of switch 25 is illustrated. Switch 25 is constructed identical to the previously described switch 15 of FIGURE 6. The signals on leads 24 and 26 are selectively coupled to the output lead 27 via DC control S E L L~
signals select ~HF (SELVHF) and select UHF (~VHF) respectively.
FIGURES lla-llh illustrate the details of SWD filter 28.
Filter 28 is comprised of a surface wave device chip 160 having an L-C input circuit 170, and an L-C output circui~ 190 for f~3~

impedance matching and phase shifting siynals to and from SWD
de~ice 160. ~eads'172-174 and 192-194 couple circuits 170 and 190 to SWD 160 as illustrated in FIGURE lla.
D,evice 160 has a magnitude-frequency characteristic as illustrated in FIGURE llb. In particular, device 160 attenuates signals which are 1.5 MHZ above the picture carrier of the desired channel by at least 65 db. At that frequency, the sound carrier of the channel adjacent to the desired channel is present. It is important to greatly attenuate this sound carrier because it is translated into the video signals of the selected channel by synchronous detector 42. That is, the sound carrier at 46.5 MHZ
is'translated to 1.5 MHZ by detector 42. In the receiver of FIGURE 1, ju~t perceptable picture lnterference occurs when the sound carrier at 46.5 MHæ is passed- through SWD filter 28 with a magnitude that is within 36 dB of the picture carrier at 45 MHZ.
Thus, SWD filter 28 with its greater than 65 dB adjacent channel rejection enables the receiver of FIGURE 1 to have ~ood picture reception even though signals at the filter input in an adjacent channel are much larger than signals in the desired channel. For example, if the signal level of the desired channel is -55 DBM at the input o~ filter 2~, then the 46.5 MHZ sound carrier at the input oE filter 28 can be as high as -26 dBm and ~he receiver of FIGURE 1 will meet the requirement of a -36dB difference between the in band and out band signals at the output of filter 28. By comparison, prior art television receivers typically have per-cepkable picture interference when the adjacent channel sound carrier is -40 dBm.
FIGURES llc and lld illustrate the physical structure of one embodiment of SWD 160. Basically, device 160 is comprised of a piezoelectric substrate 161 which in a preferred embodiment is made of quartz. Quartæ has a desirable feature in that the velocity of surface waves through quartz is practically indepen-dent of temperature. The dependence of velocity on temperature is of considerable importance since velocity effects the center frequency of the filter as described below.
Three electrically independent conductors 162~164 are ,.ti3r5~

disposed on substrate 161. Each conductor has corresponding finger electrodes 162a-164a which axe disposed on substrate 161 in a comb like fashion. Fingers 162a-164a are equally spaced.
The distance between two consecutive fingers on the same con-ductor is one wavelength of the center frequency of the filter.
The velocity of a surface wave in quartz is approximately 3,300 meters per second, and the center frequency of device 160 is approximately 330 MHZ. Thus, the distance between two consecu- -tive 162a fingers for example is approximately 10 x 10 6 meters.
Conductors 162-164 are coupled to the input circuitry 170 of FIGURE lla via leads 172-174 respectively. Input cir-cuitry 170 generates voltages on electrodes 162-164 which are 120 out of phase with each other. This phase relationship generates a unidirectional surface wave on substrate 161. That is, waves in the forward direction add constructively, while waves in the reverse direction add destructively. As a-result, device 160 has a low insertion loss for signals in the passband.

In particular, the loss through filter 160 is no more than 3.5 Pa~
~ DB. United States ~a~e*~ 3,686,518 issued August ~l 1972 to ~artmann et al and assigned to Texas Instruments Incorporated includes additional structuraI details of unidirectional surface wave filter 160.
FIGURE lld illustrates a method for shaping the impulse response of surface wave filter 160. The method therein illus-trated is known as the ~inger withdrawal method. It involved removal of groups of fingers from selected portions of substraté
161. Basically, the amplitude o the impulse response in thos~
regions from which the fingers are removed is reduced below the value which it would ha~e if the fingers had not been removed.
The method thus provides the capability to control the relative ~2-7--b3~`~

amplitude of the impulse response along the length of substrate ~61. The desired impulse response is obtained by taking the inverse Fourier transfer of the requency response of FIGURE llb;
and then fingers 162a-16:4a are selectively removed in accordance with the desired impulse response. Further details of the finger withdrawal method are contained in U. S. Patent No. 3,946,342 issued March 23, 1976 to Hartmann and assigned to Texas Instruments ; Incorporated.
FIGURES lle-llh illustrate a method for determining the value of inductors 175-177 and capacitor 178 comprising input circuitry 170. As was previously described, one of the functions of circuit 170 is to generate voltages on leads 172-174 which are ~: shifted in phase from each other by 120. FIGURE lle illustrates that this 120 phase shift can be achieved by a circuit 180 which produces a 60 phase lag between the voltace on lead 172 and the ~oltage on lead 173, in combination with a grounding of lead 174.
B This point is:~urther illustrated by the phaseS diagram of FIGURE llf.
; A circuit for producing a 60 phase lag is illustrated in FIGURE llg. The circuit consists of a pi shaped R-L-C network consisting of a capacitor 181,and inductor 182, a capacitor 183, and a resis~or 184. Included in FIGURE llg are two equations relating the angle of phase lag between leads 172 and 173 in terms of components 181-184. In the case at hand, the angle of phase is 60~, and resistor 184 is the resistance between the electrodes 162 and 163 divided by 2. Thus, utilizing the equa-tions of FIGURE llg, values ~or ¦Xl¦ and ¦Xcl can be calculated.
Impedance 183 is physically implemented by coupling . inductor 175 across leads 173 and 174; and impedance 182 is physically implemented ky coupling inductor 176 across leads 172

-2~-

3~

and 173 as illustrated in FIGURE llh~ The parallel combination of inductor 175 and the capacitance between leads 173 and 174 due to the SWD electrodes is chosen to equal impadance 183.
Similarly, the parallel combination of inductor 176 and the capacitance across leads 172 and 173 due to the SWD electrodes is chosen to equal impedance 182. Inductor 175 may typically equal 30 nanohenries, while inductor ~ may typically equal 35 nanohenries as an example.
Inductor 177 is then added between leads 172 and 174 while capacitor 178 is added between leads 172 and 29 so as to match the impedance between leads 172 and 174. Typically, l7 inductor 177 is approximately 25 nanohenries, and capacitor is 10 picofarads.
To this point, the discussion with reference to FIGURES
lla-llh has concentrated primarily on the structure of input circuitry 170 and the input transducer of SWD device 160. It will be understood ~owever, that surface wave device 160 also has an output transducer on substrate 161 of a construction similar to that of the input transducer~ Also, output circuitry 190 of F~GURE lla is constructed similar to lnput circuitry 170.
Referring next to FIGURES 12a and 12b, a circuit dia-gram of RF amplifier 29 is therein illustrated. Amplifier 29 is comprised of stages 200 and 201. Stage 200 includes a single bipolar transistor 202 as i~s amplifying element, whereas stage 201 includes several DC coupled bipolar transistors in an integrated circuit 203 as its ampiifying element. Circuit 203 is indicated in FIGURE 12a as a single circuit element, and is shown in detail in FIGURE 12b.

Amplifier 29 provides a maximum gain of 30 DB to signals on its input lead 30. A small portion of the gain is 3~ ~

provided by stage Z00 which has a relatively good noise figure, whereas the remainder of the gain is provided by stage 201. AGC
control signal AGCIF couples to stages 200 and 201 as indicated in FIGURE 12a through bias circuits 216 and 217. The magnitude of signal AGCIF varies so as to keep the output of amplifier 29 at approximately -26~5 DBM. The variation of gain versus input signal strength for amplifier 29 was previously indicated in FIGURE 4a.
The output of amplifier 29 couples to mixer 32 which is illustrated in the circuit diagram of FIGURE 13. The basic mixLng circuit element utilized therein is a bipolar transistor 220. Transistor 220 has an emitter 221 which is coupled to the output of amplifier 29. Similarly, transistor 220 has a base 223 which is coupled.to SWD oscillator 35, Oscillator 35 generates mixing signals of 285 MHZ on the base 223 of transistor 220. As a result, sum and difference fxequencies are generated on the co~lector 225 of transistor 220. Collector 225 is coupled to an LC tank circuit 226 having a resonant frequency of about 45 MHZ.
The ou~put of tank circuit 226 is coupled via a tapped transformer 227 to lead 37 and slgnals S4(f) are generated thereon.

Referring next to FIGURES 14a and 14b, a circuit diagram of oscillator 35 and a schematic diagram of acoustic surface wave resonator 36 are illustrated. Oscillator 35 includes a bipolar transistor 231 as the amplifying element. Transistor 231 has a, grounded base 232, an emitter coupled via lead 233 to the input of resonator. 36, and a collector coupled through a capacitor 234 to the output of resonator 36. An LC circuit 236 couples a DC supply . r ~
. E ; voltage V~ to the collector of transistor 231 and also provides an output signal on lead 34.
., ....

.

Surface wave resonator 36 has a relatively high resonant frequency of 285 MHZ. Thus it is relatively small in size. Device size decreases as the resonant frequency increases.
Typically, the SWD 36 of FIGURE 14~ is only approximately 0.10 inches in length. SWD 36 also has good long term frequency stability. This is because the resonator has a large Q. Typically its Q is greater than 15,Q00. Q is the ratio of energy stored to energy dissipated per cycle within the device.
SWD ~esonator 36 is comprised of a piezoelectric sub-strate 240 which in the preferred embodiment is made of quartz.
Reflective grating structures 241 and 242 are disposed at opposite ends of substrate 240. These grating structures form e s ~ 6 s ~L~ 7~e discontinuities in the surface of~240 which reflect surface waves thereon. Gratings 241 and 242 may be comprised of grooves t or alternatively bars of gold or copp~r for an example. The bars are~spaced apart by one half wavelength of the resonant frequency.
Typically, 250 to 400 bars are contained within each of the grating structures 241 and 242; The Q of the resonator increases a~ the number of bars increases. Also, as previously pointed out, the velocity o~ the surface waves in quartz is relatively insensitive to temperature change. Thus, the resonant frequency of xesonator 36 has low temperature drift. Typical~y, the resonant fre~uency varies less than 20 KHZ over~temperature range Cl f. ,o ~ QC
.
An input transducer 243 and an output transducer ,244 are disposed on substrate 240 in the space between grating structures 241 and 242. Lead 233 couples to input transducer 243, and lead 235 couples to output transducer 244. Transducers 243 and 244 are comprised of a number of interleaved fingers which are placediat the peaks of the resonating standing wave that is set up by reflecti~e grating structures 241 and 242 "ii3l5~

Typically, 60 fingers are on each txansducer. Further details of resonator 36 are included in U.S. Patent 3,886,504 issued to Hartmann et al on May 27, 1975 and assigned to Texas Instruments Incorporated.
FI~URE 15 is a circuit diagram of amplifier 38. Ampli~
fier 38 is basically comprised of a circuit 250 which was previously illustrated in detail in FIGURE 12b. Nodes A-G
of circuit 250 as illustrated in FIGU~E 15 correspond to the node A-5 as illustrated in FIGURE 12b. Signals from mixer 32 couple to the input of circuit 250 through a capacitor ; 251. The gain of circuit 250 is automatically ad~usted by a gain control signal AGCIF. Signal A5CIF is coupled to node C through an RLC circuit 252. The output of circuit 250 is coupled to lead 40 through an LC tank circuit 253.
FI5URES 16-19 are detailed circuit diagrams of the remaining portions of the channel selector of FIGURE 1. The circuits utilize conventional components and are generally ;~
self-explanatory to those with ordinary skill in the art.
Synchronous detector 42 is illustrated in FIGURE 16. Detector 42 is essentially comprised of a commercially available chip MC1596. Chip MC1596 is~described in the Linear Integrated Circuits Catalogue on pages 8-404 to 8-414 of Motorola's Semiconductor Data Library, Volume 6, Series, 1975. Details ;
o.f the circuit are given on page 8-411 in FIGURE 23 of the :
cited reference. Chip MC1596 is appropriately biased at each of its inputs and ouputs by ~LC circuits 261-269 as illustrated in FIGUP~E 16. The biasing required by the component is also described in the cited reference.
Oscillator 45 is illustrated in FIGURE 17. Oscillator 45 has a MESFET transistor 270 as the amplifying element, and frequency determining L-C feedback networks 271 and 272. Two . 315~

separate output signals are provided by oscillator 45. One of the output signals is generated on lead 47 and the other signal i5 generated on lead 44. Leads 44 and 47 are separated by an RLC
phase shifting network 273 which generates approximately a 90 phase difference between the two signals. Phase shifting network 273 insures that the picture carrier on lead 40 is in phase with the oscillator signal on lead 44. The phase detecting circuit 43, as illustrated in FIGURE lB, generates phase detection signals PDl and PD2 which lock the oscillator signal on lead 47 90 out of phase with the picture carrier on lead 40. Circuit 273 rein-serts this 90 phase shift.
The phase detector 43 of FIGURE la also utilizes chip MC1596 as was utilized ln sync detector 42. Bias networks 281-287 are applied to chip MC1596 to achieve the phase detecting function.
Ramp generator 5~ and loop filter 49 couple to outputs of chip MC1596 as illustrated in FIGURE 18. The combination generates pHase detecting signals PDl and PD2 on leads 51.
FIGURE 19 illustrates a circuit diagram of VHF-VCO 22.
V~F-VCO 22 utilizes phase signals PDl and PD2 in combination with a third signal PD3 to genexate the selectable frequency L0 signals on lead 23.. Signal PD3 is a multilevel analog signal which provides a coarse voltage level of a unique value far each channel to be selected. Signal PD3 is generated by controller 52 in response to manually operated channel selection switches 56 as is previously described in conjunction with FIGURE 1. Signals PDl-PD3 are coupled across a varactor diode 290 through summer 50. Diode 290 in combination with. capacitors 291 and a micro strip 292 form the ~re~uency determining circuit of VHF VCO 22.
Signal amplification within VCO 22 is provided by a bipolar transi.stor 293.. Transistor 293 is selectively enabled or d~ 3~i~

disabled ~y a control signal "enable VHF" (ENVHF). Signal ENVHF
couples to the base of transistor 293 through a resistor 294 and to the collector of transistor 293 through an inductor 295. It is selectively ~ in response to the position of the channel selection switches 56.
A complete television receiver which incorporates the channel selector of FIGURE 1 is illustrated in block diagram form in FIGURE 20. The channel sel~ctor has input leads 11 and 61 for receiving VHF and UHF television signals as previously described.
The channel selector also has input leads 53 for receiving coarse analog voltages indicating the selected channel. The RF
section of the channel selector frequency shifts the selected channel to approximately 330 MHZ, while the IF section of the channel selector filters the channel at 330 MHZ and frequency shifts the filtered channel to baseband.
Lead 46 is the audio output of the channel selector.
As previously described, the sound carrier of the selected channel i's generated on lead 46- at 4.5 MHZ. Lead 45 is coupled to the input of an audio demodulator 301. Demodulator 301 generates sLgnals on a lead 302 with an amplitude that is proportional to the requency of the fre~uency modulated signals on lead 46. This frequency demodulation process may be implemented by a variety of circuits that are well known in the art. The demodulated signalc on lead 302 couple to the input of a speaker 303 where they are electromechanically conver~ëd to audible sounds.
Lead 72 is the composlte video output of the channel selector. That is, s'ignals on lead 72 include frame synchronizing information and vid,eo information of the selected channel. Lead , ~ ~ ~ .

~' 3~ ~ .

72 couples to the input of a video processing unit 304, Video processor 304 separates the picture signals from the frame synchronizing signals. The picture signals are generated on a lead 305 which couples to the electron gun input of a picture tube 306. The frame synchronizing signals are generated on a lead 307 which couples to the input of a drive circuit 308.
Drive circuit 308 generates horizontal and vertical synchronizing signals on a lead 309 which couples to electron beam deflection circuitry 310 of picture tube 306. Drive circuit 308 also generates horizontal synchronizing signals on a lead 311 which couples to an input o picture tube high voltage generator 312.
Additionally, drive circuit 308 generates synchronizing signals on lead 73 which couples to th~ AGC circuitry 74 of the channel selector.
Tele~ision receiver components 301-312 have been described in detail in many prior art publications. See for example the Fundamentals o~ Display System Design by Sol Sherr, 1970, published by Wiley-Interscience. A bibliography on pages 445-46g of the cited reference also includes many additional re~erences.
FIGURE 21 is a block diagram of a second embodiment o f a channel selector constructed according to the inyention.~ A
significant portion of this second embodiment is similar in co~struction to that of the FIGURE 1 embodiment. The similar portions are indicated by the identifying reference numeral's.
One structural difference of the second embodiment is that it has only one VHF filter. That is, signals on VH~ antenna 10 are coupled to a local oscillator mixer 320 through a fixed filter 321 which passes the entire VHF band of frequencies. A

second difference is that the embodiment of FIGURE 21 contains no RF amplifier. As a result, the system has improved inter-a~
modulation distortion and cross modulation distortion but has~

, 3~6 increased noise figure. Still another difference is that the FIGURE 21 embodiment contains only one RF MESFET mixer. A two by one switch 322 is provided having one input coupled to receive VHF signals on a lead 323 and a second input coupled to receive UHF signals on a lead 324. The output of switch 322 couples via a lead 325 to MESFET mixer 320.
The embodiment of FIGURE 21 also includes a different means for generating the coarse channel selection voltages on leads 53. As FIGURE 21 illustrates, controller 52 is comprised of a phase locked loop. The loop receives reference signals of a fixed frequency from a circuit 331, and simultaneously receives feedback signals from the LO ~CO. The LO VCO signals are sent through.a variable counter 332. Counter 332 divides by anumber which is selectable via logic signals on leads 333.
The signals on .leads 333 are generated by a logic circuit 33~
in response to logic signals received from the channel selection switches 56. A~phase detector :335 compares the output signals of variable counter 332 to the reference signals, and generates phase detection signals for the ~O VCO~ Thus, a relatively high local oscillator frequency is generated when counter 332 is 5elected to divide by a relatively large n~mber and vice versa.
Various embodiments of the invention have now been described in detail. However, many changes and modifications can be made to the above details without departing from the nature and spirit of the invention. For example, the IF
frequency of the mixer output is not restricted to 330 MHZ. Other IF frequencies in the range of 300-400 MHZ
may be employed. A~ another example, a bipolar RF amplifier may be substituted for the MES~T ,R~ amplifier. This is because the mixer introduces cross modulation distortion and inter- , modulation distortion into the rece~ver to a much smaller degree than does the linear RF amplifier. Thus, utilization of a MESFET mixer yields a receiver having greatly reduced third order distortion even though the RF amplifier is bipolar.
As another example, the surface wave device resonator utilized to generate 285 MH~ mixiny signals may be comprised of a single transducer as opposed to a dual transducer. In a single transducer resonator, lead 233 couples to one set of electrodes on the transducer while lead 235 couples to the other set of electrodes. The single transducer is configured similar to transducer 243. As still another example, the channel selector of FI~URE 1 or FIGURE 21 may be readily adapted for use in systems other than television receivers.
The channel selector has application wherever one channel of frequencies is desired to be selected from a plurality of nonoverlapping frequency channels. The information contained in the channels need not be television signals. Therefore, it will be understood that many changes and modifications can be made in the above details without departing from the nature and spirit of the invention. It ~s understood that the invention is not limited to said detail except as set forth in the appended claims.

, ~
, ,, , : , ,: ,

Claims (8)

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:

1. In a channel selector for a television receiver in-cluding a radio frequency section and an intermediate frequency section, the combination comprising:
spectrum filter means disposed in the radio frequency section for receiving radio frequency signals representative of a plurality of television channels and for filtering at least one frequency spectrum of television channels from the received signals;
at least one mixing means connected to the output of said spectrum filter means, said mixing means in-cluding a metal semiconductor field effect transis-tor and having first and second inputs coupled to simultaneously receive said frequency spectrum of television channels from said spectrum filter means and mixing signals of a selected intermediate fre-quency channel of said frequency spectrum of tele-vision channels to a high intermediate frequency substantially greater than 45 MHZ;
channel selecting filter means having an input coupled to the output of said mixing means for filtering said selected frequency channel at said high inter-mediate frequency so as to pass said selected fre-quency channel;
second mixing means disposed in the intermediate frequency section and having first and second inputs coupled to simultaneously receive the filtered selected frequency channel at said high intermediate frequency from the output of said channel selecting filter means and mixing signals of a predetermined fixed frequency for frequency shifting said filtered selected frequency channel at said high intermediate frequency to a sub-stantially lower second intermediate frequency; and amplifier means having a relatively high gain disposed in said intermediate frequency section and connected to the output of said second mixing means for providing gain to said filtered selected frequency channel at the lower second intermediate frequency, the added gain provided by said amplifier means in said intermediate frequency section being substantially in excess of the total gain provided to the signal in the radio frequency section through said channel selecting filter means.

2. In a channel selector as set forth in Claim 1, further including amplifier means having a relatively low gain disposed in said radio frequency section so as to be interposed between said spectrum filter means and said at least one mixing means, said amplifier means in said radio frequency section having an input coupled to said spectrum filter means for receiving said frequency spectrum of television channels from said spectrum filter means and an output coupled to the first input of said at least one mixing means for transmitting low gain amplified signals of said frequency spectrum of television channels to said at least one mixing means.

3. In a channel selector as set forth in Claim 2, wherein the total gain provided to the signal in the radio frequency section by said amplifier means disposed therein is less than 10dB.

4. In a channel selector as set forth in Claim 1, wherein said spectrum filter means comprises at least one fixed filter having at least one predetermined frequency passband for re-ceiving radio frequency signals representative of a plurality of television channels and for filtering at least one fixed frequency spectrum of television channels from the received signals.

5. In a channel selector as set forth in Claim 1, wherein said channel selecting filter means comprises acoustic surface wave filter means.

6. In a channel selector as set forth in Claim 1, wherein the high intermediate frequency produced from said at least one mixing means is between 300 MHZ and 400 MHZ.

7. In a channel selector as set forth in Claim 6, wherein the lower second intermediate frequency produced from said second mixing means is approximately 45 MHZ.

8. In a channel selector as set forth in Claim 1, further including additional amplifier means connected between said channel selecting filter means and the first input of said second mixing means for providing gain to said filtered selected frequency channel passed by said channel selecting filter means prior to the introduction of said filtered selected frequency channel at the high intermediate frequency to the first input of said second mixing means.