CA1058291A - Protective relay device - Google Patents

Protective relay device

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Publication number
CA1058291A
CA1058291A CA240,189A CA240189A CA1058291A CA 1058291 A CA1058291 A CA 1058291A CA 240189 A CA240189 A CA 240189A CA 1058291 A CA1058291 A CA 1058291A
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CA
Canada
Prior art keywords
output
current
voltage
winding
circuit
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired
Application number
CA240,189A
Other languages
French (fr)
Inventor
Shan C. Sun
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CBS Corp
Original Assignee
Westinghouse Electric Corp
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Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02HEMERGENCY PROTECTIVE CIRCUIT ARRANGEMENTS
    • H02H3/00Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection
    • H02H3/08Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection responsive to excess current
    • H02H3/093Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection responsive to excess current with timing means
    • H02H3/0935Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection responsive to excess current with timing means the timing being determined by numerical means
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02HEMERGENCY PROTECTIVE CIRCUIT ARRANGEMENTS
    • H02H1/00Details of emergency protective circuit arrangements
    • H02H1/06Arrangements for supplying operative power
    • H02H1/063Arrangements for supplying operative power primary power being supplied by fault current
    • H02H1/066Arrangements for supplying operative power primary power being supplied by fault current and comprising a shunt regulator

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Emergency Protection Circuit Devices (AREA)
  • Measurement Of Current Or Voltage (AREA)

Abstract

ABSTRACT OF THE DISCLOSURE
An electrical device for protecting an alter-nating current network in which a static inverse time-overcurrent relay obtains network current information and relay circuit operating power from a current transformer having two secondary windings. Low transformer burden and accurate network information signals that are linearly responsive to a wide range of network current magnitudes are achieved through the use of a switching circuit that prevents the simultaneous operation of both secondary windings The relay circuit employs an RC network with a short, pre-cision time constant for curve shaping and employs digital counting techniques to provide time delay multiplication, precision time scale selection and a variable delay time variable in accordance with the magnitude of system current.

Description

13ACK(IROUND OE Tl~ L~IVEN'['~O~I
This invention rela-tes in general to e ectrical devices employed for pro-tec-tion of alternating cu:Arent networks.
In electrical quantity sensing devices ~ the prior art, the current information and relay circuit operatingpower was derived from two current transformers. Under some conditions at least one of the transforrners was designed to saturate with-in the expected range of the curren-t magnitudes. Other prior art devices utilized a current transformer in combination with a separate power source.
In static time delay relay circuits of the prior art it is common to employ a curve shaping device for pro-viding relay characteristics similar to those of prior widely 32~:~

used electromechanical devices. Curve shaping arrangements are illustrated in U.S. Patent 3,496,417 to N. D. Tennebaum dated February 17, 1970; and U.S. Patent 3,544,846 to F. T.
Thompson dated Decernber 1, 1970.
BRIEF SUMMARY OF THE INVENTION
The invention provides a novel electrical quanti-ty responsive device for the pro-tection of an alter-nating current network. In its more specific form, the invention utilizes a transformer to alternately charge a power supply and provide a control quantity having a magni-tude proportional to the magnitude of the electrical quan-tity in the alternating curren-t network which is to be sensed.
A detecting circuit is energized by the control quantity and provides a time delayed output signal which has an inverse time characteristic with respect to the magnitude of the control quantity. The time delay circuit includes RC network means and digital counting means for shaping a desired inverse time-response curve.
BRIEF DESCRIPTION OF THE DRAWINGS
Figure 1 is a block diagram of an electrical quantity responsive device protecting a three phase alterna-ting current network and embodying the invention;
Figure 2 is a schematic diagram illustrating -the input circuit shown in block form in Figure l;
Figure 3 is a schema-tic diagram illustrating -the AC-DC conversion circuit and the time delay circuit shown in block form in Figure l;
Figure 4 is a semilog plot showing the manner in which curve- shaping of a desired inverse -time response 0 curve is achieved using a superposition technique;

5~

Figure 5 is a schematic diagram useful in describing the RC network used in the time delay circuit;
Figure 6 is a schema-tic diagram illustrating the instantaneous comparator circui-t and the trip circuit shown in block form in Figure l;
Figure 7 is a schematic diagram illustrating a modified form of the input circuit illustrated in Figure 2;
Figure 8 is a schematic diagram illustrating ano-ther modified form of the input circui-t illustrated in Figure 2; and, Figure 9 is a schema-tic diagram illustrating an input overvoltage protection circuit and power supply using a transformer having a single secondary winding.
DESCRIPTION OF THE PREFERRED EMBODIMENT
Figure 1 illustrates an electrical quantity sensing device or protective relay that is associated with an electrical network to be protected. This electrical net-work may be of any type having a condition to which the sensing device is -to respond. For present purposes, it will be assumed -that the network is a three phase alternating current network operating at a frequency of 60 Hertz and represented by the line conductors Ll, L2, and L3. Thése line conductors transmit an alternating curren-t from a suitable source -to a load through a circuit breaker 4 having a trip coil 6. The circui-t breaker 4 further comprises a plurality of separable line contacts 8, 10 and 12 which are closed when the circuit breaker 4 is closed and which are open when -the circuit breaker 4 is open. Energization of trip coil 6 while circuit breaker 4 is closed results in a tripping or opening operation o:f the circuit breaker. I-t is to be dis-tinctly understood
2~

that while the invention is illustrated in connection with a polyphase network, it is equally applicable to a single phase network. It may be used to monitor total current in either a single phase network or total curren-t in a poly-phase network dependent upon de-tai]s of -the circuit which combines the output of -the sensing windings of the current -transformers.
Ln a preferred embodimen-t, the inven-tion is shown as comprising a current relay which is responsive -to the magnitudes of the line current flowing -through the line conductors Ll, L2 and L3 and will respond to the highest current magnitude flowing in the conductors Ll, L2 and L3.
When this curren-t magnitude exceeds a predetermined value for a time interval depending on the current magnitude, the relay energizes the trip coil 6 to open the circuit breaker ~ in either a substantially instantaneous manner or af-ter a pre-determined time delay depending upon the current magnitude.
As illustrated in Figure 2, an input circuit 100 comprises a plurality of substantially similar current trans-formers CTl, CT2 and CT3 with -the prirnary windings of said transformers being individually energized by the line curren-t of conductors Ll, L2 and L3 respec-tively. Each current transformer has a first ou-tpu-t winding 10~, referred to as a power winding, and a second output winding 106, referred to as an information winding. Both output windings are in-ter-linked by the same flux and the number of winding -turns Nl in each power winding is less than the number of winding turns N2 in each information winding.
To facilitate explanation, only the opera-tion of current transformer CTl and i-ts accompanying circui-try wi:Ll be _ ~ _ 1~5~3Z~

described in detail. All three transformers and thei.r accompanying circuitry operate in a similar fashion -to each provide first and second output currents, Iol and Io2, which are proportional -to the line current energizing the respective -transformer.
The current transformer CTl has its primary winding 102 energized in accordance with the line current in line conduc-tor Ll. The power winding 104 and informa-tion winding 106 are connected respec-tively to the input terminals of the full wave bridge rectifiers RE108 and REllO having outpu-t terminals 112-114 and, 116-118 respec-tively. Terminals 112 and 116 are connected to a common bus which is shown as being grounded. Terminals 114~and 118 are connected to first and second input terminals of a control circuit 120.
Control circuit 120 comprises a voltage ac-tuated device which is illustrated as being in the form of a Zener diode Z122, a first switch device T124 which is shown as a PNP
transistor, a plurality of diodes D126, D128 and D130, a second switch device SCR136 which is shown as a thyristor, capacitors C137, C166 and C168, and a plurality of resistors R138, R140, R142, R144 and R146. The control circuit 120 provides the sequential operation of the power secondary winding 104 and the informa-tion secondary winding 106.
Hereinafter, the word thyristor shall be used to designate any switching device having a first or anode terminal, a second or cathode terminal and a third or gate terminal. Once conduction begins, the device will not cease conducting unless the current flowing into the anode drops 0 and remains below a predetermined magnitude for at least a ~5~

minimum period of time which shall be designated as theminimum shutof~ time.
Following each zero crossing of the line current in conductor Ll, balancing of -the primary ampere-turns is first achieved by the current flow IGl through the power output winding 104. During this -time, the information output winding 106 rernains open circuited and ineffective due -to the non-conducting or deenergized condition of the thyristor SCR136 which acts as an open swi-tch.
I0 The current Iol is rectified by rectifier RE108 and is used to charge a power supply 1~8. A portion of current Iol flows between the terminals 112 and 11~ through a diode D126 and an energy storage device or capacitor C150 to provide a positive (with respect to ground) voltage Vs at terminal 152. A second portion of the rectified current Iol flows between the ou-tput terminals 112 and 114 through a diode D128 and an energy storage device or capacitor C162 to provide a second power supply circuit 15~ having a substantially ripple-free regulated voltage VR between ground and an output terminal 156. The second power supplying circuit 15~ is used to supply power for low current drain use. The reference voltage power supplying circuit 15~ is connected in parallel with capacitor C150 and comprises serially connected resistor R158 and rheostat R160 connected in shun-t across the capa-citor C162. The movable arm of the rheos-tat R160 is connected to terminal 156. A filter capacitor C16~ is connected be-tween the movable arm of rheostat R160 and ground.
When the voltage between terminal 11~ and ground is below the breakover vol-tage Vzll2 of a temperature compen-sated Zener diode Z122, no substantial magnitude of current ~5~2~

Iol flows into the emitter of transistor T124 or -through resistor R138. As the rectified current Iol charges the capacitors Cl50 and C162, the voltage across the capacitors C150 and C162 increases until it approaches a magnitude equal to Vzl22 which is the regula-ted ou-tput vol-tage Vs.
Apart from preven-ting the discharge of -the capacitors C150 and C162, -the diodes D126 and D128 provide temperature compensation for the regulated voltages since the diodes Dl26 and Dl28 tend to balance the base-emitter junction temperature effect of the transistor T124.
When the voltage level across capacitors C150 and C162 substantially reach Vzl22, the Zener diode Zl22 will breakover and conduct current through resistor R138.
This reduces the potential of the base of transistor T124 and base drive current will then flow causing the transistor Tl24 to conduct through its emitter and collector, diode Dl30, and resistor Rl42 to energize the gate of the Thyristor SCR136.
In the embodiment illustrated in Figure 2, a resistor Rl~
and capacitors C166 and C168 have been connected so as to prevent thyristor SCRl36 from being energized by noise signals which may occur at the gate or by -temperature related leakage current at the anode.
The energization of its gate causes -the thyristor SCRl36 to conduct and cornple-te a pa-th for the rectified current Io2 between -the output -terminals 116-118, through the anode and cathode of thyris-tor SCR136, resis-tor R146 and ground bus. This will produce a voltage across the resistor R146 having a rnagnitude equal to (Io2) (R146).
Since both the ou-tpu-t windings 104 and 106 are interlinked by the same flux, the rela-tive magni-tudes of -the ~.~5~

voltages of the windings 104 and 106 must be the same as their turn rat:io Nl to N2. Af-ter the thyristor SCR136 conducts, -the -terminals 116 and 118 will be subs-tantially at that voltage which appears across -the resistor R146 and the vol-tage level at output terminals 112 and 114 will drop from approximately Vzl22 plus the emitter to base voltage magni-tude of transistor T124 to approximately (Io2) (R1~6) (Nl).

The values of Nl, N2 and of -the resistance of R146 and of the maximum expected value of Io2 must be selected such that the magni-tude of (Io2) (R146) (N1) for all expected magnitudes of line current in conductor Ll will never exceed ~Z122 plus -the emitter to base voltage of transistor T124. A suitable, but not critical design value for the illustrated quantity sensing device is one in which the magnitude of Io2 is limited -to the value corresponding -to a maximum line curren-t magnitude in conductors L1, L2 and L3 which is equal to forty times the magnitude of the minimum pickup current. Minimum pickup current is defined as the minimum magnitude of line curren-t which produces a pickup voltage of the minimum magni-tude which will actua-te the pickup comparator circuit 402. Limi-ting means to be more clearly set out below have been included to limit the magni-tude of Io2 at line currents which exceed forty times -the pickup value.
When the -thyristor SCR136 is energized, the magnitude of the voltage across R146 wil]. be greater -than the voltage between output terminals 112 and 11~ thus back biasing the diode D130. The rnagnitude of -the voltage between terminals 112 and 114 will drop below the voltage Vzl22 and current through the Zener diode Z122 termina-tes. The voltage of the - 8 -~s~

terminals 112 and 114 will drop below the voltage across capaci-tors C150 and C162, however current will no-t discharge because of back biased diodes D126 and D128. Thus the conduction of the thyristor SCR136 effectively open-circuits the winding 104.
The gate of thyristor SCR136 loses control of the thyristor device once rec-tified current Io2 begins to flow between its anode and cathode and the thyristor SCR136 remains energized until the line current in conduc-tor L1 experiences the nex-t zero crossing. Upon deenergization of the thyristor SCR136, the information winding 106 is open circuited. At that time, balancing of the primary ampere-turns is achieved solely by the power winding 104 and the magnitude of -the voltage at output -terminals 112 and 114 jumps to a magnitude determined by the ampere-turns balance principle of the windings 102 and 104. The power winding 104 is again rendered effective to recharge the capacitors C150 and C162 as described above.
The rectified current Io2 passing through resistor R146 when the thyristor SCR136 is energized provides a suitable information voltage regardless of whether the signal is processed in RMS, peak or average form. The voltage across resistors R170 and R172 respectively provide information voltages respectively proportional -to the line currents in conductors L2 and L3.
The power supply 148 is recharged during each initial portion of each one half cycle of the line current and thereafter provides information voltages for -the remaining portion of the half cycle. The leng-th of time during each half cycle of line current in conductor Ll that power _ g ~

~s~z9~
winding 104 and information winding 106 are each effec-tive will depend on the time necessary for the vol-tages across capacitors C150 and C162 to recharge to a voltage magnitude equal to Vzl22. If -the information vol-tages are to be processed in RMS or average form, the accuracy of said processed information voltages will vary inversely wi-th the maximum charging time of capacitors C150 and C162 and the circuit components should be chosen to recharge the power supply 148 in as shor-t a time as is expedient eonsidering the acceptable burden on the network being monitored. For peak signal detection, it is only necessary to limit the charging time to less than 90 of the sinusoidal line current. ~The magnitudes of the resis-tors R1~6, R170 and R172 should be as low as is consistent with the desired sensiti~
vity to reduce the burden. A suitable value may be 50 ohms.
The polyphase overcurrent relay of Figure 2 responds to the highest magnitude of the line current in conductors Ll, L2 and L3. An auctioneering circuit 17~ provides a fi,rst auctioneered voltage signal which is responsive only to -the highest magnitude of the three voltages developed across resistors R146, R170 and R172 and comprises diodes D175, D176 and D177 which are respectively conneeted between -the undergrounded ends of the resistors R1~6, R170 and R172 and a common output conductor 178 which is connec-ted to -the averaging circui-t 200. The vol-tage aeross the one of the resistors P.1~6, R170 or R172 whieh is greatest determines the magnitude of the voltage of the eonduc-tor 178. Two of the three diodes D175, D176 and D177 will be reverse biased or blocked.
The auctioneering circui-t 17~ provides a second :
:
:
~s~z~
auctioneered voltage which is similar to -the first auc-tioneered vol-tage and comprises diodes D181, D182 and .
D183 which respectively connect the ungrounded ends of , -the resistors R145, Rl'70 and R172 -to a common output conduc-tor 184 which is connected to a limi-ting circuit 186 , whereby a second auc-tioneered voltage signal is s~plied -to the aforementioned limiting means.
, The limiting circuit 186 connects the conductor 184 to ground in parallel wi-th resistors R146, R170 and R172. A vol-tage regulating device Z188 shown as a Zener diode breaks over when the second auc-tioneered voltage has a magnitude in excess of forty times pickup current. When the device Z188 breaks over a Thyristor SCRlgO, having its ea-thode connee-ted to the eommon bus at ground potential, is rendered eonductive or energized. The shun-t circuit 186 will thus render the whole input circuit 100 ineffective by shunting , to ground subs-tantially all of current normally flowing through resis-tors R146, R170 and R172. Resistor R192 is provided to limi-t the current through the gate of Thyris-tor SCRl90 when the Zener diode Z188 is in its conduc-ting s-tate. Resis-tor R194 and capacitor C196 have been connected so as -to prevent -thyristor SCRl90 from being energized by noise signals,at the gate.
A second form of limi-ting means is aceomplished by designing the current transformers CTl, CT2 and CT3 -to saturate a-t a predetermined level of line current such -that the maximum magnitude of -the voltage between terminals 112 and 114 is limited to a value that is less than the voltage mag-nitude Vzl22 plus the emitter -to base voltage of -the transistor T124 when one or more of the windings 106 are energizing f~

their respec-tive loading resis-tors Rl45, R170 and R172.
As shown in Figure 3, the conductor 178 supplies the ~irst auctioneered voltage to a peak voltage averaging circuit 200 which may take any form bu-t which is shown comprising a ~ilter network including a ~ilter capacitor C202 and resistor R204 connected in parallel. The circuit 200 energizes the output conductor 202 with a DC voltage whose magnitude is proportional to the averages of the peak values o~ the ~irst auctioneered voltage. The conductor 202 of the averaging circuit 200 is connected to a -time de-layed detec-ting circuit 400 and to an instantaneous trip detecting circuit 500.
TIME DELAYED CIRCUIT
The time delayed circuit 400, more completely shown in Figure 3, comprises a pickup circui-t 402, a digital counting circuit 406, a resetting RC delay network 410, an inter~ace 412 and a monostable mul-tivibrator circuit 414.
At a predetermined time interval, TD, after -the magnitude o~ the DC voltage ~rom circuit 200 exceeds i-ts minimurn predetermined pickup value -the counting circui-t will supply a control signal to energize the trip circuit 600 either through the time delayed compara-tor circuit 400 or through the instantaneous detecting circuit 500 depending upon the magnitude o~ the pickup current supplying the pickup voltage signal. The length o~ the time delay TD provided by circuit 400 varies inversely wi-th -the magnitude O:r -the DC pickup voltage, except -that a ~ixed minimum time delay is provided by the time required ~or the multivibrator 414 to change its state. As will be more ~ully set out below, 0 the time delay circuit is capable o~ providing an ~058f~1 output voltage according to any desired inverse timeresponse curve depending upon the values of R and C used in the delay network 410 and the setting of the counting circui-t 406.
The pickup circuit 402 inhibits the operation of and resets the time delay circuit 400 whenever the magnitude of the pickup DC voltage from the averaging circuit 200 is less than the predetermined pickup value and will initiate the operation of the circuit 400 when -the pickup voltage exceeds the minimum magnitude. The pickup circuit 402 comprises a comparator device 408 which has its negative input terminal connected -to and energized by said conversion circuit 200 and which has its positive terminal connected to terminal 156 of the reference voltage circui-t 154. The voltage VR determines the minimum pickup voltage and may be adjusted by -the movable arm of rheostat R]60. When the voltage output of the circuit 200 is less than the value Vp, the comparator 408 does not draw current from the supply terminal 152 and -the output conduc-tor 409 is maintained at substantially the voltage Vs. When -the pickup voltage exceeds the value Vp, the compara-tor 408 effectively connects the conductor 409 to ground and the counters 444, 446 and 447 are conditioned to coun-t the pulses from the multi-vibrator 414. A suitable pickup comparator 408 is the first unit of a monolithic quad comparator in-tegrated circuit MC3302P
which is commercially available from Motorola, Inc. Device MC3302P comprises four identical comparator units arranged in a 14 lead dual in-line plastic package. The second, third fourth units of device MC3302P are conveniently u-tilized 30 elsewhere in the circuits 400 and 500.

~635~32~

Shaping of -the inverse time response curve is first provided by a passive ne-twork 410 having resistive and capacitive elements, the inter:~ace 412 and a monos-table multivibrator or one-shot device 414. The capacitive elements are deenergized each time that ou-tput voltage a-t terminal 436 exceeds the voltage VR. The transistors T414, T416, T418 and T420 are shunt connected across these capacitive elements and have their bases connected to -the output of the multivibrator 41~. Each time the multivibrator 414 is pulsed, the transistors will conduct and discharge the capacitive elements, thus permi-tting the network 410 to achieve cyclic operation to pulse the counters 444, 446 and 447.
There are two timing characteristics provided by the pulse generating circuit 404, the time delay, TDl, pro-vided by the RC network 410 which is a function of the line current magnitude and the fixed time delay, TD2, of the one-shot device 414.
The resetting delay network 410 comprises a plurali-ty of parallel branches. Each branch includes capacitors such -tha-t the charge on the capacitor of each branch is charged along a different time rate which if plotted on semilog paper results in the time curren-t relationships 426, 428, 430 and 432 shown in Figure 4. The curve 424 shows -the inverse time current relationship obtained by -the network 410 when suddenly energized by the DC voltage signal which is pro-portional to the highest peak magnitude of -the line currents flowing through conduc-tors Ll, L2, and L3. :[t will be appreci-ated tha-t curve 424 is -the sum of the time relationships 426-432. Thus the tirne required for the equivalen-t voltage 25~

response magnitude of the RC network 410 to rise from zero to a predetermined magnitude will be inversely related to the highest peak magnitude of said line currents.
Although four branches are shown in Figure 3, any number of branches may be used. Each branch of the illus-trated RC network 410 comprises two resis-tors serially connected between an input terminal 434 and an output terminal 436, and a capacitor connected from a point between the two resistors to ground potential.
Figure 5 has been included for an analysis of the RC network 410. It will be assumed that all of the capaci-tive elements are initially deenergized and that the voltage VIN between input terminal 434 and ground suddenly increases to or above the minimum pickup magnitude. The capacitors of the branches will charge at their individual rates. During this time the voltage VIN will be determined by the magni-tude of the fault current (which will be assumed to be of fixed magnitude during a given fault) causing a charging current IRCA to flow to the capacitors. A voltage VR will be main-tained at the terminal 436 of the conversion circuit 200.
It will be apprecia-ted -that -the amplifier 413 will attemp-t to maintain the po-tential VR at its input terminal 439. A
second component of charging curren-t IRCB flows to the net-work 410 from the output terminal 438 -through -the Zener diode Z440 and diode D442. When the capacitors O:r the network 410 reach their critical charge the current component IRCB will be zero. However since the voltage at the -terminal 434 will of necessity be greater, current will continue to flow to charge the capacitors and the vol-tage at terminal 436 will rise sufficiently to cause the opera-tional amplifier 413 to ~ 15 -reduce the voltage at its output terminal 438 in an endeavor to maintain the voltage magnitude VR at its negative input terminal and consequentl~ the voltage magnitude at terminal 436.
Identifying the voltage at terminal 434 as VIN~ the voltage at the terminal 436 as VR~ the voltage at the ungrounded terminal of the capac~tor as Vc~ the resistance of the resistor between the capacitor and terminal 434 as Rl, and the resi~tance o~ the resistor between the capaci~or and the terminal 436 as R2, the mag-nitude o~ the current flow to the capacitor ~rom the terminal 436 as IRCB~ and using the direction of current flow to the capacitor from the terminals 434 and 436, the follow-ing mathematical formula for IRCB in terms of time t may be derived either by calculus directly or with the use of LaPlace transform~.

~ R ~ _ 1 2 ~ ~ (~1+R2) RCB ~ ~ _ Ll~

By adding together the IRcBg ~or each or the RC
branche~ and equating the sum to zero, the time t at which the network 410 ~11 actuate the operational ampllfier 413 of the interface 412 may be d~termlnedO
m e purpose Or the Zener Dlode Z440 is to raise the voltage at the output terminal 438 of the operational amplifier 413 sufficiently above the voltage V~ to prevent the operation of the monostable vibrator 414 durlng the charglng period of the network 410.
When the amplif'ier 413 re~ponds to network 410 ~513Z91 the potential of the termlnal 4~8 thereof will almost instantaneously decrease in magnltude to provide an operating signal to actuate the monostable multivibrator 41~ which9 at the expiration of its fixed time interval, will supply a positiYe pulse to the counter 444 and to the transistor T422, When transiator T42~ conducts it cau3es the transistors T414, T4169 T~18 and T420 to conduct and discharge the capacitors of the network 410. When the capacitors discharge, the potential o~ the negative input terminal 4~9 f'alls to and is maintained at the voltage magnitude VR, At the end of ~he time delay interval TD2 of the multlvibrator 414 it will again revert to its low output condition causing the transistors T414-T422 to cease conducting and the network 410 will repeat its timing operation, The monostable multivibra$or 414 also provldes it~ positlve golng pulse to the counting circuit 406, which in turn provides an output voltage signal upon receipt of a predetermined number o~ one-æhot pulæe signals, me NE/SE 555 monolithic timlng circuit provides a sultable monostable multivibrator 414 and is commercially available from the Signetics Corporation, AB illustrated in Flgure ~, the counting circuit 406 comprises counters 444, 446 and 447, and comparator means 448 for cascading the three counter circuits in proper sequence, A large number of counts obtained with a network 410 having a relatively low RC time constant is desired for greater accuracy, A suitable RC time con~tant range for the network 410 is ,5M.S, to 5M,S. and a suitable counting range for the counter 444 is ~rom 500 to 4~,000, This ~s~z~

relationship deter~ines the time scale reference. A
binary counter 444 is particularly desirable and a suitable counter ls the RCA binary counter/di~ider CD4040AE which advances on the negative transition of the one-shot pulse and is commercially available rrom ~he RCA Corporation. The second comparator unit 448 o~ the previously descr~bed comparator device MC~302P (a first unit of which was used for the comparator 408) has its negative lnput terminal connected throu~h a selector s~itch 452 and its output connected to the count input terminal oP the decade counter 446. The binary counter 444 is arranged to provide an output pulse for each desired number of operations o~ the multivi-bratorO me desired number is different for each of the terminals o~ the switch 452. The counter 444 is reset to its initial or starting condition by the positive signal received from the comparator 408 and is released to count when the output signal o~ the comparator 408 decreases.
The counters 41~ and 447 may take the form o~
decade counters whereby the number Or output pulse~ ~rom the binary counter 444 reauired ~o actuate the tr~p circuit may be selected. These counters 446 and 447 collectlvely select the one o~ a fa~lly o~ ~ime curves w~ich con~rol~ the actuation or the trip circult 600.
me ten position selector switches 450 and 451 are respectively connected to said decade counters 446 and 447 such that one hundred di~ferent time curves based on the time scale re~erence curve may be obtained for precision ~election, ~en the counters 446 and 447 each provlde a positive output pulse at their respective selector switches 450 and 451, both ot~ the diodes D462 and D464 will be back biased 5~3Z~

to actuate the comparator 454, which may be the third unit o~
device MC~02P having the comparators 408 and 448. The decade counters may be o~ the type supplied to the trade by RCA Corporatlon as CD4017AE.
me tlme-current relationships 426-4~2 may be chosen to provide a trip curve 424 having an almost unlimited number of shapes to duplicate any existing inverse tlme current curve o~ the more conventional electro~mechanical over current rela~s such as the CO relays presently manufact-ured and sold by Westinghouse Electric Corpora~ion. mevalues o~ the required resistors and capacitors may be obtained by determining a number o~` time-current relationships on the curve to be duplicated and simultaneously solving the time~
current relationships ~or each o~ the RC branches of the network 410 in accordance with the IRCB formula set forth above.
INSTANTANEOUS COMPARATOR CIRCUIT
The instantaneous detecting circuit 500 provides an output voltage signal for tripping the brea~er 4 immediately after the magnitude o~ the current in any o~
the lines Ll, L2 or L3 and consequently the DC voltage from circuit 200 exceeds a predstermined value.
AB shown in Figure 6, ~he instantaneous comparator circult 500 consists of voltage divlding resistors R502, R504, a rheostat R506, a filter capacitor C508, a resistor R510, a comparator 512 (which may be the fourth unit of device MC~302P), and an overvoltage protection Zener dlode Z514, ~hen the level o~ the voltage across capacitor C508 reaches a magnitude e~ual to that o~ VR, the output of comparator 512 ~0 will provide a negative going signal to said trip circuit 600.

~q3S8~

me movab~e arm o~ rheostat R506 adjusts the magnitude of the DC voltage from circuit 200 which is required to actuate the instantaneous compara~or circuit 5000 TRIP CIRCUIT
A suitable trip circuit 600 for energizing the breaker trip coil in response to the ~egative going out-put signals from either the time delay circuit 400 or the lnstantaneous trip comparator circuit 500 is illustrated in Figure 60 Trip circuit 600 comprises an indicating circuit 602, a brea~er actuating circuit 604, and an indicator reset circuit 606, ~ voltage drop at the output of the instantaneous detecting circuit 500 causes base drive current to flow through a PNP transistor T608, a Zener diode Z614, a resistor R616 and the comparator 512. The ~low o~ th~s base current causes the transistor T608 to raise the potential of ~unction 618 to substantially that o~ Vs+~ Current then ~lows through three separate conductors 620, 622 and 624.
me currenk in conductor 620 flows through a diode D626, resistor R628 and the comparator 512 to ground, This ~eedback loop keeps the comparator 512 energized until the line current in conductors Ll, L2 and L3 is interrupted by the breaker 4.
The current ln conductor 622 flows through diode D630 and the breaker activating circuit 604 where it ~lows through a reslstor R6~2, a diode D6~4 and into the gate of a thyristor SCR6~6, The energized ~hyristor SCR636 when conducting completes a path ~or current flow between the positi~e termlnal 6~8 of a station battery 640 through diode D642, the thyristor SCR636, diode D644, the 52a normally ~S~29: L

closed contacts of the breaker 4, the trip coll 6 and the negative terminal 646 of battery 6400 me trip coil 6 when energized trips breaker 40 Capacitors C648 and C650 and resistor R652 are ~rovided to prevent thyristor SCR636 from energization by noise or l:ealcage currents~ Th 52a contacts are located on the brea~er ~ and are open circuited when the breaker 4 trips, thus inter~upting current through the coil 6 and the thyristor SCR636.
me breaker actuating c~rcuit 604 further comprises a resistor R654 having a negative voltage-resistance characteristic to absorb transient energy signals at terminal 638 of battery 640, and also a discharge circuit 656 con~prising resistor R658 and diode D660 ~or discharging the indicutive energy stored in trip coil 6 should 1~ not have dissipated when the breaker 4 is reclosed, me indicating circuit 602 will lndicate only a~ter the thyristor SCR636 has been energized. Indication o~ an instantaneous trip is provided by the ~low o~ current in conductor 6ZLI through diode ~662, and electrically resettable indicating device 664, a two position switch 666 haYing an indicating de~ice re~et position and a Ret pos~tionJ
diodes D668, coil 6, battery 640, diode D642, thyristor SCR636, and(the common bus 667 connected ko ground, The indicating device 664 is shown as a li~;ht re~3~ecting electromagnetic status indicator having an inherent memory.whereby it will maintain its lndicating status even though its energizing current is intensi~ied by the opening of the contacts 52a, me device 664, when once actuated, will continue to indicate until the indicator device is reset, A suitable indicator devlce 664 ~s~g~

is commercially available ~rom Ferranti~Packard Ltd.
me operation of trip circuit 600 in response to the negative going signal from the time delay circuit 400 is ~ubstantially similar. A voltage drop at the output of comparator 454 causes base drive current to flow through an PNP transistor T670, a resistor R678 and Zener diode Z676~ me base current causes the transistor to conduct current through transistor T670 to raise the potential o~ ~unction 680 to substantiall~ that o~
Vs~. Currentthen flows through conductors 682 and 684.
me current of conductor 682 ~lows through diode D686, resistor R632~ diode 634 and causes the thyristor SCR636 to conduct and energlze the trip coil 6 in the manner previously described, The current ln conductor 684 ~lows between Vs~ and ground bus 667 through diode D688, an electrically resettable indicating device 689, the selector switch 666, the diode ~668, contacts 52a, coil 6, bat~ery 640, dio~e 642~ and thyristor SCR636. Indicating device 689 is substantially similar to device 664 and when once actuated will remain actuated until reset,, The Zener d-lodes Z61~ and Z676 have been included to prevent an undesired tripping of breaker 4 during the initial period of relay operation when the voltage at termlnal 152 of the power suppl~ circuit 148 and ground has not yet reached its normal operating value Vs~, me breakover voltage magnltude~ of Zener diodes Z614 and Z676 should be high enou~h to prevent current flow duri ng this period.
A conductor 690 i~ connected to the conductor 68 ~58'~

such that the llmitlng circuit l~6 and the breaker actuating thyristor SCR636 are concurrently energiæed.
me limiting circuit 186 preven~s high voltage failure Or circul.t components during the period be~ore the line current is interrupted by khe opening o~ breaker contacts 8, lO and 12. Current flows be~ween terminal 680 and ground through diode D686, conductor 690, a resistor R691, a diode D692 and the gate of the thyristor SCRl90~
When thyristor SCRl90 conducts, it connects the conductor 184 to groundJ ~hus shunting the load resistors Rl46, Rl70, Rl72.
h reset c~rcuit 606 for resetting -the devices 664 and 689 comprises a resistor R693 and a capaci-tor C694 which is energized from battery 640, When switch 666 is in its reset position~ the capacitor C594 discharges through the indicating devices 664 and 689, r~sistors R695 and R696, a Zener diode Z698 and the common bus connected to ground potential. The Zener diode Z698 has a breakover voltage level high enough to prevent conduction through to the common bus except during reset operation of the trip circuit 600.
It is conceivable that the DC voltage signal from circuit 200 will continue to rlse a~ter it activates the time delay circuit 400 until it reacheQ a magnitude sufficient to also activate the instantaneous co~parator circuit 500. To prevent this from occurring, a diode D699 is connected between the negative input -terminal of the comparator 5l2 and the common connection o~ the resistor T678 and Zener diode Z676. When the time delayed ~3SI~Z~l detecting circuit comparator l~54 provides its negative going signal, the cathode of the diode D699 goes to a potential which is sufficiently low enough to prevent the operation of the comparer 512 of the instan-taneous detecting circuit 500. Consequently, the transistor T608 is cut off and the energization of the instantaneous trip indicator device 664 is prevented. Thus the indi-cating circuit 602 is capable of indicating which of the detecting circuits 400 or 500 initially actuated the tripping circuit 600.
Figure 7 illustrates an input circuit 700 which is a modified form of the input circuit 100 of Figure 2.
To simplify the explanation, only the apparatus associated with phase transformer CTl is shownS as the phase transformers CT2 and CT3 and their accompanying circuitry would be similar. me thyristor SCR136 and its accompanying circuitry, and two diodes of the rectifier bridge REllO have been replaced with -thyristors SCR702 and SCR70L~ and accompanying circuitr~ comprising capacitors C706, C708 and C710 and resistor R712.
The modified imput circuit 700 assures that the information winding 106 of transformer CTl will be rendered ineffective after each zero crossing of the line current in conductor Ll. As previously explained, the thyristor SCR136 shown in Figure 2 is designed to turn off when the rectified current at its anode drops to and remains below a predetermined magnitude for at least a minimum shutoff time. If the first derivative of the anode current magni-tude with respect to time is such that the current does not remain below said predeter-mined level for the minimum shutoff time, the thyristor SCR136 will never be deenergizedO If, however, a pair _ ,~y_ J~

~6~S8~1 of' thyristors S~R702 and SCR704 are substituted for the pair o~ the rectif'iers in the brid~e circuit RE110, as illustrated in the bridge circuit RF710, each receives an alternating potent~al at its anode. When energized the thyristor SCR702 provides a path ~or current between the inf'ormation winding 106 and the common bus connected to ground potential through the thyristor SCR702 and the resistor R146 when the upper terminal of the bridge RE710 is posltive. Similarly, when energized the thyristor SCR704 provides a path f'or current between the in~ormation winding 106 and the ground bus through the thyristor SCR704 and the resistor R146, The operation o~' the input circuit 700 is ln other respects similar to the input circuit 100 of Figure 1 and its operation will be apparent ~rom the description thereof, Figure 8 illustrates an input circuit 800, which is a modified ~orm o~ the input circuit 100. me input circuit 800 comprises a dual regulated power supply circuit f'or providing positive and negative regulated supply voltages to a detecting circuit 804 requiring said voltages. The input circuit 800 ~urther provides an AC in~ormation signal which is unrecti~ied and responsive to the line current through conductors Ll, L2 and L3. For simpli~ication and explanation only one phase oP the three phase design has been shown and will be described.
me current trans~ormer CT4 has a power secondary winding 806, an information ~econdary win~ing 808 and a primary winding 810. me primary winding 810 is energized in accordance with the line current ln conductor Ll. me ~0 number of' winding turns ~2 f the information winding 808 ~ s~z~
is greater than the number of winding turns Nl of the power windîng 806. A center tap 812 on the power winding 806 is connected to the common bus.
For simplification, it is assumed -that the line current in conductor Ll has just experienced a zero crossing, the power winding 806 is effective and the information winding 808 is non-conducting~ With line current flowing in conductor Ll in a direction to make the upper terminal of the winding 806 positive with respect to the lower terminal (which will be hereinaf-ter referred to as the "positive half cycle"), current to charge the capacitor C816 will flow between the grounded terminal of capacitor C816 and the grounded center tap 812 oP the winding 806 through the capacitor C816, diodes r818 and D820, and the lower half of the power winding 806. Current for charging the capacitor C826 will flow between the grounded terminal of winding 806 and capacitor C826 through the upper half of the winding 806, the diodes D822 and D824, and the capacitor C826. During the opposite or "negative" half cycle of the line current in conductor L1 current will charge the capacitors C816 and C824 from the power winding 806 through the diodes D828 and D830 in place of the diodes D820 and D8220 Stated other-wise, during the "positive'l half cycle the capacitor C816 is charged from the lower hal~ of the winding 806 and the ~apacitor C826 is charged from the upper half of the winding 8060 During the "negative" half cycle the capacitor-winding relationship is reversedO The current flowing in the two halves of the winding 806 may or may not be equal since it is merely necessary that the total ampere-turns -~6-~s~

in winding 806 e~ual the total am~ere-turns ln winding 810, and the division of the current between the two halves is dictated solely by the charged condition o~ the capacitors C816 and C826.
A control circuit 814 is provided ~or regulating said supply voltages and for controlling the sequential operatlon of secondary windings 806 and 808. The control circuit 814 co~prises a voltage regula~ing device Z832, such as a Zener diode, a switch device T8~4 such as a NPN
transistor, a capacitor C836, resistors R838~ R840, and R842, and a triac device 844. me term triac as used herein designates any switching device having a ~irst or anode terminal, a second or cathode terminal, a third or gate terminal and the fo~lowing properties. Current flowing from the gate permits the device to conduct current in either direction between the anode and cathode terminals.
Once conduction beginsg the device will not deenergize unless the curren~ ~lo~Jlng between the anode and cathode drops and remains below a predetermined magnitu~e for at least a minimum period o~ time.
As the capacitors C816 and C&6 charge, as above describe~, they will e~entually both reach a predetermined magnitude e~ual to the breakover voltage magnitude of the Zener diode Z8~2. T~en this occurs current ~ill flow through the Zener diode Z832, resistor R840 and the base of the transistor T834, me transistor T834 will conduct and current wlll flow through the gate of the triac device 844, the resistor R842, the transistor T83~, diode D820~ or D822~ and one half of the power winding 806. The conduction of translstor T834 can occur in either the "positive" or ~s~

"negative" half c~cle o~ the trans~ormer, The diodes D8l8 and D824 provlde temperature compensation of the transistor T834 and thus ~urther regulate the voltage across capacitors C816 and C826.
Diode D8l8 provides isolation ~or the capacitor C816 f`rom leakage current of the transistor T834 or Zener diode Z8~2.
me energized triac device 844 when donducting renders the information winding 808 eff`ective by providing a path for current through the in~crmation winding 808, the triac 844 and a loading resistor R8~6. This in turn induces a drop in the voltage across the powar winding 806 and .~fectively terminates the flow o~ current therethrough in the manner set ~orth above in connection with the description of Figure 2. The ma~or difference between the ~orm o~
Figure 2 and Figure 8 is that the voltage acros~ the loading resistor R846 provides an electrical quantlty responsive to the line current in ~.onductor Ll which is alternating in polarity, The triac device 844 deenergizes each hal~ cycle of the line current similarly to the thyristor SCRl36 of F~gure 2 thus rendering the information winding ~o8 inef~ective and open clrcuited and rendering the power winding 806 ef'~ectlve to recharge the capacitors C816 and C826 each half' cycle, SATURABLE CURRE~T TRANSFORMER OPERATION
Flgure 9 illustrates a form o~' the invention in which an input circuit 900 utllizes a current transformer CT5 having a single secondary winding. As illustrated~
the ~rinding-core relationship o~ the transf'ormer CT5 i6 such that the core will saturate ~ 5~Z 9 ~

within the expected range o~ network curren~ magnitudes.
Again~ for purposes o~ simplify1ng the explanation the invention is shown as energized by onl~ one phase of the input circuit, me saturable current ~ansformer CT5 has a primary winding 902 energized by a quantit~ proportional to the curren~ in conductor Ll, and a secondary winding 904, The secondary winding energizes series connected resistors R9o8 and R909 through a full wave rectifier RE906. A power supply 148 is connected through a regulator circuit 910 and an input overvoltage protection circuit 912 to one output of khe rectifier RE906 in shunt with the resistors X908 and R909. Diodes D914 and D915 prevent any po~er flow back ~rom the power supply circuit 148. me trans~ormer CT5 is designed to saturate at a current corresponding to twlce the magnitude of network current (2PU), or any desired level~
For simplicity of explanation, it will be assumed that the line current in conductor Ll has ~ust experienced a zero crossing and that ~he energy storage devices C150 and C164 o~ the power supply 148 are initially deenergizea.
me energization o~ the primary winding 902 causes currenk to flow ln the secondary winding 904 which is rectified b~
the full wave rectifier RE9o6, The output terminals 924 and 926 Or the rectifler RE906 are connecked between the ground bus and a conductor 928.
Initially (with capacitors C150 and C164 discharged) the impedance of khe power supply 148 will be less than the kokal impedance of the resistors R9o8 and R909 and substantially all o~ the rectified secondary current 1~5~2~311 flows from terminal 926 through the conductor 928 to the power supply 148~ At magnltudes o~ line current below 2PU
current will flow for one or more cycles to charge the power suppl~ 148~ The number o~ cycles ~Jhich are requlred will depend on the magnitude of the line current and the capacity of the capacitors C150 and C164. Eventually the voltage across the capacitors (and across the series connected resistors R9o8 and R909) will reach the desired ~alue to provide the Vs~ and VR outputs, ~en this occurs the regulator circuit 910 will be actuated to cause current to flow between conductor 928 and groun~ bus through the register R940 and Zsner diode Z9~6~
The regulator circult 910 includes NPN transistors T9~0, T932, and T934, the voltage regulating Zener diode Z936 and resistors R938, R940 and R942. Initially, the transistor T930 conducts to supply base current to transistor T9~2 whereby it maintains a low impedance connection between conductor 928 and the power suppl~ 148 for the flow of charging current above discussed, Staging the transistors as shown permits a large flow o~ current through transistor T932 in response to a small magnitude of current flowing into the base of transistor T9300 ~en the magnitude of the voltage of the power supply 148 reaches lts desired value the potential of the conductor 928 w111 be high enough for Zener diode Z936 to breako~er and conduct, current through resistor R940, This results ln the flow o* base current in transistor T934 causing it to conduct, l~en the transistor T934 conducts and lowers the potential of the base of the transistor Tg3o~ this reduces the conduction of the transistor T9~2~ and an equilibrium - ~0 -~L~51~2~1 conditi.on is reached such that -the current through the transistor T932 drops to a magnitude requ.ired to maintain the desired voltage magnitude of the power supply 148 as determined by the Zener diode Z936.
I~ the drain on the power supply lL~ is low in comparison with the current supplied by the transformer CT5, the major current flow is through the burden resistors R908 and R909. The resistors R908 and R909 provide a voltage dividing circuit and the relative magnitudes thereo~ depend upon the relative magnitudes o~ the voltage of the power supply 148 and the desired output voltage range of the peak voltage averaging circuit 200. The voltage across the resistor R909 provides the information signal to the auctioneering circuit 174 for providing an in~ormation signal to the peak voltage averaging circuit 200 that is responsive to the magnitude of the highest line current in any one o~ the conductors Ll, L2 and L3.
The Zener diode Z936 is a temperature compensated device and regulates the voltage of the power supply 148.
This occurs because the diodes D914 and D915 tend to cancel the temperature ef~ect o~ the transistor T938 base-emitter junction~
The input overvoltage protection circuit 912 prevents the breakdown of circuit components due to excessive voltages caused by high ~ault currents. The protection circuit 912 comprises a Zener diode Z946, NPN transistors T948 and T950, and resistors R952, R954 and R956. The transis tors T948 and T950 connect the conductor 928 to the ground bus when the Zener diode Z946 conducts. A suitable magnitude ~or the
-3~-~ [35~Z~

Zener breakover voltage VZ946 may be 200 volts.
The use of saturating current transformers which are saturated substantially throughout their normal operating range reduces the burden on the current transformer while at the same time enabling an inverse time current response at current input ranges well above the saturating current magnitude. In this regard it should be remembered that as the magnitude of the input current to the transformer increases the saturating current value is reached in a shorter time per~oa~ Since the magnitude of the current pulse in the secondary wlnding is dependent upon the rate of cha~ge of the transformer core flux (input current), an increasing peak value of the voltage across the loading resistor will occur as the input magnitude current increases. In the range below saturation this will be a straight line relationship and, of course, secondary current will floow the primary current throughout each hal~-cyc~e~ As the magnitude of the input current increases beyond the core saturation value the current will flow at an increasing rate to follow the -lncreasing rate o~ the input current but will only flow for a portion o~
each half cycle and of course the burden on the primary line conductor is greatly if not completely reduced during the time periods ~Ihen no current is flo~Jing in the secondary winding~
The operation of the current trans~ormers in the saturating range provides a constant RMS current o~tput and therefore such an arrangement utilizes the peak value magnitudes to sense the input current magnitudes in the saturating range, ~ h~le the use of ~aturating c~rrent trans~ormers 45,373 -~3S~g~L

has been discussed in connection with the apparatus of Flgure 9, it may also be used with the double secondary winding construction in which the windings are used in sequence each hal~ c~cle. When used with the two secondary winding construction, the transformer should saturate only durlng the time periods that the magnitude sensing winding is being utili&ed.

~33-

Claims (38)

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:
1. An electrical quantity responding device comprising:
a current transformer, said current transformer including a primary winding adapted to be energized by an alternating current, and first and second output windings, power supply means responsive to said first output winding, current sensing means responsive to said second output winding, and control means controlling said first and second output windings to achieve substantially complete balancing of the ampere turns of the primary winding in a selected one of said first and second output windings, and then in the other, within the same half cycle of the alternating current.
2. An electrical quantity responding device comprising:
a current transformer, said current transformer including an input winding adapted to be energized by an alternating current, and first and second output windings providing first and second electrical quantities, respectively, rectifying means rectifying said first electrical quantity, energy storage means adapted to be energized by said rectified first electrical quantity, a detecting circuit for providing an output electrical quantity when the magnitude of said second electrical quantity exceeds a predetermined value, and control means controlling said first and second output windings to provide their first and second electrical quantities during different portions of the same half cycle of the alternating current, such that the energy storage means provides no load on the current transformer when the second output winding is providing its second electrical quantity to said detecting circuit.
3. An electrical quantity responding device comprising:
a current transformer said current transformer including an input wind-ing adapted to be energized by an alternating current, and first and second output windings providing first and second electrical quantities, respectively, said first and second output windings being interlinked by the same flux path, said first output winding having N1 turns and when effect-ive providing said first electrical quantity, said second output winding having N2 turns which exceeds N1 turns, and when effective providing said second electrical quantity, rectifying means rectifying said first electrical quantity, energy storage means adapted to be energized by said rectified first electrical quantity, a detecting circuit for providing an output electri-cal quantity when the magnitude of said second electrical quantity exceeds a predetermined value, and control means controlling said first and second output windings to provide their first and second electrical quantities during different portions of the same half cycle of the alternating current, such that the energy storage means providing no load on the current transformer when the second output winding is providing its second electrical quantity to said detecting circuit.
4. The electrical quantity responding device of claim 2 wherein the control means comprises first and second switching means switchable between conductive and non-conductive states, said second switching means being connected to the second output winding, said second switching means changing to its non-conductive state at the end of each half cycle of the alternating current, to render said second output winding ineffective and said first output winding effective at the start of each half cycle of the alternating current, said first switching means being connected to the first out-put winding and to the second switching means, said first switching means changing from its non-conductive state to its conductive state during each half cycle of the alterna-ting current which causes said second switching means to switch to its conductive state, said second switching means, upon switching to its conductive state, rendering the second output winding effective and the first input winding ineffective.
5. The electrical quantity responding device of claim 4 including impedance means connected to be responsive to said second electrical quantity when the second switching means is conductive, and semiconductive means connected to limit the maximum voltage magnitude across said energy storage means to substantially a predetermined value and to operate the first switching means when said predetermined maximum voltage magnitude across said storage device is reached, to cause the first switching means to switch to its con-ductive state.
6. The electrical quantity responding device of claim 5 wherein the impedance means is a resistor connected such that the second output electrical quantity provides a voltage across said resistor when said second output winding means is rendered effective.
7. The electrical quantity responding device of claim 6 wherein the voltage applied to the energy storage means in response to the first electrical quantity drops below the predetermined value of the energy storage means when the second output winding becomes effective, and including unidirectional means connected between the first output winding and the energy storage means which effectively disconnects the energy storage means from the first output winding to render it ineffective, when the voltage of the energy storage means exceeds the voltage applied thereto in response to the first electrical quantity.
8. The electrical quantity responding device of claim 6 wherein said resistor has a magnitude such that for all expected magnitudes of the alternating current in the input winding of the transformer, the ratio of the turns in the first output winding to the turns in the second output winding multiplied by the magnitude of the voltage across said resistor provides a magnitude less than the magnitude of the predetermined maximum voltage across the energy storage means.
9. The electrical quantity responding device of claim 5 wherein said switch device is a thyristor.
10. An electrical quantity responding device, comprising:
a current transformer, said current transformer having an input winding adapted for energization by an alternating current, and first and second output windings providing first and second electrical quantities, respectively, energy storage means, said energy storage means including first and second capacitors, diode means inter-connecting said first and second capacitors with said first output winding such that said first capacitor provides a negative regulated voltage supply, and said second capacitor provides a positive regulated voltage supply, detector means providing an output electrical quantity when the magnitude of said second electrical quantity exceeds a predetermined value, and control means controlling said first and second output windings to provide their first and second electrical quantities during different portions of a half cycle of the alternating current, such that said energy storage means provides no load on said current transformer when the second output winding is providing its second electrical quantity to said detector means.
11. The electrical quantity responding device of claim 5 wherein the semiconductive means comprises at least one Zener diode.
12. An electrical quantity responding device comprising:
a current transformer, said current transformer including an input winding adapted to be energized from an alternating current network, and a first and second output winding means that are interlinked by the same flux path, said first output winding means when effective providing a first output current and having a fewer number of winding turns N1 than the number of winding turns N2 in said second output winding means, said second output winding means when effective providing a second output current having a magnitude proportional to the magnitude of the current in said alternating current network, control means controlling said first and second output winding means such that substantial magnitudes of said first and second output currents are provided by said output winding means in a sequential manner, an energy storage device to be energized by said first output current, a detecting circuit providing an output electrical quantity when the magnitude of said second output current exceeds a predetermined value, and limiting means for desensitizing said electrical quantity responding device when the magnitude of said current in said alternating current network exceeds a predetermined value.
13. The electrical quantity responding device of claim 12 wherein a resistor is connected such that when said second output winding means is effective said second output current flows through said resistor and provides a voltage across said resistor having a magnitude propor-tional to the magnitude of said current in said alternating current network.
14. The electrical quantity responding device of claim 12 wherein the detector circuit includes a resistor, and wherein the limiting means comprises a shunt circuit for shunting the second output current from said resistor when the current in the alternating current network exceeds the predetermined value of the limiting means.
15. The electrical quantity responding device of claim 14 wherein the shunt circuit comprises a voltage responsive switch device and means connecting said switch device in parallel with said resistor such that when the voltage across the resistor exceeds a predetermined magnitude, the second output current will be shunted from the resistor and through said switch device.
16. The electrical quantity responding device of claim 15 wherein the voltage responsive switch device comprises a Zener diode, a thyristor and connecting means connecting said Zener diode to said thyristor such that said second output current flows through said thyristor when the voltage magnitude across said Zener diode exceeds the breakover voltage of said Zener diode.
17. A detecting circuit for an electrical quantity sensing device for the protection of an alternating current network, said detecting circuit comprising:
an input circuit, said input circuit providing a DC voltage having a magnitude responsive to the magnitude of the electrical quantity in said alternating current network, and a time delay circuit connected with said input circuit such that an output electrical quantity is provided by said time delay circuit at a predetermined time TD after the magnitude of said DC input voltage exceeds a predetermined pickup value, said time delay circuit comprising circuit activating means, an RC network having at least one branch connected between input and output means, pulse generating means and counting means, said circuit activating means inhibiting and resetting the operation of said time delay circuit until the magnitude of said DC input voltage exceeds and pickup value, said RC network having its input means connected to be responsive to the DC voltage provided by said input circuit, said RC network providing an output at its output means responsive to the DC voltage said pulse generating means providing a pulse when the output of said RC network reaches a predetermined value, with said pulse advancing said counting means and also resetting and freeing said RC network to again be responsive to the DC input voltage, said counting means providing said output electrical quantity upon receipt of a predetermined number of the pulses from said pulse generating means.
18. The detecting circuit of claim 17 wherein said pulse generating means comprises a monostable multi-vibrator and triggering means, said triggering means gener-ating a pulse when the output of the RC network reaches the predetermined level, with the voltage pulse having a magnitude sufficient to energize said monostable multivibrator, said monostable multivibrator providing, in response to said voltage pulse a rectangular pulse shaped voltage having a predetermined pulse width TD2, said rectangular pulse shaped voltage advancing the counting means and resetting the RC
network.
19. The detecting circuit of claim 18 wherein the RC network includes resistive and capacitive elements, and reset means for momentarily deenergizing all of said capacitive elements, said RC network requiring, after said capacitive elements have been momentarily deenergized, a predetermined time period TD1 to generate an RC network voltage signal having a magnitude large enough to energize the triggering means, said time delay TD1 varying inversely with the magnitude of the DC input voltage.
20. The detecting circuit of claim 19 wherein the RC network comprises a plurality of parallel branches, each of said branches having resistive and capacitive elements, connecting means connecting said branches such that each of said branches provides a predetermined branch voltage having an inverse time characteristic, with the summation of said branch voltages providing the output of the RC network.
21. A polyphase electrical quantity sensing device for use with a polyphase alternating current network, said sensing device comprising:
an input circuit, said input circuit providing a DC voltage having a magnitude responsive to the highest magnitude of the phase currents in said network, and a detecting circuit comprising a time delay circuit, said time delay circuit comprising an RC circuit having an input and output means, circuit activating means, pulse generating means and counting means, said circuit activating means inhibiting and resetting the operation of said time delay circuit until the magnitude of said DC
input voltage exceeds a predetermined value) said RC network having its input means connected to be responsive to the DC
voltage provided by said input circuit, said RC network providing an output at its output means responsive to the DC voltage, said pulse generating means providing a pulse when the output of said RC circuit exceeds a predetermined value, with said pulse advancing said counting means and resetting said RC circuit to again be responsive to the DC
voltage, said counting means generating an output electrical quantity upon receipt of a predetermined number of pulses from said pulse generating means, to multiply the time delay of the RC circuit.
22. The electrical quantity responding device of claim 1 wherein the control means includes switching means switchable between conductive and non-conductive states, said switching means being connected to activate and de-activate the second output winding in response to the con-ductive state of the switching means, and means for control-ling the conductive state of said switching means.
23. The electrical quantity responding device of claim 1 wherein the control means includes regulator means for regulating the power supply means, switching means switchable between conductive and non-conductive states, said switching means being connected to activate and deactivate the second output winding in response to the conductive state of the switching means, and means responsive to said regulator means for controlling the conductive state of said switching means.
24. The electrical quantity responding device of claim 1 including first and second rectifier means for pro-viding unidirectional signals for the power supply means and for the current sensing means, responsive to the outputs of the first and second output windings, respectively.
25. An electrical quantity responding device comprising:
a current transformer, said current transformer including primary winding adapted to be energized by an alternating current, and first and second output windings, power supply means responsive to said first output winding, current sensing means responsive to said second output winding, first and second rectifier means for providing unidirectional signals for the power supply means and for the current sensing means, responsive to the outputs of the first and second output windings, respectively, and control means controlling said first and second output windings to achieve substantially complete balancing of the ampere turns of the primary winding in a selected one of said first and second output windings, and then in the other, within the same half cycle of the alter-nating current, said control means including unidirectional means connected between the first rectifier means and the power supply means, switching means connected to the second rectifier means to activate and deactivate the second output winding in response to its conductive state, and means respon-sive to said power supply means for controlling the conductive state of said switching means, said unidirectional means effectively disconnecting the power supply means from the first rectifier means when said switching means activates the second output winding, to deactivate the first output winding.
26. An electrical quantity responding device comprising:
a current transformer, said current transformer including primary winding adapted to be energized by an alternating current, and first and second output windings, power supply means responsive to said first output winding, said power supply means including energy storage means, current sensing means responsive to said second output winding, first and second rectifier means for providing unidirectional signals for the power supply means and for the current sensing means, responsive to the outputs of the first and second output windings, respectively, and control means controlling said first and second output windings to achieve substantially complete balancing of the ampere turns of the primary winding in a selected one of said first and second output windings, and then in the other, within the same half cycle of the alternating current, said control means including unidirectional means connected between the first rectifier means and the power supply means, regulator means connected to regulate the voltage of said energy storage means, switching means connected to the second rectifier means to activate and deactivate the second output winding in response to its conductive state, and means responsive to said regulator means for controlling the conductive state of said switching means, said unidirectional means effect-ively disconnecting the power supply means from the first rectifier means when said switching means activates the second output winding, to deactivate the first output winding.
27. An electrical quantity responding device comprising:
a current transformer, said current transformer including a primary winding adapted to be energized by an alternating current and first and second output windings, power supply means connected to be charged by said first output winding, current sensing means responsive to said second output winding, and control means controlling said first and second output windings to achieve substantially complete balancing of the ampere turns of the primary winding in a selected one of said first and second output windings, and then in the other, within the same half cycle of the alternating current, said control means including regulator means responsive to the power supply means for providing a signal when the power supply means is charged by the first output winding, and including a controllable bridge rectifier connected to the second output winding, said controllable bridge rectifier being switched to a conductive state in response to the signal from said regulator means, to allow current flow through the second output winding and provide a unidirectional signal for the current sensing means.
28. The detecting circuit of claim 17 wherein the counting means includes means for selecting the number of output pulses from the pulse generating means which will cause the counting means to provide the output electrical quantity.
29. The detecting circuit of claim 17 wherein the counting means includes binary counting means responsive to the output pulses from the pulse generating means, decade counting means connected to be responsive to said binary counting means, and selector switch means connected to said decade counting means for selecting a desired time scale reference curve.
30. An overcurrent relay, comprising:
a saturable current transformer having a primary winding adapted for energization by an alternating current, and a single secondary winding, rectifier and impedance means connected to said secondary winding, to provide a unidirectional voltage responsive to the current flowing in said primary winding, current sensing means connected to said impedance means, power supply means connected to be charged by said impedance means, control means connected to be responsive to the condition of said power supply means, said control means having a first condition which enables substantial current flow from said rectifier means when said power supply means requires charging, and a second condition which inhibits said substantial current flow when said power supply means has been charged.
31. The overcurrent relay of claim 30 including protective means responsive to the voltage across the impedance means, said protective means including switching means which is switched to a state which enables substantial current flow from the rectifier means to ground when the voltage across the impedance means reaches a predetermined value.
32. An electrical circuit for providing a pre-determined current versus time response, comprising:
RC network means having at least one RC branch connected between input and output terminal means, said input terminal means applying a DC input voltage to said input terminal means, means applying a DC reference voltage to the output terminal means of said RC network means, said DC
reference voltage causing current to flow in a predetermined direction relative to said output terminal means when the DC
reference voltage exceeds the DC input voltage, said RC network means causing a predetermined change in said current in response to the DC input voltage exceeding said DC reference voltage, with the time required to cause said predetermined change following the exceeding of the DC reference voltage by the DC input voltage being inversely proportional to the magnitude of said DC input voltage, and detector means providing an output signal in response to the occurrence of said predetermined change in said current.
33. The electrical circuit of claim 32 wherein the detector means provides its output signal when the current flow into the output terminal means drops to zero.
34. The electrical circuit of claim 32 wherein the predetermined direction of the current caused by the DC reference voltage is into the output terminal means of the RC network means, with the detector means providing its output signal when the current into the output terminal means drops to zero.
35. The electrical circuit of claim 32 including a circuit breaker having trip setting means and trip actuating means, with the means providing the DC reference voltage being responsive to the trip setting means, and within the output of the detector means being applied to said trip actuating means.
36. The electrical circuit of claim 32 including a circuit breaker having trip setting means and trip actuating means, pulse generating means, counting means, reset means for the RC network means, and decoding means, with the means providing the DC reference voltage being responsive to said trip setting means, said pulse generating means providing a pulse when the detector means provides its output signal, said pulse advancing said counting means and causing said reset means to reset the RC network means and free it to again be responsive to the DC input voltage, said decoding means providing a trip signal for said trip actuating means in response to a predetermined count on said counting means, to multiply the time delay provided by the RC network means.
37. The electrical circuit of claim 32 wherein the RC network means includes a plurality of RC branches connected in parallel between the input and output terminal means.
38. The electrical circuit of claim 37 wherein each RC branch has a different RC time constant.
CA240,189A 1974-11-26 1975-11-21 Protective relay device Expired CA1058291A (en)

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US52746074A 1974-11-26 1974-11-26

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JP (2) JPS5544532B2 (en)
AR (1) AR211110A1 (en)
AU (1) AU504554B2 (en)
BR (1) BR7507737A (en)
CA (1) CA1058291A (en)
DE (1) DE2552536A1 (en)
ES (1) ES443016A1 (en)
FR (1) FR2293093A1 (en)
GB (3) GB1539385A (en)
IN (1) IN146005B (en)
IT (1) IT1056349B (en)
SE (1) SE418787B (en)
SU (1) SU667175A3 (en)

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4476511A (en) * 1980-04-15 1984-10-09 Westinghouse Electric Corp. Circuit interrupter with front panel numeric display
GB2123627A (en) * 1982-04-08 1984-02-01 David Alan Dolbey Jones Electrical circuit interruption
GB2140633A (en) * 1983-04-29 1984-11-28 Plessey Co Plc Load tripping circuits
IT1197388B (en) * 1985-10-18 1988-11-30 Westinghouse Electric Corp RELEASE CONTROL WITH SENSOR FOR NUCLEAR REACTOR
JPH0327885Y2 (en) * 1985-11-01 1991-06-17
DE3638935A1 (en) * 1985-11-22 1987-05-27 Gen Electric SIGNAL ADJUSTMENT CIRCUIT FOR ELECTRONIC RELEASE SWITCHES
US4809125A (en) * 1987-02-20 1989-02-28 Westinghouse Electric Corp. Circuit interrupter apparatus with a style saving rating plug
US4751606A (en) * 1987-02-20 1988-06-14 Westinghouse Electric Corp. Circuit interrupter apparatus with a battery backup and reset circuit
FR2630594B1 (en) * 1988-04-22 1990-07-06 Alsthom POWER SUPPLY SYSTEM FOR DIGITAL PROTECTION EQUIPMENT IN ELECTRICITY DISTRIBUTION
FR2648176B1 (en) * 1989-06-07 1991-08-30 Hispano Suiza Sa HINGE ASSOCIATED WITH A MOBILE HOOD, IN PARTICULAR FOR AN AIRCRAFT NACELLE
US20080055795A1 (en) * 2006-08-25 2008-03-06 Miller Theodore J Power supply start-up circuit for a trip unit and circuit interrupter including the same
DE102007005748A1 (en) * 2007-02-01 2008-08-07 Siemens Ag Device for monitoring discharge processes in a conductor of a medium or high voltage system

Also Published As

Publication number Publication date
JPS5544532B2 (en) 1980-11-12
GB1539387A (en) 1979-01-31
GB1539386A (en) 1979-01-31
JPS5850499B2 (en) 1983-11-10
BR7507737A (en) 1976-08-10
SE418787B (en) 1981-06-22
SU667175A3 (en) 1979-06-05
AU504554B2 (en) 1979-10-18
FR2293093A1 (en) 1976-06-25
AR211110A1 (en) 1977-10-31
JPS5534897A (en) 1980-03-11
JPS5176543A (en) 1976-07-02
IN146005B (en) 1979-02-03
DE2552536A1 (en) 1976-08-12
FR2293093B1 (en) 1978-05-19
IT1056349B (en) 1982-01-30
GB1539385A (en) 1979-01-31
ES443016A1 (en) 1977-07-01
SE7513207L (en) 1976-05-28
AU8653775A (en) 1977-05-19

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