WO2024073955A1 - Gfsk通信模式内的双比特组解调方法及解调器 - Google Patents

Gfsk通信模式内的双比特组解调方法及解调器 Download PDF

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WO2024073955A1
WO2024073955A1 PCT/CN2022/141995 CN2022141995W WO2024073955A1 WO 2024073955 A1 WO2024073955 A1 WO 2024073955A1 CN 2022141995 W CN2022141995 W CN 2022141995W WO 2024073955 A1 WO2024073955 A1 WO 2024073955A1
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symbol
demodulated
phase rotation
demodulation
interval
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PCT/CN2022/141995
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English (en)
French (fr)
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吴政勋
许诒翔
钱海锋
崔国宇
门长有
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杭州万高科技股份有限公司
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/10—Frequency-modulated carrier systems, i.e. using frequency-shift keying
    • H04L27/14—Demodulator circuits; Receiver circuits

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  • the invention belongs to the field of communication baseband signal processing, and in particular relates to a double-bit group demodulation method and a demodulator in a GFSK communication mode.
  • the original signal demodulation method is to process one symbol at a time, and then process the next received symbol after demodulation is completed, and there is no demodulation correlation between symbols.
  • the GFSK modulation method effectively reduces the use of communication bandwidth through Gaussian filters to save hardware costs, the cost is the generation of inter-symbol interference (ISI).
  • ISI inter-symbol interference
  • This inter-symbol interference created for bandwidth control will have a certain negative impact on the receiver demodulation performance.
  • the use of digital filters can reduce the inter-symbol interference in the symbol sequence, perform symbol decisions based on the filtered symbol sequence, and use the symbol to bit mapper to directly obtain the demodulated data bits.
  • the methods for removing ISI at the receiver end can be roughly divided into matched filter (MF) methods and differential phase methods.
  • MF matched filter
  • the matched filter provides a phase reference for the receiver and detects the data from the MF output to complete the demodulation work.
  • MI modulation index
  • the differential method calculates the phase difference between consecutive symbols to obtain the symbol frequency, and directly performs further demodulation processing on the symbol frequency.
  • this direct demodulation processing mechanism often limits the performance of the receiver. When receiving weak signals close to the sensitivity limit, the probability of obtaining erroneous demodulation data is higher.
  • ISI inter-symbol interference
  • AFE Analog front end
  • IoT Internet of Things
  • the technical problem to be solved by the present invention is to provide a two-bit group demodulation method in a GFSK communication mode in view of the deficiencies in the prior art.
  • a first aspect discloses a two-bit group demodulation method in a GFSK communication mode, comprising the following steps:
  • Step 1 receiving a GFSK modulated signal and converting the GFSK modulated signal into a complex valued baseband sample
  • Step 2 calculating the phase of the complex-valued baseband sample to obtain the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated;
  • Step 3 determining the phase rotation amount of each symbol sent by the transmitter and setting the demodulation interval
  • Step 4 Obtain the current symbol to be demodulated according to the demodulation interval.
  • determining the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter in step 3 includes: inter-symbol interference introduced by Gaussian filtering at the transmitter makes the phase rotation amount ⁇ n of the symbol Sn affected by its previous symbol Sn -1 and the next symbol Sn +1 ,
  • phase rotation amount ⁇ n of the symbol Sn is ⁇ ⁇ L ;
  • phase rotation amount ⁇ n of the symbol Sn is ⁇ ⁇ M ;
  • ⁇ H , ⁇ M and ⁇ L are determined by the GFSK modulation index and the bandwidth-symbol-time product, and ⁇ H > ⁇ M > ⁇ L .
  • the demodulation interval in step 3 is set according to the phase rotation amount ⁇ n of each symbol S n sent by the transmitter and the total phase rotation amount ⁇ dbg of two consecutive symbols (i.e., two-bit groups), including:
  • the partition threshold of the demodulation interval is configured to be the middle value between two adjacent subsets to obtain the first partition threshold and the second partition threshold - ⁇ dbg , the demodulation interval is divided into three intervals with - ⁇ dbg and ⁇ dbg as boundaries.
  • the demodulation interval in step 3 is set based on the previous demodulation symbol Dn -1 , the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter, and the total phase rotation amount ⁇ dbg of two consecutive symbols of the transmitter, including:
  • the partition threshold of the demodulation interval is configured to be the middle value between two adjacent subsets to obtain the first partition threshold and the second partition threshold in and
  • the demodulation interval is and It is divided into three intervals.
  • ⁇ ′′ dbg represents the sum of the phase rotation amounts of the current demodulated symbol and the previous demodulated symbol
  • ⁇ ambi represents the ambiguity interval threshold, which determines the range of the ambiguity interval
  • ⁇ represents whether ⁇ ′′ dbg falls within this ambiguity interval
  • ⁇ ambi (3 ⁇ M - ⁇ L )/2; the demodulation interval is and It is divided into three intervals.
  • step 4 obtains the current to-be-demodulated symbol Dn according to the demodulation interval:
  • ⁇ ' dbg represents the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the method of determining the current demodulation symbol D n under the condition that the second partition threshold ⁇ ⁇ ' dbg ⁇ the first partition threshold
  • ⁇ 'n represents the phase rotation amount of the current symbol to be demodulated
  • ⁇ 'n +1 represents the phase rotation amount of the next symbol to be demodulated
  • step 4 obtains the current to-be-demodulated symbol Dn according to the demodulation interval:
  • ⁇ ' dbg represents the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the method of determining the current demodulation symbol D n under the condition that the second partition threshold ⁇ ⁇ ' dbg ⁇ the first partition threshold
  • ⁇ 'n represents the phase rotation amount of the current symbol to be demodulated
  • ⁇ 'n -1 represents the phase rotation amount of the next symbol to be demodulated
  • a dual-bit group demodulator in a GFSK communication mode comprising a symbol buffer module, an addition module and a symbol decision module, wherein the symbol buffer module is used to store a phase rotation amount of a symbol to be demodulated,
  • the adding module is used to add the phase rotation amount of the current symbol to be demodulated and the phase rotation amount of the next symbol to be demodulated to obtain the sum of the phase rotation amounts of the two-bit group;
  • the symbol decision module is used to obtain the current to-be-demodulated symbol according to the demodulation interval.
  • the partition threshold of the demodulation interval is set according to the phase rotation amount of each symbol sent by the transmitter and the sum of the phase rotation amounts of two consecutive symbols.
  • the partition threshold of the demodulation interval is set based on the previous demodulation symbol, the phase rotation amount of each symbol sent by the transmitter, and the sum of the phase rotation amounts of two consecutive symbols of the transmitter;
  • the dual-bit group demodulator further includes a previous symbol decision buffer module and a dynamic threshold decision module, wherein the previous symbol decision buffer module is used to store a previous demodulated symbol.
  • the dynamic threshold determination module is used to select a partition threshold of the demodulation interval according to a previous demodulation symbol.
  • the two-bit group demodulator further includes a previous symbol ambiguity buffer module and an ambiguity correction module, wherein the previous symbol ambiguity buffer module is used to store the result of whether the sum of the phase rotation amounts of the current to-be-demodulated symbol and the previous demodulated symbol falls within the ambiguity interval;
  • the ambiguity correction module is used to determine whether the sum of the phase rotation amounts of the current symbol to be demodulated and the previous demodulated symbol falls within the ambiguity interval, and store the result in the previous symbol ambiguity buffer module, and correct the partition threshold of the demodulation interval of the current symbol to be demodulated according to the result.
  • the present invention uses two consecutive symbols (i.e., two-bit groups) to perform demodulation.
  • the two consecutive symbols here are the current target demodulation symbol and the next symbol on the time axis. According to the possible results of the two consecutive symbols, a judgment result with high reliability is summarized and analyzed. Based on this new demodulation technology, the probability of demodulation error is reduced, and the accuracy (reliability) of GFSK in demodulation is improved, thereby improving the actual communication performance.
  • FIG1 is a schematic structural diagram of a receiving end of a two-bit group demodulation method in a GFSK communication mode provided in an embodiment of the present application.
  • FIG. 2 is a first schematic diagram of the structure of a two-bit group demodulator in a GFSK communication mode provided in an embodiment of the present application.
  • FIG. 3 is a schematic diagram of the sum of phase rotations within a dibit group when the transmitting end sends two identical symbols in the dibit group demodulation method within the GFSK communication mode provided in an embodiment of the present application.
  • FIG4 is a schematic diagram of the sum of phase rotations within a dibit group when the transmitting end sends two different symbols in the dibit group demodulation method within the GFSK communication mode provided in an embodiment of the present application.
  • FIG5 is a schematic diagram showing the relative relationship of the ascending order sorting of the phase rotation sums within a dibit group in the dibit group demodulation method within the GFSK communication mode provided in an embodiment of the present application.
  • FIG6 is a schematic diagram of determining a phase rotation sum threshold ⁇ dbg within a dibit group in a dibit group demodulation method within a GFSK communication mode provided in an embodiment of the present application.
  • FIG7 is a schematic diagram of dual-symbol phase rotation based on the previous symbol D n-1 symbol decision in the dual-bit group demodulation method in the GFSK communication mode provided by an embodiment of the present application.
  • FIG8 is a schematic diagram showing the relative relationship of dual-symbol phase rotations in a dual-bit group demodulation method in a GFSK communication mode provided in an embodiment of the present application.
  • FIG. 9 is a schematic diagram of determining dual-symbol phase rotation sum thresholds ⁇ dbg,p and ⁇ dbg,n based on the previous symbol D n-1 symbol decision in the dual-bit group demodulation method in the GFSK communication mode provided by an embodiment of the present application.
  • FIG10 is a schematic diagram of an ambiguous interval in a two-bit group demodulation method in a GFSK communication mode provided in an embodiment of the present application.
  • FIG. 11 is a second schematic diagram of the structure of a two-bit group demodulator in the GFSK communication mode provided in an embodiment of the present application.
  • FIG. 12 is a third schematic diagram of the structure of a two-bit group demodulator in the GFSK communication mode provided in an embodiment of the present application.
  • FIG13 is a schematic diagram showing the relationship between the signal-to-noise ratio and the error rate of the two-bit group demodulation method in the GFSK communication mode provided in an embodiment of the present application and the demodulation method in the prior art.
  • the two-bit group demodulation method in the GFSK communication mode provided in the present application can be applied to Bluetooth chips and power line communication chips, and is widely used in wireless two-way application products such as IoT products requiring low power consumption, smart home/security, remote meter reading, industrial/agricultural controllers, etc.
  • FSK modulation uses different frequencies to carry bit information. For example, when BFSK sends a 0 bit, the carrier frequency corresponding to the symbol 0 is sent in a predetermined symbol period, and another carrier frequency is sent as the symbol of bit 1. In the symbol period, an almost constant frequency is always sent. With the switching of the transmitted bit information, the carrier frequency will also have a sudden change. This rapid change will bring out unnecessary spectral lines, and also cause the increase of transmission bandwidth and the divergence of transmission energy. Therefore, the rapid switching between the two frequencies not only increases the design complexity, but also reduces the spectrum efficiency.
  • Gaussian FSK modulation Data is encoded in FSK, which is a variation of FSK.
  • the modulator used is the same as that used for FSK modulation.
  • the pulses pass through a Gaussian filter to reduce the bandwidth before entering the pulse modulator.
  • the Gaussian filter is a time-domain pulse shaper that is used to smooth out rapid changes between consecutive pulse values.
  • the transfer function of the Gaussian low-pass filter is
  • f represents the frequency
  • parameter ⁇ is related to the 3dB bandwidth B of the baseband Gaussian shaping filter. It is usually expressed as the normalized 3dB bandwidth-symbol time product BTs:
  • Ts represents the symbol period.
  • ⁇ increases, the spectrum occupancy of the Gaussian filter decreases, and the impulse response spreads over adjacent symbols, causing a substantial increase in the ISI seen by the receiver. Since multiple GFSK symbols are transmitted, both the previous symbol and the next symbol have an impact on the current symbol, which is called ISI.
  • the magnitude of ISI depends not only on the channel, but also on the accuracy of the BT and GFSK modulation index (MI). If the receiver can fully detect the phase of the received signal, ISI can be removed and will not cause any problems. However, in actual implementations, since the phase is unknown, ISI often affects the demodulation performance.
  • digital filters can be used to reduce inter-symbol interference (ISI) in the symbol sequence, and symbol decisions are performed based on the filtered symbol sequence.
  • ISI inter-symbol interference
  • the demodulated data bits are directly obtained using a symbol-to-bit mapper.
  • the methods for removing ISI at the receiver can be roughly divided into matched filter (MF) methods and differential phase methods.
  • MF matched filter
  • the matched filter provides a phase reference for the receiver and detects the data from the MF output to complete the demodulation. This method is more like a coherent demodulator, which requires higher computational complexity and is sensitive to phase noise or MI accuracy.
  • the differential method calculates the phase difference between consecutive symbols to obtain the symbol frequency, and directly performs further demodulation on the symbol frequency.
  • the dibit group demodulation method in the GFSK communication mode proposed in the first embodiment of the present application is a new GFSK demodulation method, which provides good demodulation performance for GFSK, and includes the following steps:
  • Step 1 receiving a GFSK modulated signal and converting the GFSK modulated signal into a complex valued baseband sample
  • Step 2 calculating the phase of the complex-valued baseband sample to obtain the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated;
  • Step 3 determining the phase rotation amount of each symbol sent by the transmitter and setting the demodulation interval
  • Step 4 Obtain the current symbol to be demodulated according to the demodulation interval.
  • FIG. 1 it is a schematic diagram of the structure of the receiving end of this embodiment.
  • the receiver end it includes an analog front end and a digital front end.
  • the analog front end is used to receive the GFSK modulated signal and convert it into a digital signal, and filter and down-convert the carrier through the digital front end to obtain complex-valued baseband samples.
  • step 2 includes calculating the phase of the complex-valued baseband sample, estimating the phase rotation of all to-be-demodulated symbols according to the phase of the complex-valued baseband sample and the symbol timing recovery circuit, and obtaining the sum of the phase rotation of the current to-be-demodulated symbol and the next to-be-demodulated symbol. Then, the to-be-demodulated symbol is obtained by a dibit group demodulator, and the dibit group demodulator executes steps 3 and 4.
  • determining the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter in step 3 includes: a two-bit group at the transmitter includes two consecutive GFSK symbols [ Sn , Sn +1 ].
  • the current symbol index is n, and the index n+1 refers to the next symbol after the current symbol. That is, symbol n+1 is needed to demodulate the current symbol.
  • n+1 is needed to demodulate the current symbol.
  • the indexes n and n+1 are retained and explained in terms of the representation of a non-causal system.
  • the step 3 of setting the demodulation interval is set according to the phase rotation amount ⁇ n of each symbol S n sent by the transmitter and the sum of the phase rotation amounts ⁇ dbg of two consecutive symbols, including:
  • phase rotation sum ⁇ dbg ⁇ n + ⁇ n +1 within a dibit group can be one of the values of ⁇ dbg ⁇ ⁇ 2 ⁇ M , ⁇ 2 ⁇ H , ⁇ ( ⁇ M + ⁇ H ) ⁇ , as shown in FIG3.
  • ⁇ dbg The values of ⁇ dbg are sorted in ascending order to obtain the first subset ⁇ -2 ⁇ H ,- ⁇ H - ⁇ M ,-2 ⁇ M ⁇ , the second subset ⁇ -( ⁇ M - ⁇ L ),0, ⁇ M - ⁇ L ⁇ and the third subset ⁇ 2 ⁇ M , ⁇ M + ⁇ H ,2 ⁇ H ⁇ , as shown in FIG5 .
  • the dibit group demodulator receives the phase rotation sum ⁇ ' dbg of consecutive GFSK symbols, it determines which subset (that is, which dibit group) ⁇ ' dbg belongs to by comparing ⁇ ' dbg with the partition threshold of the demodulation interval.
  • the optimal setting of the partition threshold of the demodulation interval is to configure it to the middle value between each subset, and obtain the first partition threshold and the second partition threshold - ⁇ dbg , the demodulation interval is divided into three intervals with - ⁇ dbg and ⁇ dbg as boundaries, as shown in FIG6 .
  • the step 4 obtains the current to-be-demodulated symbol Dn according to the demodulation interval:
  • ⁇ ' dbg represents the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the method of determining the current demodulation symbol D n under the condition of - ⁇ dbg ⁇ ' dbg ⁇ dbg .
  • Q( ⁇ ' n , ⁇ ' n+1 ) is obtained by comparing the symbol phase rotation amount ⁇ ' n of the current symbol to be demodulated with the symbol phase rotation amount ⁇ ' n+1 of the next symbol to be demodulated, and the expression is as follows:
  • Q( ⁇ ' n , ⁇ ' n+1 ) only considers the phase rotation amount ⁇ ' n of the current symbol to be demodulated, and the expression is as follows:
  • the demodulation interval in step 3 is set based on the previous demodulation symbol Dn -1 , the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter, and the total phase rotation amount ⁇ dbg of two consecutive symbols of the transmitter, including:
  • the optimal solution for the two-bit group partition is to make the partition threshold in the middle of two adjacent subsets. It should be noted that the partition threshold is no longer symmetrical around 0.
  • the partition threshold of the demodulation interval is configured at the middle value between two adjacent subsets to obtain the first partition threshold and the second partition threshold in and
  • the demodulation interval is and As the boundary, it is divided into three intervals, as shown in Figure 9.
  • the step 4 obtains the current to-be-demodulated symbol Dn according to the demodulation interval:
  • ⁇ ' dbg represents the sum of the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated.
  • Q( ⁇ 'n , ⁇ 'n +1 ) is obtained by comparing the symbol phase rotation amount ⁇ 'n of the current symbol to be demodulated with the phase rotation amount ⁇ 'n +1 of the next symbol to be demodulated, and the expression is as follows:
  • Q( ⁇ ' n , ⁇ ' n+1 ) only considers the phase rotation amount ⁇ ' n of the current symbol to be demodulated, and the expression is as follows:
  • the distance between the dibit group subsets is substantially extended. This provides a more robust decision interval for the dibit groups, and thus correspondingly better demodulation performance.
  • the improvement in demodulation performance is achieved by taking advantage of the fact that we know the bits carried by symbol n-1. However, there are still possible errors in the decoding of symbol n-1. If erroneous demodulation of symbol n-1 occurs, the partition threshold of the dynamic two-bit grouping adopted for the current symbol will no longer be appropriate, and then the threshold for subset determination becomes incorrect, which may lead to erroneous demodulation of the current symbol n. This is called the error propagation effect.
  • an additional ambiguity detection scheme is proposed, that is, if the sum of the phase rotation amounts of the current symbol to be demodulated and the previous demodulated symbol (the sum of the phase rotation amounts of the two-bit group) ⁇ ′′ dbg is too close to the partition threshold of the two-bit group, that is, too close to the first partition threshold and the second partition threshold.
  • the decision of the previous demodulation symbol is determined to be an ambiguous decision, and the demodulation interval needs to be ambiguously corrected for demodulation of the current symbol to be demodulated.
  • the demodulation interval is set in step 3 based on the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter and the total phase rotation amount ⁇ dbg of two consecutive symbols; if the following expression is false, the symbol decision of the n-1th symbol is determined to be a reliable decision, and the demodulation interval of the n-1th symbol does not need to be corrected, that is, the demodulation interval is set in step 3 based on the previous demodulation symbol Dn -1 , the phase rotation amount ⁇ n of each symbol Sn sent by the transmitter, and the total phase rotation amount ⁇ dbg of two consecutive symbols of the transmitter.
  • ⁇ ′′ dbg represents the sum of the phase rotations of the current demodulated symbol and the previous demodulated symbol
  • ⁇ ambi represents the ambiguity interval threshold, which determines the range of the ambiguity interval
  • the value range of the ambiguity interval threshold ⁇ ambi is between 0 and ( ⁇ H - ⁇ L ), preferably ⁇ indicates whether ⁇ ′′ dbg falls within this ambiguous interval, as shown in Figure 10;
  • ⁇ ambi (3 ⁇ M - ⁇ L )/2; the demodulation interval is and It is divided into three intervals.
  • the step 4 obtains the current to-be-demodulated symbol D n according to the demodulation interval, that is, the symbol decision as shown in FIG2 :
  • ⁇ ' dbg represents the sum of the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated
  • ⁇ 'n represents the phase rotation amount of the current symbol to be demodulated
  • ⁇ 'n +1 represents the phase rotation amount of the next symbol to be demodulated
  • the step 4 obtains the current to-be-demodulated symbol Dn according to the demodulation interval:
  • ⁇ ' dbg represents the sum of the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the phase rotations of the current symbol to be demodulated and the next symbol to be demodulated
  • ⁇ 'n represents the phase rotation amount of the current symbol to be demodulated
  • ⁇ 'n +1 represents the phase rotation amount of the next symbol to be demodulated
  • the second embodiment of the present application discloses a dual-bit group demodulator in a GFSK communication mode, as shown in FIG2 , comprising a symbol buffer module, an addition module and a symbol decision module, wherein the symbol buffer module is used to store the phase rotation amount of the symbol to be demodulated,
  • the adding module is used to add the phase rotation amount of the current symbol to be demodulated and the phase rotation amount of the next symbol to be demodulated to obtain the sum of the phase rotation amounts of the two-bit group;
  • the symbol decision module is used to obtain the current to-be-demodulated symbol Dn according to the demodulation interval.
  • ⁇ ' dbg represents the sum of the phase rotation amounts of the current symbol to be demodulated and the next symbol to be demodulated
  • Q( ⁇ ' n , ⁇ ' n+1 ) represents the method of determining the current demodulated symbol D n under the condition that the second partition threshold ⁇ ' dbg ⁇ the first partition threshold.
  • Q( ⁇ ' n , ⁇ ' n+1 ) is obtained by comparing the symbol phase rotation amount ⁇ ' n of the current symbol to be demodulated with the phase rotation amount ⁇ ' n+1 of the next symbol to be demodulated, and the expression is as follows:
  • Q( ⁇ ' n , ⁇ ' n+1 ) only considers the phase rotation amount ⁇ ' n of the current symbol to be demodulated, and the expression is as follows:
  • the partition threshold of the demodulation interval is set according to the phase rotation amount of each symbol sent by the transmitter and the sum of the phase rotation amounts of two consecutive symbols, as shown in FIG6 .
  • the partition threshold of the demodulation interval is configured to be the middle value between two adjacent subsets to obtain the first partition threshold and the second partition threshold - ⁇ dbg , the demodulation interval is divided into three intervals with - ⁇ dbg and ⁇ dbg as boundaries.
  • the partition threshold of the demodulation interval is set based on the previous demodulation symbol, the phase rotation amount of each symbol sent by the transmitter, and the sum of the phase rotation amounts of two consecutive symbols of the transmitter, as shown in FIG9 ;
  • the partition threshold of the demodulation interval is configured to be the middle value between two adjacent subsets to obtain the first partition threshold and the second partition threshold in and
  • the demodulation interval is and It is divided into three intervals.
  • the two-bit group demodulator further includes a previous symbol decision buffer module and a dynamic threshold decision module.
  • the previous symbol decision buffer module is used to store the previous demodulated symbol.
  • the dynamic threshold determination module is used to select a partition threshold of the demodulation interval according to a previous demodulation symbol.
  • the two-bit group demodulator further includes a previous symbol ambiguity buffer module and an ambiguity correction module, wherein the previous symbol ambiguity buffer module is used to store the result of whether the sum of the phase rotation amounts of the current to-be-demodulated symbol and the previous demodulated symbol falls within the ambiguity interval;
  • the ambiguity correction module is used to determine whether the sum of the phase rotation amounts of the current demodulated symbol and the previous demodulated symbol falls within the ambiguity interval, store the result in the previous symbol ambiguity buffer module, and correct the partition threshold of the demodulation interval of the current demodulated symbol according to the result.
  • ⁇ ′′ dbg represents the sum of the phase rotations of the current demodulated symbol and the previous demodulated symbol
  • ⁇ ambi represents the ambiguity interval threshold, which determines the range of the ambiguity interval
  • the value range of the ambiguity interval threshold ⁇ ambi is between 0 and ( ⁇ H - ⁇ L ), preferably ⁇ indicates whether ⁇ ′′ dbg falls within this ambiguous interval, as shown in Figure 10;
  • ⁇ ambi (3 ⁇ M - ⁇ L )/2; the demodulation interval is and It is divided into three intervals.
  • the two-bit group demodulation method and demodulator in the GFSK communication mode provided by the embodiment of the present application have a lower bit error rate (BER) than the single symbol demodulation method and demodulator in the prior art, that is, under the same channel conditions, the two-bit group demodulation method and demodulator provided by the embodiment of the present application are less prone to errors; and when the signal-to-noise ratio gradually increases, the bit error rate of the two-bit group demodulation method and demodulator provided by the embodiment of the present application is lower than the bit error rate target. Therefore, the two-bit group demodulation method and demodulator in the GFSK communication mode provided by the embodiment of the present application reduce the demodulation error probability, improve the accuracy (reliability) of GFSK in demodulation, and thus improve the actual communication performance.
  • SNR signal-to-noise ratio
  • the present application provides a computer storage medium and a corresponding data processing unit, wherein the computer storage medium can store a computer program, and when the computer program is executed by the data processing unit, the invention content of the double-bit group demodulation method in a GFSK communication mode provided by the present invention and some or all of the steps in each embodiment can be executed.
  • the storage medium can be a disk, an optical disk, a read-only memory (ROM) or a random access memory (RAM), etc.
  • the technical solutions in the embodiments of the present invention can be implemented by means of computer programs and their corresponding general hardware platforms. Based on such an understanding, the technical solutions in the embodiments of the present invention are essentially or partly contributed to the prior art can be embodied in the form of a computer program, i.e., a software product, which can be stored in a storage medium and includes several instructions for enabling a device including a data processing unit (which can be a personal computer, a server, a single-chip microcomputer, a MUU or a network device, etc.) to execute the methods described in various embodiments of the present invention or certain parts of the embodiments.
  • a data processing unit which can be a personal computer, a server, a single-chip microcomputer, a MUU or a network device, etc.
  • the present invention provides a two-bit group demodulation method and demodulator in a GFSK communication mode.

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Abstract

本发明公开一种GFSK通信模式内的双比特组解调方法及解调器，所述双比特组解调方法包括步骤1，接收GFSK调制信号并将所述GFSK调制信号转化为复数值基带样本；步骤2，计算所述复数值基带样本的相位，获得当前待解调符号和下一个待解调符号的相位旋转量总和；步骤3，确定发送端发送的每个符号的相位旋转量，设置解调区间；步骤4，根据解调区间获得当前待解调符号。该方法采用处理两个连续的符号来进行解调，降低了解调错误机率，提升GFSK在解调上的精准度。

Description

GFSK通信模式内的双比特组解调方法及解调器 技术领域
本发明属于通信基带信号处理领域，尤其涉及一种GFSK通信模式内的双比特组解调方法及解调器。
背景技术
在GFSK的通信架构下，原本信号解调方式为一次处理一个符号，解调完成后再处理下一个接收符号，而符号与符号之间并无解调相关性。GFSK的调变方式，虽说透过高斯滤波器，来有效的减少使用通信带宽，以达到节省硬件成本，但其代价是符号间干扰(ISI)的产生，这种为了带宽控制而制造的符号间干扰对于接收器解调性能上会有一定的负面影响。概念上，使用数字滤波器可以来减少符号序列中的符号间干扰，根据滤波后的符号序列执行符号决策，使用符号到比特映像器来直接得到解调的数据比特。一般来说，在接收机端去除ISI的方法可以大致分为匹配滤波器(MF)方法和差分相位方法。匹配滤波器是为接收器提供相位参考并检测来自MF输出的数据来完成解调的工作，这种方法更像是一种相干解调器，它需要更高的计算复杂度，且对相位噪声或调制指数(MI)精度很敏感。而差分方法通过计算连续符号之间的相位差得出符号频率，并且直接对符号频率作进一步的解调处理，但是这种直接解调处理机制往往会让接收器的效能受限，在接收接近灵敏度(sensitivity)极限的弱信号时，得到错误解调资料的概率较大。
术语解释：
FSK(Frequency-shift keying modulation)，移频键控
BFSK(binary FSK modulation)，二进制频移键控
GFSK(Gaussian FSK modulation)，高斯频移键控
BTs(bandwidth-symbol time product)，带宽-符号时间积
ISI(inter-symbol interference)，符号间干扰
MI(Modulation Index)，调制指数
MF(matched filter)，匹配滤波器
DFE(Digital front end)，数字前端
AFE(Analog front end)，模拟前端
STR(Symbol timing recovery，符号定时恢复
DBG(Dual-bit group)，双比特组
IoT(Internet of Things)，物联网
SNR(signal-to-noise ratio)，信噪比
BER(bit error rate)，误码率
发明内容
发明目的：本发明所要解决的技术问题是针对现有技术的不足，提供一种GFSK通信模式内的双比特组解调方法。
为了解决上述技术问题，第一方面公开了一种GFSK通信模式内的双比特组解调方法，包括以下步骤：
步骤1，接收GFSK调制信号并将所述GFSK调制信号转化为复数值基带样本；
步骤2，计算所述复数值基带样本的相位，获得当前待解调符号和下一个待解调符号的相位旋转量总和；
步骤3，确定发送端发送的每个符号的相位旋转量，设置解调区间；
步骤4，根据解调区间获得当前待解调符号。
进一步地，所述步骤3中确定发送端发送的每个符号S n的相位旋转量φ n包括：发送端高斯滤波 引入的符号间干扰使得符号S n的相位旋转量φ n受到其前一个符号S n-1和下一个符号S n+1的影响，
当S n≠S n-1，S n≠S n+1时，符号S n的相位旋转量φ n为±θ L；
当S n＝S n-1，S n≠S n+1或者S n≠S n-1，S n＝S n+1时，符号S n的相位旋转量φ n为±θ M；
当S n＝S n-1＝S n+1时，符号S n的相位旋转量φ n为±θ H；
其中，θ H、θ M和θ L由GFSK调制指数和带宽符号时间积确定，且θ H>θ M>θ L。
进一步地，所述步骤3中设置解调区间是根据发送端发送的每个符号S n的相位旋转量φ n以及连续两个符号(即双比特组)的相位旋转量总和Φ dbg进行设置，包括：
连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，当S n＝S n+1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{±2θ M,±2θ H,±(θ M+θ H)}；
当S n≠S n+1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{±(θ M-θ L),0}；
将上述Φ dbg的取值按升序排序，获得第一子集{-2θ H,-θ H-θ M,-2θ M}、第二子集{-(θ M-θ L),0,θ M-θ L}和第三子集{2θ M,θ M+θ H,2θ H}；
将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000001
Figure PCTCN2022141995-appb-000002
和第二分区阈值-Ψ dbg，解调区间以-Ψ dbg和Ψ dbg为界，分为三个区间。
进一步地，所述步骤3中设置解调区间是基于前一个解调符号D n-1、发送端发送的每个符号S n的相位旋转量φ n以及发送端连续两个符号的相位旋转量总和Φ dbg进行设置，包括：
发送端连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，当D n-1＝0时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-2θ H,-θ H-θ M,-(θ M-θ L),0,2θ M,θ H+θ M}，将上述Φ dbg的取值按升序排序，获得第四子集{-2θ H,-θ H-θ M}、第五子集{-(θ M-θ L),0}和第六子集{2θ M,θ H+θ M}；
当D n-1＝1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-θ H-θ M,-2θ M,0,θ M-θ L,θ H+θ M,2θ H}，将上述Φ dbg的取值按升序排序，获得第七子集{-θ H-θ M,-2θ M}、第八子集{0,θ M-θ L}和第九子集{θ H+θ M,2θ H}；
将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000003
和第二分区阈值
Figure PCTCN2022141995-appb-000004
其中
Figure PCTCN2022141995-appb-000005
和
Figure PCTCN2022141995-appb-000006
Figure PCTCN2022141995-appb-000007
如果D n-1为0
Figure PCTCN2022141995-appb-000008
如果D n-1为1
解调区间以
Figure PCTCN2022141995-appb-000009
和
Figure PCTCN2022141995-appb-000010
为界，分为三个区间。
进一步地，所述步骤3中设置解调区间时，若如下表达式为真，
Figure PCTCN2022141995-appb-000011
其中，Φ″ dbg表示当前待解调符号和前一个解调符号的相位旋转量总和，τ ambi表示歧义区间阈值，决定歧义区间的范围，Λ表示Φ″ dbg是否落在此歧义区间；
则修正解调区间的分区阈值，将第一分区阈值修正为
Figure PCTCN2022141995-appb-000012
第二分区阈值修正为
Figure PCTCN2022141995-appb-000013
Figure PCTCN2022141995-appb-000014
其中，Ψ ambi＝(3θ M-θ L)/2；解调区间以
Figure PCTCN2022141995-appb-000015
和
Figure PCTCN2022141995-appb-000016
为界，分为三个区间。
进一步地，所述步骤4根据解调区间获得当前待解调符号D n：
Figure PCTCN2022141995-appb-000017
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在第二分区阈值≤Φ' dbg≤第一分区阈值这个条件下决定当前解调符号D n的方式，表达式如下：
Figure PCTCN2022141995-appb-000018
其中，φ' n表示当前待解调符号的相位旋转量，φ' n+1表示下一个待解调符号的相位旋转量。
进一步地，所述步骤4根据解调区间获得当前待解调符号D n：
Figure PCTCN2022141995-appb-000019
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在第二分区阈值≤Φ' dbg≤第一分区阈值这个条件下决定当前解调符号D n的方式，表达式如下：
Figure PCTCN2022141995-appb-000020
其中，φ' n表示当前待解调符号的相位旋转量，φ' n-1表示下一个待解调符号的相位旋转量。
第二方面，公开了一种GFSK通信模式内的双比特组解调器，包括符号缓冲模块、相加模块和符号决策模块，所述符号缓冲模块，用于存储待解调符号的相位旋转量，
所述相加模块，用于将当前待解调符号的相位旋转量和下一个待解调符号的相位旋转量相加，获得双比特组的相位旋转量总和；
所述符号决策模块，用于根据解调区间获得当前待解调符号。
进一步地，所述解调区间的分区阈值是根据发送端发送的每个符号的相位旋转量以及连续两个符号的相位旋转量总和进行设置的。
进一步地，所述解调区间的分区阈值是基于前一个解调符号、发送端发送的每个符号的相位旋转量以及发送端连续两个符号的相位旋转量总和进行设置的；
所述双比特组解调器还包括前一符号决策缓冲模块和动态阈值决定模块，所述前一符号决策缓冲模块，用于存储前一个解调符号，
所述动态阈值决定模块，用于根据前一个解调符号，选择解调区间的分区阈值。
进一步地，所述双比特组解调器还包括前一符号歧义缓冲模块和歧义修正模块，所述前一符号歧义缓冲模块，用于存储当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间的结果；
所述歧义修正模块，用于判断当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间，并将结果存储至前一符号歧义缓冲模块，以及根据结果修正当前待解调符号的解调区间 的分区阈值。
有益效果：
本发明采用处理两个连续的符号(即双比特组)来进行解调，这里的两个连续符号分别为当下目标解调符号和时间轴上接续的下一个符号，根据两个连续符号的可能结果，汇整并分析出可靠度高的判断结果，依据该新的解调技术，降低了解调错误机率，提升GFSK在解调上的精准度(可靠度)，从而提升了实质上的通信效能。
附图说明
下面结合附图和具体实施方式对本发明做更进一步的具体说明，本发明的上述和/或其他方面的优点将会变得更加清楚。
图1为本申请实施例提供的GFSK通信模式内的双比特组解调方法接收端的结构示意图。
图2为本申请实施例提供的GFSK通信模式内的双比特组解调器结构示意图一。
图3为本申请实施例提供的GFSK通信模式内的双比特组解调方法中发送端发送2个相同符号时双比特组内的相位旋转总和示意图。
图4为本申请实施例提供的GFSK通信模式内的双比特组解调方法中发送端发送2个不相同符号时双比特组内的相位旋转总和示意图。
图5为本申请实施例提供的GFSK通信模式内的双比特组解调方法中双比特组内的相位旋转总和升序排序的相对关系示意图。
图6为本申请实施例提供的GFSK通信模式内的双比特组解调方法中双比特组内的相位旋转总和阈值Ψ dbg确定示意图。
图7为本申请实施例提供的GFSK通信模式内的双比特组解调方法中基于前一个符号D n-1符号判决基础上的双符号相位旋转示意图。
图8为本申请实施例提供的GFSK通信模式内的双比特组解调方法中双符号相位旋转的相对关系示意图。
图9为本申请实施例提供的GFSK通信模式内的双比特组解调方法中基于前一个符号D n-1符号判决基础上的双符号相位旋转总和阈值Ψ dbg,p和Ψ dbg,n的确定示意图。
图10为本申请实施例提供的GFSK通信模式内的双比特组解调方法中歧义区间示意图。
图11为本申请实施例提供的GFSK通信模式内的双比特组解调器结构示意图二。
图12为本申请实施例提供的GFSK通信模式内的双比特组解调器结构示意图三。
图13为本申请实施例提供的GFSK通信模式内的双比特组解调方法和现有技术解调方法的信噪比与错误率关系示意图。
具体实施方式
下面将结合附图，对本发明的实施例进行描述。
本申请提供的GFSK通信模式内的双比特组解调方法可以应用于蓝牙芯片和电力线通信芯片，广泛适用于需求低功耗的IoT产品、智能家庭/安防、远传抄表、工业/农业控制器等无线双向应用产品。
FSK调制使用不同的频率来承载比特信息。例如，当BFSK发送一个0比特时，在预定的符号周期内发送符号0对应的载波频率，另一个载波频率则作为比特1的符号发送。在符号周期内，总是发送一个几乎恒定的频率。随着发射比特信息的切换，载波频率也会有着突然的变化，此一快速的变化将带出不需要的谱线，也造成传输带宽的增加以及传输能量的发散。因此，两个频率之间的快速切换不仅增加了设计复杂性，而且降低了频谱效率。
高斯FSK调制(GFSK)数据以FSK编码，是一种FSK的变体形式。使用的调制器与用于FSK调制的调制器相同。然而，脉冲在进入脉冲调制器之前会通过一个高斯滤波器以减小带宽。高斯滤波器是一种时域脉冲整形器，用于平滑连续脉冲值之间的快速变化。高斯低通滤波器的传递 函数为
H(f)＝exp(-α 2f 2)
f表示频率，参数α与基带高斯整形滤波器的3dB带宽B有关。它通常用归一化的3dB带宽-符号时间积BTs表示：
Figure PCTCN2022141995-appb-000021
T s表示符号周期，随着α的增加，高斯滤波器的频谱占用率降低，并且脉冲响应会在相邻符号上扩展开来，导致接收器看到的ISI实质增加。由于传输了多个GFSK符号，前一个符号和下一个符号都会对当前符号产生影响，这被称为ISI。ISI的大小不仅取决于信道，还取决于BT和GFSK调制指数(MI)的精度。如果接收器能充分侦测接收信号的相位，则可移除ISI并且不会造成任何问题。但是，在实际实现中，由于相位未知，ISI往往会影响解调性能。
概念上，使用数字滤波器可以来减少符号序列中的符号间干扰(ISI)，根据滤波后的符号序列执行符号决策，使用符号到比特映像器来直接得到解调的数据比特。一般来说，在接收机端去除ISI的方法可以大致分为匹配滤波器(MF)方法和差分相位方法。匹配滤波器是为接收器提供相位参考并检测来自MF输出的数据来完成解调的工作，这种方法更像是一种相干解调器，它需要更高的计算复杂度，且对相位噪声或MI精度很敏感。而差分方法通过计算连续符号之间的相位差得出符号频率，并且直接对符号频率作进一步的解调处理。
为了提高接收器性能，降低接收器解调错误率，Masamura等人提出了具有非冗余单纠错的MSK差分检测[1]。它遵循卷积纠错码的概念，利用双时隙差分检测器的输出以及传统差分检测器的输出，单个错误可以通过一个简单的电路来纠正，而无需添加冗余位。在[2]中，提出了将非冗余双纠错应用于差分MSK。
[1]T.Masamura,S.Samejima、Y.Morihiro和H.Fuketa，“具有非冗余纠错的MSK差分检测”，IEEE Trans。通信，卷。COM-27，页。912，1979年6月。
[2]T.Masamura，“通过非冗余纠错降低差分MSK的符号间干扰”，IEEE车辆技术汇刊，第一卷。39，1990年2月。
本申请第一实施例提出的GFSK通信模式内的双比特组解调方法是一种新的GFSK解调方法，为GFSK提供了良好的解调性能，包括以下步骤：
步骤1，接收GFSK调制信号并将所述GFSK调制信号转化为复数值基带样本；
步骤2，计算所述复数值基带样本的相位，获得当前待解调符号和下一个待解调符号的相位旋转量总和；
步骤3，确定发送端发送的每个符号的相位旋转量，设置解调区间；
步骤4，根据解调区间获得当前待解调符号。
如图1所示为本实施例接收端的的结构示意图，在接收器端，包括模拟前端和数字前端，模拟前端用于接收GFSK调制信号并转化为数字信号，并通过数字前端进行滤波和载波下变频，获得复数值基带样本。
本实施例中，步骤2包括计算所述复数值基带样本的相位，根据所述复数值基带样本的相位以及符号时序恢复电路估计所有待解调符号的相位旋转量，获得当前待解调符号和下一个待解调符号的相位旋转量总和。再通过双比特组解调器获得待解调符号，双比特组解调器执行步骤3和步骤4。
本实施例中，所述步骤3中确定发送端发送的每个符号S n的相位旋转量φ n包括：发送端一个双比特组包含2个连续的GFSK符号[S n,S n+1]。记当前符号索引为n，索引n+1指的是当前符号之后的下一个符号。也就是说，需要符号n+1来解调当前符号。对于因果系统，在实际上，我们必须直到接收到符号n后才能解调符号n-1。但是为简单起见，保留n和n+1的索引，以非因果系统 的表示进行说明。
假设GFSK的BT＝0.5的有效高斯滤波跨度为3个符号。发送端高斯滤波引入的符号间干扰使得符号S n的相位旋转量φ n受到其前一个符号S n-1和下一个符号S n+1的影响。具体如下表所示：
Figure PCTCN2022141995-appb-000022
在一些实施例中，所述步骤3设置解调区间是根据发送端发送的每个符号S n的相位旋转量φ n以及连续两个符号的相位旋转量总和Φ dbg进行设置，包括：
连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，
考虑发送端发送2个相同符号的情况(即S n＝S n+1)，这包括[S n,S n+1]＝[0,0]和[S n,S n+1]＝[1,1]的情况。双比特组内的相位旋转总和Φ dbg≡φ n+φ n+1可以是Φ dbg∈{±2θ M,±2θ H,±(θ M+θ H)}的值之一，如图3所示。
相反的，在S n≠S n+1的情况下，Φ dbg∈{±(θ M-θ L),0}。可能的组合如图4所示。
将上述Φ dbg的取值按升序排序，获得第一子集{-2θ H,-θ H-θ M,-2θ M}、第二子集{-(θ M-θ L),0,θ M-θ L}和第三子集{2θ M,θ M+θ H,2θ H}，如图5所示。
在接收端，一旦双比特组解调器接收到连续GFSK符号的相位旋转和Φ' dbg，通过将Φ' dbg与解调区间的分区阈值进行比较来确定Φ' dbg属于哪个子集(也就是哪个双比特组)。解调区间的分区阈值的最优设置是将其配置在每个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000023
和第二分区阈值-Ψ dbg，解调区间以-Ψ dbg和Ψ dbg为界，分为三个区间，如图6所示。
所述步骤4根据解调区间获得当前待解调符号D n：
Figure PCTCN2022141995-appb-000024
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在-Ψ dbg≤Φ' dbg≤Ψ dbg这个条件下决定当前解调符号D n的方式。
对于Φ' dbg落在-Ψ dbg≤Φ' dbg≤Ψ dbg这区间的情况，在一种可选的实现方式中，Q(φ' n,φ' n+1)通过比较当前待解调符号的相位旋转量φ' n和下一个待解调符号的相位旋转量φ' n+1之间的符号相位旋转量获得，表达式如下：
Figure PCTCN2022141995-appb-000025
在另一种可选的实现方式中，Q(φ' n,φ' n+1)只考虑当前待解调符号的相位旋转量φ' n，表达式如下：
Figure PCTCN2022141995-appb-000026
在另一些实施例中，所述步骤3中设置解调区间是基于前一个解调符号D n-1、发送端发送的每 个符号S n的相位旋转量φ n以及发送端连续两个符号的相位旋转量总和Φ dbg进行设置，包括：
发送端连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，如图7和图8所示，当D n-1＝0时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-2θ H,-θ H-θ M,-(θ M-θ L),0,2θ M,θ H+θ M}，将上述Φ dbg的取值按升序排序，获得第四子集{-2θ H,-θ H-θ M}、第五子集{-(θ M-θ L),0}和第六子集{2θ M,θ H+θ M}；
当D n-1＝1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-θ H-θ M,-2θ M,0,θ M-θ L,θ H+θ M,2θ H}，将上述Φ dbg的取值按升序排序，获得第七子集{-θ H-θ M,-2θ M}、第八子集{0,θ M-θ L}和第九子集{θ H+θ M,2θ H}；
同样，双比特组分区的最优解是使分区阈值在相邻两个子集的中间。需要注意的是，分区阈值不再围绕0对称，将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000027
和第二分区阈值
Figure PCTCN2022141995-appb-000028
其中
Figure PCTCN2022141995-appb-000029
和
Figure PCTCN2022141995-appb-000030
Figure PCTCN2022141995-appb-000031
如果D n-1为0
Figure PCTCN2022141995-appb-000032
如果D n-1为1
解调区间以
Figure PCTCN2022141995-appb-000033
和
Figure PCTCN2022141995-appb-000034
为界，分为三个区间，如图9所示。
所述步骤4根据解调区间获得当前待解调符号D n：
Figure PCTCN2022141995-appb-000035
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在
Figure PCTCN2022141995-appb-000036
这个条件下决定当前解调符号D n的方式。
对于Φ' dbg落在
Figure PCTCN2022141995-appb-000037
这区间的情况，在一种可选的实现方式中，Q(φ' n,φ' n+1)通过比较当前待解调符号的相位旋转量φ' n和下一个待解调符号的相位旋转量φ' n+1之间的符号相位旋转量获得，表达式如下：
Figure PCTCN2022141995-appb-000038
在另一种可选的实现方式中，Q(φ' n,φ' n+1)只考虑当前待解调符号的相位旋转量φ' n，表达式如下：
Figure PCTCN2022141995-appb-000039
通过利用已知的解调符号n-1的结果(D n-1)，实质上扩展双比特组子集之间的距离。这为双比特组提供了更稳健的决策区间，因此相应地提供了更好的解调性能。
解调性能的提高是利用我们知道符号n-1携带的比特来获得的。但是，符号n-1译码仍存在着可能的错误。如果发生符号n-1的错误解调，当前符号采用的动态双比特分组的分区阈值将不再合适，进而子集判定的阈值变得不正确，可能会导致当前符号n的错误解调。这称为错误传播效应。为了解决这个问题，在另一些实施例中，提出一个额外的歧义检测方案，即如果当前待解调符号和前一个解调符号的相位旋转量总和(双比特组的相位旋转量总和)Φ″ dbg太接近双比特组的分 区阈值，即太接近第一分区阈值
Figure PCTCN2022141995-appb-000040
和第二分区阈值
Figure PCTCN2022141995-appb-000041
则判定前一个解调符号的决策为一个具有歧义的决策，需要对解调区间进行歧义修正以用于当前待解调符号的解调。
所述步骤3中设置解调区间时，若如下表达式为真，则判定第n-1个符号的符号决策为不可靠的决策，需要修正第n个符号的解调区间，所述步骤3中设置解调区间是根据发送端发送的每个符号S n的相位旋转量φ n以及连续两个符号的相位旋转量总和Φ dbg进行设置；若如下表达式为假，则判定第n-1个符号的符号决策为可靠的决策，不需要修正第n个符号的解调区间，即所述步骤3中设置解调区间是基于前一个解调符号D n-1、发送端发送的每个符号S n的相位旋转量φ n以及发送端连续两个符号的相位旋转量总和Φ dbg进行设置。
Figure PCTCN2022141995-appb-000042
其中，Φ″ dbg表示当前待解调符号和前一个解调符号的相位旋转量总和，τ ambi表示歧义区间阈值，决定歧义区间的范围，歧义区间阈值τ ambi的取值范围为0到(θ H-θ L)之间，优选
Figure PCTCN2022141995-appb-000043
Λ表示Φ″ dbg是否落在此歧义区间，如图10所示；
修正解调区间的分区阈值，即将第一分区阈值修正为
Figure PCTCN2022141995-appb-000044
第二分区阈值修正为
Figure PCTCN2022141995-appb-000045
Figure PCTCN2022141995-appb-000046
其中，Ψ ambi＝(3θ M-θ L)/2；解调区间以
Figure PCTCN2022141995-appb-000047
和
Figure PCTCN2022141995-appb-000048
为界，分为三个区间。
在一种可选的实现方式中，所述步骤4根据解调区间获得当前待解调符号D n，即如图2所示的符号决策：
Figure PCTCN2022141995-appb-000049
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在
Figure PCTCN2022141995-appb-000050
这个条件下决定当前解调符号D n的方式，表达式如下：
Figure PCTCN2022141995-appb-000051
其中，φ' n表示当前待解调符号的相位旋转量，φ' n+1表示下一个待解调符号的相位旋转量。
在另一种可选的实现方式中，所述步骤4根据解调区间获得当前待解调符号D n：
Figure PCTCN2022141995-appb-000052
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在
Figure PCTCN2022141995-appb-000053
这个条件下决定当前解调符号D n的方式，表达式如下：
Figure PCTCN2022141995-appb-000054
其中，φ' n表示当前待解调符号的相位旋转量，φ' n+1表示下一个待解调符号的相位旋转量。
本申请第二实施例公开一种GFSK通信模式内的双比特组解调器，如图2所示，包括符号缓冲模块、相加模块和符号决策模块，所述符号缓冲模块，用于存储待解调符号的相位旋转量，
所述相加模块，用于将当前待解调符号的相位旋转量和下一个待解调符号的相位旋转量相加，获得双比特组的相位旋转量总和；
所述符号决策模块，用于根据解调区间获得当前待解调符号D n。
Figure PCTCN2022141995-appb-000055
其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在第二分区阈值≤Φ' dbg≤第一分区阈值这个条件下决定当前解调符号D n的方式，在一种可选的实现方式中，Q(φ' n,φ' n+1)通过比较当前待解调符号的相位旋转量φ' n和下一个待解调符号的相位旋转量φ' n+1之间的符号相位旋转量获得，表达式如下：
Figure PCTCN2022141995-appb-000056
在另一种可选的实现方式中，Q(φ' n,φ' n+1)只考虑当前待解调符号的相位旋转量φ' n，表达式如下：
Figure PCTCN2022141995-appb-000057
进一步地，所述解调区间的分区阈值是根据发送端发送的每个符号的相位旋转量以及连续两个符号的相位旋转量总和进行设置的，如图6所示。
将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000058
Figure PCTCN2022141995-appb-000059
和第二分区阈值-Ψ dbg，解调区间以-Ψ dbg和Ψ dbg为界，分为三个区间。
进一步地，所述解调区间的分区阈值是基于前一个解调符号、发送端发送的每个符号的相位旋转量以及发送端连续两个符号的相位旋转量总和进行设置的，如图9所示；
将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
Figure PCTCN2022141995-appb-000060
和第二分区阈值
Figure PCTCN2022141995-appb-000061
其中
Figure PCTCN2022141995-appb-000062
和
Figure PCTCN2022141995-appb-000063
Figure PCTCN2022141995-appb-000064
如果D n-1为0
Figure PCTCN2022141995-appb-000065
如果D n-1为1
解调区间以
Figure PCTCN2022141995-appb-000066
和
Figure PCTCN2022141995-appb-000067
为界，分为三个区间。
如图11所示，所述双比特组解调器还包括前一符号决策缓冲模块和动态阈值决定模块，所述前一符号决策缓冲模块，用于存储前一个解调符号，
所述动态阈值决定模块，用于根据前一个解调符号，选择解调区间的分区阈值。
进一步地，如图12所示，所述双比特组解调器还包括前一符号歧义缓冲模块和歧义修正模块，所述前一符号歧义缓冲模块，用于存储当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间的结果；
所述歧义修正模块，用于判断当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间，并将结果存储至前一符号歧义缓冲模块，以及根据结果修正当前待解调符号的解调区间的分区阈值。
Figure PCTCN2022141995-appb-000068
其中，Φ″ dbg表示当前待解调符号和前一个解调符号的相位旋转量总和，τ ambi表示歧义区间阈值，决定歧义区间的范围，歧义区间阈值τ ambi的取值范围为0到(θ H-θ L)之间，优选
Figure PCTCN2022141995-appb-000069
Λ表示Φ″ dbg是否落在此歧义区间，如图10所示；
修正解调区间的分区阈值，即将第一分区阈值修正为
Figure PCTCN2022141995-appb-000070
第二分区阈值修正为
Figure PCTCN2022141995-appb-000071
Figure PCTCN2022141995-appb-000072
其中，Ψ ambi＝(3θ M-θ L)/2；解调区间以
Figure PCTCN2022141995-appb-000073
和
Figure PCTCN2022141995-appb-000074
为界，分为三个区间。
如图13所示，在相同信噪比(SNR)下本申请实施例提供的GFSK通信模式内的双比特组解调方法及解调器比现有技术单个符号解调方法及解调器的误码率(BER)低，即在同样的通道条件下，本申请实施例提供的双比特组解调方法及解调器比较不容易出错；且当信噪比逐渐增加，本申请实施例提供的双比特组解调方法及解调器的误码率低于误码率目标。因此，本申请实施例提供的GFSK通信模式内的双比特组解调方法及解调器降低了解调错误机率，提升GFSK在解调上的精准度(可靠度)，从而提升了实质上的通信效能。
具体实现中，本申请提供计算机存储介质以及对应的数据处理单元，其中，该计算机存储介质能够存储计算机程序，所述计算机程序通过数据处理单元执行时可运行本发明提供的一种GFSK通信模式内的双比特组解调方法的发明内容以及各实施例中的部分或全部步骤。所述的存储介质可为磁碟、光盘、只读存储记忆体(read-only memory，ROM)或随机存储记忆体(random access memory，RAM)等。
本领域的技术人员可以清楚地了解到本发明实施例中的技术方案可借助计算机程序以及其对应的通用硬件平台的方式来实现。基于这样的理解，本发明实施例中的技术方案本质上或者说对现有技术做出贡献的部分可以以计算机程序即软件产品的形式体现出来，该计算机程序软件产品可以存储在存储介质中，包括若干指令用以使得一台包含数据处理单元的设备(可以是个人计算机，服务器，单片机，MUU或者网络设备等)执行本发明各个实施例或者实施例的某些部分所述的方法。
本发明提供了一种GFSK通信模式内的双比特组解调方法及解调器，具体实现该技术方案的方法和途径很多，以上所述仅是本发明的具体实施方式，应当指出，对于本技术领域的普通技术人员来说，在不脱离本发明原理的前提下，还可以做出若干改进和润饰，这些改进和润饰也应视为本发明的保护范围。本实施例中未明确的各组成部分均可用现有技术加以实现。

Claims (11)

  1. 一种GFSK通信模式内的双比特组解调方法，其特征在于，包括以下步骤：
    步骤1，接收GFSK调制信号并将所述GFSK调制信号转化为复数值基带样本；
    步骤2，计算所述复数值基带样本的相位，获得当前待解调符号和下一个待解调符号的相位旋转量总和；
    步骤3，确定发送端发送的每个符号的相位旋转量，设置解调区间；
    步骤4，根据解调区间获得当前待解调符号。
  2. 根据权利要求1所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤3中确定发送端发送的每个符号S n的相位旋转量φ n包括：发送端高斯滤波引入的符号间干扰使得符号S n的相位旋转量φ n受到其前一个符号S n-1和下一个符号S n+1的影响，
    当S n≠S n-1，S n≠S n+1时，符号S n的相位旋转量φ n为±θ L；
    当S n＝S n-1，S n≠S n+1或者S n≠S n-1，S n＝S n+1时，符号S n的相位旋转量φ n为±θ M；
    当S n＝S n-1＝S n+1时，符号S n的相位旋转量φ n为±θ H；
    其中，θ H、θ M和θ L由GFSK调制指数和带宽符号时间积确定，且θ H>θ M>θ L。
  3. 根据权利要求2所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤3中设置解调区间是根据发送端发送的每个符号S n的相位旋转量φ n以及连续两个符号的相位旋转量总和Φ dbg进行设置，包括：
    连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，当S n＝S n+1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{±2θ M,±2θ H,±(θ M+θ H)}；
    当S n≠S n+1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{±(θ M-θ L),0}；
    将上述Φ dbg的取值按升序排序，获得第一子集{-2θ H,-θ H-θ M,-2θ M}、第二子集{-(θ M-θ L),0,θ M-θ L}和第三子集{2θ M,θ M+θ H,2θ H}；
    将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
    Figure PCTCN2022141995-appb-100001
    Figure PCTCN2022141995-appb-100002
    和第二分区阈值-Ψ dbg，解调区间以-Ψ dbg和Ψ dbg为界，分为三个区间。
  4. 根据权利要求2所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤3中设置解调区间是基于前一个解调符号D n-1、发送端发送的每个符号S n的相位旋转量φ n以及发送端连续两个符号的相位旋转量总和Φ dbg进行设置，包括：
    发送端连续两个符号的相位旋转量总和Φ dbg≡φ n+φ n+1，当D n-1＝0时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-2θ H,-θ H-θ M,-(θ M-θ L),0,2θ M,θ H+θ M}，将上述Φ dbg的取值按升序排序，获得第四子集{-2θ H,-θ H-θ M}、第五子集{-(θ M-θ L),0}和第六子集{2θ M,θ H+θ M}；
    当D n-1＝1时，根据[S n-1,S n,S n+1,S n+2]的所有组合获得Φ dbg∈{-θ H-θ M,-2θ M,0,θ M-θ L,θ H+θ M,2θ H}，将上述Φ dbg的取值按升序排序，获得第七子集{-θ H-θ M,-2θ M}、第八子集{0,θ M-θ L}和第九子集{θ H+θ M,2θ H}；
    将解调区间的分区阈值配置在相邻两个子集之间的中间值，获得第一分区阈值
    Figure PCTCN2022141995-appb-100003
    和第二分区阈值
    Figure PCTCN2022141995-appb-100004
    其中
    Figure PCTCN2022141995-appb-100005
    和
    Figure PCTCN2022141995-appb-100006
    Figure PCTCN2022141995-appb-100007
    如果D n-1为0
    Figure PCTCN2022141995-appb-100008
    如果D n-1为1
    解调区间以
    Figure PCTCN2022141995-appb-100009
    和
    Figure PCTCN2022141995-appb-100010
    为界，分为三个区间。
  5. 根据权利要求4所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤3中设置解调区间时，若如下表达式为真，
    Figure PCTCN2022141995-appb-100011
    其中，Φ″ dbg表示当前待解调符号和前一个解调符号的相位旋转量总和，τ ambi表示歧义区间阈值，决定歧义区间的范围，Λ表示Φ″ dbg是否落在此歧义区间；
    则修正解调区间的分区阈值，将第一分区阈值修正为
    Figure PCTCN2022141995-appb-100012
    第二分区阈值修正为
    Figure PCTCN2022141995-appb-100013
    Figure PCTCN2022141995-appb-100014
    其中，Ψ ambi＝(3θ M-θ L)/2；解调区间以
    Figure PCTCN2022141995-appb-100015
    和
    Figure PCTCN2022141995-appb-100016
    为界，分为三个区间。
  6. 根据权利要求3、4或5任一项所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤4根据解调区间获得当前待解调符号D n：
    Figure PCTCN2022141995-appb-100017
    其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在第二分区阈值≤Φ' dbg≤第一分区阈值这个条件下决定当前解调符号D n的方式，表达式如下：
    Figure PCTCN2022141995-appb-100018
    其中，φ' n表示当前待解调符号的相位旋转量，φ' n+1表示下一个待解调符号的相位旋转量。
  7. 根据权利要求3、4或5任一项所述的一种GFSK通信模式内的双比特组解调方法，其特征在于，所述步骤4根据解调区间获得当前待解调符号D n：
    Figure PCTCN2022141995-appb-100019
    其中，Φ' dbg表示当前待解调符号和下一个待解调符号的相位旋转量总和，Q(φ' n,φ' n+1)表示在第二分区阈值≤Φ' dbg≤第一分区阈值这个条件下决定当前解调符号D n的方式，表达式如下：
    Figure PCTCN2022141995-appb-100020
    其中，φ' n表示当前待解调符号的相位旋转量，φ' n+1表示下一个待解调符号的相位旋转量。
  8. 一种GFSK通信模式内的双比特组解调器，其特征在于，包括符号缓冲模块、相加模块和符号决策模块，所述符号缓冲模块，用于存储待解调符号的相位旋转量，
    所述相加模块，用于将当前待解调符号的相位旋转量和下一个待解调符号的相位旋转量相加， 获得双比特组的相位旋转量总和；
    所述符号决策模块，用于根据解调区间获得当前待解调符号。
  9. 根据权利要求8所述的一种GFSK通信模式内的双比特组解调器，其特征在于，所述解调区间的分区阈值是根据发送端发送的每个符号的相位旋转量以及连续两个符号的相位旋转量总和进行设置的。
  10. 根据权利要求8所述的一种GFSK通信模式内的双比特组解调器，其特征在于，所述解调区间的分区阈值是基于前一个解调符号、发送端发送的每个符号的相位旋转量以及发送端连续两个符号的相位旋转量总和进行设置的；
    所述双比特组解调器还包括前一符号决策缓冲模块和动态阈值决定模块，所述前一符号决策缓冲模块，用于存储前一个解调符号，
    所述动态阈值决定模块，用于根据前一个解调符号，选择解调区间的分区阈值。
  11. 根据权利要求10所述的一种GFSK通信模式内的双比特组解调器，其特征在于，还包括前一符号歧义缓冲模块和歧义修正模块，所述前一符号歧义缓冲模块，用于存储当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间的结果；
    所述歧义修正模块，用于判断当前待解调符号和前一个解调符号的相位旋转量总和是否落在歧义区间，并将结果存储至前一符号歧义缓冲模块，以及根据结果修正当前待解调符号的解调区间的分区阈值。
PCT/CN2022/141995 2022-10-08 2022-12-26 Gfsk通信模式内的双比特组解调方法及解调器 WO2024073955A1 (zh)

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Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20040218696A1 (en) * 2003-04-29 2004-11-04 Texas Instruments Incorporated Gaussian frequency shift keying digital demodulator
CN101047677A (zh) * 2006-03-31 2007-10-03 捷顶微电子（上海）有限公司 一种低复杂度、高性能的gfsk信号多比特解调法
CN104702550A (zh) * 2015-03-12 2015-06-10 灵芯微电子科技（苏州）有限公司 用于fsk调制系统的数字检测纠错算法
CN104980177A (zh) * 2015-06-12 2015-10-14 清华大学 一种用于零中频gfsk解调器中的位同步电路
CN105812303A (zh) * 2016-03-15 2016-07-27 苏州卓智创芯电子科技有限公司 一种gfsk基带数字接收机及其基带同步及解调方法
CN108141420A (zh) * 2015-09-23 2018-06-08 高通股份有限公司 用于gfsk中的联合解调和解映射的系统和方法

Family Cites Families (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1439658A1 (en) * 2003-01-17 2004-07-21 Telefonaktiebolaget LM Ericsson (publ) A signal processing apparatus and method for decision directed symbol synchronisation
US8625722B2 (en) * 2010-07-30 2014-01-07 Sensus Usa Inc. GFSK receiver architecture and methodology
CN104935538B (zh) * 2015-06-17 2018-02-27 江苏卓胜微电子股份有限公司 低复杂度的gfsk符号间干扰抵消处理方法及装置
CN108989256B (zh) * 2018-09-04 2021-03-19 泰凌微电子(上海)股份有限公司 一种fsk/gfsk解调方法及装置
CN113507296B (zh) * 2021-09-13 2022-01-11 北京思凌科半导体技术有限公司 通信方法、装置、存储介质及电子设备
CN114640562B (zh) * 2022-03-16 2023-05-30 中山大学 一种cpfsk/gfsk信号非相干解调方法

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20040218696A1 (en) * 2003-04-29 2004-11-04 Texas Instruments Incorporated Gaussian frequency shift keying digital demodulator
CN101047677A (zh) * 2006-03-31 2007-10-03 捷顶微电子（上海）有限公司 一种低复杂度、高性能的gfsk信号多比特解调法
CN104702550A (zh) * 2015-03-12 2015-06-10 灵芯微电子科技（苏州）有限公司 用于fsk调制系统的数字检测纠错算法
CN104980177A (zh) * 2015-06-12 2015-10-14 清华大学 一种用于零中频gfsk解调器中的位同步电路
CN108141420A (zh) * 2015-09-23 2018-06-08 高通股份有限公司 用于gfsk中的联合解调和解映射的系统和方法
CN105812303A (zh) * 2016-03-15 2016-07-27 苏州卓智创芯电子科技有限公司 一种gfsk基带数字接收机及其基带同步及解调方法

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