WO2022056761A1 - 一种光伏系统、谐振开关电容变换器及控制方法 - Google Patents

一种光伏系统、谐振开关电容变换器及控制方法 Download PDF

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Publication number
WO2022056761A1
WO2022056761A1 PCT/CN2020/115801 CN2020115801W WO2022056761A1 WO 2022056761 A1 WO2022056761 A1 WO 2022056761A1 CN 2020115801 W CN2020115801 W CN 2020115801W WO 2022056761 A1 WO2022056761 A1 WO 2022056761A1
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Prior art keywords
rscc
current
angle
phase
bridge arm
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PCT/CN2020/115801
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English (en)
French (fr)
Inventor
王朝辉
王均
石磊
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华为数字能源技术有限公司
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Application filed by 华为数字能源技术有限公司 filed Critical 华为数字能源技术有限公司
Priority to AU2020468167A priority Critical patent/AU2020468167A1/en
Priority to EP20953619.2A priority patent/EP4160896A4/en
Priority to CN202080013591.XA priority patent/CN114586270B/zh
Priority to PCT/CN2020/115801 priority patent/WO2022056761A1/zh
Priority to BR112023002411A priority patent/BR112023002411A2/pt
Publication of WO2022056761A1 publication Critical patent/WO2022056761A1/zh
Priority to US18/185,976 priority patent/US20230231481A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0095—Hybrid converter topologies, e.g. NPC mixed with flying capacitor, thyristor converter mixed with MMC or charge pump mixed with buck
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/06—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
    • H02M3/07—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • H02M3/1586—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel switched with a phase shift, i.e. interleaved
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E10/00—Energy generation through renewable energy sources
    • Y02E10/50—Photovoltaic [PV] energy
    • Y02E10/56—Power conversion systems, e.g. maximum power point trackers

Definitions

  • the present application relates to the technical field of photovoltaic power generation, and in particular, to a photovoltaic system, a resonant switched capacitor converter and a control method.
  • SCC switched capacitor circuits
  • the current SCC works in an open-loop control mode, and therefore has poor flexibility.
  • the current of each SCC is generally uncontrollable.
  • the present application provides a photovoltaic system, a resonant switched capacitor converter and a control method, which can ensure current sharing among multiple SCC circuits.
  • the embodiments of the present application provide a photovoltaic power generation system, including: a DC/DC converter, a resonant switched capacitor converter, an inverter and a controller; an input end of the DC/DC converter is connected to a photovoltaic array; The first input end is connected to the positive output end of the DC/DC converter; the second input end of the resonant switched capacitor converter is connected to the negative output end of the DC/DC converter; the first output end of the resonant switched capacitor converter The neutral line of the inverter is connected, and the second output end of the resonant switched capacitor converter is connected to the negative bus of the inverter.
  • the resonant switched capacitor converter is used to provide a negative voltage required by the inverter between the neutral line and the negative input terminal of the inverter, and the resonant switched capacitor converter realizes the conversion of DC voltage to DC voltage.
  • DC converters and resonant switched capacitor converters have high power conversion efficiency.
  • the capacitor and inductor in the resonant switched capacitor converter are connected in series to form an LC resonant circuit.
  • the resonant switched capacitor converter includes at least two RSCCs connected in parallel, and adjusts the phase shift angle between the corresponding driving signals of the two RSCCs according to the current difference of the two RSCCs, thereby achieving equal currents of the two RSCCs, that is, current sharing.
  • the start-up time of the resonant cavity of the LC resonant circuit can be changed, and different start-up times cause the voltage difference between the input and output filter capacitor voltages to be different, so that the currents of the two RSCCs can be consistent.
  • Current sharing control makes the energy of each RSCC fully utilized, and avoids damage to a certain RSCC circuit due to overloading. Since the solution is to adjust the driving signal between the two independent RSCCs for phase shifting, it does not affect the soft switching characteristics of the switch tube in the single RSCC, thereby reducing switch damage and improving power conversion efficiency.
  • the resonant switched capacitor converter includes a parallel multi-channel resonant switched capacitor converter RSCC, such as at least two parallel RSCCs: a first RSCC and a second RSCC; the controller is based on the first current of the first RSCC and the second RSCC of the second RSCC.
  • the current difference of the current adjust the phase shift angle between the first driving signal of the first RSCC and the second driving signal of the second RSCC, so that the first current is consistent with the second current, that is, the control of the two RSCCs
  • the currents are equal to realize current sharing among multiple RSCCs connected in parallel.
  • the first current of the first RSCC can be obtained by measuring the current of the LC resonance circuit of the first RSCC, and similarly, the second current of the second RSCC can be obtained by measuring the current of the LC resonance circuit of the second RSCC.
  • the phase shift angle is positively correlated with the current difference, that is, the larger the current difference between the two RSCCs, the larger the phase shift angle between the driving signals corresponding to the two RSCCs.
  • closed-loop adjustment of the current difference can be performed to realize the adjustment of the phase shift angle, thereby realizing that the currents of the two RSCCs are equal.
  • the current difference between the first current and the second current can be obtained, and the current difference can be adjusted by proportional and integral PI to obtain a dynamically adjustable angle in the phase shift angle, and the dynamic adjustable angle is positively correlated with the difference.
  • the specific phase shift angle can be generated by using the phase shift angle generator according to the result of PI adjustment.
  • the phase shift angle generator can be realized by changing the initial value of the carrier, or by adjusting the value of the comparison value. In this embodiment is not limited.
  • the controller adjusts the phase of at least one of the first drive signal and the second drive signal to adjust the phase shift angle between the first drive signal and the second drive signal.
  • the controller only adjusts the phase of the first driving signal, and the phase of the second driving signal is fixed to adjust the phase shift angle.
  • the controller only adjusts the phase of the second driving signal, and the phase of the first driving signal is fixed to adjust the phase shift angle.
  • the controller can also adjust the phases of the first driving signal and the second driving signal to move in opposite directions respectively, so as to realize the adjustment of the phase shift angle. This embodiment does not limit a specific phase shift manner.
  • the phase shift angle between the two may be 0, that is, the drive signals of the two RSCCs are controlled in phase.
  • the phase shift angle is the sum of the preset fixed angle and the dynamic adjustable angle, and the preset fixed angle is 0; in this case, the phase shift angle is equal to the dynamic adjustable angle, and the controller adjusts the dynamic adjustable angle according to the current difference Adjustable angle to adjust the phase shift angle.
  • the controller when the phase shift angle is equal to the dynamic adjustable angle, the controller, when the second current is smaller than the first current, is specifically configured to control the phase of the second drive signal to lead the phase of the first drive signal
  • the dynamically adjustable angle of the phase when the second current is greater than the first current, it is specifically used to control the phase of the second drive signal to lag the phase of the first drive signal by the dynamically adjustable angle.
  • the above describes the situation that the driving signals of the controllable switches at the same position on the first bridge arm and the third bridge arm are controlled in the same phase when they are not phase-shifted.
  • the following describes the phases on the first bridge arm and the third bridge arm.
  • the case where the driving signals of the controllable switch tubes of the position are interleaved. Since the switches in the two RSCCs are controlled by interleaving, the interleaving control can effectively reduce the current on the input filter capacitor and the output filter capacitor. Therefore, a smaller filter capacitor can be used to reduce the volume occupied by the filter capacitor.
  • the phase shift angle is the sum of a preset fixed angle and a dynamically adjustable angle
  • the preset fixed angle is 360°/N, where N is the number of the RSCCs connected in parallel, and N is an integer greater than 1
  • control The controller adjusts the phase shift angle by adjusting the dynamic adjustable angle on the basis of the preset fixed angle according to the current difference. For example, when two RSCCs are connected in parallel, before the phase shift angle adjustment of the corresponding driving signals of the two RSCCs, the phase shift angle between the two driving signals is 180 degrees. On the basis of the phase shift angle, the dynamic adjustable angle is adjusted to realize the equal current of the two RSCCs.
  • the controller when the driving signals corresponding to the multi-channel RSCCs are controlled by interleaving, the controller, when the second current is smaller than the first current, is specifically configured to control the phase of the second driving signal to lag the first driving signal the dynamically adjustable angle of the phase; when the second current is greater than the first current, it is specifically used to control the phase of the second drive signal to lead the phase of the first drive signal by the dynamically adjustable angle .
  • N channels of RSCC When N channels of RSCC are connected in parallel, it is necessary to detect the current of the resonant inductors of each channel of RSCC, and obtain the current average value of the N channels of RSCC circuits through arithmetic averaging, that is, the controller obtains the current average value of the resonant circuits of the N channels of RSCC circuits; fixed; For the phase of the driving signal of one of the RSCC circuits, the currents of the remaining N-1 resonant circuits are compared with the current average value, and the respective dynamic adjustable angles are obtained according to the respective comparison results.
  • the dynamically adjustable angle shifts the phase of its drive signal. That is, the N-1 channel RSCC performs closed-loop control according to the difference between the current of its own resonant inductance and the average value of the current, so as to realize the current sharing control between the N channels of RSCC.
  • the phase of the driving signal of one channel of RSCC can continue to be fixed, and the phase-shift control of the driving signals of the remaining N-1 channels of RSCC can be performed.
  • the current of the resonant circuit is compared with the average current value, and the corresponding difference value of each channel is obtained, and the corresponding closed-loop control is performed on each channel according to the difference value of each channel, that is, by dynamically adjusting the drive in RSCC-B to RSCC-N
  • the dynamic adjustable angle of the signal realizes the current sharing control between each RSCC.
  • the dynamic adjustable angle when the dynamic adjustable angle is increased to a certain extent, the current difference of the two RSCCs basically reaches a limit value. If the dynamic adjustable angle is further increased, the currents between the two RSCCs may change in opposite directions, resulting in The control appears non-monotonic, and thus the ability to control is lost. Therefore, in practical applications, the dynamic adjustable angle can be limited, that is, the maximum value of the dynamic adjustable angle needs to be limited.
  • the controller is also used to control the phase difference between the preset fixed angle and the preset threshold when the dynamic adjustable angle is greater than the preset threshold angle
  • the sum of the angles, the preset threshold angle is the maximum upper limit value of the preset dynamic adjustable angle.
  • the controller is further configured to control the dynamically adjustable angle to be the preset threshold angle when the dynamically adjustable angle is greater than a preset threshold angle.
  • the preset threshold angle is less than or equal to 30°.
  • the preset threshold angle is less than or equal to 15°.
  • the embodiments of the present application do not specifically limit the specific position of the LC resonant circuit, and the specific architectures of the resonant switched capacitor converters corresponding to two different LC resonant circuits are provided below:
  • the first RSCC includes: a first bridge arm, a second bridge arm and a first LC resonance circuit;
  • the second RSCC includes: a third bridge arm, a fourth bridge arm and a second LC resonance circuit; the first bridge arm The first end of the first bridge arm and the first end of the third bridge arm are both connected to the first input end of the resonant switched capacitor converter, the second end of the first bridge arm and the second end of the third bridge arm The terminals are both connected to the second input terminal of the resonant switched capacitor converter; the first terminal of the second bridge arm and the first terminal of the fourth bridge arm are both connected to the first output of the resonant switched capacitor converter The second end of the second bridge arm and the second end of the fourth bridge arm are both connected to the second output end of the resonant switched capacitor converter; the first LC resonant circuit is connected to the second end of the first LC resonant circuit. Between the midpoint of a bridge arm and the midpoint of the second bridge arm, the second LC resonance circuit is connected
  • the first RSCC includes: a first bridge arm, a second bridge arm and a first LC resonance circuit;
  • the second RSCC includes: a third bridge arm, a fourth bridge arm and a second LC resonance circuit; the first bridge arm The first end of the third bridge arm and the first end of the third bridge arm are both connected to the first input end of the resonant switched capacitor converter, and the second end of the first bridge arm is connected to the first end of the second bridge arm.
  • the second end of the third bridge arm is connected to the first end of the fourth bridge arm, and both the second end of the second bridge arm and the second end of the fourth bridge arm are connected to the resonance the second output end of the switched capacitor converter;
  • the resonant capacitor of the first LC resonant circuit is connected between the midpoint of the first bridge arm and the midpoint of the second bridge arm, and the second LC resonates
  • the resonant capacitor of the circuit is connected between the midpoint of the third bridge arm and the midpoint of the fourth bridge arm;
  • the resonant inductance of the first LC resonant circuit is connected to the second end of the first bridge arm and the second input end of the resonant switched capacitor converter;
  • the resonant inductance of the second LC resonant circuit is connected to the second end of the third bridge arm and the second input of the resonant switched capacitor converter between the ends.
  • the switching devices of all bridge arms are controllable switching transistors, that is, the first bridge arm includes at least a series of first bridge arms. a switch tube and a second switch tube, the third bridge arm at least includes a third switch tube and a fourth switch tube connected in series, and the second bridge arm at least includes a fifth switch tube and a sixth switch tube connected in series; the The fourth bridge arm includes at least a seventh switch tube and an eighth switch tube connected in series;
  • the second bridge arm and the fourth bridge arm may include diodes. That is, the first bridge arm includes the first switch tube and the second switch tube connected in series, the third bridge arm includes the third switch tube and the fourth switch tube connected in series, and the second bridge arm at least includes the first switch tube and the fourth switch tube connected in series. a diode and a second diode, and the fourth bridge arm includes at least a third diode and a fourth diode connected in series.
  • An embodiment of the present application provides a resonant switched capacitor converter, including a controller and at least two of the following resonant switched capacitor circuits RSCC connected in parallel: a first RSCC and a second RSCC; a first input of the resonant switched capacitor converter The terminal is connected to the positive output terminal of the DC power supply; the second input terminal of the resonant switched capacitor converter is connected to the negative output terminal of the DC power supply; the resonant switched capacitor converter is used to convert the voltage of the DC power supply
  • the controller is configured to adjust the first drive signal of the first RSCC and the first drive signal of the first RSCC according to the current difference between the first current of the first RSCC and the second current of the second RSCC The phase shift angle between the second driving signals of the two RSCCs, so that the first current is consistent with the second current.
  • the first current of the first RSCC can be obtained by measuring the current of the LC resonance circuit of the first RSCC
  • the second current of the second RSCC can be obtained by measuring the current of the LC resonance circuit of the second RSCC.
  • the resonant switched capacitor converter can be applied to the photovoltaic field, and can also be applied to other scenarios. For example, other scenarios that require 1:1 voltage conversion.
  • the DC power supply can be the output voltage of the previous-stage DC/DC converter, and the input end of the previous-stage DC/DC converter is connected to the photovoltaic array.
  • the capacitor and inductor in the resonant switched capacitor converter are connected in series to form an LC resonant circuit.
  • the resonant switched capacitor converter includes at least two RSCCs connected in parallel, and adjusts the phase shift angle between the corresponding driving signals of the two RSCCs according to the current difference of the two RSCCs, thereby achieving equal currents of the two RSCCs, that is, current sharing.
  • the start-up time of the resonant cavity of the LC resonant circuit can be changed, and different start-up times cause the voltage difference between the input and output filter capacitor voltages to be different, so that the currents of the two RSCCs can be consistent.
  • Current sharing control makes the energy of each RSCC fully utilized, and avoids damage to a certain RSCC circuit due to overloading. Since the solution is to adjust the driving signal between the two independent RSCCs for phase shifting, it does not affect the soft switching characteristics of the switch tube in the single RSCC, thereby reducing switch damage and improving power conversion efficiency.
  • the phase shift angle is positively correlated with the current difference, that is, the larger the current difference between the two RSCCs, the larger the phase shift angle between the driving signals corresponding to the two RSCCs.
  • closed-loop adjustment of the current difference can be performed to realize the adjustment of the phase shift angle, thereby realizing that the currents of the two RSCCs are equal.
  • a controller is specifically configured to adjust the phase shift angle between the first driving signal and the second driving signal according to the current difference, so that the first current and the second The currents are consistent; the phase shift angle is positively correlated with the current difference.
  • the controller is specifically configured to adjust the phase of at least one of the first drive signal and the second drive signal to adjust the phase shift angle.
  • the phase shift angle is the sum of a preset fixed angle and a dynamically adjustable angle, and the preset fixed angle is 0; the controller is specifically configured to adjust the dynamically adjustable angle according to the current difference angle to adjust the phase shift angle.
  • the controller when the second current is smaller than the first current, the controller is specifically configured to control the phase of the second drive signal to lead the phase of the first drive signal by the dynamically adjustable angle;
  • the second current is specifically used to control the phase of the second driving signal to lag the phase of the first driving signal by the dynamically adjustable angle.
  • the phase shift angle is the sum of a preset fixed angle and a dynamically adjustable angle
  • the preset fixed angle is 360°/N, where N is the number of the RSCCs connected in parallel, and the N is greater than 1
  • the controller is specifically configured to adjust the phase shift angle by adjusting the dynamic adjustable angle on the basis of the preset fixed angle according to the current difference.
  • the controller when the second current is smaller than the first current, the controller is specifically configured to control the phase of the second drive signal to lag the phase of the first drive signal by the dynamically adjustable angle;
  • the second current is specifically used to control the phase of the second driving signal to lead the phase of the first driving signal by the dynamic adjustable angle.
  • the controller controls the dynamically adjustable angle to be the preset threshold angle.
  • the embodiments of the present application also provide a current sharing control method, which is applied to a photovoltaic system, where the photovoltaic system includes: a DC/DC converter, a resonant switched capacitor converter, and an inverter; an input end of the DC/DC converter connected to the photovoltaic array; the first input end of the resonant switched capacitor converter is connected to the positive output end of the DC/DC converter; the second input end of the resonant switched capacitor converter is connected to the DC/DC converter negative output terminal; the first output terminal of the resonant switched capacitor converter is connected to the neutral line of the inverter, and the second output terminal of the resonant switched capacitor converter is connected to the negative bus of the inverter; the resonant switched capacitor converter
  • the switched capacitor converter includes the following at least two resonant switched capacitor circuits RSCC connected in parallel: a first RSCC and a second RSCC; the method includes: obtaining a first current of the first RSCC, obtaining a first current of
  • the switching devices on each bridge arm can be all controllable switches, and when all are controllable switches, the bidirectional movement of energy can be realized, that is, the energy transfer can be realized from the input end to the output end, or the energy can be transferred from the output end. Energy transfer to the input. If it is a unidirectional energy transfer, the switching devices on the second bridge arm and the fourth bridge arm can be diodes, that is, uncontrollable devices, which can be unidirectionally conductive.
  • the first current of the first RSCC can be obtained by measuring the current of the LC resonance circuit of the first RSCC, and similarly, the second current of the second RSCC can be obtained by measuring the current of the LC resonance circuit of the second RSCC.
  • the phase shift angle is positively correlated with the current difference, that is, the greater the current difference between the two RSCCs, the greater the phase shift angle between the driving signals corresponding to the two RSCCs.
  • closed-loop adjustment of the current difference can be performed to realize the adjustment of the phase shift angle, thereby realizing that the currents of the two RSCCs are equal.
  • the capacitor and inductor in the resonant switched capacitor converter are connected in series to form an LC resonant circuit.
  • the resonant switched capacitor converter includes at least two RSCCs connected in parallel, and adjusts the phase shift angle between the corresponding driving signals of the two RSCCs according to the current difference of the two RSCCs, thereby achieving equal currents of the two RSCCs, that is, current sharing.
  • the start-up time of the resonant cavity of the LC resonant circuit can be changed, and different start-up times cause the voltage difference between the input and output filter capacitor voltages to be different, so that the currents of the two RSCCs can be consistent.
  • Current sharing control makes the energy of each RSCC fully utilized, and avoids damage to a certain RSCC circuit due to overloading. Since the solution is to adjust the driving signal between the two independent RSCCs for phase shifting, it does not affect the soft switching characteristics of the switch tube in the single RSCC, thereby reducing switch damage and improving power conversion efficiency.
  • the phase shift angle is the sum of a preset fixed angle and a dynamically adjustable angle, and the preset fixed angle is 0; adjust the first driving signal of the first RSCC and the second driving signal of the second RSCC
  • the phase shift angle between the drive signals specifically includes: adjusting the dynamically adjustable angle between the first drive signal of the first RSCC and the second drive signal of the second RSCC.
  • the adjusting the dynamically adjustable angle between the first driving signal of the first RSCC and the second driving signal of the second RSCC specifically includes: when the second current is smaller than the first When the current is one current, adjust the phase of the second driving signal to advance the phase of the first driving signal by the dynamic adjustable angle; when the second current is greater than the first current, adjust the second driving signal The phase lags the phase of the first drive signal by the dynamically adjustable angle.
  • the phase shift angle is the sum of a preset fixed angle and a dynamically adjustable angle
  • the preset fixed angle is 360°/N, where N is the number of the RSCCs connected in parallel, and the N is greater than 1
  • Adjusting the phase shift angle between the first drive signal of the first RSCC and the second drive signal of the second RSCC specifically includes: on the basis of the preset fixed angle, adjusting the first The phase shift angle is adjusted by the dynamically adjustable angle between the first drive signal of an RSCC and the second drive signal of the second RSCC.
  • the adjusting the phase shift angle by adjusting the dynamic adjustable angle between the first driving signal of the first RSCC and the second driving signal of the second RSCC specifically includes: when When the second current is less than the first current, adjusting the phase of the second driving signal to lag the phase of the first driving signal by the dynamic adjustable angle; when the second current is greater than the first current When , the phase of the second drive signal is adjusted to lead the phase of the first drive signal by the dynamically adjustable angle.
  • the method further includes: when the dynamically adjustable angle is greater than a preset threshold angle, controlling the dynamically adjustable angle to be the preset threshold angle.
  • the embodiments of the present application have the following advantages:
  • the photovoltaic system includes a resonant switched capacitor converter.
  • the resonant switched capacitor converter is connected between the output end of the ordinary DC/DC converter and the input end of the inverter, and is generally connected between the output end of the DC/DC converter and the inverter. Before the neutral line and negative bus bar of the inverter, it is used to convert the output voltage of the DC/DC converter into negative voltage and provide the neutral line and negative bus bar of the inverter to realize voltage conversion.
  • the capacitor and inductor in the resonant switched capacitor converter are connected in series to form an LC resonant circuit.
  • the resonant switched capacitor converter includes at least two RSCCs connected in parallel, and adjusts the phase shift angle between the corresponding driving signals of the two RSCCs according to the current difference of the two RSCCs, thereby achieving equal currents of the two RSCCs, that is, current sharing.
  • the driving signal of the RSCC is phase-shifted, the start-up time of the resonant cavity of the LC resonant circuit can be changed, and different start-up times cause the voltage difference between the input and output filter capacitor voltages to be different, so that the currents of the two RSCCs can be consistent.
  • Current sharing control makes the energy of each RSCC fully utilized, and avoids damage to a certain RSCC circuit due to overloading.
  • the converter includes a resonant inductor, which can effectively reduce the current impact in the switching process and protect each electrical component in the converter.
  • the resonant switched capacitor converter can realize parallel use of multiple RSCCs through phase-shift control, thereby increasing the power processing capability of the entire converter.
  • FIG. 1 is a schematic diagram of a resonant switched capacitor circuit according to an embodiment of the present application
  • FIG. 2 is a timing diagram of the driving signal and the resonant inductor current corresponding to FIG. 1;
  • Fig. 3 is the current schematic diagram of the two-way resonant circuit corresponding to the control sequence of Fig. 2;
  • FIG. 4 is a schematic diagram of a photovoltaic system provided by an embodiment of the present application.
  • FIG. 5 provides a schematic diagram of charging an LC resonant circuit according to an embodiment of the present application
  • FIG. 7 provides a circuit diagram of a resonant switched capacitor converter according to an embodiment of the present application.
  • FIG. 8 provides a sequence diagram corresponding to FIG. 7 for an embodiment of the present application.
  • FIG. 9 provides a timing diagram in which the phase of the drive signal of S1A leads the phase of the drive signal of S1B according to an embodiment of the present application
  • FIG. 10 is a diagram of a phase-shift closed-loop control model provided by an embodiment of the present application.
  • FIG. 11 provides a model diagram for controlling only one channel of phase shifting for an embodiment of the present application.
  • FIG. 13 is a schematic diagram of another resonant switched capacitor converter provided by an embodiment of the present application.
  • FIG. 14 is a schematic diagram of the second bridge arm and the fourth bridge arm provided by an embodiment of the application being diodes;
  • FIG. 15 is a schematic diagram of the first bridge arm and the third bridge arm being diodes according to an embodiment of the application;
  • 16 is a timing diagram of two-way RSCC circuit sampling complementary drive signals provided by an embodiment of the application.
  • 17 is a diagram of a current sharing control model corresponding to the phase out-of-phase control provided by this embodiment.
  • FIG. 18 is a timing diagram of the RSCC-B advance phase shift provided by the present embodiment.
  • FIG. 19 is a timing diagram of the RSCC-B lag phase shift provided by the present embodiment.
  • 21 is a schematic diagram of a bidirectionally converted resonant switched capacitor converter provided by an embodiment of the application.
  • 22 is a schematic diagram of a resonant switched capacitor converter formed by multiple RSCCs provided by an embodiment of the present application;
  • FIG. 23 is a diagram of a current sharing control model corresponding to FIG. 22 provided by an embodiment of the present application.
  • FIG. 24 is a flowchart of a current sharing control method for a converter according to an embodiment of the present application.
  • the semiconductor switching device in the SCC directly switches between the capacitor and the voltage source.
  • the mismatch between the capacitor voltage and the power supply voltage causes serious current surges and the circuit noise is very large.
  • the semiconductor switching device is referred to as a switching device hereinafter.
  • an embodiment of the present application provides a resonant switched capacitor circuit (Resonant Switched Capacitor Circuit, RSCC).
  • RSCC is the introduction of a small-capacity resonant inductor into the SCC, which can significantly suppress the current impact during the switching process, and at the same time realize the soft switching of the switching device, reduce the switching loss of the switching device, improve the conversion efficiency, and reduce circuit noise at the same time.
  • the RSCC circuit is to convert the DC input voltage into a DC output voltage with a preset ratio, which is different from the traditional Buck and Boost circuits.
  • the inductance of the resonant inductor is small, resulting in poor current control capability of the circuit.
  • open-loop control is usually used to realize the voltage conversion of a fixed ratio.
  • the RSCC circuit can be used as a DC/DC converter to connect the input end to the photovoltaic array, and the output end to connect to the inverter.
  • the RSCC can be located in the combiner box to complete the function of DC/DC conversion.
  • RSCC can be applied in other scenarios that require a DC/DC conversion function, such as the field of communication power supply, etc.
  • the specific application scenarios of the RSCC circuit are not specifically limited in the embodiments of the present application.
  • N can be an integer greater than or equal to 2.
  • the resonant switched capacitor converter includes two RSCCs connected in parallel, namely RSCC-A and RSCC-B.
  • RSCC-A includes: a first bridge arm, a second bridge arm and a first LC resonant circuit; the first bridge arm includes two series-connected switch tubes S1A and S2A, and the second bridge arm includes two series-connected switch tubes S3A and S4A; S1A and S2A are connected in series between the positive bus BUS+ and the neutral line BUSN; S3A and S4A are connected in series between BUSN and the negative bus BUS-.
  • the first LC resonant circuit includes a resonant capacitor Cra and a resonant inductor Lra connected in series. Cra and Lra are connected in series between the midpoint of the first bridge arm and the midpoint of the second bridge arm.
  • the midpoint refers to the common terminal of S1A and S2A
  • the midpoint of the second bridge arm refers to the common terminal of S3A and S4A.
  • the resonant current of the first LC resonant circuit is iLra.
  • BUS+ and BUSN are the first input end and the second input end of the converter, respectively, and BUSN and BUS- are the first output end and the second output end of the converter, respectively. That is, the converter can convert the DC voltage input from the first input terminal and the second input terminal and output it from the first output terminal and the second output terminal.
  • RSCC-B includes: a first bridge arm, a second bridge arm and a first LC resonant circuit; the first bridge arm includes two series-connected switch tubes S1B and S2B, and the second bridge arm includes two series-connected switch tubes S3B and S4B; S1B and S2B are connected in series between the positive bus BUS+ and the neutral line BUSN; S3B and S4B are connected in series between BUSN and the negative bus BUS-.
  • the second LC resonant circuit includes a resonant capacitor Crb and a resonant inductor Lrb connected in series. Crb and Lrb are connected in series at the midpoint of the third bridge arm and the midpoint of the fourth bridge arm, respectively.
  • the midpoint refers to the common terminal of S1B and S2B
  • the midpoint of the fourth bridge arm refers to the common terminal of S3B and S4B.
  • the resonant current of the second LC resonant circuit is iLrb.
  • Capacitor C1a is connected in parallel with both ends of the first bridge arm, and is the input filter capacitor of RSCC-A.
  • Capacitor C2a is connected in parallel with both ends of the second bridge arm, and is the output filter capacitor of RSCC-A.
  • Capacitor C1b is connected in parallel with both ends of the third bridge arm, and is the input filter capacitor of RSCC-B.
  • Capacitor C2b is connected in parallel with both ends of the fourth bridge arm, and is the output filter capacitor of RSCC-B.
  • each switching device is driven open-loop with a duty cycle of 50%, S1 and S2 of the first bridge arm are driven complementary, and S3 and S2 of the second bridge arm are driven S4 is driven complementary, while S1 and S3 are driven synchronously, and S2 and S4 are driven synchronously.
  • Lr and Cr resonate in series, and the inductor current exhibits sinusoidal characteristics.
  • the resonant inductor current iLra of RSCC-A is significantly larger than the resonant inductor current iLrb of RSCC-B.
  • the two parallel RSCC circuits do not share the current, and the two may deviate several times, resulting in the over-power operation of the high-current RSCC circuit, which may seriously exceed the working margin of the switching device, causing the circuit to burn out, while the low-current RSCC is under-powered. Power works, underutilized.
  • an embodiment of the present application provides a photovoltaic system including a resonant switched capacitor converter, which can realize the parallel connection of multiple channels in the resonant switched capacitor converter. current sharing between RSCCs.
  • the following describes the system embodiment. When the system embodiment is introduced, the implementation manner of the fused resonant switched capacitor converter is introduced together.
  • FIG. 4 this figure is a schematic diagram of a photovoltaic system provided by an embodiment of the present application.
  • the photovoltaic power generation system includes: a resonant switched capacitor converter 300, an MPPT DC/DC converter 200 connected to the resonant switched capacitor converter 300, an inverter 2000, and a controller (not shown in the figure); Also included is the MPPT DC/DC converter 100 directly connected to the input of the inverter 2000.
  • the DC/DC converter 200 having the Maximum Power Point Tracking (MPPT, Maximum Power Point Tracking) function is taken as an example for introduction.
  • MPPT Maximum Power Point Tracking
  • it may be an ordinary DC/DC converter, that is, a DC/DC converter without the MPPT function, which is not specifically limited in this embodiment.
  • the output terminals of the two DC/DC converters 100 are connected in parallel, and the output terminals of the two DC/DC converters 200 are connected in parallel as an example. Of course, more The outputs of the DC/DC converters of the circuit are connected in parallel.
  • the input end of the DC/DC converter 100 and the input end of the DC/DC converter 200 are both connected to the photovoltaic array PV;
  • the first input terminal of the resonant switched capacitor converter 300 is connected to the positive output terminal of the DC/DC converter 200, namely BUS+; the second input terminal of the resonant switched capacitor converter 300 is connected to the negative output terminal of the DC/DC converter 200, namely BUSN;
  • the first output terminal of the resonant switched capacitor converter 300 is connected to the neutral line of the inverter 2000, namely BUSN, and the second output terminal of the resonant switched capacitor converter 300 is connected to the negative bus of the inverter 2000, that is, BUS-.
  • the resonant switched capacitor converter 300 includes the following at least two resonant switched capacitor circuits RSCC connected in parallel: a first RSCC and a second RSCC;
  • the controller adjusts the phase shift angle between the first driving signal of the first RSCC and the second driving signal of the second RSCC according to the current difference between the first current of the first RSCC and the second current of the second RSCC, so that the first current is consistent with the second current.
  • FIG. 4 the introduction is made by taking the photovoltaic system including the combiner box 1000 as an example.
  • the resonant switched capacitor converter 300 is set in the combiner box 1000 .
  • FIG. 4 is only an illustration. Only the MPPT DC/DC converter 100 can be connected between it and the neutral line BUSN. For between the photovoltaic array and the neutral line BUSN of the inverter 2000 and the negative input terminal (ie BUS-), not only the MPPT DC/DC converter 200 is connected, but also the resonant switched capacitor converter 300 is connected.
  • the resonant switched capacitor converter 300 is used to convert the output voltage of the MPPT DC/DC converter 200 into a corresponding voltage between the neutral line and the negative bus of the inverter 2000 .
  • the positive bus BUS+ connected to the first input end of the resonant switched capacitor converter 300 is different from the bus bar connected to the positive input end of the inverter 2000 .
  • the neutral line of the inverter 2000 and the neutral line of the resonant switched capacitor converter 300 are the same, they are connected together and belong to the same reference potential.
  • the photovoltaic array PV is connected to the input end of the MPPT DC/DC converter 200, and its output end is connected to the input end of the resonant switched capacitor converter 300.
  • the resonant switched capacitor converter 300 includes a multi-channel parallel RSCC circuit, The output ends of the two MPPT DC/DC200s are connected in parallel, and connected in parallel between the neutral line and the negative bus of the input end of the inverter 2000.
  • the photovoltaic system provided in the embodiments of the present application may further include an energy storage circuit, which can realize energy storage, that is, integration of photovoltaic and storage, while realizing grid-connected power generation.
  • the photovoltaic system provided in this embodiment uses a resonant switched capacitor converter to realize DC-to-DC voltage conversion.
  • the resonant switched capacitor converter 300 can convert the output voltage of the MPPT DC/DC converter 200 to 1
  • a negative voltage of :1 is provided to the inverter 2000 , that is, a negative voltage is provided between the neutral line and the negative bus bar of the inverter 2000 .
  • the voltage between the neutral line and the positive input terminal of the inverter 2000 is positive, and the voltage between the neutral line and the negative input terminal of the inverter 2000 is negative.
  • each RSCC in the resonant switched capacitor converter Since the currents of each RSCC in the resonant switched capacitor converter are shared, the energy of each RSCC circuit can be more fully utilized to avoid damage to a certain RSCC circuit due to overloading in the case of uneven current flow. Since the solution is to adjust the driving signal between the two independent RSCCs for phase shifting, it does not affect the soft switching characteristics of the switch tube in the single RSCC, thereby reducing switch damage and improving power conversion efficiency.
  • this figure is a schematic diagram of charging the LC resonant circuit according to the embodiment of the present application.
  • RSCC-A is taken as an example for introduction, wherein RSCC-B is connected in parallel with RSCC-A, and the working principle is the same as that of RSCC-A, and the working principle of RSCC-B is not repeated here.
  • the charging process is described below, for the energy transfer between BUS+ and BUSN to the LC resonant circuit.
  • the switch S1A in Figure 5 is turned on, S3A is turned on, S2A is turned off, and S4A is turned off.
  • the path of the charging current is: BUS+ to S1A to Cra to Lra to S3A to BUSN.
  • FIG. 6 is a schematic diagram of discharging an LC resonant circuit according to an embodiment of the present application.
  • the process of discharge is that the energy of the LC resonant circuit is transferred between BUSN and BUS-.
  • the charging and discharging process of the LC resonant circuit completes the transfer of voltage from the energy of the first bus to the second bus.
  • the first bus bar is the positive bus bar BUS+
  • the second bus bar is the negative bus bar BUS-.
  • FIG. 7 is a circuit diagram of a resonant switched capacitor converter provided by an embodiment of the present application.
  • the resonant switched capacitor converter provided by this embodiment includes: a controller and the following at least two resonant switched capacitor circuits RSCC connected in parallel: a first RSCC and a second RSCC; namely, RSCC-A and RSCC in FIG. 7 , respectively -B.
  • both the second bridge arm and the fourth bridge arm use uncontrollable diodes, only a unidirectional flow of energy can be realized, that is, the transfer from the busbar corresponding to the filter capacitor C1a to the busbar corresponding to the filter capacitor C2a.
  • the first RSCC includes: a first bridge arm (S1A and S2A connected in series), a second bridge arm (D1A and D2A connected in series) and a first LC resonance circuit (Cra and Lra connected in series); the first LC resonance circuit A circuit (Cra and Lra in series) is connected between the midpoint Ma of the first bridge arm and the midpoint Na of the second bridge arm.
  • the second RSCC includes: a third bridge arm (S1B and S2B connected in series), a fourth bridge arm (D1B and D2B connected in series) and a second LC resonance circuit (Cra and Crb connected in series); the second LC resonance circuit A circuit (Cra and Crb in series) is connected between the midpoint Mb of the third arm and the midpoint Nb of the fourth arm.
  • Both S1A and S2A are controllable switches
  • S1B and S2B are both controllable switches
  • D1A and D2A are diodes
  • D1B and D2B are diodes.
  • RSCC-A and RSCC-B Due to the discreteness of parameters in RSCC-A and RSCC-B, for example, the size of the resonant inductance is different or the size of the resonant capacitor is different, the current in the two resonant circuits will be different, and the difference may be several times, resulting in a large current.
  • Overpower operation of RSCC may cause circuit damage, while RSCC with low current works under power and cannot be fully utilized. Therefore, in order to solve the technical problem, the technical solutions provided by the embodiments of the present application can realize the consistent resonant current in the parallel multi-channel RSCC circuits, so that each RSCC circuit can be fully utilized, and circuits with large currents can be prevented from being damaged.
  • the controller (not shown in the figure) adjusts the difference between the first driving signal and the second driving signal according to the current difference between the first current of the first LC resonance circuit and the second current of the second LC resonance circuit The phase shift angle; so that the first current and the second current are consistent.
  • first current and the second current are consistent, which theoretically means that the first current and the second current are equal, but in actual control, there are generally errors, the absolute value of the difference between the first current and the second current Within the preset error range, that is, the control realizes that the first current and the second current are consistent, it is considered that the first current and the second current are equal, that is, the current sharing of the two RSCC circuits is realized.
  • the phase shift angle between the first drive signal of the first RSCC and the second drive signal of the second RSCC can be adjusted according to the current difference between the first current and the second current. proportional.
  • the phase shift angle may include a preset fixed angle and a dynamically adjustable angle, that is, the phase shift angle is the sum of the preset fixed angle and the dynamically adjustable angle.
  • the dynamic adjustable angle is not required, that is, the dynamic adjustable angle is 0.
  • the preset fixed angle has nothing to do with the size of the resonant currents of the two resonant circuits. It is a fixed angle between the driving signals corresponding to the two RSCC circuits set in advance, and can be fixed once set. For example, the preset fixed angle may be 0. Ideally, when the dynamic adjustable angle is 0, the driving signals of the two RSCCs are synchronized, that is, the two RSCCs are controlled in phase.
  • the dynamic adjustable angle is concerned, that is, the controller adjusts the dynamic adjustable angle in the phase shift angle between the first driving signal and the second driving signal, so that the first current and the second current are consistent.
  • the preset fixed angle may also be set to 360°/N, where N is the number of the RSCCs connected in parallel, and the N is an integer greater than 1. For example, when N is 2, that is, when two RSCCs are connected in parallel, the preset fixed angle is 180 degrees. When N is 3, that is, when three RSCCs are connected in parallel, the preset fixed angle is 120 degrees. And so on, and will not be illustrated one by one here.
  • the controller adjusts the dynamic adjustable angle on the basis of the preset fixed angle to adjust the phase shift angle, so that the first current and the second current are consistent.
  • the controller controls the phase difference between the first drive signal and the second drive signal to be the phase shift angle, and specifically adjusts the phase of at least one of the first drive signal and the second drive signal to achieve the phase difference.
  • the phase of one of the driving signals can be fixed and the phase of the other driving signal can be adjusted.
  • the phases of the two driving signals can also be adjusted, for example, the phases of the two driving signals are adjusted in opposite directions to achieve the above phase difference. Since the phase difference between the two driving signals is a preset fixed angle before the current sharing, the current sharing of the two RSCCs can be realized by adjusting the dynamic adjustable angle during actual adjustment.
  • FIG. 8 this figure is a sequence diagram corresponding to FIG. 6 provided by an embodiment of the present application.
  • the preset fixed phase between the driving signals of the two RSCC circuits is 0 for description. That is, the preset fixed phase between the driving signals used by the switch tubes in the same position of the two RSCC circuits is 0, that is, if the dynamic adjustable angle between the two RSCC circuits is not controlled, then the phase positions of the switch tubes in the two RSCC circuits are not controlled.
  • the phases of the drive signals are the same. That is, when the dynamic adjustable angle is 0, S1A and S1B are turned on and off at the same time, and S2A and S2B are turned on and off at the same time.
  • S1A and S2A are complementarily turned on, that is, the two are not turned on at the same time, and in actual control, there will be a certain amount of difference between the two.
  • Dead time that is, after S1A is turned off for a preset time, S2A is turned on.
  • S1B and S2B conduct complementary conduction.
  • each RSCC circuit S1A and S2A are driven complementarily with a duty cycle of 50%; S1B and S2B are driven complementarily with a duty cycle of 50%.
  • the duty ratio of 50% is a theoretical value. In practical applications, the dead zone between the switches on the same bridge arm needs to be considered to ensure reliable commutation. Generally, the duty ratio is slightly lower than 50%.
  • a controller (not shown in the figure) for obtaining a dynamically adjustable angle according to the current difference between the first current iLra of the first LC resonant circuit (Cra and Lra) and the second current iLrb of the second LC resonant circuit ⁇ ;
  • the dynamic adjustable angle between the first bridge arm and the second bridge arm is controlled to be ⁇ , specifically, the phase difference between the first drive signal of the first bridge arm and the second drive signal of the second bridge arm is
  • the angle ⁇ is dynamically adjustable to make the first current equal to the second current.
  • a certain dynamic adjustable angle ⁇ is introduced between different RSCC circuits.
  • FIG. 8 takes an example in which the phase of the driving signal of RSCC-B leads the phase of the driving signal of RSCC-A.
  • phase of the driving signal of S1B leads the driving signal of S1A by a dynamic adjustable angle ⁇ .
  • the phase of the driving signal of S2B leads the driving signal of S2A by a dynamic adjustable angle ⁇ .
  • S1A and S2A occupy The duty cycle is the same, and the duty cycle of S2A and S2B is the same.
  • the phase of the driving signal of the switch tube is phase-shifted, the current in the corresponding resonant circuit can be phase-shifted accordingly, but the soft-switching characteristics of a single RSCC circuit are not changed, and the switch tube can continue to achieve zero-current switching, thereby ensuring high Efficient power conversion. Due to the phase-shift control between each RSCC circuit, the start-up time of the resonant circuit is changed, and at the same time, the voltage difference between the filter capacitor and the switch capacitor is different due to different start-up times. Current sharing control.
  • the dynamic adjustable angle and phase shift direction between different RSCC circuits can be determined according to closed-loop control.
  • the dynamic adjustable angle is related to the difference between the resonant currents of the two RSCC circuits, so it is not a fixed angle.
  • the dynamic adjustable angle is positively correlated with the absolute value of the difference between the resonant currents corresponding to the two resonant circuits, that is, the greater the absolute value of the difference between the two resonant currents, the greater the corresponding dynamic adjustable angle. .
  • phase of the drive signal of S1A lags the phase of the drive signal of S1B.
  • the phase of the drive signal of S1A can be controlled to lead the phase of the drive signal of S1B as required.
  • FIG. 9 it is a timing diagram in which the phase of the drive signal of S1A leads the phase of the drive signal of S1B.
  • the phase of the drive signal of S1B in the RSCC-B circuit lags the phase of the drive signal of S1A in the RSCC-A.
  • FIG. 9 only illustrates the timing sequence of the driving signals controlled by the phase shift between different RSCC circuits.
  • phase difference between the driving signals corresponding to RSCC-A and RSCC-B is a dynamically adjustable angle
  • FIG. 10 is a diagram of a phase-shift closed-loop control model provided by this embodiment of the present application.
  • the first type one driving signal is fixed, and the other driving signal is controlled to shift the phase.
  • the first current of the resonant inductor in RSCC-A is detected
  • the second current of the resonant inductor in RSCC-B is detected
  • the first current and the second current are closed-loop adjusted to obtain a dynamically adjustable angle in the phase shift angle.
  • the current difference between the first current and the second current can be obtained, and the current difference can be adjusted by proportional and integral PI to obtain a dynamically adjustable angle in the phase shift angle, and the dynamic adjustable angle is positively correlated with the difference.
  • the specific phase shift angle can be generated by using the phase shift angle generator according to the result of PI adjustment.
  • the phase shift angle generator can be realized by changing the initial value of the carrier, or by adjusting the value of the comparison value. In this embodiment is not limited.
  • the embodiments of the present application do not specifically limit the specific implementation manner of detecting the current on the resonant inductor, for example, a Hall sensor or the like may be used to perform current detection.
  • the controller controls the phase of the driving signal corresponding to RSCC-A to remain unchanged, and controls the driving signal corresponding to RSCC-B to shift the phase. That is, the controller controls the phase of the first drive signal to be fixed, and controls the phase shift of the second drive signal to dynamically adjust the angle. Since RSCC-A and RSCC-B are connected in parallel, the controller can also control the phase of the driving signal corresponding to RSCC-B to remain unchanged, and control the driving signal corresponding to RSCC-A to shift the phase.
  • the controller controls the phase of the first driving signal in RSCC-A to shift the phase in the first direction by a first angle, and controls the phase of the second driving signal in RSCC-B to shift the phase in the second direction by a second angle, so
  • the sum of the first angle and the second angle is the dynamic adjustable angle, and the first direction and the second direction are opposite. That is, since the phase-shifting directions of the two driving signals are opposite, the more the phase-shifting is, the larger the phase difference between the two driving signals will be, and the phase-shifting will stop until the phase difference becomes a dynamically adjustable angle.
  • this figure is a model diagram for controlling only one channel of phase shifting provided by the embodiment of the present application.
  • the driving signal of RSCC-A is fixed and the driving signal of RSCC-B is controlled to be phase-shifted as an example.
  • the dynamic adjustable angle ⁇ is obtained by closed-loop control for the two resonant currents
  • the driving signal of RSCC-B is shifted in advance by an angle of ⁇ , that is, the phase of the driving signal of control S1B is ahead of the phase of the driving signal of S1A by an angle of ⁇ . ;
  • phase-shifted RSCC-B drive signal ⁇ angle that is, the phase of the control S1B drive signal lags the phase ⁇ angle of the S1A drive signal.
  • the abscissa represents the phase shift angle of the RSCC-B drive signal relative to the RSCC-A, in degrees, and a positive value means that the RSCC-B drive signal lags the RSCC-A drive signal; the ordinate represents the current effective value of the resonant inductor , the unit is A.
  • the dotted line represents the trend of the current of the resonant circuit of RSCC-B with the dynamic adjustable angle.
  • the adjustment angle gradually increases and gradually decreases.
  • the solid line represents the change trend of the current of the resonant circuit of RSCC-A with the dynamic adjustable angle. It can be seen that the current of the resonant circuit of RSCC-A gradually increases with the increase of the lag phase shift angle of the RSCC-B drive signal. Increase. The total current of the resonant circuits of RSCC-A and RSCC-B remains basically unchanged, indicating that the total power processed remains unchanged.
  • Figure 12 shows that the discrete parameters of the two RSCCs are different, for example, when the resonant parameter deviation is +10% to -10%, that is, the resonant inductance Lra and switched capacitor Cra of RSCC-A are greater than 10% of the rated value, and the resonant inductance of RSCC-B is Lrb and switched capacitor Crb are less than 10% of the rated value.
  • the deviation of the resonance parameters in the two RSCCs is not the above value, the relationship between the current and the phase shift angle of the resonance circuit is slightly different from that in Fig. 11, but still maintains a monotonic variation relationship.
  • the effective values of the currents of the resonant inductors of RSCC-A and RSCC-B are 6.8A and 24A respectively, and the absolute value difference is 17.2A.
  • the current of the resonant inductors of RSCC-B is RSCC-A 3.5 times the current of the resonant inductor, the difference is very significant.
  • the phase of the driving signal corresponding to RSCC-A is fixed, and the phase of the driving signal corresponding to RSCC-B is gradually increased to make it lag the phase ⁇ of the driving signal of RSCC-A (the phase of RSCC-A is Compared with RSCC-B, it is ahead of ⁇ ), and the currents of the resonant inductors of the two RSCCs are gradually consistent.
  • the dynamic adjustable angle is ⁇ to 12.5°
  • the resonant currents of the two channels are basically the same, that is, the currents of the resonant inductors of the two channels of RSCC-A and RSCC-B are equal, and the current sharing of the two channels of RSCC is realized.
  • the currents of the two RSCCs are represented by detecting the currents of the resonant inductors. Since the currents of the resonant inductors are relatively convenient to detect, for example, a detection circuit or a sensor for detecting the currents of magnetic devices can be implemented.
  • the relationship between the corresponding resonant current and the dynamic adjustable angle can be obtained according to the actual application scenario, the parameters of the resonant capacitor and the resonant inductance, and the application of the RSCC circuit.
  • the intersection of the two curves is when the currents of the two RSCCs are equal, and the dynamically adjustable angle corresponding to the intersection is the phase difference between the driving signals of the two RSCCs.
  • the controller is also used to control the phase difference between the preset fixed angle and the preset threshold when the dynamic adjustable angle is greater than the preset threshold angle
  • the sum of the angles, the preset threshold angle is the maximum upper limit value of the preset dynamic adjustable angle.
  • the preset threshold angle may be tested according to specific application scenarios to obtain empirical values.
  • the preset threshold angle may be set to 30°, and the embodiment of the present application does not specifically limit the obtaining method.
  • the corresponding dynamic adjustable angle can be obtained according to the resonant inductor currents of the two RSCCs, thereby controlling the phase difference between the driving signals corresponding to the two RSCCs to be a preset fixed angle and a dynamically adjustable angle. Therefore, the current sharing of the two RSCCs can be realized, and the effective parallel connection of multiple RSCC circuits can be realized under the premise of current sharing, and the power processing capability of the entire converter can be increased.
  • this scheme controls the phase shift between two independent RSCCs, and implements open-loop control for the driving signal of a single RSCC, it does not affect the soft switching characteristics of the switch in a single RSCC, thereby reducing switch damage and improving power. conversion efficiency.
  • this figure is a schematic diagram of another resonant switched capacitor converter according to an embodiment of the present application.
  • the first RSCC includes: a first bridge arm, a second bridge arm and a first LC resonance circuit;
  • the second RSCC includes: a third bridge arm, a fourth bridge arm and a second LC resonance circuit;
  • the first end of the first bridge arm and the first end of the third bridge arm are both connected to the first input end of the resonant switched capacitor converter, namely BUS+, and the second end of the first bridge arm is connected to the The first end of the second bridge arm, the second end of the third bridge arm are connected to the first end of the fourth bridge arm, the second end of the second bridge arm and the first end of the fourth bridge arm Both ends are connected to the second output end of the resonant switched capacitor converter, namely BUS-;
  • the resonant capacitor Cra of the first LC resonant circuit is connected between the midpoint of the first bridge arm and the midpoint of the second bridge arm, and the resonant capacitor Crb of the second LC resonant circuit is connected to the third bridge arm. between the midpoint of the bridge arm and the midpoint of the fourth bridge arm;
  • the resonant inductor Lra of the first LC resonant circuit is connected between the second end O1a of the first bridge arm and the second input end BUSN (ie O2a) of the resonant switched capacitor converter; the second LC The resonant inductance Lrb of the resonant circuit is connected between the second terminal O1b of the third bridge arm and the second input terminal BUSN (ie, O2b) of the resonant switched capacitor converter.
  • the second end of the first bridge arm is connected to BUSN, but not in FIG. 13 , but a resonant inductor Lra is connected between the second end of the first bridge arm and BUSN.
  • connection manner of the resonant inductor in the resonant circuit introduced in this embodiment is applicable to all other embodiments in this application.
  • each bridge arm of the two RSCC circuits provided in the above embodiments are described by taking the controllable switch tube as an example, and the following describes the implementation of the lower bridge arm, that is, the switch modules of the output bridge arm are diodes.
  • FIG. 14 is a schematic diagram of the second bridge arm and the fourth bridge arm provided by an embodiment of the present application being diodes.
  • the first bridge arm between BUS+ and BUSN in FIG. 14 is the controllable switch transistors S1A and S2A, and the third bridge arm is similarly the controllable switch transistor S1B and S2B.
  • the second bridge arm between BUSN and BUS- includes a first diode and a second diode connected in series, namely diodes D1A and D2A, the energy is transferred from BUS+ to BUS-, then D1A and D2A form a freewheeling loop , that is, the positive pole of D1A is connected to the negative pole of D2A, the negative pole of D1A is connected to the common point of the first bridge arm and the second bridge arm; the positive pole of D2A is connected to BUS-.
  • the fourth bridge arm between BUSN and BUS- includes a third diode and a fourth diode connected in series, namely D1B and D2B, and the energy is transferred from BUS+ to BUS-, that is, C1a transfers energy to C2a.
  • D1B and D2B form a freewheeling loop, that is, the positive pole of D1B is connected to the negative pole of D2B, the negative pole of D1B is connected to the common point of the first bridge arm and the second bridge arm; the positive pole of D2B is connected to BUS-.
  • the switch modules on the second bridge arm and the fourth bridge arm introduced in this embodiment are both diodes, which is suitable for the transfer of energy from BUS+ to BUS-, and if the energy is transferred from BUS- to BUS+, That is, the energy transfer from C2a to C1a needs to be reversed, that is, the switch module of the first bridge arm and the switch module of the third bridge arm can be diodes, while the switch module of the second bridge arm and the switch module of the fourth bridge arm need to be reversed.
  • the switch module of the first bridge arm and the switch module of the third bridge arm can be diodes, while the switch module of the second bridge arm and the switch module of the fourth bridge arm need to be reversed.
  • FIG. 15 is a schematic diagram of the first bridge arm and the third bridge arm provided by an embodiment of the present application being diodes.
  • RSCC-A is used as an example for introduction, and the same is true for RSCC-B.
  • S2A When charging, S2A is closed, S1A is open, and the energy between BUS- and BUSN is transferred to the LC resonance circuit, that is, the LC resonance circuit is charged.
  • the two switch modules of the bridge arm corresponding to the energy output end are diodes, namely D1A and D2A.
  • the bridge arm of the output end is uniformly defined as the second bridge arm, that is, the two switch modules on the first bridge arm need to be controllable switch tubes, and the bridge arm corresponding to the energy output end is only for freewheeling, and the switch module on it Can be an uncontrollable diode.
  • the switch modules on all bridge arms need to be set as controllable switch tubes.
  • the negative pole of D1A is connected to BUS+
  • the positive pole of D1A is connected to the negative pole of D2A
  • the positive pole of D2A is connected to BUSN.
  • the output bridge arms corresponding to RSCC-B include D1B and D2B.
  • the negative pole of D1B is connected to BUS+
  • the positive pole of D1B is connected to the negative pole of D2B
  • the positive pole of D2B is connected to BUSN.
  • controllable switch transistor in all the embodiments of the present application may be an IGBT or a MOS transistor, that is, a gate controllable switch transistor, and the specific implementation form is not limited.
  • the above describes the control of the dynamic adjustable angle when the preset fixed angle between the first driving signal of the first RSCC and the second driving signal of the second RSCC is 0.
  • the following describes the first driving signal and the second driving signal In the case where the preset fixed angle between them is 360°/N, continue to take N as 2, that is, two-way RSCC as an example, that is, the preset fixed angle is 180°.
  • the switches in the two RSCCs are controlled by 180° interleaving, the current on the filter capacitors (C1a, C2a, C1b, C2b) can be effectively reduced. Therefore, a smaller filter capacitor can be used to reduce the volume occupied by the filter capacitor. .
  • this figure is a timing diagram of a two-way RSCC circuit using interleaved driving signals according to an embodiment of the present application.
  • the switches in the same position in RSCC-A and RSCC-B are driven by complementary driving signals, as shown in the figure
  • the first bridge arm includes a first switch S1A and a second switch S2A
  • the third bridge arm includes a third switch S1B and a fourth switch S2B;
  • the drive signal of the first switch S1A and the drive signal of the second switch S2A are complementary, and the drive signal of the third switch S1B and the drive signal of the fourth switch S2B are complementary;
  • FIG. 17 is a diagram of a current sharing control model corresponding to the phase out-of-phase control provided in this embodiment.
  • the same control strategy as in Figure 10 can be used, that is, one is to fix the driving signal of one of the RSCCs, and control the driving signal of the other RSCC to shift the phase. .
  • the other is to phase-shift the driving signals of the two RSCCs in opposite directions.
  • phase shift direction in FIG. 17 is just opposite.
  • the driving signal of RSCC-B will be phase-shifted by keeping the driving signal of RSCC-A unchanged.
  • the RSCC circuit with a small control current needs to delay the phase shift, or the RSCC circuit with a large control current needs to shift the phase ahead.
  • the driving signal of the lead-phase-shifted RSCC-B can dynamically adjust the angle ⁇ .
  • the controller controls the phase of the first driving signal to be fixed during interleaving control, and when the second current is smaller than the first current, controlling the phase lag of the second driving signal to dynamically shift the phase Adjusting the angle; when the second current is greater than the first current, the phase of the second drive signal is controlled to advance and shift the phase to dynamically adjust the angle.
  • the drive signal in the hysteresis phase-shifted RSCC-B can be dynamically adjusted by the angle ⁇ .
  • the phase of the driving signal of RSCC-A is fixed, the phase of RSCC-B is shifted, and the phase of control RSCC-B is shifted.
  • the phase effect is the same.
  • the driving signal of RSCC-A can be dynamically adjusted by the angle ⁇ . If the resonant inductor current of RSCC-A is smaller than the resonant inductor current of RSCC-B, the driving signal in the phase-shifted RSCC-A is dynamically adjustable by the angle ⁇ .
  • the preset fixed angle between the first driving signal and the second driving signal is 180°, so the phase difference between the first driving signal and the second driving signal is 180°+ ⁇ .
  • FIG. 20 is a graph of the resonant current and the dynamically adjustable angle during the interleaving control provided by the embodiment of the present application.
  • the abscissa is the dynamic adjustable angle of the phase lag of the drive signal of RSCC-B relative to the drive signal of RSCC-A, in degrees, and the ordinate is the effective value of the resonant current, in A.
  • the effective value of the current of the resonant inductor of RSCC-A is 19.1A
  • the resonant inductor of RSCC-B is 19.1A.
  • the effective value of the current is 9.1A
  • the difference between the two is 10A
  • the difference is lower than the 17.2A under the non-interleaved control, but the difference between the two is still very different. Comparing Figure 20 and Figure 21 at the same time, the effects of interleaving control and non-interleaving control on the currents of the two RSCCs are just opposite.
  • the lead and lag of the drive signal are relative concepts, which essentially control the dynamic adjustable angle between the drive signals of the two RSCCs in parallel, and dynamically adjust according to the current detection of the resonant inductor, In order to achieve the purpose of closed-loop automatic adjustment.
  • the driving signal of the RSCC-B channel can also be fixed to shift the driving signal of the RSCC-A channel.
  • the DC/DC converter provided in the embodiment of the present application may be a bidirectional converter, that is, the energy can flow in the reverse direction, that is, from the negative bus BUS- is transferred to positive bus BUS+.
  • the DC/DC converter is bidirectional, the corresponding switching devices on all the bridge arms need to be controllable switches, that is, the energy flow in different directions can be realized by controlling the switching states thereof.
  • this figure is a schematic diagram of a bidirectionally converted resonant switched capacitor converter provided by an embodiment of the present application.
  • two RSCCs are used as an example for description.
  • the first bridge arm of RSCC-A includes controllable switches S1A and S2A
  • the second bridge arm of RSCC-A includes controllable switches S1A and S2A
  • the bridge arm includes controllable switch tubes S3A and S4A, and all four controllable switch tubes include anti-parallel diodes.
  • the third bridge arm of RSCC-B includes controllable switches S1B and S2B
  • the fourth bridge arm of RSCC-B includes controllable switches S3B and S4B
  • the four controllable switches also include anti-parallel diode.
  • energy can be transferred both from C1a to C2a and from C2a to C1a.
  • energy can be transferred both from C1b to C2b and from C2b to C1b. Since RSCC-A and RSCC-B are connected in parallel, the directions of the energy transfer of the two RSCCs are the same.
  • the above embodiments are all introduced by taking a two-level resonant switched capacitor converter as an example.
  • the following describes a multi-level resonant switched capacitor converter.
  • the current sharing control methods introduced in the above embodiments are also applicable to multi-level resonant switched capacitor converters.
  • Resonant switched capacitor converters The following continues to take the parallel connection of two RSCCs as an example for introduction.
  • this figure is a schematic diagram of a resonant switched capacitor converter formed by multiple RSCCs provided in an embodiment of the present application.
  • the resonant switched capacitor converter provided in this embodiment includes N channels of RSCCs connected in parallel, namely RSCC-A, RSCC-B and RSCC-N.
  • N is an integer greater than or equal to 3.
  • RSCC-A and RSCC-B are exactly the same as those shown in FIG. 5 and FIG. 6 , which will not be repeated here.
  • RSCC-N the structure and internal connection relationship of RSCC-N are also the same as those of RSCC-A.
  • the following mainly introduces the current sharing control when N channels of RSCC are connected in parallel.
  • this figure is a current sharing control model diagram corresponding to FIG. 22 .
  • N channels of RSCC When N channels of RSCC are connected in parallel, it is necessary to detect the current of the resonant inductors of each channel of RSCC, and obtain the current average value of the N channels of RSCC circuits through arithmetic averaging, that is, the controller obtains the current average value of the resonant circuits of the N channels of RSCC circuits; fixed; For the phase of the driving signal of one of the RSCC circuits, the currents of the remaining N-1 resonant circuits are compared with the current average value, and the respective dynamic adjustable angles are obtained according to the respective comparison results.
  • the dynamically adjustable angle shifts the phase of its drive signal. That is, the N-1 channel RSCC performs closed-loop control according to the difference between the current of its own resonant inductance and the average value of the current, so as to realize the current sharing control between the N channels of RSCC.
  • the phase of the driving signal of one channel of RSCC can continue to be fixed, and the phase-shift control of the driving signals of the remaining N-1 channels of RSCC can be performed.
  • the current of the resonant circuit is compared with the average current value, and the corresponding difference value of each channel is obtained, and the corresponding closed-loop control is performed on each channel according to the difference value of each channel, that is, by dynamically adjusting the drive in RSCC-B to RSCC-N
  • the dynamic adjustable angle of the signal realizes the current sharing control between each RSCC.
  • the current sharing control of multiple channels of RSCC in parallel also includes the two types of control introduced in the above embodiment, that is, the driving signals between each channel adopt non-interleaved control or interleaved control, which can be determined according to whether the non-interleaved control or the interleaved control is used.
  • the interleaving control is used to select the dynamic adjustable angle corresponding to the lead or the lag, and the specific implementation is similar to that of the above embodiment, which will not be repeated here. It should be noted that, when N circuits are connected in parallel, the interleaving control is often implemented in a 360°/N phase-staggered manner.
  • the embodiments of the present application further provide a current sharing control method, which will be described in detail below with reference to the accompanying drawings.
  • FIG. 24 is a flowchart of a current sharing control method for a resonant switched capacitor converter provided by an embodiment of the present application.
  • the current sharing control method provided in this embodiment is applied to the resonant switched capacitor converter provided in the above embodiment.
  • the method includes:
  • S2701 Obtain the first current of the first RSCC, and obtain the second current of the second RSCC;
  • Obtaining the first current of the first RSCC may be achieved by obtaining the first current of the first LC resonance circuit, and obtaining the second current of the second RSCC may be achieved by obtaining the second current of the second LC resonance circuit.
  • This step does not limit the sequence of obtaining the first current and the second current. Since each RSCC circuit is independent, the obtaining of the respective currents can be accomplished by the respective current sampling circuits or current sensors without affecting each other.
  • the current of the resonant circuit is represented by the current of the resonant inductor in the embodiments of the present application, and the specific manner of obtaining the current of the resonant inductor is not limited, and any method of obtaining the current of the magnetic device can be used to obtain it.
  • S2702 Obtain a current difference between the first current of the first RSCC and the second current of the second RSCC.
  • the phase shift angle includes a dynamically adjustable angle; wherein, the dynamically adjustable angle is positively correlated with the current difference.
  • phase shift angle can be obtained in the following ways:
  • the first current and the second current are obtained, and the closed-loop adjustment control is performed on the first current and the second current to obtain a dynamically adjustable angle in the phase shift angle.
  • the difference between the first current and the second current is obtained, and closed-loop control is performed on the difference to obtain a dynamically adjustable angle in the phase shift angle
  • the greater the absolute value of the difference between the first current and the second current, the greater the dynamic adjustable angle, and the effective value of the resonant inductor current can be obtained in this embodiment.
  • the embodiment of the present application does not specifically limit whether it is the first current minus the second current, or the second current minus the first current, because two RSCC circuits are connected in parallel, the first and the second are only a name, and there is no actual The meaning of sorting can be reversed, and the effect is exactly the same.
  • the closed-loop control is the difference between the resonant currents of the two RSCCs, that is, in order to achieve the same resonant currents of the two RSCCs, the phase shift angle represents the relative phase shift of the drive signals between the two.
  • S2703 Adjust the difference between the first driving signal of the first RSCC and the second driving signal of the second RSCC according to the current difference between the first current of the first RSCC and the second current of the second RSCC The phase shift angle between them is so that the first current is consistent with the second current.
  • the two are considered to be consistent, that is, the two are considered to be equal.
  • the first current and the second current are equal, and may be equal to effective current, equal to average current, or equal to peak current, which is not limited in this embodiment, and current sampling and closed-loop control may be performed according to actual needs.
  • the dynamic adjustable angle is not required, that is, the dynamic adjustable angle is 0.
  • the preset fixed angle has nothing to do with the magnitudes of the resonant currents of the two resonant circuits. It is a fixed angle between the driving signals corresponding to the two RSCC circuits set in advance, and can be fixed once set. For example, the preset fixed angle may be 0. Ideally, when the dynamic adjustable angle is 0, the driving signals of the two RSCCs are synchronized.
  • the dynamic adjustable angle is concerned, that is, the controller adjusts the dynamic adjustable angle in the phase shift angle between the first driving signal and the second driving signal, so that the first current and the second current are consistent.
  • the current consistency between each RSCC is achieved by controlling the dynamic adjustable angle.
  • the preset fixed angle may also be set to 360°/N, where N is the number of the RSCCs connected in parallel, and the N is an integer greater than 1. For example, when N is 2, that is, when two RSCCs are connected in parallel, the preset fixed angle is 180 degrees. When N is 3, that is, when three RSCCs are connected in parallel, the preset fixed angle is 120 degrees. And so on, and will not be illustrated one by one here.
  • the controller controls the phase difference between the first drive signal and the second drive signal to be the phase shift angle, and specifically adjusts the phase of at least one of the first drive signal and the second drive signal to achieve the phase difference.
  • the phase of one of the driving signals can be fixed and the phase of the other driving signal can be adjusted.
  • the phases of the two driving signals can also be adjusted, for example, the phases of the two driving signals are adjusted in opposite directions to achieve the above phase difference. Since the phase difference between the two driving signals is a preset fixed angle before the current sharing, the current sharing of the two RSCCs can be realized by adjusting the dynamic adjustable angle during actual adjustment.
  • the preset fixed angle between the drive signals of the two RSCCs is 0 for introduction, in order to make the phase between the first drive signal of the first bridge arm and the second drive signal of the second bridge arm The difference is the phase shift angle. Since the preset fixed angle is 0, the phase difference between the two drive signals is controlled to be a dynamically adjustable angle, which can include the following two implementations.
  • the first type one driving signal is fixed, and the other driving signal is controlled to shift the phase.
  • the phase of the first drive signal is controlled to be fixed, and the phase of the second drive signal is controlled to be shifted by the phase shift angle.
  • the phase of the drive signal corresponding to RSCC-A is controlled to remain unchanged, and the phase of the drive signal corresponding to RSCC-B is controlled to be shifted. That is, the phase of the first driving signal is controlled to be fixed, and the phase of the second driving signal is controlled to be shifted and the phase is dynamically adjustable by an angle. Since RSCC-A and RSCC-B are connected in parallel, the phase of the driving signal corresponding to RSCC-B can also be controlled to remain unchanged, and the driving signal corresponding to RSCC-A can be controlled to shift the phase.
  • the phase of the first driving signal in RSCC-A is controlled to be shifted in the first direction by a first angle
  • the phase of the second driving signal in RSCC-B is controlled to be shifted in the second direction by a second angle.
  • the sum of the first angle and the second angle is the dynamically adjustable angle
  • the first direction and the second direction are opposite. That is, since the phase-shifting directions of the two driving signals are opposite, the more the phase-shifting is, the larger the phase difference between the two driving signals is, and the phase-shifting is stopped until the phase difference becomes a dynamically adjustable angle.
  • controlling the phase of the first drive signal to be fixed, and controlling the phase of the second drive signal to shift the phase to the dynamic adjustable angle specifically includes:
  • phase of the first drive signal to be fixed, and control the phase of the second drive signal to advance by a dynamic adjustable angle when the second current is less than the first current; when the second current is greater than When the first current is used, the phase lag phase shift of the second drive signal is controlled to be dynamically adjustable by an angle.
  • the phase of the corresponding drive signal is out of phase by 180°.
  • the phase of the first drive signal is controlled to be fixed, and the phase of the second drive signal is controlled to be shifted by the phase shift angle, specifically including:
  • the phase of the first drive signal is controlled to be fixed, and when the second current is less than the first current, the phase lag of the second drive signal is controlled by a dynamic adjustable angle of phase lag; when the second current is greater than the first current When the first current is controlled, the phase of the second driving signal is controlled to be advanced and shifted by a dynamic adjustable angle.
  • the phase difference between the first drive signal of the first bridge arm and the second drive signal of the second bridge arm is controlled to be a preset fixed angle and a preset threshold angle Sum.
  • the controller controls the phase difference to be the sum of the preset fixed angle and the preset threshold angle.
  • the preset angle may be tested according to a specific application scenario to obtain an empirical value, and the embodiment of the present application does not specifically limit the obtaining method.
  • the present application can obtain the corresponding dynamic adjustable angle according to the difference between the resonant inductor currents of the two RSCCs, so as to control the phase difference between the driving signals corresponding to the two RSCCs as the phase shift angle, and realize the realization of the two RSCCs. Therefore, the effective parallel connection of two RSCC circuits can be realized, and the power handling capacity of the whole converter can be increased.
  • the current sharing control method introduced above is introduced by taking two RSCCs as an example.
  • the following describes a scenario where N RSCCs are connected in parallel, and N is greater than or equal to 3.
  • N is an integer greater than or equal to 3; the current sharing control specifically includes:
  • the dynamic adjustable angle of the phase shifts its driving signal.
  • the phase of the driving signal of one channel of RSCC can continue to be fixed, and the phase-shift control of the driving signals of the remaining N-1 channels of RSCC can be performed.
  • the current of the resonant circuit is compared with the average current value, and the corresponding difference value of each channel is obtained, and the corresponding closed-loop control is performed on each channel according to the difference value of each channel, that is, by dynamically adjusting the drive in RSCC-B to RSCC-N
  • the dynamic adjustable angle of the signal realizes the current sharing control between each RSCC.
  • the method provided by the embodiment of the present application can obtain the corresponding dynamic adjustable angle according to the resonant inductor current of the two RSCCs, thereby controlling the phase difference between the driving signals corresponding to the two RSCCs to be between the preset fixed angle and the dynamic adjustable angle Therefore, the current sharing of the two RSCCs can be realized, and the effective parallel connection of multiple RSCC circuits can be realized under the premise of current sharing, and the power processing capability of the entire converter can be increased.
  • this scheme controls the phase shift between two independent RSCCs, and implements open-loop control for the driving signal of a single RSCC, it does not affect the soft switching characteristics of the switch in a single RSCC, thereby reducing switch damage and improving power. conversion efficiency.
  • the methods provided in the above embodiments are not only applicable to the specific topologies of the resonant switched capacitor converters provided in the above embodiments, but also to the topologies of resonant switched capacitor converters of other topologies, for example, including other topologies and connection relationships of multiple parallel RSCCs circuits are available.
  • the above embodiments are only described by taking an example that one RSCC includes two bridge arms, and each bridge arm includes one switching device.
  • the current sharing method provided above is suitable for resonant switched capacitor converters with other voltage proportional conversions, as long as the resonant switched capacitor converters include multiple RSCCs in parallel.
  • At least one (item) refers to one or more, and "a plurality” refers to two or more.
  • “And/or” is used to describe the relationship between related objects, indicating that there can be three kinds of relationships, for example, “A and/or B” can mean: only A, only B, and both A and B exist , where A and B can be singular or plural.
  • the character “/” generally indicates that the associated objects are an “or” relationship.
  • At least one item(s) below” or similar expressions refer to any combination of these items, including any combination of single item(s) or plural items(s).
  • At least one (a) of a, b or c can mean: a, b, c, "a and b", “a and c", “b and c", or "a and b and c" ", where a, b, c can be single or multiple.

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Abstract

一种光伏系统、谐振开关电容变换器及控制方法，包括：DC/DC变换器（100，200）、谐振开关电容变换器（300）、逆变器（2000）和控制器；DC/DC变换器（100，200）的输入端连接光伏阵列（PV）；谐振开关电容变换器（300）的第一输入端连接DC/DC变换器（200）的正输出端；谐振开关电容变换器（300）的第二输入端连接DC/DC变换器（200）的负输出端；谐振开关电容变换器（300）的第一输出端连接逆变器（2000）的中线，谐振开关电容变换器（300）的第二输出端连接逆变器（2000）的负母线；谐振开关电容变换器（300）包括至少两路并联的谐振开关电容电路RSCC：第一RSCC和第二RSCC；控制器根据第一RSCC的第一电流和第二RSCC的第二电流的电流差值，调整第一RSCC的第一驱动信号和第二RSCC的第二驱动信号之间的移相角，以使第一电流与第二电流一致。

Description

一种光伏系统、谐振开关电容变换器及控制方法 技术领域
本申请涉及光伏发电技术领域，尤其涉及一种光伏系统、谐振开关电容变换器及控制方法。
背景技术
传统的直流/直流DC/DC变换器包括Boost和Buck等电路，但是，这些电路的电能转换效率较低。
目前，为了提高DC/DC变换器的电能转换效率，越来越多的领域应用开关电容电路(Switched Capacitor Circuit,SCC)，SCC是一种DC/DC转换电路，通过采用半导体开关器件和低损耗电容储能元件，实现固定比例的电压转换。SCC相对于传统的Boost等DC/DC转换电路来说，SCC的电能转换效率较高。
但是，目前的SCC工作时属于开环控制模式，因此，灵活性较差，例如多路SCC并联形成的DC/DC变换器，各路SCC的电流一般不可控。
申请内容
本申请提供了一种光伏系统、谐振开关电容变换器及控制方法，能够保证多个SCC电路之间实现均流。
本申请实施例提供一种光伏发电系统，包括：DC/DC变换器、谐振开关电容变换器、逆变器和控制器；DC/DC变换器的输入端连接光伏阵列；谐振开关电容变换器的第一输入端连接所述DC/DC变换器的正输出端；谐振开关电容变换器的第二输入端连接所述DC/DC变换器的负输出端；谐振开关电容变换器的第一输出端连接所述逆变器的中线，所述谐振开关电容变换器的第二输出端连接所述逆变器的负母线。即利用谐振开关电容变换器为逆变器的中线和负输入端之间提供逆变器需要的一个负电压，谐振开关电容变换器实现直流电压到直流电压的转换，相比于传统的DC/DC变换器，谐振开关电容变换器的电能转换效率较高。
为了降低开关损耗，实现软开关，谐振开关电容变换器中的电容与电感串联形成LC谐振电路。谐振开关电容变换器包括至少两路并联的RSCC，根据两路RSCC的电流差值调整两个RSCC对应的驱动信号之间的移相角，进而实现两路RSCC的电流相等，即均流。当RSCC的驱动信号移相时，可以改变LC谐振电路的谐振腔的起振时刻，而不同的起振时刻造成输入输出滤波电容电压的压差不同，进而可以实现两路RSCC的电流一致，实现均流控制，使每个RSCC的能量被充分利用，而且避免某个RSCC电路过载而损坏。由于该方案是调整两路独立的RSCC之间的驱动信号进行移相，因此，不影响单路RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。谐振开关电容变换器包括并联的多路谐振开关电容变换器RSCC，例如至少两路并联的RSCC：第一RSCC和第二RSCC；控制器根据第一RSCC的第一电流和第二RSCC的第二电流的电流差值，调整第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使第一电流与第二电流一致，即控制两路RSCC的电流相等，实现并联的多路RSCC之间均流。
其中，第一RSCC的第一电流可以通过测量第一RSCC的LC谐振电路的电流来获得， 同理，第二RSCC的第二电流可以通过测量第二RSCC的LC谐振电路的电流来获得。
优选地，移相角与电流差值正相关，即两路RSCC之间的电流差值越大，则两路RSCC对应的驱动信号之间的移相角越大。具体实现时，可以对电流差值进行闭环调节，实现对移相角的调整，进而实现两路RSCC的电流相等。例如具体可以为获得第一电流和第二电流的电流差值，对电流差值进行比例积分PI调整获得移相角中的动态可调角度，所述动态可调角度与所述差值正相关。具体移相角的大小可以利用移相角生成器来根据PI调整的结果来生成，移相角生成器可通过改变载波的初始值实现，亦可通过调整比较值的数值实现，在本实施例中不做限定。
优选地，控制器调整第一驱动信号和所述第二驱动信号中的至少一个的相位，来调整所述第一驱动信号和所述第二驱动信号之间的所述移相角。例如，控制器仅调整第一驱动信号的相位，第二驱动信号的相位固定不变来调整移相角。另外，控制器仅调整第二驱动信号的相位，第一驱动信号的相位固定不变来调整移相角。另外，控制器还可以调整第一驱动信号和第二驱动信号的相位分别向相反方向移动，来实现移相角的调整。本实施例不限定具体的相位移动方式。
优选地，在未调整第一驱动信号和第二驱动信号之前，两者之间的移相角可以为0，即两路RSCC的驱动信号采用同相控制。移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；此种情况下，移相角就等于动态可调角度，控制器根据电流差值调整所述动态可调角度来对移相角进行调整。
优选地，当移相角就等于动态可调角度时，控制器，当第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
以上介绍的是第一桥臂和第三桥臂上相同位置的可控开关管的驱动信号在非移相时，同相位进行控制的情景，下面介绍第一桥臂和第三桥臂上相位位置的可控开关管的驱动信号交错控制的情况。由于两路RSCC中的开关管采用交错控制，而交错控制可以有效降低输入滤波电容和输出滤波电容上的电流，因此，可以使用较小的滤波电容，降低滤波电容所占用的体积。优选地，移相角为预设固定角度和动态可调角度之和，预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；控制器根据所述电流差值在所述预设固定角度的基础上，调整所述动态可调角度来对所述移相角进行调整。例如两路RSCC并联时，两路RSCC对应的驱动信号在移相角调整之前，两个驱动信号之间的移相角为180度，当两路RSCC的电流不相等时，控制器在180°移相角的基础上再调节动态可调角度来实现两路RSCC电流相等。
优选地，当多路RSCC对应的驱动信号采用交错控制时，控制器，当第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
当N路RSCC并联时，需要检测各路RSCC的谐振电感的电流，并通过算数平均获得 N路RSCC的电流平均值，即控制器获得所述N路RSCC电路的谐振电路的电流平均值；固定其中一路RSCC电路的驱动信号的相位，将剩余的N-1路的谐振电路的电流分别与所述电流平均值进行比较，根据各自的比较结果获得各自的动态可调角度，根据所述各自的动态可调角度对其驱动信号进行移相。即N-1路RSCC根据自身的谐振电感的电流与电流平均值的差值进行闭环控制，从而实现N路的RSCC之间的均流控制。
具体控制时，可以继续采用固定一路RSCC的驱动信号的相位，对其余N-1路RSCC的驱动信号进行移相控制，例如固定RSCC-A的驱动信号的相位，对RSCC-B至RSCC-N的谐振电路的电流分别与平均电流值进行比较，获得各路对应的差值，根据各路的差值分别对各路进行对应的闭环控制，即通过动态调整RSCC-B至RSCC-N中驱动信号的动态可调角度，实现各路RSCC之间的均流控制。
优选地，当动态可调角度增加到一定程度时，两路RSCC的电流差异性基本上达到极限值，如果进一步增加动态可调角度，两路RSCC之间的电流可能呈现相反方向的变化，导致控制出现非单调性，从而失去控制能力。因此在实际应用中，可以对动态可调角度进行限幅，即需要限制动态可调角度的最大值。当动态可调角度达到预设阈值角度时，则保持在预设阈值角度，即控制器还用于在动态可调角度大于预设阈值角度时，控制相位差为预设固定角度与预设阈值角度之和，预设阈值角度是预先设定的动态可调角度的最大上限值。控制器，还用于当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。当所述控制器调整所述第一驱动信号和所述第二驱动信号中的一个驱动信号的相位来调整所述动态可调角度时，所述预设阈值角度小于等于30°。优选地，当所述控制器调整所述第一驱动信号的相位和所述第二驱动信号的相位来调整所述动态可调角度时，所述预设阈值角度小于等于15°。
本申请实施例不具体限定LC谐振电路的具体位置，下面提供两种不同LC谐振电路对应的谐振开关电容变换器的具体架构：
第一种:
第一RSCC包括：第一桥臂、第二桥臂和第一LC谐振电路；所述第二RSCC包括：第三桥臂、第四桥臂和第二LC谐振电路；所述第一桥臂的第一端和所述第三桥臂的第一端均连接所述谐振开关电容变换器的第一输入端，所述第一桥臂的第二端和所述第三桥臂的第二端均连接所述谐振开关电容变换器的第二输入端；所述第二桥臂的第一端和所述第四桥臂的第一端均连接所述谐振开关电容变换器的第一输出端，所述第二桥臂的第二端和所述第四桥臂的第二端均连接所述谐振开关电容变换器的第二输出端；所述第一LC谐振电路连接在所述第一桥臂的中点和所述第二桥臂的中点之间，所述第二LC谐振电路连接在所述第三桥臂和所述第四桥臂的中点之间。
第二种:
第一RSCC包括：第一桥臂、第二桥臂和第一LC谐振电路；所述第二RSCC包括：第三桥臂、第四桥臂和第二LC谐振电路；所述第一桥臂的第一端和所述第三桥臂的第一端均连接所述谐振开关电容变换器的第一输入端，所述第一桥臂的第二端连接所述第二桥臂的第一端，所述第三桥臂的第二端连接所述第四桥臂的第一端，所述第二桥臂的第二端 和所述第四桥臂的第二端均连接所述谐振开关电容变换器的第二输出端；所述第一LC谐振电路的谐振电容连接在所述第一桥臂的中点和所述第二桥臂的中点之间，所述第二LC谐振电路的谐振电容连接在所述第三桥臂的中点和所述第四桥臂的中点之间；所述第一LC谐振电路的谐振电感连接在所述第一桥臂的第二端和所述谐振开关电容变换器的第二输入端之间；所述第二LC谐振电路的谐振电感连接在所述第三桥臂的第二端和所述谐振开关电容变换器的第二输入端之间。
优选地，为了能量可以双向移动，即从正母线向负母线转移，或从负母线向正母线转移，所有桥臂的开关器件为可控开关管，即第一桥臂至少包括串联的第一开关管和第二开关管，所述第三桥臂至少包括串联的第三开关管和第四开关管，所述第二桥臂至少包括串联的第五开关管和第六开关管；所述第四桥臂至少包括串联的第七开关管和第八开关管；
另外一种实现方式是，除了所有桥臂为可控开关管以外，第二桥臂和第四桥臂可以包括二级管。即所述第一桥臂包括串联的第一开关管和第二开关管，所述第三桥臂包括串联的第三开关管和第四开关管，所述第二桥臂至少包括串联的第一二极管和第二二极管，所述第四桥臂至少包括串联的第三二极管和第四二极管。
本申请实施例提供一种谐振开关电容变换器，包括控制器和以下至少两路并联在一起的谐振开关电容电路RSCC：第一RSCC和第二RSCC；所述谐振开关电容变换器的第一输入端连接直流电源的正输出端；所述谐振开关电容变换器的第二输入端连接所述直流电源的负输出端；所述谐振开关电容变换器，用于将所述直流电源的电压进行变换后输出；所述控制器，用于根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。其中，第一RSCC的第一电流可以通过测量第一RSCC的LC谐振电路的电流来获得，同理，第二RSCC的第二电流可以通过测量第二RSCC的LC谐振电路的电流来获得。
需要说明的是，该谐振开关电容变换器可以应用于光伏领域，也可以应用于其他场景。例如，其他需要进行1:1电压转换的场景。当应用于光伏领域时，直流电源可以为前一级DC/DC变换器的输出电压，前一级DC/DC变换器的输入端连接光伏阵列。
为了降低开关损耗，实现软开关，谐振开关电容变换器中的电容与电感串联形成LC谐振电路。谐振开关电容变换器包括至少两路并联的RSCC，根据两路RSCC的电流差值调整两个RSCC对应的驱动信号之间的移相角，进而实现两路RSCC的电流相等，即均流。当RSCC的驱动信号移相时，可以改变LC谐振电路的谐振腔的起振时刻，而不同的起振时刻造成输入输出滤波电容电压的压差不同，进而可以实现两路RSCC的电流一致，实现均流控制，使每个RSCC的能量被充分利用，而且避免某个RSCC电路过载而损坏。由于该方案是调整两路独立的RSCC之间的驱动信号进行移相，因此，不影响单路RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。
优选地，移相角与电流差值正相关，即两路RSCC之间的电流差值越大，则两路RSCC对应的驱动信号之间的移相角越大。具体实现时，可以对电流差值进行闭环调节，实现对移相角的调整，进而实现两路RSCC的电流相等。
优选地，控制器，具体用于根据所述电流差值调整所述第一驱动信号和所述第二驱动信号之间的所述移相角，以使所述第一电流与所述第二电流一致；所述移相角与所述电流差值正相关。
优选地，所述控制器，具体用于调整所述第一驱动信号和所述第二驱动信号中的至少一个的相位，来调整所述移相角。
优选地，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；所述控制器，具体用于根据所述电流差值调整所述动态可调角度来对所述移相角进行调整。
优选地，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
优选地，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；所述控制器，具体用于根据所述电流差值在所述预设固定角度的基础上，调整所述动态可调角度来对所述移相角进行调整。
优选地，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
优选地，控制器当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。
本申请实施例还提供一种均流控制方法，应用于光伏系统，所述光伏系统包括：DC/DC变换器、谐振开关电容变换器和逆变器；所述DC/DC变换器的输入端连接光伏阵列；所述谐振开关电容变换器的第一输入端连接所述DC/DC变换器的正输出端；所述谐振开关电容变换器的第二输入端连接所述DC/DC变换器的负输出端；所述谐振开关电容变换器的第一输出端连接所述逆变器的中线，所述谐振开关电容变换器的第二输出端连接所述逆变器的负母线；所述谐振开关电容变换器包括以下至少两路并联在一起的谐振开关电容电路RSCC：第一RSCC和第二RSCC；该方法包括：获得所述第一RSCC的第一电流，获得所述第二RSCC的第二电流；根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。
该方法应用于以上实施例提供的谐振开关电容变换器，包括多路并联的RSCC电路，不具体限定并联的路数，N为大于等于2的整数。并且，每个桥臂上的开关器件可以全部为可控开关管，当全部为可控开关管时，可以实现能量的双向移动，即从输入端到输出端实现能量转移，也可以从输出端向输入端实现能量转移。如果是单方向的能量转移，则第二桥臂和第四桥臂上的开关器件可以为二极管，即不可控器件，单向导通即可。
其中，第一RSCC的第一电流可以通过测量第一RSCC的LC谐振电路的电流来获得， 同理，第二RSCC的第二电流可以通过测量第二RSCC的LC谐振电路的电流来获得。
所述移相角与所述电流差值正相关，即两路RSCC之间的电流差值越大，则两路RSCC对应的驱动信号之间的移相角越大。具体实现时，可以对电流差值进行闭环调节，实现对移相角的调整，进而实现两路RSCC的电流相等。
为了降低开关损耗，实现软开关，谐振开关电容变换器中的电容与电感串联形成LC谐振电路。谐振开关电容变换器包括至少两路并联的RSCC，根据两路RSCC的电流差值调整两个RSCC对应的驱动信号之间的移相角，进而实现两路RSCC的电流相等，即均流。当RSCC的驱动信号移相时，可以改变LC谐振电路的谐振腔的起振时刻，而不同的起振时刻造成输入输出滤波电容电压的压差不同，进而可以实现两路RSCC的电流一致，实现均流控制，使每个RSCC的能量被充分利用，而且避免某个RSCC电路过载而损坏。由于该方案是调整两路独立的RSCC之间的驱动信号进行移相，因此，不影响单路RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。
优选地，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，具体包括：调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度。
优选地，所述调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度，具体包括：当所述第二电流小于所述第一电流时，调整所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，调整所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
优选地，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，具体包括：在所述预设固定角度的基础上，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度来对所述移相角进行调整。
优选地，所述调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度来对所述移相角进行调整，具体包括：当所述第二电流小于所述第一电流时，调整所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，调整所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
优选地，还包括：当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。
从以上技术方案可以看出，本申请实施例具有以下优点：
该光伏系统包括谐振开关电容变换器，谐振开关电容变换器连接在普通DC/DC变换器的输出端和逆变器的输入端之间，一般是连接在DC/DC变换器的输出端和逆变器的中线和负母线之前，用于将DC/DC变换器的输出电压转换为负压提供为逆变器的中线和负母线， 实现电压的变换。为了降低开关损耗，实现软开关，谐振开关电容变换器中的电容与电感串联形成LC谐振电路。谐振开关电容变换器包括至少两路并联的RSCC，根据两路RSCC的电流差值调整两个RSCC对应的驱动信号之间的移相角，进而实现两路RSCC的电流相等，即均流。当RSCC的驱动信号移相时，可以改变LC谐振电路的谐振腔的起振时刻，而不同的起振时刻造成输入输出滤波电容电压的压差不同，进而可以实现两路RSCC的电流一致，实现均流控制，使每个RSCC的能量被充分利用，而且避免某个RSCC电路过载而损坏。由于该方案是调整两路独立的RSCC之间的驱动信号进行移相，因此，不影响单路RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。另外，该变换器中包括谐振电感，可以有效降低开关过程中的电流冲击，保护变换器中的各个电气元件。该谐振开关电容变换器通过移相控制可实现多路RSCC的并联使用，从而增加整个变换器的功率处理能力。
附图说明
图1为本申请实施例提供的一种谐振开关电容电路的示意图；
图2为图1对应的驱动信号和谐振电感电流的时序图；
图3为图2的控制时序对应的两路谐振电路的电流示意图；
图4为本申请实施例提供的一种光伏系统的示意图；
图5为本申请实施例提供一种给LC谐振电路充电的示意图；
图6为本申请实施例提供一种给LC谐振电路放电的示意图；
图7为本申请实施例提供一种谐振开关电容变换器的电路图；
图8为本申请实施例提供与图7对应的一种时序图；
图9为本申请实施例提供S1A的驱动信号的相位超前S1B的驱动信号的相位的时序图；
图10为本申请实施例提供的移相闭环控制模型图；
图11为本申请实施例提供仅控制一路移相的模型图；
图12为本申请实施例提供同相控制时两路RSCC谐振电流与移相角之间的关系曲线图；
图13为本申请实施例提供的另一种谐振开关电容变换器的示意图；
图14为本申请实施例提供的第二桥臂和第四桥臂为二极管的示意图；
图15为本申请实施例提供的第一桥臂和第三桥臂为二极管的示意图；
图16为本申请实施例提供的两路RSCC电路采样互补驱动信号的时序图；
图17为本实施例提供的错相控制时对应的均流控制模型图；
图18为本实施例提供的RSCC-B超前移相的时序图；
图19为本实施例提供的RSCC-B滞后移相的时序图；
图20为本申请实施例提供的交错控制时谐振电流与移相角的曲线图；
图21为本申请实施例提供的双向变换的谐振开关电容变换器的示意图；
图22为本申请实施例提供的多路RSCC形成的谐振开关电容变换器的示意图；
图23为本申请实施例提供的图22对应的均流控制模型图；
图24为本申请实施例提供的针对变换器的均流控制方法的流程图。
具体实施方式
SCC中半导体开关器件直接在电容和电压源之间切换，电容电压和电源电压的不匹配导致严重的电流冲击，电路噪声很大。为了描述方便，以下简称半导体开关器件为开关器件。
为了抑制上述电流冲击，本申请实施例提供一种谐振开关电容电路(Resonant Switched Capacitor Circuit,RSCC)。RSCC是在SCC中引入小容量的谐振电感，可显著抑制开关过程中的电流冲击，同时实现开关器件的软开关，降低开关器件的开关损耗，提升转换效率，同时降低电路噪声。
为了将RSCC电路应用在大功率变换中，受限于单个开关器件的容量及无源器件容量和工艺，需要将多个RSCC电路并联使用。
RSCC电路是为了将直流输入电压转换为预设比例的直流输出电压，区别于传统的Buck、Boost电路，在RSCC电路中，谐振电感的感量较小，导致电路的电流控制能力差，所以在传统的RSCC电路中，通常采用开环控制，实现固定比例的电压变换。当RSCC电路应用在光伏发电系统时，RSCC作为DC/DC变换器可以输入端连接光伏阵列，输出端连接逆变器。另外对于有的光伏发电系统，RSCC可以位于汇流箱中，完成DC/DC转换的功能。除了光伏发电领域以外，RSCC可以应用在其他需要DC/DC变换功能的场景，例如通信电源供电领域等，本申请实施例中不具体限定RSCC电路的具体应用场景。
为了使本领域技术人员更好地理解本申请实施例提供的技术方案，下面先以两路RSCC电路并联为例介绍谐振开关电容变换器的工作原理，本申请不具体限定并联的RSCC电路的路数，例如并联N路，N可以为大于等于2的整数。
如图1所示，该谐振开关电容变换器包括并联的两路RSCC，分别为RSCC-A和RSCC-B。
其中，RSCC-A包括：第一桥臂、第二桥臂和第一LC谐振电路；第一桥臂包括两个串联的开关管S1A和S2A，第二桥臂包括两个串联的开关管S3A和S4A；其中S1A和S2A串联后连接在正母线BUS+和中线BUSN之间；S3A和S4A串联后连接在BUSN和负母线BUS-之间。
RSCC-A中，第一LC谐振电路包括串联的谐振电容Cra和谐振电感Lra，Cra和Lra串联后连接第一桥臂的中点和第二桥臂的中点之间，第一桥臂的中点是指S1A和S2A的公共端，第二桥臂的中点是指S3A和S4A的公共端。第一LC谐振电路的谐振电流为iLra。
将谐振开关电容变换器作为一个整体，BUS+和BUSN分别为该变换器的第一输入端和第二输入端，BUSN和BUS-分别为该变换器的第一输出端和第二输出端。即该变换器可以将第一输入端和第二输入端输入的直流电压转换后从第一输出端和第二输出端输出。
同理，RSCC-B包括：第一桥臂、第二桥臂和第一LC谐振电路；第一桥臂包括两个串联的开关管S1B和S2B，第二桥臂包括两个串联的开关管S3B和S4B；其中S1B和S2B串联后连接在正母线BUS+和中线BUSN之间；S3B和S4B串联后连接在BUSN和负母线BUS-之间。
RSCC-B中，第二LC谐振电路包括串联的谐振电容Crb和谐振电感Lrb，Crb和Lrb串联后分别连接在第三桥臂的中点和第四桥臂的中点，第三桥臂的中点是指S1B和S2B的公共端，第四桥臂的中点是指S3B和S4B的公共端。第二LC谐振电路的谐振电流为iLrb。
电容C1a并联在第一桥臂的两端，为RSCC-A的输入滤波电容。电容C2a并联在第二桥臂的两端，为RSCC-A的输出滤波电容。电容C1b并联在第三桥臂的两端，为RSCC-B的输 入滤波电容。电容C2b并联在第四桥臂的两端，为RSCC-B的输出滤波电容。
传统的RSCC电路中，通常采用开环控制，如图2所示，各开关器件以50％的占空比开环驱动，第一桥臂的S1和S2互补驱动，第二桥臂的S3和S4互补驱动，而S1和S3同步驱动，S2和S4同步驱动。Lr和Cr串联谐振，电感电流呈现正弦特性。当开关频率和谐振频率相同时，所有开关器件实现零电流开关，有效降低开关损耗。
当多个RSCC电路并联时，由于电感和电容均存在一定的容差，典型值为-10％至+10％，不同的RSCC电路会出现严重的不均流现象。例如图3所示，RSCC-A的谐振电感偏低10％，而RSCC-B的谐振电感偏高10％，两个RSCC电路中的Cr相同，即Cra等于Crb。由于两个RSCC电路同步开关，而RSCC-A的谐振腔阻抗低于RSCC-B的谐振腔阻抗，导致RSCC-A的谐振电感电流iLra明显大于RSCC-B的谐振电感电流iLrb。
因此，导致两个并联的RSCC电路不均流，两者可能偏差数倍，导致大电流的RSCC电路过功率工作，可能严重超过开关器件的工作裕量，导致电路烧毁，而小电流的RSCC欠功率工作，未充分利用。
为了解决以上谐振开关电容变换器中并联的多路RSCC之间不均流的问题，本申请实施例提供一种光伏系统，包括谐振开关电容变换器，可以实现谐振开关电容变换器中多路并联的RSCC之间的均流。下面介绍系统实施例，在介绍系统实施例时，将融合谐振开关电容变换器的实现方式一并介绍。
系统实施例：
参见图4，该图为本申请实施例提供的一种光伏系统的示意图。
本实施例提供的光伏发电系统，包括：谐振开关电容变换器300、与谐振开关电容变换器300连接的MPPT DC/DC变换器200、逆变器2000和控制器(图中未示出)；还包括直接与逆变器2000的输入端连接的MPPT DC/DC变换器100。
本实施中以DC/DC变换器200具有最大功率追踪(MPPT，Maximum Power Point Tracking)功能为例进行介绍。当然，可以为普通DC/DC变换器，即不具有MPPT功能的DC/DC变换器，本实施例中不做具体限定。
可以理解的是，为了提高输出能力，以两路DC/DC变换器100的输出端并联在一起，两路DC/DC变换器200的输出端并联在一起为例进行介绍，当然也可以更多路的DC/DC变换器的输出端并联在一起。
DC/DC变换器100的输入端和DC/DC变换器200的输入端均连接光伏阵列PV；
谐振开关电容变换器300的第一输入端连接DC/DC变换器200的正输出端，即BUS+；谐振开关电容变换器300的第二输入端连接DC/DC变换器200的负输出端，即BUSN；
谐振开关电容变换器300的第一输出端连接逆变器2000的中线，即BUSN，谐振开关电容变换器300的第二输出端连接逆变器2000的负母线，即BUS-。
谐振开关电容变换器300包括以下至少两路并联在一起的谐振开关电容电路RSCC：第一RSCC和第二RSCC；
控制器根据第一RSCC的第一电流和第二RSCC的第二电流的电流差值，调整第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电 流与所述第二电流一致。
图4中是以光伏系统中包括汇流箱1000为例进行的介绍，汇流箱1000中设置谐振开关电容变换器300，图4仅是一种示意，对于光伏阵列与逆变器2000的正输入端和中线BUSN之间仅连接MPPT DC/DC变流器100即可。对于光伏阵列与逆变器2000的中线BUSN和负输入端(即BUS-)之间，不仅连接MPPT DC/DC变流器200，还连接谐振开关电容变换器300。谐振开关电容变换器300用于将MPPT DC/DC变流器200的输出电压转换为逆变器2000的中线和负母线之间对应的电压。
需要说明的是，谐振开关电容变换器300的第一输入端连接的正母线BUS+与逆变器2000的正输入端连接的母线有所区别。但是，逆变器2000的中线与谐振开关电容变换器300的中线是相同的，两者连接在一起，属于等参考电位。
如图4所示，光伏阵列PV连接在MPPT DC/DC变流器200的输入端，其输出端连接谐振开关电容变换器300的输入端，谐振开关电容变换器300包括多路并联RSCC电路，两个MPPT DC/DC200的输出端并联在一起，并联后连接在逆变器2000的输入端的中线和负母线之间。
另外，本申请实施例提供的光伏系统还可以包括储能电路，实现并网发电的同时，也可以实现储能，即光储一体化。
本实施例提供的光伏系统，利用谐振开关电容变换器实现直流到直流的电压转换，如图4所示，谐振开关电容变换器300可以将MPPT DC/DC变流器200的输出电压转换为1:1的负压提供给逆变器2000，即为逆变器2000的中线和负母线之间提供一个负电压。一般情况下，逆变器2000的中线和正输入端之间为正电压，逆变器2000的中线和负输入端之间为负电压。由于该谐振开关电容变换器中各路RSCC的电流均流，因此，可以更充分地利用每个RSCC电路的能量，避免在不均流情况下某个RSCC电路过载而损坏。由于该方案是调整两路独立的RSCC之间的驱动信号进行移相，因此，不影响单路RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。
下面结合附图详细介绍本申请实施例提供的光伏系统中谐振型开关电容的工作原理。
参见图5，该图为本申请实施例提供的给LC谐振电路充电的示意图。
为了方便描述，以RSCC-A为例进行介绍，其中RSCC-B与RSCC-A并联，工作原理与RSCC-A相同，在此不再赘述RSCC-B的工作原理。
下面介绍充电的过程，为BUS+和BUSN之间的能量转移到LC谐振电路。
充电时，图5中的开关S1A导通，S3A导通，S2A断开，S4A断开，充电电流的路径为：BUS+到S1A到Cra到Lra到S3A到BUSN。
下面结合图6介绍LC谐振电路放电的工作原理。
参见图6，该图为本申请实施例提供的给LC谐振电路放电的示意图。
放电的过程是LC谐振电路的能量转移到BUSN和BUS-之间。
放电时，图6中的S1A断开，S3A断开，S2A闭合，S4A闭合，放电电流的路径为：Cra到S2A到C2a到S4A到Lra。
从以上分析可知，LC谐振电路的充放电过程完成了电压从第一母线能量向第二母线的 转移。图6中，第一母线为正母线BUS+，第二母线为负母线BUS-。并且转移过程中，由于开关电容C的储能作用，从而完成了电压变换。
变换器实施例一：
下面结合附图详细介绍本申请实施例提供的谐振开关电容变换器实现两路甚至多路均流的工作原理。
参见图7，该图为本申请实施例提供的一种谐振开关电容变换器的电路图。
本实施例提供的谐振开关电容变换器，包括：控制器和以下至少两路并联在一起的谐振开关电容电路RSCC：第一RSCC和第二RSCC；即分别为图7中的RSCC-A和RSCC-B。
图7中由于第二桥臂和第四桥臂均采用了不可控二极管，仅能实现能量的单方向流动，即从滤波电容C1a对应的母线向滤波电容C2a对应的母线转移。
所述第一RSCC包括：第一桥臂(串联的S1A和S2A)、第二桥臂(串联的D1A和D2A)和第一LC谐振电路(串联的Cra和Lra)；所述第一LC谐振电路(串联的Cra和Lra)连接在所述第一桥臂的中点Ma和所述第二桥臂的中点Na之间。
所述第二RSCC包括：第三桥臂(串联的S1B和S2B)、第四桥臂(串联的D1B和D2B)和第二LC谐振电路(串联的Cra和Crb)；所述第二LC谐振电路(串联的Cra和Crb)连接在所述第三桥臂的中点Mb和所述第四桥臂的中点Nb之间。
S1A和S2A均为可控开关管，S1B和S2B均为可控开关管，D1A和D2A均为二极管，D1B和D2B均为二极管。
由于RSCC-A和RSCC-B中参数的离散性，例如，谐振电感的大小不同或谐振电容的大小不同，会导致两个谐振电路中的电流不同，而且有可能相差数倍，导致大电流的RSCC过功率工作，可能导致电路损坏，而小电流的RSCC欠功率工作，无法充分利用。因此，为了解决该技术问题，本申请实施例提供的技术方案可以实现并联的多路RSCC电路中的谐振电流一致，使每个RSCC电路被充分利用，而且避免电流大的电路被损坏。
具体地，控制器(图中未示出)根据第一LC谐振电路的第一电流和所述第二LC谐振电路的第二电流的电流差值调整第一驱动信号和第二驱动信号之间的移相角；以使第一电流与第二电流一致。
需要说明的是，第一电流和第二电流一致，理论上是指第一电流和第二电流相等，但是实际控制中，一般都存在误差，第一电流和第二电流的差值的绝对值在预设误差范围内，即控制实现了第一电流和第二电流一致，则认为第一电流与第二电流相等，即实现了两路RSCC电路的均流。
具体控制时，第一RSCC的第一驱动信号和第二RSCC的第二驱动信号之间的移相角可以根据第一电流和第二电流的电流差值来调整，移相角与电流差值成正比。
移相角可以包括预设固定角度和动态可调角度，即移相角为预设固定角度与动态可调角度之和。
其中，理想情况下，当第一RSCC和第二RSCC的离散参数完全一致时，即两个谐振电路的谐振电流相等，不需要动态可调角度，即动态可调角度为0。
预设固定角度与两个谐振电路的谐振电流的大小没有任何关系，是提前设置的两路 RSCC电路对应的驱动信号之间的固定角度，一旦设定便可以固定不变。例如预设固定角度可以为0，理想情况下，当动态可调角度为0时，两路RSCC的驱动信号同步，即两路RSCC采用同相控制。
本申请实施例中关注的是动态可调角度，即控制器调整第一驱动信号和第二驱动信号之间的移相角中的动态可调角度，以使第一电流与第二电流一致。
通过调整动态可调角度来实现各路RSCC之间的电流一致。
另外，预设固定角度还可以设置为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数。例如，当N为2时，即两路RSCC并联时，预设固定角度为180度。当N为3时，即三路RSCC并联时，预设固定角度为120度。以此类推，在此不再一一举例说明。当预设固定角度为360°/N时，控制器在预设固定角度的基础上，调整动态可调角度来实现对移相角进行调整，进而使第一电流和第二电流一致。
实际实施时，控制器控制第一驱动信号和所述第二驱动信号之间的相位差为所述移相角，具体调整第一驱动信号和第二驱动信号中的至少一个的相位，来达到所述相位差。
具体地，可以固定其中一个驱动信号的相位不变，调整另外一个驱动信号的相位。也可以调整两个驱动信号的相位，例如向相反方向调整两个驱动信号的相位，来实现以上的相位差。由于在均流之前，两个驱动信号之间的相位差就是预设固定角度，因此，实际调整时可以通过调整动态可调角度来实现两路RSCC的均流。
参见图8，该图为本申请实施例提供的与图6对应的一种时序图。
本实施例中以两个RSCC电路的驱动信号之间的预设固定相位为0来进行介绍。即两路RSCC相同位置的开关管采用的驱动信号之间的预设固定相位为0，即如果不控制两个RSCC电路之间的动态可调角度，则两个RSCC电路中相位位置的开关管的驱动信号的相位相同。即，在动态可调角度为0时，S1A和S1B同时导通和关断，S2A和S2B同时导通和关断。由于同一个桥臂的两个开关管的驱动信号需要互补，因此，S1A与S2A互补导通，即两个不存在同时导通的情况，而且实际控制中，两者之间还会存在一定的死区时间，即S1A关断了预设时间后，S2A才导通。同理，S1B与S2B互补导通。
其中在各个RSCC电路中，S1A与S2A以50％的占空比互补驱动；S1B与S2B以50％的占空比互补驱动。占空比50％为理论值，在实际应用中需要考虑同一桥臂上的开关管之间的死区以保证可靠换流，一般占空比略低于50％。
控制器(图中未示出)，用于根据第一LC谐振电路(Cra和Lra)的第一电流iLra和所述第二LC谐振电路的第二电流iLrb的电流差值获得动态可调角度Φ；控制第一桥臂和第二桥臂之间的动态可调角度为Φ，具体可以为第一桥臂的第一驱动信号和第二桥臂的第二驱动信号之间的相位差为动态可调角度Φ，以使所述第一电流与所述第二电流相等。
为了控制各RSCC电路之间的电流均衡，在不同的RSCC电路之间引入一定的动态可调角度Φ。
图8以RSCC-B的驱动信号的相位超前RSCC-A的驱动信号的相位为例进行介绍。
即S1B的驱动信号的相位超前S1A的驱动信号的相位动态可调角度Φ，由于互补导通，S2B的驱动信号的相位超前S2A的驱动信号的相位动态可调角度Φ，其中，S1A和S2A占 空比相同，S2A和S2B占空比相同。
由于开关管的驱动信号的相位发生移相，因此可以使对应谐振电路中的电流进行相应的移相，但是不改变单个RSCC电路的软开关特性，开关管继续可以实现零电流开关，从而保证高效率的功率变换。由于各RSCC电路之间的移相控制，改变了谐振电路的起振时刻，同时由于不同的起振时刻导致滤波电容和开关电容之间的压差不同，进而可以实现各路RSCC电路之间的均流控制。
不同RSCC电路之间的动态可调角度及移相方向可以根据闭环控制来决定，动态可调角度与两个RSCC电路的谐振电流之间的差值有关，因此不是固定的角度。一般来说，动态可调角度与两个谐振电路对应的谐振电流的差值的绝对值正相关，即两路的谐振电流的差值的绝对值越大，则对应的动态可调角度越大。
图8中是S1A的驱动信号的相位滞后S1B的驱动信号的相位，实际控制中可以根据需要，控制S1A的驱动信号的相位超前S1B的驱动信号的相位。
参见图9，该图为S1A的驱动信号的相位超前S1B的驱动信号的相位的时序图。
RSCC-B电路中的S1B的驱动信号的相位滞后RSCC-A中的S1A的驱动信号的相位。
由于同一RSCC电路中，同一桥臂上的开关管的驱动信号互补，图9中仅示意了不同RSCC电路之间的移相控制的驱动信号的时序。
为了实现RSCC-A与RSCC-B对应的驱动信号之间相位差为动态可调角度，可以包括以下两种实现方式：
具体可以参见图10，该图为本申请实施例提供的移相闭环控制模型图。
第一种：一个驱动信号固定不变，控制另一个驱动信号移相。
检测RSCC-A中谐振电感的第一电流，检测RSCC-B中谐振电感的第二电流，对第一电流和第二电流进行闭环调整获得移相角中的动态可调角度。例如具体可以为获得第一电流和第二电流的电流差值，对电流差值进行比例积分PI调整获得移相角中的动态可调角度，所述动态可调角度与所述差值正相关。具体移相角的大小可以利用移相角生成器来根据PI调整的结果来生成，移相角生成器可通过改变载波的初始值实现，亦可通过调整比较值的数值实现，在本实施例中不做限定。
本申请实施例中不具体限定检测谐振电感上电流的具体实现方式，例如可以利用霍尔传感器等进行电流检测。
例如，控制器控制RSCC-A对应的驱动信号的相位保持不变，控制RSCC-B对应的驱动信号进行移相。即控制器控制第一驱动信号的相位固定，控制第二驱动信号的相位移相动态可调角度。由于RSCC-A与RSCC-B两路并联，因此，控制器也可以控制RSCC-B对应的驱动信号的相位保持不变，控制RSCC-A对应的驱动信号进行移相。
第二种：两个驱动信号向相反的方向移相。
具体地，控制器控制RSCC-A中的第一驱动信号的相位向第一方向移相第一角度，控制RSCC-B中的第二驱动信号的相位向第二方向移相第二角度，所述第一角度和所述第二角度之和为所述动态可调角度，所述第一方向和所述第二方向相反。即由于两个驱动信号的移相方向相反，因此，越移相，则两个驱动信号之间的相位差越大，直到相位差为动态 可调角度为止，停止移相。
下面结合附图详细介绍对于第一种移相控制的实现方式。
参见图11，该图为本申请实施例提供的仅控制一路移相的模型图。
本实施例中以固定RSCC-A的驱动信号，控制RSCC-B的驱动信号移相为例，当然也可以反过来，固定RSCC-B的驱动信号，移相RSCC-A的驱动信号。
与图10所示的相同，对两个谐振电流通过闭环控制获得动态可调角度Φ；
如果RSCC-B的谐振电感的电流小于RSCC-A的谐振电感的电流，则超前移相RSCC-B的驱动信号Φ角度，即控制S1B的驱动信号的相位超前S1A的驱动信号的相位Φ角度；
如果RSCC-B的谐振电感的电流大于RSCC-A的谐振电感的电流，则滞后移相RSCC-B的驱动信号Φ角度，即控制S1B的驱动信号的相位滞后S1A的驱动信号的相位Φ角度。
为了直观理解两路RSCC的谐振电流与动态可调角度之间的关系，可以参见图11，该图为本申请实施例提供的两路RSCC谐振电流与动态可调角度之间的关系曲线图。
图12中横坐标表示RSCC-B驱动信号相对RSCC-A的移相角，单位为度，正值表示RSCC-B的驱动信号滞后RSCC-A的驱动信号；纵坐标表示谐振电感的电流有效值，单位为A。
从图12中可以看出，虚线表示RSCC-B的谐振电路的电流随着动态可调角度的变化趋势，可以看出，RSCC-B的谐振电路的电流随着RSCC-B驱动信号滞后动态可调角度的逐渐增大而逐渐降低。
实线表示RSCC-A的谐振电路的电流随着动态可调角度的变化趋势，可以看出，RSCC-A的谐振电路的电流随着RSCC-B驱动信号滞后移相角的逐渐增大而逐渐增加。RSCC-A和RSCC-B的谐振电路总电流基本保持不变，说明处理的总功率不变。
综上，RSCC-A和RSCC-B中的谐振电路的电流均与动态可调角度之间呈现单调的变化关系。
图12是以两路RSCC的离散参数不同，例如谐振参数偏差+10％到-10％时，即其中RSCC-A的谐振电感Lra和开关电容Cra大于额定值10％，RSCC-B的谐振电感Lrb和开关电容Crb小于额定值10％。可以理解的是，当两个RSCC中的谐振参数的偏差不是以上的数值时，谐振电路的电流与移相角的关系曲线与图11略有差异，但是仍然保持单调性的变化关系。
在两路RSCC的驱动信号同步时，RSCC-A和RSCC-B的谐振电感的电流的有效值分别为6.8A和24A，绝对值相差17.2A，RSCC-B的谐振电感的电流为RSCC-A的谐振电感的电流的3.5倍，差异十分显著。
从图12可以看出，固定RSCC-A对应的驱动信号的相位固定不变，逐步增加RSCC-B对应的驱动信号的相位，使其滞后RSCC-A的驱动信号的相位Φ(RSCC-A相比RSCC-B则为超前Φ)，两路RSCC的谐振电感的电流逐步一致。当动态可调角度Φ到12.5°时，两路的谐振电流基本一致，即实现RSCC-A与RSCC-B两路的谐振电感的电流相等，实现两路RSCC的均流。本实施例中通过检测谐振电感的电流来表征两路RSCC的电流，由于谐振电感的电流比较方便检测，例如检测磁器件的电流的检测电路或者传感器均可以实现。
以上的数值仅是一种示例，具体实施时，可以根据实际的应用场景，谐振电容和谐振电感的参数以及RSCC电路的应用场合来获得对应的谐振电流与动态可调角度的关系。从图中可以看出，两个曲线的交点处便是两路RSCC的电流相等时，该交点对应的动态可调角度便是两路RSCC的驱动信号之间的相位差。
另外，从图12中还可以看出，偏离两个电流曲线的交点之后，再逐渐增加动态可调角度，两路RSCC的谐振电感电流的有效值的差值会向相反方向增加。当动态可调角度增加到30°时，两路的电流差异性基本上达到极限值，如果进一步增加动态可调角度，两路RSCC之间的谐振电感电流可能呈现相反方向的变化，导致控制出现非单调性，从而失去控制能力。因此在实际应用中，可以对动态可调角度进行限幅，即需要限制动态可调角度的最大值。当动态可调角度达到预设阈值角度时，则保持在预设阈值角度，即控制器还用于在动态可调角度大于预设阈值角度时，控制相位差为预设固定角度与预设阈值角度之和，预设阈值角度是预先设定的动态可调角度的最大上限值。
预设阈值角度可以根据具体应用场景来进行测试，获得经验值，比如本实施例中预设阈值角度可以取为30°，本申请实施例不具体限定其获得方式。以上预设阈值角度取值时，仅是以两路RSCC为例，固定其中一路RSCC的驱动信号，控制另一路RSCC的驱动信号进行移相。如果两路RSCC的驱动信号均移相时，预设阈值角度可以取为30°/2＝15°。
综上所述，本申请实施例可以根据两路RSCC的谐振电感电流获得对应的动态可调角度，从而控制两路RSCC对应的驱动信号之间的相位差为预设固定角度和动态可调角度之和，从而实现两路RSCC的均流，在均流的前提下真正实现多路RSCC电路的有效并联，增加整个变换器的功率处理能力。另外，由于该方案是控制两路独立的RSCC之间移相，同时对于单个RSCC的驱动信号实行开环控制，因此，不影响单个RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。
变换器实施例二：
以上介绍的是两路RSCC电路中LC谐振电路连接在两个桥臂的中点之间，下面介绍另一种实现方式。
参见图13，该图为本申请实施例提供的另一种谐振开关电容变换器的示意图。
所述第一RSCC包括：第一桥臂、第二桥臂和第一LC谐振电路；所述第二RSCC包括：第三桥臂、第四桥臂和第二LC谐振电路；
第一桥臂的第一端和所述第三桥臂的第一端均连接所述谐振开关电容变换器的第一输入端，即BUS+，所述第一桥臂的第二端连接所述第二桥臂的第一端，所述第三桥臂的第二端连接所述第四桥臂的第一端，所述第二桥臂的第二端和所述第四桥臂的第二端均连接所述谐振开关电容变换器的第二输出端，即BUS-；
第一LC谐振电路的谐振电容Cra连接在所述第一桥臂的中点和所述第二桥臂的中点之间，所述第二LC谐振电路的谐振电容Crb连接在所述第三桥臂的中点和所述第四桥臂的中点之间；
所述第一LC谐振电路的谐振电感Lra连接在所述第一桥臂的第二端O1a和所述谐振 开关电容变换器的第二输入端BUSN(即O2a)之间；所述第二LC谐振电路的谐振电感Lrb连接在所述第三桥臂的第二端O1b和所述谐振开关电容变换器的第二输入端BUSN(即O2b)之间。
在图7中是第一桥臂的第二端连接与BUSN，但是在图13中并不是，而是第一桥臂的第二端与BUSN之间连接有谐振电感Lra。
需要说明的是，本实施例介绍的谐振电路中谐振电感的连接方式适用于本申请中其他所有的实施例。
虽然图13所示的变换器，谐振电感的连接方式发生了变化，但是不影响LC谐振电路的充放电路径，与图5和图6所示的充放电路径相同，在此不再赘述。
变换器实施例三：
以上实施例提供的两路RSCC电路均各个桥臂的开关模块均以可控开关管为例进行的介绍，下面介绍下桥臂，即输出桥臂的开关模块均为二极管的实现方式。
参见图14，该图为本申请实施例提供的第二桥臂和第四桥臂为二极管的示意图。
图14中BUS+和BUSN之间的第一桥臂为可控开关管S1A和S2A，同理第三桥臂为可控开关管S1B和S2B。
BUSN与BUS-之间的所述第二桥臂包括串联的第一二极管和第二二极管，即二极管D1A和D2A，能量从BUS+向BUS-转移，则D1A和D2A形成续流回路，即D1A的正极连接D2A的负极，D1A的负极连接第一桥臂和第二桥臂的公共点；D2A的正极连接BUS-。
同理，BUSN与BUS-之间的所述第四桥臂包括串联的第三二极管和第四二极管，即D1B和D2B，能量从BUS+向BUS-转移，即C1a向C2a进行能量转移，则D1B和D2B形成续流回路，即D1B的正极连接D2B的负极，D1B的负极连接第一桥臂和第二桥臂的公共点；D2B的正极连接BUS-。
需要说明的是，本实施例介绍的第二桥臂和第四桥臂上的开关模块均为二极管的实现方式适用于能量从BUS+向BUS-转移，从如果是能量从BUS-向BUS+转移，即C2a向C1a进行能量转移，则需要颠倒，即第一桥臂的开关模块和第三桥臂的开关模块均可以为二极管，而第二桥臂的开关模块和第四桥臂的开关模块需要为可控开关管。
参见图15，该图为本申请实施例提供的第一桥臂和第三桥臂为二极管的示意图。
下面为了介绍方便，仅以RSCC-A为例进行介绍，RSCC-B同理。
充电时，S2A闭合，S1A断开，BUS-和BUSN之间的能量向LC谐振电路转移，即给LC谐振电路充电。
放电时，S2A断开，S1A闭合，LC谐振电路的能量向BUS+和BUSN之间转移，即LC谐振电路放电。
此时，能量输出端对应的桥臂的两个开关模块为二极管，即D1A和D2A，为了与以上实施例统一理解，可以将能量输入端对应的桥臂统一定义为第一桥臂，而能量输出端的桥臂统一定义为第二桥臂，即第一桥臂上的两个开关模块需要为可控开关管，而能量输出端对应的桥臂仅是为了实现续流，其上的开关模块可以为不可控的二极管。但是如果为了实 现能量能够双向流动，则所有桥臂上的开关模块需要均设置为可控开关管。
如图所示，D1A的负极连接BUS+，D1A的正极连接D2A的负极，D2A的正极连接BUSN。同理，RSCC-B对应的输出桥臂包括D1B和D2B。D1B的负极连接BUS+，D1B的正极连接D2B的负极，D2B的正极连接BUSN。
本申请所有实施例中的可控开关管可以为IGBT，也可以为MOS管，即门极可控开关管即可，具体的实现形式不作限定。
变换器实施例四：
以上介绍的是第一RSCC的第一驱动信号和第二RSCC的第二驱动信号之间的预设固定角度为0时对于动态可调角度的控制，下面介绍第一驱动信号和第二驱动信号之间的预设固定角度为360°/N的情况，继续以N为2，即两路RSCC为例进行介绍，即预设固定角度为180°。
由于两路RSCC中的开关管采用180°的交错控制，可以有效降低滤波电容(C1a、C2a、C1b、C2b)上的电流，因此，可以使用较小的滤波电容，降低滤波电容所占用的体积。
参见图16，该图为本申请实施例提供的两路RSCC电路采用交错驱动信号的时序图。
继续结合图7介绍本实施例，当两路RSCC的驱动信号之间的动态可调角度Φ为0时，RSCC-A与RSCC-B中相同位置的开关管采用互补驱动信号进行驱动，如图16所示，第一桥臂包括第一开关管S1A和第二开关管S2A，所述第三桥臂包括第三开关管S1B和第四开关管S2B；
第一开关管S1A的驱动信号和第二开关管S2A的驱动信号互补，第三开关管S1B的驱动信号和第四开关管S2B的驱动信号互补；
从图16中可以看出，S1A与S1B对应的驱动信号正好反相，即错相180°，S1A闭合时，S1B断开；S1A断开时，S1B导通。而且两个谐振电路的电流方向也反向，即iLra和iLrb的方向相反。比较两个谐振电感的电流是指比较两个谐振电感的电流的峰值或有效值或平均值，本实施例不做限定，取决于实际控制需求。
以上也是以50％的占空比为例进行的介绍。如果对于各个开关管采用开环控制模式进行驱动。但是，如果电路的谐振参数之间存在差异性，例如谐振电感不同或谐振电容不同，则各个RSCC电路的谐振电感的电流仍会显著不同，具体可以如图16所示，两者的电流的绝对值差异较大。
下面结合附图详细介绍交错驱动时对应的均流控制策略。
参见图17，该图为本实施例提供的错相控制时对应的均流控制模型图。
为了解决两路RSCC之间电流的差异性，使其实现均流，可以采用与图10相同的控制策略，即一种是固定其中一路RSCC的驱动信号，控制另一路RSCC的驱动信号进行移相。另一种是将两路RSCC的驱动信号分别向相反的方向进行移相。
区别于图11中的移相方向，图17的移相方向正好相反。
下面继续以RSCC-A的驱动信号固定不变，对RSCC-B的驱动信号进行移相。
对于交错控制时，需要控制电流小的RSCC电路滞后移相，或者控制电流大的RSCC 电路超前移相。
如图18所示，RSCC-B超前移相的时序图。
如果RSCC-B的谐振电感电流大于RSCC-A的谐振电感电流，则超前移相RSCC-B的驱动信号动态可调角度Φ。
如图19所示，RSCC-B滞后移相的时序图。
对于交错控制时，即交错控制时控制器控制所述第一驱动信号的相位固定，在所述第二电流小于所述第一电流时，控制所述第二驱动信号的相位滞后移相动态可调角度；在所述第二电流大于所述第一电流时，控制所述第二驱动信号的相位超前移相动态可调角度。
例如，如果RSCC-B的谐振电感电流小于RSCC-A的谐振电感电流，则滞后移相RSCC-B中的驱动信号动态可调角度Φ。
对于交错180°控制时，由于RSCC-A与RSCC-B并联，因此固定RSCC-A的驱动信号的相位，移相RSCC-B的相位，与控制RSCC-B的相位，移相RSCC-A的相位效果是相同的。例如，当固定RSCC-B的相位时，如果RSCC-A的谐振电感电流大于RSCC-B的谐振电感电流，则超前移相RSCC-A的驱动信号动态可调角度Φ。如果RSCC-A的谐振电感电流小于RSCC-B的谐振电感电流，则滞后移相RSCC-A中的驱动信号动态可调角度Φ。
需要说明的是，图18和图19中的超前和滞后均是指的在交错180度的基础上的移相动态可调角度Φ。
第一驱动信号和第二驱动信号之间的预设固定角度为180°，因此第一驱动信号和第二驱动信号之间的相位差为180°+Φ。
下面结合附图介绍对于错相控制时，谐振电流与动态可调角度的单调关系。
参见图20，该图为本申请实施例提供的交错控制时谐振电流与动态可调角度的曲线图。
其中，横坐标为RSCC-B的驱动信号相对RSCC-A的驱动信号的相位滞后的动态可调角度，单位为度，纵坐标为谐振电流的有效值，单位为A。
在与实施例一相同的离散参数下，RSCC-A的驱动信号与RSCC-B的驱动信号交错180°控制时，RSCC-A的谐振电感的电流有效值为19.1A，RSCC-B的谐振电感的电流有效值为9.1A，两者相差10A，差异低于非交错控制下的17.2A，但两者相差仍十分悬殊。同时对比图20和图21，交错控制和非交错控制对两路RSCC电流的影响刚好相反。
当增加RSCC-A的驱动信号与RSCC-B的驱动信号之间的动态可调角度时，滞后移相RSCC-B(或超前移相RSCC-A)到16°附近时，两者的谐振电感的电流有效值基本相同，实现均流控制。
以上所有实施例中，驱动信号的超前和滞后均是相对概念，本质上是控制并联的两路RSCC的驱动信号之间的动态可调角度，并且根据谐振电感的电流检测情况进行动态的调整，以达到闭环自动调整的目的。当固定一路的驱动信号的相位，移相另一路的驱动信号的相位时，也可以固定RSCC-B路的驱动信号，来移相RSCC-A路的驱动信号。
需要说明的是，交错控制时的均流适用于以上其他电路的拓扑，例如输出桥臂对应二极管的实现形式，如图7所示。同理，也适用于谐振电感位置变化的实现形式，如图13所示。
变换器实施例五：
以上介绍的实施例均是能量从正母线BUS+向负母线BUS-转移，另外，本申请实施例提供的DC/DC变换器，可以为双向变换器，即能量能够反向流动，即从负母线BUS-向正母线BUS+转移。
但是，既然该DC/DC变换器为双向的，对应的所有桥臂上的开关器件需要为可控开关管，即可以通过控制其开关状态来实现不同方向的能量流动。
参见图21，该图为本申请实施例提供的双向变换的谐振开关电容变换器的示意图。
本实施例中继续以两路RSCC为例进行说明。
由于能量可以双向流动，因此，所有桥臂的所有开关器件均为可控开关管，如图所示，RSCC-A的第一桥臂包括可控开关管S1A和S2A，RSCC-A的第二桥臂包括可控开关管S3A和S4A，并且四个可控开关管均包括反并联二极管。
同理，RSCC-B的第三桥臂包括可控开关管S1B和S2B，RSCC-B的第四桥臂包括可控开关管S3B和S4B，并且四个可控开关管也均包括反并联的二极管。
图21所示的拓扑，能量既可以从C1a向C2a转移，又可以从C2a向C1a转移。同理，能量既可以从C1b向C2b转移，又可以从C2b向C1b转移。由于RSCC-A与RSCC-B并联，因此两路RSCC能量转移时的方向是相同的。
以上实施例介绍均是以两电平的谐振开关电容变换器为例进行的介绍，下面介绍多电平的谐振开关电容变换器，以上实施例介绍的均流控制方式同样适用于多电平的谐振开关电容变换器。下面继续以两路RSCC并联为例进行介绍。
变换器实施例六：
以上介绍的实施例均是以两路RSCC并联为例时的均流控制，下面介绍多路RSCC并联时的均流控制。
参见图22，该图为本申请实施例提供的多路RSCC形成的谐振开关电容变换器的示意图。
本实施例提供的谐振开关电容变换器包括并联的N路RSCC，分别为RSCC-A、RSCC-B直到RSCC-N。N为大于等于3的整数。
其中RSCC-A与RSCC-B的结构和连接关系与图5和图6所示的完全相同，在此不再赘述。而且RSCC-N的结构和内部连接关系与RSCC-A也相同。
下面主要介绍N路RSCC并联时的均流控制。
参见图23，该图为与图22对应的均流控制模型图。
继续以谐振电感的电流来表征谐振电路的电流。
当N路RSCC并联时，需要检测各路RSCC的谐振电感的电流，并通过算数平均获得N路RSCC的电流平均值，即控制器获得所述N路RSCC电路的谐振电路的电流平均值；固定其中一路RSCC电路的驱动信号的相位，将剩余的N-1路的谐振电路的电流分别与所 述电流平均值进行比较，根据各自的比较结果获得各自的动态可调角度，根据所述各自的动态可调角度对其驱动信号进行移相。即N-1路RSCC根据自身的谐振电感的电流与电流平均值的差值进行闭环控制，从而实现N路的RSCC之间的均流控制。
具体控制时，可以继续采用固定一路RSCC的驱动信号的相位，对其余N-1路RSCC的驱动信号进行移相控制，例如固定RSCC-A的驱动信号的相位，对RSCC-B至RSCC-N的谐振电路的电流分别与平均电流值进行比较，获得各路对应的差值，根据各路的差值分别对各路进行对应的闭环控制，即通过动态调整RSCC-B至RSCC-N中驱动信号的动态可调角度，实现各路RSCC之间的均流控制。
需要说明的是，对于多路RSCC并联的均流控制，也包括以上实施例介绍的两大类控制，即各路之间的驱动信号采用非交错控制，还是交错控制，可以根据非交错控制还是交错控制来选择超前还是滞后对应的动态可调角度，具体实现方式与以上实施例的类似，在此不再赘述。需要说明的是，当N路并联时，交错控制的常采用360°/N错相的方式实现。
方法实施例一：
基于以上实施例提供的一种谐振开关电容变换器、光伏设备及光伏发电系统，本申请实施例还提供一种均流控制方法，下面结合附图进行详细介绍。
参见图24，该图为本申请实施例提供的谐振开关电容变换器的均流控制方法流程图。
本实施例提供的均流控制方法，应用于以上实施例提供的谐振开关电容变换器，具体可以参见图5-图7等所示的电路图。
该方法包括：
S2701：获得第一RSCC的第一电流，获得所述第二RSCC的第二电流；
获得第一RSCC的第一电流可以通过获得第一LC谐振电路的第一电流来实现，获得第二RSCC的第二电流可以通过第二LC谐振电路的第二电流来实现。
本步骤不限定获得第一电流和第二电流的先后顺序，由于各个RSCC电路独立，所以获得各自的电流可以由各自对应的电流采样电路或电流传感器来完成，之间互不影响。
需要说明的是，本申请实施例中以谐振电感的电流表征谐振电路的电流，并且不限定获得谐振电感的电流的具体方式，可以应用获得磁器件的电流的任何方式来获得。
S2702：获得第一RSCC的第一电流和第二RSCC的第二电流的电流差值。
根据所述第一LC谐振电路的第一电流和所述第二LC谐振电路的第二电流的电流差值获得移相角；
移相角包括动态可调角度；其中，动态可调角度与电流差值正相关。
控制器，具体用于调整所述第一驱动信号和所述第二驱动信号之间的所述移相角中的所述动态可调角度，以使所述第一电流与所述第二电流一致。具体可以采用如下方式获得移相角：
获得第一电流和第二电流，对于第一电流和第二电流进行闭环调整控制，获得移相角中的动态可调角度。
具体地，获得所述第一电流和所述第二电流的差值，对所述差值进行闭环控制获得所 述移相角中的动态可调角度，
一般情况下，第一电流和第二电流的差值的绝对值越大，则动态可调角度越大，本实施例中可以获得谐振电感电流的有效值。另外，本申请实施例中不具体限定是第一电流减去第二电流，还是第二电流减去第一电流，因为两个RSCC电路并联，第一和第二仅是一个名称，没有实际的排序意义，可以互相颠倒，效果是完全相同的。闭环控制的是两路RSCC的谐振电流的差值，即为了实现两路RSCC的谐振电流相等，移相角表示的是两路之间的驱动信号的相对相位位移。
S2703：根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。
第一电流和第二电流的电流差值的绝对值在一定的误差范围内，则认为两者一致，即视为两者相等。
第一电流和第二电流相等，可为有效电流相等、平均电流相等或峰值电流相等，本实施例中不做限定，可根据实际需求进行电流的采样和闭环控制。
其中，理想情况下，当第一RSCC和第二RSCC的离散参数完全一致时，即两个谐振电路的谐振电流相等，不需要动态可调角度，即动态可调角度为0。
预设固定角度与两个谐振电路的谐振电流的大小没有任何关系，是提前设置的两路RSCC电路对应的驱动信号之间的固定角度，一旦设定便可以固定不变。例如预设固定角度可以为0，理想情况下，当动态可调角度为0时，两路RSCC的驱动信号同步。
本申请实施例中关注的是动态可调角度，即控制器调整第一驱动信号和第二驱动信号之间的移相角中的动态可调角度，以使第一电流与第二电流一致。
通过控制动态可调角度来实现各路RSCC之间的电流一致。
另外，预设固定角度还可以设置为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数。例如，当N为2时，即两路RSCC并联时，预设固定角度为180度。当N为3时，即三路RSCC并联时，预设固定角度为120度。以此类推，在此不再一一举例说明。
实际实施时，控制器控制第一驱动信号和所述第二驱动信号之间的相位差为所述移相角，具体调整第一驱动信号和第二驱动信号中的至少一个的相位，来达到所述相位差。
具体地，可以固定其中一个驱动信号的相位不变，调整另外一个驱动信号的相位。也可以调整两个驱动信号的相位，例如向相反方向调整两个驱动信号的相位，来实现以上的相位差。由于在均流之前，两个驱动信号之间的相位差就是预设固定角度，因此，实际调整时可以通过调整动态可调角度来实现两路RSCC的均流。
为了方便理解，下面以两个RSCC的驱动信号之间的预设固定角度为0为了进行介绍，为了使第一桥臂的第一驱动信号和第二桥臂的第二驱动信号之间的相位差为移相角，由于预设固定角度为0，因此控制两个驱动信号之间的相位差为动态可调角度，可以包括以下两种实现方式。
第一种：一个驱动信号固定不变，控制另一个驱动信号移相。
控制所述第一驱动信号的相位固定，控制所述第二驱动信号的相位移相所述移相角。
例如，控制RSCC-A对应的驱动信号的相位保持不变，控制RSCC-B对应的驱动信号进行移相。即控制第一驱动信号的相位固定，控制第二驱动信号的相位移相动态可调角度。由于RSCC-A与RSCC-B两路并联，因此，也可以控制RSCC-B对应的驱动信号的相位保持不变，控制RSCC-A对应的驱动信号进行移相。
具体地，控制RSCC-A中的第一驱动信号的相位向第一方向移相第一角度，控制RSCC-B中的第二驱动信号的相位向第二方向移相第二角度，所述第一角度和所述第二角度之和为所述动态可调角度，所述第一方向和所述第二方向相反。即由于两个驱动信号的移相方向相反，因此，越移相，则两个驱动信号之间的相位差越大，直到相位差为动态可调角度为止，停止移相。
当第一驱动信号和第二驱动信号采用非交错控制时，即RSCC-A和RSCC-B相同位置的开关管在不移相的前提下，对应的驱动信号同相位。此种情况控制所述第一驱动信号的相位固定，控制所述第二驱动信号的相位移相所述动态可调角度，具体包括：
控制所述第一驱动信号的相位固定，在所述第二电流小于所述第一电流时，控制所述第二驱动信号的相位超前移相动态可调角度；在所述第二电流大于所述第一电流时，控制所述第二驱动信号的相位滞后移相动态可调角度。
当所述第一驱动信号和所述第二驱动信号交错控制时，即RSCC-A和RSCC-B相同位置的开关管在不移相的前提下，以两路RSCC为例，N为2时，对应的驱动信号的相位错相180°，此种情况控制所述第一驱动信号的相位固定，控制所述第二驱动信号的相位移相所述移相角，具体包括：
控制所述第一驱动信号的相位固定，在所述第二电流小于所述第一电流时，控制所述第二驱动信号的相位滞后移相动态可调角度；在所述第二电流大于所述第一电流时，控制所述第二驱动信号的相位超前移相动态可调角度。
以上控制的更具体的细节可以参见变换器实施例中的详细描述，在此不再赘述。
第二种：两个驱动信号向相反的方向移相。
控制所述第一驱动信号的相位向第一方向移相第一角度；控制所述第二驱动信号的相位向第二方向移相第二角度；所述第一角度和所述第二角度之和为动态可调角度，所述第一方向和所述第二方向相反。
另外当动态可调角度大于预设阈值角度时，控制所述第一桥臂的第一驱动信号和第二桥臂的第二驱动信号之间的相位差为预设固定角度与预设阈值角度之和。
通过谐振电流与移相角的曲线图可以看出，偏离两个电流曲线的交点之后，再逐渐增加动态可调角度，两路RSCC的谐振电感电流的有效值的差值会向相反方向增加。当继续增加动态可调角度到，两路的电流差异性基本上达到极限值，如果进一步增加动态可调角度，两路RSCC之间的谐振电感电流可能呈现相反方向的变化，导致控制出现非单调性，从而失去控制能力。因此在实际应用中，可以对动态可调角度进行限幅，即需要限制动态可调角度的最大值。当动态可调角度达到上限值时，则取值上限值即可，将上限值设为预设阈值角度。此时，控制器在动态可调角度大于预设阈值角度时，控制相位差为预设固定 角度与预设阈值角度之和。
预设角度可以根据具体应用场景来进行测试，获得经验值，本申请实施例不具体限定其获得方式。
综上所述，本申请可以根据两路RSCC的谐振电感电流的差值获得对应的动态可调角度，从而控制两路RSCC对应的驱动信号之间的相位差为移相角，实现两路RSCC的均流，从而实现两路RSCC电路的有效并联，增加整个变换器的功率处理能力。
以上介绍的均流控制方法是以两路RSCC为例进行的介绍，下面介绍包括N路RSCC并联，N大于等于3的场景。
当包括N路RSCC电路并联时；N为大于等于3的整数；均流控制具体包括：
获得所述N路RSCC电路的谐振电路的电流平均值；同理，可以获得N路RSCC的LC谐振电路的谐振电感的电流。
固定其中一路RSCC电路的驱动信号的相位，将剩余的N-1路的谐振电路的电流分别与所述电流平均值进行比较，根据各自的比较结果获得各自的动态可调角度，根据所述各自的动态可调角度对其驱动信号进行移相。
具体控制时，可以继续采用固定一路RSCC的驱动信号的相位，对其余N-1路RSCC的驱动信号进行移相控制，例如固定RSCC-A的驱动信号的相位，对RSCC-B至RSCC-N的谐振电路的电流分别与平均电流值进行比较，获得各路对应的差值，根据各路的差值分别对各路进行对应的闭环控制，即通过动态调整RSCC-B至RSCC-N中驱动信号的动态可调角度，实现各路RSCC之间的均流控制。
本申请实施例提供的方法，可以根据两路RSCC的谐振电感电流获得对应的动态可调角度，从而控制两路RSCC对应的驱动信号之间的相位差为预设固定角度和动态可调角度之和，从而实现两路RSCC的均流，在均流的前提下真正实现多路RSCC电路的有效并联，增加整个变换器的功率处理能力。另外，由于该方案是控制两路独立的RSCC之间移相，同时对于单个RSCC的驱动信号实行开环控制，因此，不影响单个RSCC中开关管的软开关特性，从而降低开关损坏，提高功率转换效率。
以上实施例提供的方法不仅适用于以上实施例提供的谐振开关电容变换器的具体拓扑，还适用于其他拓扑的谐振开关电容变换器的拓扑，例如，包括其他拓扑和连接关系多路并联的RSCC电路均可以。以上实施例仅是以一路RSCC包括两个桥臂，每个桥臂包括一个开关器件为例进行的说明。以上提供的均流方法适用于具有其他电压比例转换的谐振开关电容变换器，只要谐振开关电容变换器包括多路RSCC并联即可。
方法实施例的其他具体工作原理可以参见以上对于变换器实施例的描述，在此不再赘述，方法实施例适用于的变换器的具体拓扑结构也参见以上变换器实施例对应的各种示意图。
应当理解，在本申请中，“至少一个(项)”是指一个或者多个，“多个”是指两个或两个以上。“和/或”，用于描述关联对象的关联关系，表示可以存在三种关系，例如，“A和/或B”可以表示：只存在A，只存在B以及同时存在A和B三种情况，其中A，B可以是单数或者复数。字符“/”一般表示前后关联对象是一种“或”的关系。“以下至少一项(个)”或其 类似表达，是指这些项中的任意组合，包括单项(个)或复数项(个)的任意组合。例如，a，b或c中的至少一项(个)，可以表示：a，b，c，“a和b”，“a和c”，“b和c”，或“a和b和c”，其中a，b，c可以是单个，也可以是多个。
以上所述，以上实施例仅用以说明本申请的技术方案，而非对其限制；尽管参照前述实施例对本申请进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本申请各实施例技术方案的精神和范围。

Claims (28)

  1. 一种光伏发电系统，其特征在于，包括：DC/DC变换器、谐振开关电容变换器、逆变器和控制器；
    所述DC/DC变换器的输入端连接光伏阵列；
    所述谐振开关电容变换器的第一输入端连接所述DC/DC变换器的正输出端；所述谐振开关电容变换器的第二输入端连接所述DC/DC变换器的负输出端；
    所述谐振开关电容变换器的第一输出端连接所述逆变器的中线，所述谐振开关电容变换器的第二输出端连接所述逆变器的负母线；所述谐振开关电容变换器包括以下至少两路并联在一起的谐振开关电容电路RSCC：第一RSCC和第二RSCC；
    所述控制器，用于根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。
  2. 根据权利要求1所述的系统，其特征在于，所述移相角与所述电流差值正相关。
  3. 根据权利要求1所述的系统，其特征在于，所述控制器，具体用于调整所述第一驱动信号和所述第二驱动信号中的至少一个的相位，来调整所述第一驱动信号和所述第二驱动信号之间的所述移相角。
  4. 根据权利要求2或3所述的系统，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；
    所述控制器，具体用于根据所述电流差值调整所述动态可调角度来对所述移相角进行调整。
  5. 根据权利要求4所述的系统，其特征在于，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
  6. 根据权利要求2或3所述的系统，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；
    所述控制器，具体用于根据所述电流差值在所述预设固定角度的基础上，调整所述动态可调角度来对所述移相角进行调整。
  7. 根据权利要求6所述的系统，其特征在于，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
  8. 根据权利要求5或7所述的系统，其特征在于，所述控制器，还用于当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。
  9. 根据权利要求8所述的系统，其特征在于，当所述控制器调整所述第一驱动信号和所述第二驱动信号中的一个驱动信号的相位来调整所述动态可调角度时，所述预设阈值角 度小于等于30°。
  10. 根据权利要求8所述的系统，其特征在于，当所述控制器调整所述第一驱动信号的相位和所述第二驱动信号的相位来调整所述动态可调角度时，所述预设阈值角度小于等于15°。
  11. 根据权利要求1-10任一项所述的系统，其特征在于，所述第一RSCC包括：第一桥臂、第二桥臂和第一LC谐振电路；所述第二RSCC包括：第三桥臂、第四桥臂和第二LC谐振电路；
    所述第一桥臂的第一端和所述第三桥臂的第一端均连接所述谐振开关电容变换器的第一输入端，所述第一桥臂的第二端和所述第三桥臂的第二端均连接所述谐振开关电容变换器的第二输入端；
    所述第二桥臂的第一端和所述第四桥臂的第一端均连接所述谐振开关电容变换器的第一输出端，所述第二桥臂的第二端和所述第四桥臂的第二端均连接所述谐振开关电容变换器的第二输出端；
    所述第一LC谐振电路连接在所述第一桥臂的中点和所述第二桥臂的中点之间，所述第二LC谐振电路连接在所述第三桥臂和所述第四桥臂的中点之间。
  12. 根据权利要求1-10任一项所述的系统，其特征在于，所述第一RSCC包括：第一桥臂、第二桥臂和第一LC谐振电路；所述第二RSCC包括：第三桥臂、第四桥臂和第二LC谐振电路；
    所述第一桥臂的第一端和所述第三桥臂的第一端均连接所述谐振开关电容变换器的第一输入端，所述第一桥臂的第二端连接所述第二桥臂的第一端，所述第三桥臂的第二端连接所述第四桥臂的第一端，所述第二桥臂的第二端和所述第四桥臂的第二端均连接所述谐振开关电容变换器的第二输出端；
    所述第一LC谐振电路的谐振电容连接在所述第一桥臂的中点和所述第二桥臂的中点之间，所述第二LC谐振电路的谐振电容连接在所述第三桥臂的中点和所述第四桥臂的中点之间；
    所述第一LC谐振电路的谐振电感连接在所述第一桥臂的第二端和所述谐振开关电容变换器的第二输入端之间；所述第二LC谐振电路的谐振电感连接在所述第三桥臂的第二端和所述谐振开关电容变换器的第二输入端之间。
  13. 根据权利要求1-10任一项所述的系统，其特征在于，所述第一桥臂至少包括串联的第一开关管和第二开关管，所述第三桥臂至少包括串联的第三开关管和第四开关管，所述第二桥臂至少包括串联的第五开关管和第六开关管；所述第四桥臂至少包括串联的第七开关管和第八开关管；
    或，
    所述第一桥臂包括串联的第一开关管和第二开关管，所述第三桥臂包括串联的第三开关管和第四开关管，所述第二桥臂至少包括串联的第一二极管和第二二极管，所述第四桥臂至少包括串联的第三二极管和第四二极管。
  14. 一种谐振开关电容变换器，其特征在于，包括控制器和以下至少两路并联在一起 的谐振开关电容电路RSCC：第一RSCC和第二RSCC；
    所述谐振开关电容变换器的第一输入端连接直流电源的正输出端；所述谐振开关电容变换器的第二输入端连接所述直流电源的负输出端；
    所述谐振开关电容变换器，用于将所述直流电源的电压进行变换后输出；
    所述控制器，用于根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。
  15. 根据权利要求14所述的变换器，其特征在于，所述控制器，具体用于根据所述电流差值调整所述第一驱动信号和所述第二驱动信号之间的所述移相角，以使所述第一电流与所述第二电流一致；所述移相角与所述电流差值正相关。
  16. 根据权利要求15所述的变换器，其特征在于，所述控制器，具体用于调整所述第一驱动信号和所述第二驱动信号中的至少一个的相位，来调整所述移相角。
  17. 根据权利要求15或16所述的变换器，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；
    所述控制器，具体用于根据所述电流差值调整所述动态可调角度来对所述移相角进行调整。
  18. 根据权利要求17所述的变换器，其特征在于，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
  19. 根据权利要求15或16所述的变换器，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；
    所述控制器，具体用于根据所述电流差值在所述预设固定角度的基础上，调整所述动态可调角度来对所述移相角进行调整。
  20. 根据权利要求19所述的变换器，其特征在于，所述控制器，当所述第二电流小于所述第一电流时，具体用于控制所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，具体用于控制所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
  21. 根据权利要求15-20任一项所述的变换器，其特征在于，所述控制器，还用于当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。
  22. 一种均流控制方法，其特征在于，应用于光伏系统，所述光伏系统包括：DC/DC变换器、谐振开关电容变换器和逆变器；所述DC/DC变换器的输入端连接光伏阵列；所述谐振开关电容变换器的第一输入端连接所述DC/DC变换器的正输出端；所述谐振开关电容变换器的第二输入端连接所述DC/DC变换器的负输出端；所述谐振开关电容变换器的第一输出端连接所述逆变器的中线，所述谐振开关电容变换器的第二输出端连接所述逆变器的负母线；所述谐振开关电容变换器包括以下至少两路并联在一起的谐振开关电容电路RSCC： 第一RSCC和第二RSCC；
    该方法包括：
    获得所述第一RSCC的第一电流，获得所述第二RSCC的第二电流；
    根据所述第一RSCC的第一电流和所述第二RSCC的第二电流的电流差值，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，以使所述第一电流与所述第二电流一致。
  23. 根据权利要求22所述的方法，其特征在于，所述移相角与所述电流差值正相关。
  24. 根据权利要求23所述的方法，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为0；
    调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，具体包括：
    调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度。
  25. 根据权利要求24所述的方法，其特征在于，所述调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度，具体包括：
    当所述第二电流小于所述第一电流时，调整所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，调整所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度。
  26. 根据权利要求23所述的方法，其特征在于，所述移相角为预设固定角度和动态可调角度之和，所述预设固定角度为360°/N，其中N为并联的所述RSCC的数量，所述N为大于1的整数；
    调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的移相角，具体包括：
    在所述预设固定角度的基础上，调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度来对所述移相角进行调整。
  27. 根据权利要求26所述的方法，其特征在于，所述调整所述第一RSCC的第一驱动信号和所述第二RSCC的第二驱动信号之间的所述动态可调角度来对所述移相角进行调整，具体包括：
    当所述第二电流小于所述第一电流时，调整所述第二驱动信号的相位滞后所述第一驱动信号的相位所述动态可调角度；在所述第二电流大于所述第一电流时，调整所述第二驱动信号的相位超前所述第一驱动信号的相位所述动态可调角度。
  28. 根据权利要求23-27任一项所述的方法，其特征在于，还包括：当所述动态可调角度大于预设阈值角度时，控制所述动态可调角度为所述预设阈值角度。
PCT/CN2020/115801 2020-09-17 2020-09-17 一种光伏系统、谐振开关电容变换器及控制方法 WO2022056761A1 (zh)

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