WO2018215754A1 - A dc-dc converter - Google Patents

A dc-dc converter Download PDF

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Publication number
WO2018215754A1
WO2018215754A1 PCT/GB2018/051383 GB2018051383W WO2018215754A1 WO 2018215754 A1 WO2018215754 A1 WO 2018215754A1 GB 2018051383 W GB2018051383 W GB 2018051383W WO 2018215754 A1 WO2018215754 A1 WO 2018215754A1
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WO
WIPO (PCT)
Prior art keywords
switching frequency
converter
frequency
switching
output voltage
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/GB2018/051383
Other languages
French (fr)
Inventor
Sarang Vadnerkar
Ullas PAZHAYAVEETIL
Theodore Burk
Jeffrey May
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Cirrus Logic International Semiconductor Ltd
Original Assignee
Cirrus Logic International Semiconductor Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Cirrus Logic International Semiconductor Ltd filed Critical Cirrus Logic International Semiconductor Ltd
Priority to CN201880047787.3A priority Critical patent/CN110945767B/en
Priority to GB1918612.1A priority patent/GB2578379B/en
Publication of WO2018215754A1 publication Critical patent/WO2018215754A1/en
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0222—Continuous control by using a signal derived from the input signal
    • H03F1/0227—Continuous control by using a signal derived from the input signal using supply converters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0244—Stepped control
    • H03F1/025—Stepped control by using a signal derived from the input signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/181—Low-frequency amplifiers, e.g. audio preamplifiers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0012—Control circuits using digital or numerical techniques
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • a DC-DC CONVERTER Technical Field Embodiments disclosed herein relate to methods and apparatus for adapting DC-DC converter switching frequencies to maintain operation outside of predefined disallowed frequency bands.
  • Portable devices are becoming more common in every aspect of our lives. For example, many consumers listen to their music from portable audio players. These portable devices become easier and more enjoyable to use when they are provided in small packages, offering the same capabilities while still having long lasting battery charge.
  • the operating voltage of many of the components inside is often reduced to reduce the power consumption.
  • this reduction in operating voltage may affect the sound output of the portable device as the volume level of an audio signal is proportional to the output voltage.
  • a boost converter may therefore be used to increase the operating voltage for select components within a portable device that would benefit from higher operating voltages, for example, audio amplifiers which may require high voltages to provide high volumes.
  • FIG. 1 illustrates a conventional audio amplifier comprising a boost converter.
  • the audio amplifier comprises a battery supply 102 which is configured to provide a voltage V p to a boost converter 104.
  • the boost converter 104 increases the voltage V p to a boost voltage Vbst.
  • a speaker amplifier 108 is configured to receive the boost voltage Vbst and output an analog audio signal sig from a digital to analog converter (DAC).
  • DAC digital to analog converter
  • the speaker amplifier increases the low power signal sig with the power received from the boost converter to generate a signal to drive a speaker 1 10.
  • the boost voltage Vbst is higher than the supply voltage V p which allows the speaker amplifier 108 to provide louder audio signals through the speaker 1 10 than would be possible using only the supply voltage V p .
  • Portable devices are becoming increasingly complex systems where the ability of a given integrated circuit to co-exist with a neighbouring integrated circuit (IC) on a printed circuit board or in a physically adjacent area may become increasingly important to the system or device designer. Interference between the operation of any two ICs may occur not necessarily due to a direct connection between the two, but due to electromagnetic interference produced by one in a frequency or frequency range where operation of the other is susceptible to interference.
  • a boost converter may for instance create a large switching voltage as a part the Vbst generation process. This switching voltage can often cause electromagnetic interference with other nearby ICs or other sub-systems on the portable device.
  • the boost converter may also create noise in the battery supply (V p ) each time it switches.
  • This switching can cause noise to be injected onto a power supply which may be shared by other ICs in a portable device. Some ICs are more susceptible than others to noise on the power supply and performance can suffer as a result. As the portable devices become increasingly complex, so do the requirements to avoid multiple frequency bands of operation.
  • a method for maintaining an output voltage of a DC-DC converter configured to operate in a discontinuous conduction mode, within a predetermined voltage range.
  • the method comprises adjusting a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
  • a DC-DC converter configured to in a discontinuous conduction mode and to maintain an output voltage within a predetermined voltage range.
  • the DC-DC converter comprises a controller, wherein the controller comprises an input configured to receive an indication of the output voltage; and an adjustment block configured to adjust a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
  • Figure 1 is an example of a conventional audio amplifier comprising a boost converter in accordance with the prior art
  • Figure 2 is an example block diagram of a DC-DC converter in accordance with the present disclosure
  • Figure 3a is an example timing diagram illustrating the operation of a boost converter in a continuous conduction mode, CCM, in accordance with embodiments of the present disclosure
  • Figure 3b is an example timing diagram illustrating the operation of a boost converter in a continuous conduction mode, CCM, in accordance with embodiments of the present disclosure
  • FIG. 4 is an example timing diagram illustrating the operation of a boost converter in discontinuous conduction mode, DCM, in accordance with embodiments of the present disclosure
  • Figure 5 is an example timing diagram illustrating the operation of the DC-DC converter in DCM, in accordance with embodiments of the present disclosure
  • Figure 6 is an example flow chart diagram illustrating a method for maintaining an output voltage of a DC-DC converter in a predetermined voltage range, while operating in a discontinuous conduction mode, in accordance with embodiments of the present disclosure
  • Figure 7 is an example timing diagram illustrating the operation of a DC-DC converter when switching from a first switching frequency to a second switching frequency, in accordance with embodiments of the present disclosure
  • Figure 8 is an example flow chart illustrating a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure
  • Figure 9 is an example flow chart illustrating a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure
  • Figure 10 is an example flow chart illustrating a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure
  • Figure 11 is an example flow chart illustrating a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure
  • Figure 12 is an example timing diagram illustrating the operation of a DC-DC converter in accordance with embodiments of the present disclosure
  • Figure 13 is an example timing diagram illustrating the NFET 206 conducting and non- conducting states as the duty cycle is decreased in accordance with embodiments of the present disclosure
  • Figure 14 is an example timing diagram illustrating the energy as a function of frequency for different frequencies using the same duty cycle in accordance with embodiments of the present disclosure
  • Figure 15 is an example graph illustrating the switching frequencies available to the boost converter when unconstrained by any predefined disallowed frequency band containing disallowed frequencies in accordance with embodiments of the present disclosure
  • Figure 16 is a graph illustrating an example where two predefined disallowed frequency bands have been set containing disallowed frequencies in accordance with the present disclosure
  • Figure 17 is a graph illustrating an example comprising a single predefined disallowed frequency band in accordance with the present disclosure
  • Figure 18 is graph illustrating an example comprising three predefined disallowed frequency bands in accordance with the present disclosure.
  • Figure 19 is graph illustrating an example comprising four predefined disallowed frequency bands in accordance with the present disclosure.
  • Figure 20 is an example block diagram of a DC-DC converter 1800 in accordance with the present disclosure.
  • FIG. 2 illustrates an example DC-DC converter 200 in accordance with some embodiments of the present disclosure.
  • the DC-DC converter 200 comprises a battery supply 202 which provides a supply voltage VP to the DC-DC converter 200.
  • the DC-DC converter 200 comprises an inductor 204 connected to the battery supply 202.
  • the DC- DC converter 200 further comprises two switches 206 and 208 and a capacitor 210.
  • the switches 206 and 208 are in this example implemented as an n-channel field effect transistor (NFET) 206 and a p-channel field effect transistor (PFET) 208.
  • NFET n-channel field effect transistor
  • PFET p-channel field effect transistor
  • the inductor 204 is coupled to the drain of the PFET 208 and the drain of the NFET 206.
  • the drain of the PFET 208 and the drain of the NFET 206 are also coupled together.
  • the capacitor 210 is coupled between the source of the PFET 208 and ground.
  • the source of the NFET 206 is coupled to ground.
  • Both the gate of the PFET 208 and the gate of the NFET 206 are coupled to a level shifting and driving block 214 which controls whether the PFET and NFET are in conducting or non-conducting states.
  • An amplifier or other signal processing circuitry may be connected to receive the boost voltage from the source of the PFET 208.
  • the NFET 206 may be switched on to a conducting state which couples the inductor 204 to ground, while the PFET 208 is switched off to a non-conducting state.
  • This switching configuration causes the supply voltage VP to fall across the inductor 204 and therefore cause current to flow through the inductor 204 thereby storing energy in the inductor.
  • the NFET 206 may be switched off into a non-conducting state while the PFET 208 may be switched on to a conducting state.
  • This switching configuration causes the inductor to couple to ground through the capacitor 210.
  • the energy stored in the inductor 204 during the first time period is transferred to the capacitor 210 during the second time period.
  • the voltage across the capacitor increases and allows the boost voltage Vbst to increase beyond the supply voltage Vp.
  • a controller 222 may control the duration of the first and second time periods in order to regulate the transfer of power from the inductor to the capacitor and thus regulate the level of the boost voltage Vbst.
  • FIG. 3a illustrates the operation of a boost converter in a continuous conduction mode, CCM.
  • the boost converter operates with two switching states Si and S2 per switching cycle.
  • Si NFET 206 is in a conducting state
  • PFET 208 is in a non-conducting state.
  • the current is drawn across the inductor storing energy in the conductor.
  • switching state S2 as described previously, the PFET 208 is in a conducting state and the NFET 206 is in a non- conducting state.
  • the energy stored in the inductor is transferred to the capacitor and the current through the inductor drops.
  • the controller may control the length of the duty cycle in order to maintain the output voltage within a predetermined range. In other words, if Vbst goes too high, the time spent in state Si may be reduced and if Vbst drops below a minimum voltage V m in, then the time spent in state Si may be increased.
  • the switching state is switched from state Si to state S2 when the current across the inductor reaches a maximum value.
  • the switching state may be switched from state S2 to state Si at the beginning of each clock cycle as illustrated by the begin switching frequency cycle pulses in Figure 3a.
  • the boost converter may therefore be operated in discontinuous conduction mode (DCM).
  • DCM discontinuous conduction mode
  • States Si and S2 are the same as illustrated for CCM in Figures 3a and 3b, but in third state S3, both the NFET 206 and the PFET 208 are off.
  • the DC-DC converter is configured to maintain the boost voltage VBST within the predetermined voltage range between a maximum voltage VM 3X and a minimum voltage VMIN. Therefore, when the voltage goes above the maximum voltage VMAX, the DC-DC converter may decrease the duty cycle of the DC-DC converter. In other words, the controller 222 illustrated in Figure 2 may decrease the amount of time spent in switching state Si . This time decrease will reduce the energy stored in the inductor during the switching period, and therefore reduce the energy transferred to the capacitor, thereby reducing the boost voltage VBST. Equivalently, when the voltage drops below the minimum voltage VMIN, the DC-DC converter may increase the duty cycle of the DC-DC converter. In other words, the controller 222 illustrated in Figure 2 may increase the amount of time spent in switching state Si . This time increase will increase the energy stored in the inductor during the switching period, and therefore increase the energy transferred to the capacitor, thereby increasing the boost voltage VBST.
  • reduce the switching frequency By reducing the switching frequency, the time spent in the state S3 can be increased, thereby saving more power by not wasting current across the inductor.
  • Figure 6 illustrates a method for maintaining an output voltage of a DC-DC converter in a predetermined voltage range, while operating in a discontinuous conduction mode, for example as illustrated in Figure 4.
  • step 601 the DC-DC converter determines the value of the output voltage of the DC- DC converter.
  • the DC-DC converter adjusts a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range.
  • the duty cycle of the DC-DC converter may be adjusted by switching between a first switching frequency and a second switching frequency.
  • the first switching frequency and the second switching frequency may be selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
  • at least one predefined disallowed frequency band may be in place which comprises frequencies which should not be used by the DC-DC converter.
  • one or more frequencies in the at least one predefined disallowed frequency band may be used for integrated circuits with some other application within a device comprising the DC-DC converter.
  • the electromagnetic interference (EMI) with switching frequencies within the predefined disallowed frequency bands is unpredictable, and therefore it may be desirable to avoid using the frequencies within the predefined disallowed frequency bands. Therefore, the embodiments disclosed herein allow the boost converter's switching frequency and duty cycle to be manipulated in order to avoid electromagnetic interference or power supply noise injection which can be problematic for other sub-systems or ICs within the portable device.
  • EMI electromagnetic interference
  • the frequencies may also be selected such that the time period of the switching frequencies are multiples of the time period of the clock frequency used by the DC-DC converter. In this way, the switching periods of the lower frequencies will comprise an integer number of clock periods.
  • a DC-DC converter utilizing a clock frequency of X kHz. This high frequency may be used as the highest switching frequency available to the DC-DC converter.
  • the second switching frequency may then be X/2 kHz.
  • a third switching frequency of X/3 kHz may also be available, and so on.
  • the time period of the first switching frequency which is equal to the clock frequency, will be 1/X seconds.
  • the time period of the second frequency will be 2/X seconds, and the time period of the third switching frequency will be 3/X seconds, and so on.
  • These switching frequencies may therefore comprise an integer number of clock cycles.
  • the DC-DC converter effectively skips a switching period as illustrated in Figure 7.
  • Figure 7 illustrates the operation of a DC-DC converter when switching from a first switching frequency equal to the clock frequency, to a second switching frequency.
  • the DC-DC converter is operating with the first switching frequency.
  • the DC- DC converter decides to lower the switching frequency, potentially due to the boost voltage VBST being above the maximum voltage threshold VMAX as illustrated in Figure 5.
  • the DC-DC converter may then switch to the second switching frequency for the 3 RD to the 6 TH clock cycles. It can be seen that the second switching frequency effectively skips the 4 TH clock cycle and the 6 TH clock cycle by staying the third switching state S3 and pinning the inductor current to zero.
  • Figure 8 illustrates a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
  • the embodiment shown in figure 8 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first decreasing the duty cycle by decreasing the on time of the DC- DC converter until a minimum duty cycle for the first switching frequency is reached, and responsive to the output voltage remaining above the predetermined voltage range when using the first switching frequency, switching to the second switching frequency.
  • step 801 the DC-DC converter determines whether the boost voltage VBST is above the maximum voltage threshold VMAX. If the boost voltage VBST is below the threshold VMAX the method returns to the start. If in step 801 the DC-DC converter determines that the boost voltage VBST is above the maximum threshold VMAX, the method passes to step 802 in which the DC-DC converter determines whether an on time Tsi that the DC-DC converter in the state Si , is greater than a minimum value, TSI MIN.
  • An on time of the DC-DC converter may be considered a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time may be understood as the period that the DC-DC convertor remains in the state Si , i.e. the time for which the NFET is in a conducting state.
  • the minimum on time TSI MIN may be a minimum duty cycle for the frequency at which the DC-DC converter is operating. If in step 802 the DC-DC converter determines that the on time Tsi of the DC-DC converter is greater than TSI MIN, then the method passes to step 803 in which the DC- DC converter reduces the on time of the DC-DC converter. The method will then return to the start. In some embodiments, the on time of the DC-DC converter may be reduced by a predetermined increment.
  • step 802 If in step 802 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not greater than the minimum on time TSI MIN, the method then passes to step 804 in which the DC-DC converter determines whether the switching frequency Fs is greater than a minimum switching frequency FSMIN.
  • step 804 determines that the switching frequency Fs is greater than the minimum switching frequency FSMIN
  • the method passes to step 805 in which the DC-DC converter reduces the switching frequency.
  • the switching frequency may be decreased by increasing the time period of the switching frequency by one clock period. The method may then pass back to the start. In other words, by decreasing the switching frequency the DC-DC convertor switches from the first switching frequency to the second switching frequency. If in step 804 the DC-DC converter determines that the switching frequency F s is not greater than the minimum switching frequency, the method may end in step 806.
  • steps 801 to 805 may be performed iteratively until either the voltage has dropped below the maximum threshold VMAX or both the minimum on time TSI MIN and the minimum switching frequency FSMIN have been reached.
  • Figure 9 illustrates a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
  • the embodiment illustrated in figure 9 performs the steps of, responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, first increasing the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the second switching frequency is reached, and responsive to the output voltage remaining below the predetermined voltage range when using the second switching frequency, switching to the first switching frequency.
  • step 901 the DC-DC converter determines whether the boost voltage VBST is below the minimum voltage threshold VMIN. If the boost voltage VBST is above the minimum threshold VMIN the method returns to the start.
  • step 901 the DC-DC converter determines that the boost voltage VBST is below the minimum threshold VMIN
  • the method passes to step 902 in which the DC-DC converter determines whether an on time of the DC-DC converter, Tsi , is less than a maximum value, TSI MAX.
  • An on time of the DC-DC converter may be defined as a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time may be understood as the period during which the DC-DC converter is in state Si , i.e. the time for which the NFET is in a conducting state.
  • the maximum on time TSI MAX may be a maximum duty cycle for the frequency at which the DC-DC converter is operating.
  • step 902 determines that the on time Tsi of the DC-DC converter is less than TSI MAX
  • the method passes to step 903 in which the DC-DC converter increases the on time Tsi of the DC-DC converter.
  • the method will then return to the start.
  • the on time Tsi of the DC-DC converter may be increased by a predetermined increment. If in step 902 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not less than the maximum on time Tsi MAX, the method then passes to step 904 in which the DC-DC converter determines whether the switching frequency Fs is less than a maximum switching frequency FSMAX.
  • step 904 the DC-DC converter determines that the switching frequency Fs is less than the maximum switching frequency FSMAX, the method passes to step 805 in which the DC-DC converter increases the switching frequency.
  • the switching frequency may be increased by decreasing the time period of the switching frequency by one clock period. The method may then pass back to the start. In other words, by increasing the switching frequency, the DC-DC convertor switches from the second switching frequency to the first switching frequency.
  • step 904 the DC-DC converter determines that the switching frequency F s is not less than the maximum switching frequency FSMAX, the method may end in step 806. It will be appreciated that the steps 901 to 905 may be performed iteratively until either the voltage has risen above the minimum threshold VMIN or both the maximum on time Tsi MAX and the maximum switching frequency FSMAX have been reached.
  • Figure 10 illustrates a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
  • the embodiment shown in Figure 10 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switching to the second switching frequency; and responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
  • step 1001 the DC-DC converter determines whether the boost voltage VBST is above the maximum voltage threshold VMAX. If the boost voltage VBST is below the threshold VMAx. the method returns to the start. If in step 1001 the DC-DC converter determines that the boost voltage VBST is above the maximum threshold VMAX, the method passes to step 1002 in which the DC-DC converter determines whether the switching frequency Fs is greater than a minimum switching frequency FSMIN.
  • step 1002 the DC-DC converter determines that the switching frequency Fs is greater than the minimum switching frequency FSMIN, the method passes to step 1003 in which the DC-DC converter reduces the switching frequency.
  • the switching frequency may be decreased by increasing the time period of the switching frequency by one clock period. In other words, by reducing the switching frequency the DC-DC convertor switches from using the first switching frequency to using a second switching frequency. The method may then pass back to the start.
  • step 1004 the DC-DC converter determines whether an on time Tsi during which the DC-DC converter is in state Si , is greater than a minimum value, TSI MIN.
  • An on time Tsi of the DC-DC converter may be considered a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time Tsi may be understood as the period during which the DC-DC converter is in state Si, i.e. the time for which the NFET is in a conducting state.
  • the minimum on time TSI MIN may be a minimum duty cycle for the frequency at which the DC-DC converter is operating.
  • step 1004 the DC-DC converter determines that the on time Tsi of the DC-DC converter is greater than TSI MIN, then the method passes to step 1005 in which the DC- DC converter reduces the on time of the DC-DC converter. The method will then return to the start.
  • the on time of the DC-DC converter may be reduced by a predetermined increment.
  • step 1004 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not greater the TSI MIN, the method may end in step 1006. It will be appreciated that the steps 1001 to 1005 may be performed iteratively until either the voltage has dropped below the maximum threshold VMAX or both the minimum on time TsiMiN and the minimum switching frequency FSMIN have been reached. In some embodiments the switching frequency may be decreased from a starting frequency F ST ART by increasing N by one in the following equation:
  • Figure 11 illustrates a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
  • the embodiment shown in Figure 1 1 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switching to the second switching frequency; and responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
  • step 1 101 the DC-DC converter determines whether the boost voltage VBST is below the minimum voltage threshold VMIN. If the boost voltage VBST is above the minimum threshold VMIN, the method returns to the start.
  • step 1 101 If in step 1 101 the DC-DC converter determines that the boost voltage VBST is below the minimum threshold VMIN, the method passes to step 1 102 in which the DC-DC converter determines whether the switching frequency Fs is less than a maximum switching frequency FSMAX.
  • step 1 102 the DC-DC converter determines that the switching frequency Fs is less than the maximum switching frequency FSMAX, the method passes to step 1 103 in which the DC-DC converter increases the switching frequency.
  • the switching frequency may be increased by decreasing the time period of the switching frequency by one clock period. In other words, by increasing the switching frequency the DC-DC converter switches from using the first switching frequency to the second switching frequency. The method may then passes back to the start.
  • step 1 102 the method passes to step 1104 in which the DC-DC converter the DC-DC converter determines whether an on time Tsi during which the DC-DC converter is in state Si , is less than a maximum value, TSI MAX.
  • An on time Tsi of the DC-DC converter may be defined as a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this may be understood as the period during which the DC-DC converter is in state Si, i.e. the time for which the NFET is in a conducting state.
  • the maximum on time TSI MAX may be a maximum duty cycle for the frequency at which the DC-DC converter is operating.
  • step 1104 the DC-DC converter determines that the on time Tsi of the DC-DC converter is less than TSI MAX, then the method passes to step 1105 in which the DC-DC converter increases the on time Tsi of the DC-DC converter. The method will then return to the start. In some embodiments, the on time Tsi of the DC-DC converter may be increased by a predetermined increment.
  • step 1104 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not less than the maximum on time Tsi MAX, the method may end in step 1 106.
  • the steps 1101 to 1105 may be performed iteratively until either the voltage has risen above the minimum threshold VMIN or both the maximum on time Tsi MAX and the maximum switching frequency FSMAX have been reached.
  • the switching frequency may be increased from a starting frequency F ST ART by decreasing N by one in the following equation:
  • FIG. 12 illustrates an example of the operation of a DC-DC converter according to some embodiments.
  • the DC-DC converter monitors the output voltage VBST, for example, as illustrated in Figure 2, the controller 222 may receive an indication of the output voltage VBST via an analog-to-digital converter 220. The controller 222 may then adjust the duty cycle of the DC-DC converter accordingly.
  • the controller 222 if at a time ti the boost voltage VBST goes above a maximum threshold voltage VMAX, the controller 222 be configured to determine whether to decrease the duty cycle.
  • the DC-DC converter is configured to decrease the duty cycle responsive to the boost voltage VBST remaining above the maximum voltage VMAX for a predetermined time.
  • the controller 222 may comprise a filter counter configured to count clock cycles from when the voltage first passes above the maximum threshold VMAX at a time ti , until the amplitude passes back below the maximum threshold at a time t2.
  • the filter counter has not yet counted a predetermined number X of clock cycles. The DC-DC converter may therefore not decrease the duty cycle. This manner of control avoids unnecessary decreasing of the duty cycle due to transient peaks in the boost voltage VBST.
  • the boost voltage VBST passes back above the maximum threshold VMAX, this time for long enough for the filter counter to count X clock cycles twice.
  • a wait counter is also implemented where the filter counter waits a predetermined time between counting each set of X clock cycles. The controller therefore decreases the duty cycle twice after each time the filter counter counts X clock cycles.
  • the boost voltage passes below the maximum threshold at time before the filter counter can count X clock cycles for a third time, and therefore the duty cycle is not decreased a third time.
  • the duty cycle may be decreased as described with respect to Figure 8 or 10 by decreasing the on time of the DC-DC converter and/or the switching frequency.
  • the DC-DC converter may be equivalent ⁇ configured to increase the duty cycle responsive to the boost voltage remaining below the minimum voltage VMIN for a predetermined time.
  • the controller 222 may comprise a filter counter configured to count clock cycles from when the voltage first passes below the minimum threshold VMIN to when the voltage passes back above the threshold VMIN. Again, if the filter counter does not reach a predetermined number of clock cycles Y (where Y may or may not be equal to X) in this time, the controller may be configured not to increase the duty cycle. This manner of control avoids unnecessary increasing of the duty cycle due to transient troughs in the boost voltage VBST.
  • the duty cycle may be increased as described with respect to Figure 9 or 1 1 by increasing the on time of the DC-DC converter and/or the switching frequency.
  • Figure 13 illustrates the NFET 206 conducting and non-conducting states as the duty cycle is decreased.
  • the duty cycle is decreased according to the method described in Figure 8, but it will be appreciated that the duty cycle may be increased or decreased according to any of the methods described above.
  • the duty cycle is 3x/Ti .
  • the duty cycle is then decreased to 2x/Ti by decreasing the on time of the DC-DC converter from 3x to 2x.
  • the duty cycle is then decreased to x/Ti by decreasing the on time of the DC-DC converter from 2x to x.
  • the duty cycle is then decreased to x/(2Ti) by decreasing the switching frequency from 1 Ti to 1/(2Ti). Again, the duty cycle may be further decreased from x/(2Ti) to x/(3Ti) by decreasing the switching frequency from 1/(2Ti) to 1/(3Ti).
  • Figure 14 illustrates the energy as a function of frequency for different frequencies using the same duty cycle.
  • FIG. 14a Another possible reason to alter the switching frequency of the DC-DC converter may be to change the concentration of the energy within the frequency band as illustrated in Figure 14.
  • Figure 14b the NFET on and off times are illustrated.
  • the duty cycle is x/Ti and the switching frequency is 1/ ⁇ .
  • the graph 1300 illustrates how the energy is therefore concentrated around the frequency 1 Ti .
  • the duty cycle is 2x/2Ti which is equivalent to x/Ti , in other words, the same as in Figure 13a.
  • the switching frequency is 1/(2Ti).
  • the graph 1401 therefore illustrates how the energy is now concentrated around the lower frequency of
  • the energy in the frequency band may be altered by changing the switching frequency, as illustrated in Figure 14. If, for example, a concentration of energy is desired that is concentrated around a frequency 1 Ti but in this example 1 Ti is not an allowed switching frequency, in other words the frequency 1 Ti falls within the predefined disallowed frequency band, then two or more allowable switching frequencies which do not fall within the predefined disallowed frequency band, falling either side of the desired switching frequency, may be utilized to create the same energy distribution around the switching frequency 1 Ti .
  • Figure 15 illustrates the switching frequencies available to the boost converter when unconstrained by any predefined disallowed frequency band containing disallowed frequencies.
  • the boost converter has a minimum and maximum possible switching frequency.
  • MINO is the minimum switching frequency of the boost converter when allowed to operate without any predefined disallowed frequency bands containing disallowed frequencies.
  • fMAxo is the maximum switching frequency of the boost converter when allowed to operate without any predefined disallowed frequency bands containing disallowed frequencies.
  • Figure 16 illustrates an example where two predefined disallowed frequency bands have been set containing disallowed frequencies.
  • a first predefined disallowed frequency band between MINO and a frequency fiuim is introduced and a second predefined disallowed frequency band between fi «Axi and fiviAxo is introduced.
  • the values fiMiNi and fiMAxi are then set as maximum and minimum switching frequencies for the boost converter in order to ensure that the boost converter does not use switching frequencies in either of the predefined disallowed frequency bands.
  • the predefined disallowed frequency bands sit at either end of the allowed switching frequencies.
  • Figure 17 illustrates an example comprising a single predefined disallowed frequency band has been set containing disallowed frequencies.
  • the predefined disallowed frequency band falls between the frequencies fi «Axi and fiuim .
  • the frequencies between fiuAxi and fiuiNi may therefore be considered as disallowed.
  • Two frequency bands are provided that the boost converter may use between MINO and fiuAxi and MIN2 and fMAxo. In this example therefore, the boost converter may select a frequency between fiuiNo and fiMAxi and fMiN2 and fMAxo.
  • the boost converter may select a frequency between fiuiNo and fiMAxi and a different frequency between MIN2 and fiviAxo and may then alternate between the two switching frequencies in such a way that the energy may be concentrated between the frequencies fiuAxi and fMiN2.
  • the DC-DC converter may be configured to alternate between a first switching frequency and a second switching frequency wherein the first switching frequency is higher than the predefined disallowed frequency band and the second switching frequency is lower than the predefined disallowed frequency band.
  • Figure 18 illustrates an example comprising three predefined disallowed frequency bands.
  • a first predefined disallowed frequency band falls between fiuiNo and a frequency fiuiNi and a second frequency bend falls between fMAX2 and fMAxo.
  • These disallowed frequency bands are equivalent to those illustrated in Figure 16 in that they are at the ends of the possible frequencies that the boost converter is capable of using.
  • the predefined disallowed frequency band between fiuAxi and fMiN2 is also disallowed. In this example, therefore only the frequencies between fiuim and fiMAxi and the frequencies between fiuiN2 and fMAX2 may be used by the boost converter.
  • the boost converter may alternate between a switching frequency between fiuiNi and fiuAxi and a switching frequency between fiuiN2 and fMAX2 in order to concentrate the energy between the frequencies fiuAxi
  • Figure 19 illustrates an example comprises four predefined disallowed frequency bands.
  • the frequencies between fiuiNo and f MINI between fi «Axi and fiwiN2, between fiMAX2 and fMiN3 and between fMAX3 and fMiN4 are disallowed.
  • the boost convertor may alternate between frequencies outside of these predefined disallowed frequency bands containing disallowed frequencies to concentrate the energy within the predefined disallowed frequency bands without actually using the frequencies within those predefined disallowed frequency bands.
  • FIG. 20 illustrates a DC-DC converter 2000 according to some embodiments.
  • the DC- DC converter 2000 comprises a controller 2001.
  • the controller 2001 comprises an input 2002 configured to receive an indication of the output voltage.
  • the controller 2001 further comprises an adjustment block 2003 configured to adjust a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range.
  • the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency.
  • the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
  • the at least one predefined disallowed frequency band may comprise disallowed frequencies.

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Abstract

A DC-DC converter and a corresponding method for maintaining an output voltage of the DC-DC converter, wherein the DC-DC converter is configured to operate in a discontinuous conduction mode, within a predetermined voltage range. The method comprises adjusting a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.

Description

A DC-DC CONVERTER Technical Field Embodiments disclosed herein relate to methods and apparatus for adapting DC-DC converter switching frequencies to maintain operation outside of predefined disallowed frequency bands.
Background
Portable devices are becoming more common in every aspect of our lives. For example, many consumers listen to their music from portable audio players. These portable devices become easier and more enjoyable to use when they are provided in small packages, offering the same capabilities while still having long lasting battery charge. To support these longer lasting devices, the operating voltage of many of the components inside is often reduced to reduce the power consumption. However, this reduction in operating voltage may affect the sound output of the portable device as the volume level of an audio signal is proportional to the output voltage. A boost converter may therefore be used to increase the operating voltage for select components within a portable device that would benefit from higher operating voltages, for example, audio amplifiers which may require high voltages to provide high volumes.
Figure 1 illustrates a conventional audio amplifier comprising a boost converter. The audio amplifier comprises a battery supply 102 which is configured to provide a voltage Vp to a boost converter 104. The boost converter 104 increases the voltage Vp to a boost voltage Vbst. In this example, a speaker amplifier 108 is configured to receive the boost voltage Vbst and output an analog audio signal sig from a digital to analog converter (DAC). The speaker amplifier increases the low power signal sig with the power received from the boost converter to generate a signal to drive a speaker 1 10. The boost voltage Vbst is higher than the supply voltage Vp which allows the speaker amplifier 108 to provide louder audio signals through the speaker 1 10 than would be possible using only the supply voltage Vp.
Portable devices are becoming increasingly complex systems where the ability of a given integrated circuit to co-exist with a neighbouring integrated circuit (IC) on a printed circuit board or in a physically adjacent area may become increasingly important to the system or device designer. Interference between the operation of any two ICs may occur not necessarily due to a direct connection between the two, but due to electromagnetic interference produced by one in a frequency or frequency range where operation of the other is susceptible to interference. A boost converter may for instance create a large switching voltage as a part the Vbst generation process. This switching voltage can often cause electromagnetic interference with other nearby ICs or other sub-systems on the portable device. The boost converter may also create noise in the battery supply (Vp) each time it switches. This switching can cause noise to be injected onto a power supply which may be shared by other ICs in a portable device. Some ICs are more susceptible than others to noise on the power supply and performance can suffer as a result. As the portable devices become increasingly complex, so do the requirements to avoid multiple frequency bands of operation.
Summary
According to embodiments described herein there is provided a method for maintaining an output voltage of a DC-DC converter, configured to operate in a discontinuous conduction mode, within a predetermined voltage range. The method comprises adjusting a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
According to some embodiments there is provided a DC-DC converter configured to in a discontinuous conduction mode and to maintain an output voltage within a predetermined voltage range. The DC-DC converter comprises a controller, wherein the controller comprises an input configured to receive an indication of the output voltage; and an adjustment block configured to adjust a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band. Brief Description of the Drawings For a better understanding of the present invention, and to show how it may be put into effect, reference will now be made, by way of example only, to the accompanying drawings, in which:-
Figure 1 is an example of a conventional audio amplifier comprising a boost converter in accordance with the prior art;
Figure 2 is an example block diagram of a DC-DC converter in accordance with the present disclosure Figure 3a is an example timing diagram illustrating the operation of a boost converter in a continuous conduction mode, CCM, in accordance with embodiments of the present disclosure;
Figure 3b is an example timing diagram illustrating the operation of a boost converter in a continuous conduction mode, CCM, in accordance with embodiments of the present disclosure;
Figure 4 is an example timing diagram illustrating the operation of a boost converter in discontinuous conduction mode, DCM, in accordance with embodiments of the present disclosure;
Figure 5 is an example timing diagram illustrating the operation of the DC-DC converter in DCM, in accordance with embodiments of the present disclosure; Figure 6 is an example flow chart diagram illustrating a method for maintaining an output voltage of a DC-DC converter in a predetermined voltage range, while operating in a discontinuous conduction mode, in accordance with embodiments of the present disclosure; Figure 7 is an example timing diagram illustrating the operation of a DC-DC converter when switching from a first switching frequency to a second switching frequency, in accordance with embodiments of the present disclosure; Figure 8 is an example flow chart illustrating a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure;
Figure 9 is an example flow chart illustrating a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure;
Figure 10 is an example flow chart illustrating a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure;
Figure 11 is an example flow chart illustrating a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range, in accordance with embodiments of the present disclosure;
Figure 12 is an example timing diagram illustrating the operation of a DC-DC converter in accordance with embodiments of the present disclosure;
Figure 13 is an example timing diagram illustrating the NFET 206 conducting and non- conducting states as the duty cycle is decreased in accordance with embodiments of the present disclosure;
Figure 14 is an example timing diagram illustrating the energy as a function of frequency for different frequencies using the same duty cycle in accordance with embodiments of the present disclosure;
Figure 15 is an example graph illustrating the switching frequencies available to the boost converter when unconstrained by any predefined disallowed frequency band containing disallowed frequencies in accordance with embodiments of the present disclosure; Figure 16 is a graph illustrating an example where two predefined disallowed frequency bands have been set containing disallowed frequencies in accordance with the present disclosure; Figure 17 is a graph illustrating an example comprising a single predefined disallowed frequency band in accordance with the present disclosure;
Figure 18 is graph illustrating an example comprising three predefined disallowed frequency bands in accordance with the present disclosure;
Figure 19 is graph illustrating an example comprising four predefined disallowed frequency bands in accordance with the present disclosure;
Figure 20 is an example block diagram of a DC-DC converter 1800 in accordance with the present disclosure.
Description
The description below sets forth example embodiments according to this disclosure. Further example embodiments and implementations will be apparent to those having ordinary skill in the art. Further, those having ordinary skill in the art will recognize that various equivalent techniques may be applied in lieu of, or in conjunction with, the embodiments discussed below, and all such equivalents should be deemed as being encompassed by the present disclosure.
Figure 2 illustrates an example DC-DC converter 200 in accordance with some embodiments of the present disclosure. The DC-DC converter 200 comprises a battery supply 202 which provides a supply voltage VP to the DC-DC converter 200. The DC-DC converter 200 comprises an inductor 204 connected to the battery supply 202. The DC- DC converter 200 further comprises two switches 206 and 208 and a capacitor 210. The switches 206 and 208 are in this example implemented as an n-channel field effect transistor (NFET) 206 and a p-channel field effect transistor (PFET) 208.
In this example implementation, the inductor 204 is coupled to the drain of the PFET 208 and the drain of the NFET 206. The drain of the PFET 208 and the drain of the NFET 206 are also coupled together. The capacitor 210 is coupled between the source of the PFET 208 and ground. The source of the NFET 206 is coupled to ground. Both the gate of the PFET 208 and the gate of the NFET 206 are coupled to a level shifting and driving block 214 which controls whether the PFET and NFET are in conducting or non-conducting states. An amplifier or other signal processing circuitry may be connected to receive the boost voltage from the source of the PFET 208.
For example, during a first time period, the NFET 206 may be switched on to a conducting state which couples the inductor 204 to ground, while the PFET 208 is switched off to a non-conducting state. This switching configuration causes the supply voltage VP to fall across the inductor 204 and therefore cause current to flow through the inductor 204 thereby storing energy in the inductor. During a second time period, the NFET 206 may be switched off into a non-conducting state while the PFET 208 may be switched on to a conducting state. This switching configuration causes the inductor to couple to ground through the capacitor 210. Thus, the energy stored in the inductor 204 during the first time period is transferred to the capacitor 210 during the second time period. As energy is stored in the capacitor 210, the voltage across the capacitor increases and allows the boost voltage Vbst to increase beyond the supply voltage Vp.
A controller 222 may control the duration of the first and second time periods in order to regulate the transfer of power from the inductor to the capacitor and thus regulate the level of the boost voltage Vbst.
It will be appreciated that such a boost converter may operate in two distinct modes: continuous conduction mode, CCM and discontinuous conduction mode, DCM. Figure 3a illustrates the operation of a boost converter in a continuous conduction mode, CCM. In this state, the boost converter operates with two switching states Si and S2 per switching cycle. As previously described during state Si , NFET 206 is in a conducting state and PFET 208 is in a non-conducting state. During this period, the current is drawn across the inductor storing energy in the conductor. In switching state S2, as described previously, the PFET 208 is in a conducting state and the NFET 206 is in a non- conducting state. During this period, the energy stored in the inductor is transferred to the capacitor and the current through the inductor drops.
The controller may control the length of the duty cycle in order to maintain the output voltage within a predetermined range. In other words, if Vbst goes too high, the time spent in state Si may be reduced and if Vbst drops below a minimum voltage Vmin, then the time spent in state Si may be increased.
In some examples, the switching state is switched from state Si to state S2 when the current across the inductor reaches a maximum value. The switching state may be switched from state S2 to state Si at the beginning of each clock cycle as illustrated by the begin switching frequency cycle pulses in Figure 3a.
However, when the load current decreases, in other words a higher resistance is coupled to receive the boost voltage Vbst, the inductor current also decreases. In this case, if the boost converter is left to operate in continuous conduction mode, then the inductor current may become negative in each switching period, as illustrated in Figure 3b. This negative current may waste power by passing unnecessary current across the inductor. The boost converter may therefore be operated in discontinuous conduction mode (DCM). In DCM, there may be, for example, three different switching states. States Si and S2 are the same as illustrated for CCM in Figures 3a and 3b, but in third state S3, both the NFET 206 and the PFET 208 are off. As the PFET 208 is switched off, this switch deactivation stops any current flowing back through the inductor in a negative direction, thereby pinning or maintaining the inductor current at OA as illustrated in Figure 4. This mode thereby avoids the power wasted by the negative current through the inductor illustrated in Figure 3b. Figure 5 illustrates an example of the operation of the DC-DC converter in DCM.
In this example, the DC-DC converter is configured to maintain the boost voltage VBST within the predetermined voltage range between a maximum voltage VM3X and a minimum voltage VMIN. Therefore, when the voltage goes above the maximum voltage VMAX, the DC-DC converter may decrease the duty cycle of the DC-DC converter. In other words, the controller 222 illustrated in Figure 2 may decrease the amount of time spent in switching state Si . This time decrease will reduce the energy stored in the inductor during the switching period, and therefore reduce the energy transferred to the capacitor, thereby reducing the boost voltage VBST. Equivalently, when the voltage drops below the minimum voltage VMIN, the DC-DC converter may increase the duty cycle of the DC-DC converter. In other words, the controller 222 illustrated in Figure 2 may increase the amount of time spent in switching state Si . This time increase will increase the energy stored in the inductor during the switching period, and therefore increase the energy transferred to the capacitor, thereby increasing the boost voltage VBST.
According to some embodiments, it may also be desirable, for low enough loads, to reduce the switching frequency in order to further reduce the power consumption of the boost converter. By reducing the switching frequency, the time spent in the state S3 can be increased, thereby saving more power by not wasting current across the inductor.
Furthermore, a lower switching frequency reduces the power consumption across the capacitor. Figure 6 illustrates a method for maintaining an output voltage of a DC-DC converter in a predetermined voltage range, while operating in a discontinuous conduction mode, for example as illustrated in Figure 4.
In step 601 , the DC-DC converter determines the value of the output voltage of the DC- DC converter.
In step 602, the DC-DC converter adjusts a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range. For example, the duty cycle of the DC-DC converter may be adjusted by switching between a first switching frequency and a second switching frequency. The first switching frequency and the second switching frequency may be selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band. For example, in some embodiments at least one predefined disallowed frequency band may be in place which comprises frequencies which should not be used by the DC-DC converter. For example, one or more frequencies in the at least one predefined disallowed frequency band may be used for integrated circuits with some other application within a device comprising the DC-DC converter. In some embodiments, the electromagnetic interference (EMI) with switching frequencies within the predefined disallowed frequency bands is unpredictable, and therefore it may be desirable to avoid using the frequencies within the predefined disallowed frequency bands. Therefore, the embodiments disclosed herein allow the boost converter's switching frequency and duty cycle to be manipulated in order to avoid electromagnetic interference or power supply noise injection which can be problematic for other sub-systems or ICs within the portable device.
When changing switching frequencies to decrease or increase the duty cycle of the DC- DC converter, the frequencies may also be selected such that the time period of the switching frequencies are multiples of the time period of the clock frequency used by the DC-DC converter. In this way, the switching periods of the lower frequencies will comprise an integer number of clock periods.
For example, consider a DC-DC converter utilizing a clock frequency of X kHz. This high frequency may be used as the highest switching frequency available to the DC-DC converter. The second switching frequency may then be X/2 kHz. A third switching frequency of X/3 kHz may also be available, and so on.
The time period of the first switching frequency, which is equal to the clock frequency, will be 1/X seconds. The time period of the second frequency will be 2/X seconds, and the time period of the third switching frequency will be 3/X seconds, and so on.
These switching frequencies may therefore comprise an integer number of clock cycles.
Therefore, when switching from the first switching frequency to the second switching frequency, the DC-DC converter effectively skips a switching period as illustrated in Figure 7.
Figure 7 illustrates the operation of a DC-DC converter when switching from a first switching frequency equal to the clock frequency, to a second switching frequency. In the first and second illustrated clock periods, the DC-DC converter is operating with the first switching frequency. However, mid-way through the second clock cycle, the DC- DC converter decides to lower the switching frequency, potentially due to the boost voltage VBST being above the maximum voltage threshold VMAX as illustrated in Figure 5.
The DC-DC converter may then switch to the second switching frequency for the 3RD to the 6TH clock cycles. It can be seen that the second switching frequency effectively skips the 4TH clock cycle and the 6TH clock cycle by staying the third switching state S3 and pinning the inductor current to zero.
Figure 8 illustrates a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
In particular, the embodiment shown in figure 8 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first decreasing the duty cycle by decreasing the on time of the DC- DC converter until a minimum duty cycle for the first switching frequency is reached, and responsive to the output voltage remaining above the predetermined voltage range when using the first switching frequency, switching to the second switching frequency.
In step 801 the DC-DC converter determines whether the boost voltage VBST is above the maximum voltage threshold VMAX. If the boost voltage VBST is below the threshold VMAX the method returns to the start. If in step 801 the DC-DC converter determines that the boost voltage VBST is above the maximum threshold VMAX, the method passes to step 802 in which the DC-DC converter determines whether an on time Tsi that the DC-DC converter in the state Si , is greater than a minimum value, TSI MIN. An on time of the DC-DC converter may be considered a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time may be understood as the period that the DC-DC convertor remains in the state Si , i.e. the time for which the NFET is in a conducting state.
The minimum on time TSI MIN may be a minimum duty cycle for the frequency at which the DC-DC converter is operating. If in step 802 the DC-DC converter determines that the on time Tsi of the DC-DC converter is greater than TSI MIN, then the method passes to step 803 in which the DC- DC converter reduces the on time of the DC-DC converter. The method will then return to the start. In some embodiments, the on time of the DC-DC converter may be reduced by a predetermined increment.
If in step 802 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not greater than the minimum on time TSI MIN, the method then passes to step 804 in which the DC-DC converter determines whether the switching frequency Fs is greater than a minimum switching frequency FSMIN.
If in step 804 the DC-DC converter determines that the switching frequency Fs is greater than the minimum switching frequency FSMIN, the method passes to step 805 in which the DC-DC converter reduces the switching frequency. In some embodiments, the switching frequency may be decreased by increasing the time period of the switching frequency by one clock period. The method may then pass back to the start. In other words, by decreasing the switching frequency the DC-DC convertor switches from the first switching frequency to the second switching frequency. If in step 804 the DC-DC converter determines that the switching frequency Fs is not greater than the minimum switching frequency, the method may end in step 806.
It will be appreciated that the steps 801 to 805 may be performed iteratively until either the voltage has dropped below the maximum threshold VMAX or both the minimum on time TSI MIN and the minimum switching frequency FSMIN have been reached.
In some embodiments the switching frequency may be decreased from a starting frequency F ST ART by increasing N by one in the following equation: FN = (FSTART)/(N+1 )
Figure 9 illustrates a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments. The embodiment illustrated in figure 9 performs the steps of, responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, first increasing the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the second switching frequency is reached, and responsive to the output voltage remaining below the predetermined voltage range when using the second switching frequency, switching to the first switching frequency.
In step 901 , the DC-DC converter determines whether the boost voltage VBST is below the minimum voltage threshold VMIN. If the boost voltage VBST is above the minimum threshold VMIN the method returns to the start.
If in step 901 , the DC-DC converter determines that the boost voltage VBST is below the minimum threshold VMIN, the method passes to step 902 in which the DC-DC converter determines whether an on time of the DC-DC converter, Tsi , is less than a maximum value, TSI MAX. An on time of the DC-DC converter may be defined as a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time may be understood as the period during which the DC-DC converter is in state Si , i.e. the time for which the NFET is in a conducting state.
The maximum on time TSI MAX may be a maximum duty cycle for the frequency at which the DC-DC converter is operating.
If in step 902 the DC-DC converter determines that the on time Tsi of the DC-DC converter is less than TSI MAX, then the method passes to step 903 in which the DC-DC converter increases the on time Tsi of the DC-DC converter. The method will then return to the start. In some embodiments, the on time Tsi of the DC-DC converter may be increased by a predetermined increment. If in step 902 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not less than the maximum on time Tsi MAX, the method then passes to step 904 in which the DC-DC converter determines whether the switching frequency Fs is less than a maximum switching frequency FSMAX. If in step 904 the DC-DC converter determines that the switching frequency Fs is less than the maximum switching frequency FSMAX, the method passes to step 805 in which the DC-DC converter increases the switching frequency. In some embodiments, the switching frequency may be increased by decreasing the time period of the switching frequency by one clock period. The method may then pass back to the start. In other words, by increasing the switching frequency, the DC-DC convertor switches from the second switching frequency to the first switching frequency.
If in step 904 the DC-DC converter determines that the switching frequency Fs is not less than the maximum switching frequency FSMAX, the method may end in step 806. It will be appreciated that the steps 901 to 905 may be performed iteratively until either the voltage has risen above the minimum threshold VMIN or both the maximum on time Tsi MAX and the maximum switching frequency FSMAX have been reached.
In some embodiments the switching frequency may be increased from a starting frequency F ST ART by decreasing N by one in the following equation:
FN = (FSTART)/(N+1)
Figure 10 illustrates a method of reducing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
In particular, the embodiment shown in Figure 10 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switching to the second switching frequency; and responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
In step 1001 the DC-DC converter determines whether the boost voltage VBST is above the maximum voltage threshold VMAX. If the boost voltage VBST is below the threshold VMAx. the method returns to the start. If in step 1001 the DC-DC converter determines that the boost voltage VBST is above the maximum threshold VMAX, the method passes to step 1002 in which the DC-DC converter determines whether the switching frequency Fs is greater than a minimum switching frequency FSMIN.
If in step 1002 the DC-DC converter determines that the switching frequency Fs is greater than the minimum switching frequency FSMIN, the method passes to step 1003 in which the DC-DC converter reduces the switching frequency. In some embodiments, the switching frequency may be decreased by increasing the time period of the switching frequency by one clock period. In other words, by reducing the switching frequency the DC-DC convertor switches from using the first switching frequency to using a second switching frequency. The method may then pass back to the start.
If in step 1002 the DC-DC converter determines that the switching frequency Fs is not greater than the minimum switching frequency, the method pass to step 1004 in which the DC-DC converter the DC-DC converter determines whether an on time Tsi during which the DC-DC converter is in state Si , is greater than a minimum value, TSI MIN. An on time Tsi of the DC-DC converter may be considered a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this on time Tsi may be understood as the period during which the DC-DC converter is in state Si, i.e. the time for which the NFET is in a conducting state.
The minimum on time TSI MIN may be a minimum duty cycle for the frequency at which the DC-DC converter is operating.
If in step 1004 the DC-DC converter determines that the on time Tsi of the DC-DC converter is greater than TSI MIN, then the method passes to step 1005 in which the DC- DC converter reduces the on time of the DC-DC converter. The method will then return to the start. In some embodiments, the on time of the DC-DC converter may be reduced by a predetermined increment.
If in step 1004 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not greater the TSI MIN, the method may end in step 1006. It will be appreciated that the steps 1001 to 1005 may be performed iteratively until either the voltage has dropped below the maximum threshold VMAX or both the minimum on time TsiMiN and the minimum switching frequency FSMIN have been reached. In some embodiments the switching frequency may be decreased from a starting frequency F ST ART by increasing N by one in the following equation:
FN = (FSTART)/(N+1 ). Figure 11 illustrates a method of increasing the duty cycle of a DC-DC converter to maintain the boost voltage within a predetermined voltage range according to some embodiments.
In particular, the embodiment shown in Figure 1 1 performs the steps of responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switching to the second switching frequency; and responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
In step 1 101 the DC-DC converter determines whether the boost voltage VBST is below the minimum voltage threshold VMIN. If the boost voltage VBST is above the minimum threshold VMIN, the method returns to the start.
If in step 1 101 the DC-DC converter determines that the boost voltage VBST is below the minimum threshold VMIN, the method passes to step 1 102 in which the DC-DC converter determines whether the switching frequency Fs is less than a maximum switching frequency FSMAX.
If in step 1 102 the DC-DC converter determines that the switching frequency Fs is less than the maximum switching frequency FSMAX, the method passes to step 1 103 in which the DC-DC converter increases the switching frequency. In some embodiments, the switching frequency may be increased by decreasing the time period of the switching frequency by one clock period. In other words, by increasing the switching frequency the DC-DC converter switches from using the first switching frequency to the second switching frequency. The method may then passes back to the start.
If in step 1 102 the DC-DC converter determines that the switching frequency Fs is not less than the maximum switching frequency FSMAX, the method passes to step 1104 in which the DC-DC converter the DC-DC converter determines whether an on time Tsi during which the DC-DC converter is in state Si , is less than a maximum value, TSI MAX. An on time Tsi of the DC-DC converter may be defined as a period of time during which the current across the inductor of the DC-DC converter is being increased. In the example illustrated in Figure 4, this may be understood as the period during which the DC-DC converter is in state Si, i.e. the time for which the NFET is in a conducting state.
The maximum on time TSI MAX may be a maximum duty cycle for the frequency at which the DC-DC converter is operating.
If in step 1104 the DC-DC converter determines that the on time Tsi of the DC-DC converter is less than TSI MAX, then the method passes to step 1105 in which the DC-DC converter increases the on time Tsi of the DC-DC converter. The method will then return to the start. In some embodiments, the on time Tsi of the DC-DC converter may be increased by a predetermined increment.
If in step 1104 the DC-DC converter determines that the on time Tsi of the DC-DC converter is not less than the maximum on time Tsi MAX, the method may end in step 1 106.
It will be appreciated that the steps 1101 to 1105 may be performed iteratively until either the voltage has risen above the minimum threshold VMIN or both the maximum on time Tsi MAX and the maximum switching frequency FSMAX have been reached. In some embodiments the switching frequency may be increased from a starting frequency F ST ART by decreasing N by one in the following equation:
FN = (FSTART)/(N+1). Figure 12 illustrates an example of the operation of a DC-DC converter according to some embodiments.
The DC-DC converter monitors the output voltage VBST, for example, as illustrated in Figure 2, the controller 222 may receive an indication of the output voltage VBST via an analog-to-digital converter 220. The controller 222 may then adjust the duty cycle of the DC-DC converter accordingly.
For example, as illustrated in Figure 12, if at a time ti the boost voltage VBST goes above a maximum threshold voltage VMAX, the controller 222 be configured to determine whether to decrease the duty cycle.
In some embodiments, the DC-DC converter is configured to decrease the duty cycle responsive to the boost voltage VBST remaining above the maximum voltage VMAX for a predetermined time. For example, the controller 222 may comprise a filter counter configured to count clock cycles from when the voltage first passes above the maximum threshold VMAX at a time ti , until the amplitude passes back below the maximum threshold at a time t2. In this example, at the time t2 the filter counter has not yet counted a predetermined number X of clock cycles. The DC-DC converter may therefore not decrease the duty cycle. This manner of control avoids unnecessary decreasing of the duty cycle due to transient peaks in the boost voltage VBST.
However, at time t3 the boost voltage VBST passes back above the maximum threshold VMAX, this time for long enough for the filter counter to count X clock cycles twice. In this example, a wait counter is also implemented where the filter counter waits a predetermined time between counting each set of X clock cycles. The controller therefore decreases the duty cycle twice after each time the filter counter counts X clock cycles. The boost voltage passes below the maximum threshold at time before the filter counter can count X clock cycles for a third time, and therefore the duty cycle is not decreased a third time.
It will be appreciated that the duty cycle may be decreased as described with respect to Figure 8 or 10 by decreasing the on time of the DC-DC converter and/or the switching frequency. It will be appreciated that the DC-DC converter may be equivalent^ configured to increase the duty cycle responsive to the boost voltage remaining below the minimum voltage VMIN for a predetermined time. For example, the controller 222 may comprise a filter counter configured to count clock cycles from when the voltage first passes below the minimum threshold VMIN to when the voltage passes back above the threshold VMIN. Again, if the filter counter does not reach a predetermined number of clock cycles Y (where Y may or may not be equal to X) in this time, the controller may be configured not to increase the duty cycle. This manner of control avoids unnecessary increasing of the duty cycle due to transient troughs in the boost voltage VBST.
Again, it will be appreciated that the duty cycle may be increased as described with respect to Figure 9 or 1 1 by increasing the on time of the DC-DC converter and/or the switching frequency. Figure 13 illustrates the NFET 206 conducting and non-conducting states as the duty cycle is decreased. In this example, the duty cycle is decreased according to the method described in Figure 8, but it will be appreciated that the duty cycle may be increased or decreased according to any of the methods described above. At first, in this example, the duty cycle is 3x/Ti . The duty cycle is then decreased to 2x/Ti by decreasing the on time of the DC-DC converter from 3x to 2x. The duty cycle is then decreased to x/Ti by decreasing the on time of the DC-DC converter from 2x to x.
As x is the minimum on time for the DC-DC converter, the duty cycle is then decreased to x/(2Ti) by decreasing the switching frequency from 1 Ti to 1/(2Ti). Again, the duty cycle may be further decreased from x/(2Ti) to x/(3Ti) by decreasing the switching frequency from 1/(2Ti) to 1/(3Ti).
Figure 14 illustrates the energy as a function of frequency for different frequencies using the same duty cycle.
Another possible reason to alter the switching frequency of the DC-DC converter may be to change the concentration of the energy within the frequency band as illustrated in Figure 14. In Figure 14a, the NFET on and off times are illustrated. The duty cycle is x/Ti and the switching frequency is 1/ΤΊ . The graph 1300 illustrates how the energy is therefore concentrated around the frequency 1 Ti . In Figure 14b, the duty cycle is 2x/2Ti which is equivalent to x/Ti , in other words, the same as in Figure 13a. However, the switching frequency is 1/(2Ti). The graph 1401 therefore illustrates how the energy is now concentrated around the lower frequency of
As the energy in the frequency band may be altered by changing the switching frequency, as illustrated in Figure 14. If, for example, a concentration of energy is desired that is concentrated around a frequency 1 Ti but in this example 1 Ti is not an allowed switching frequency, in other words the frequency 1 Ti falls within the predefined disallowed frequency band, then two or more allowable switching frequencies which do not fall within the predefined disallowed frequency band, falling either side of the desired switching frequency, may be utilized to create the same energy distribution around the switching frequency 1 Ti .
Figure 15 illustrates the switching frequencies available to the boost converter when unconstrained by any predefined disallowed frequency band containing disallowed frequencies. As illustrated, the boost converter has a minimum and maximum possible switching frequency. In this example, MINO is the minimum switching frequency of the boost converter when allowed to operate without any predefined disallowed frequency bands containing disallowed frequencies. In this example, fMAxo is the maximum switching frequency of the boost converter when allowed to operate without any predefined disallowed frequency bands containing disallowed frequencies.
Figure 16 illustrates an example where two predefined disallowed frequency bands have been set containing disallowed frequencies. In this example, a first predefined disallowed frequency band between MINO and a frequency fiuim is introduced and a second predefined disallowed frequency band between fi«Axi and fiviAxo is introduced. The values fiMiNi and fiMAxi are then set as maximum and minimum switching frequencies for the boost converter in order to ensure that the boost converter does not use switching frequencies in either of the predefined disallowed frequency bands. In this example, the predefined disallowed frequency bands sit at either end of the allowed switching frequencies. Figure 17 illustrates an example comprising a single predefined disallowed frequency band has been set containing disallowed frequencies. In this example, the predefined disallowed frequency band falls between the frequencies fi«Axi and fiuim . The frequencies between fiuAxi and fiuiNi may therefore be considered as disallowed. Two frequency bands are provided that the boost converter may use between MINO and fiuAxi and MIN2 and fMAxo. In this example therefore, the boost converter may select a frequency between fiuiNo and fiMAxi and fMiN2 and fMAxo. In circumstances where the desired energy concentration is between fiuAxi and fiwiN2, the boost converter may select a frequency between fiuiNo and fiMAxi and a different frequency between MIN2 and fiviAxo and may then alternate between the two switching frequencies in such a way that the energy may be concentrated between the frequencies fiuAxi and fMiN2.
In some embodiments therefore, the DC-DC converter may be configured to alternate between a first switching frequency and a second switching frequency wherein the first switching frequency is higher than the predefined disallowed frequency band and the second switching frequency is lower than the predefined disallowed frequency band.
Figure 18 illustrates an example comprising three predefined disallowed frequency bands. In this example, a first predefined disallowed frequency band falls between fiuiNo and a frequency fiuiNi and a second frequency bend falls between fMAX2 and fMAxo. These disallowed frequency bands are equivalent to those illustrated in Figure 16 in that they are at the ends of the possible frequencies that the boost converter is capable of using. However, in this example, the predefined disallowed frequency band between fiuAxi and fMiN2 is also disallowed. In this example, therefore only the frequencies between fiuim and fiMAxi and the frequencies between fiuiN2 and fMAX2 may be used by the boost converter. Similarly to as described with regards to Figure 17, the boost converter may alternate between a switching frequency between fiuiNi and fiuAxi and a switching frequency between fiuiN2 and fMAX2 in order to concentrate the energy between the frequencies fiuAxi
Figure 19 illustrates an example comprises four predefined disallowed frequency bands. In this example, the frequencies between fiuiNo and f MINI , between fi«Axi and fiwiN2, between fiMAX2 and fMiN3 and between fMAX3 and fMiN4 are disallowed. As previously, the boost convertor may alternate between frequencies outside of these predefined disallowed frequency bands containing disallowed frequencies to concentrate the energy within the predefined disallowed frequency bands without actually using the frequencies within those predefined disallowed frequency bands.
Figure 20 illustrates a DC-DC converter 2000 according to some embodiments. The DC- DC converter 2000 comprises a controller 2001. The controller 2001 comprises an input 2002 configured to receive an indication of the output voltage.
The controller 2001 further comprises an adjustment block 2003 configured to adjust a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range. The duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency. The first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band. As described previously, the at least one predefined disallowed frequency band may comprise disallowed frequencies.
There is therefore provided methods and apparatus for maintaining an output voltage of a DC-DC converter, configured to operate in a discontinuous conduction mode, within a predetermined voltage range.
It should be noted that the above-mentioned embodiments illustrate rather than limit the invention, and that those skilled in the art will be able to design many alternative embodiments without departing from the scope of the appended claims. The word "comprising" does not exclude the presence of elements or steps other than those listed in the claim, "a" or "an" does not exclude a plurality, and a single feature or other unit may fulfill the functions of several units recited in the claims. Any reference numerals or labels in the claims shall not be construed so as to limit their scope. Terms such as amplify or gain include possible applying a scaling factor or less than unity to a signal.

Claims

1. A method for maintaining an output voltage of a DC-DC converter, configured to operate in a discontinuous conduction mode, within a predetermined voltage range, the method comprising:
adjusting a duty cycle of the DC-DC converter based on an output voltage to maintain the output voltage within a predetermined voltage range; wherein
the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and
the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
2. The method of claim 1 wherein the second switching frequency is equal to 1/N of the first switching frequency.
3. The method of claim 1 wherein the DC-DC converter is configured for use in a device and the at least one predefined disallowed frequency band comprises frequencies reserved for use with other components in the device.
4. The method of claim 1 further comprising:
adjusting the duty cycle of the DC-DC converter by decreasing an on time of the DC-DC converter.
5. The method of claim 4 further comprising: responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the first switching frequency is reached, and
responsive to the output voltage remaining above the predetermined voltage range when using the first switching frequency, switching to the second switching frequency.
6. The method of claim 5 further comprising:
responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, switching to a third switching frequency wherein the third frequency is 1/N+1 times the second frequency.
7. The method of claim 4 further comprising:
responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, first increasing the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the second switching frequency is reached, and
responsive to the output voltage remaining below the predetermined voltage range when using the second switching frequency, switching to the first switching frequency.
8. The method of claim 7 further comprising:
responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, switching to a third switching frequency wherein the third frequency is greater than the first frequency.
9. The method of claim 4 further comprising: responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switching to the second switching frequency; and
responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decreasing the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
10. The method of claim 9 further comprising:
responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, switching to a third switching frequency wherein the third frequency is 1/N+1 times the second frequency.
1 1. The method of claim 4 further comprising:
responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, switching to the first switching frequency and
responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, increasing the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the first switching frequency is reached.
12. The method of claim 11 further comprising: responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, switching to a third switching frequency wherein the third frequency is greater than the first frequency.
13. The method of claim 1 further comprising:
alternating between the first switching frequency and the second switching frequency, wherein the first switching frequency is higher than the predefined disallowed frequency band and the second switching frequency is lower than the predefined disallowed frequency band.
14. The method of claim 1 wherein the at least one predefined disallowed frequency band comprises two or more separate predefined disallowed frequency bands.
15. A DC-DC converter configured to in a discontinuous conduction mode and to maintain an output voltage within a predetermined voltage range, the DC-DC converter comprising a controller, wherein the controller comprises:
an input configured to receive an indication of the output voltage; and
an adjustment block configured to adjust a duty cycle of the DC-DC converter based on the output voltage to maintain the output voltage within a predetermined voltage range; wherein
the duty cycle of the DC-DC converter is adjusted by switching between a first switching frequency to a second switching frequency, and
the first switching frequency and the second switching frequency are selected such that the first switching frequency and the second switching frequency fall outside of at least one predefined disallowed frequency band.
16. The DC-DC converter of claim 15 wherein the second switching frequency is equal to 1/N of the first switching frequency.
17. The DC-DC converter of claim 15 wherein the DC-DC converter is configured for use in a device and the at least one predefined disallowed frequency band comprises frequencies reserved for use with other components in the device.
18. The DC-DC converter of claim 15 wherein the adjustment block is further configured to adjust the duty cycle of the DC-DC converter by decreasing an on time of the DC-DC converter.
19. The DC-DC converter of claim 18 wherein the adjustment block is further configured to: responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first decrease the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the first switching frequency is reached, and
responsive to the output voltage remaining above the predetermined voltage range when using the first switching frequency, switch to the second switching frequency.
20. The DC-DC converter of claim 19 wherein the adjustment block is further configured to: responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, switch to a third switching frequency wherein the third frequency is 1/N+1 times the second frequency.
21. The DC-DC converter of claim 18 wherein the adjustment block is further configured to: responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, first increase the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the second switching frequency is reached, and
responsive to the output voltage remaining below the predetermined voltage range when using the second switching frequency, switch to the first switching frequency.
22. The DC-DC converter of claim 21 wherein the adjustment block is further configured to: responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, switch to a third switching frequency wherein the third frequency is greater than the first frequency.
23. The DC-DC converter of claim 18 wherein the adjustment block is further configured to: responsive to the output voltage being above the predetermined voltage range when using the first switching frequency, first switch to the second switching frequency; and
responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, decrease the duty cycle by decreasing the on time of the DC-DC converter until a minimum duty cycle for the second switching frequency is reached.
24. The DC-DC converter of claim 23 wherein the adjustment block is further configured to: responsive to the output voltage remaining above the predetermined voltage range when using the second switching frequency, switch to a third switching frequency wherein the third frequency is 1/N+1 times the second frequency.
25. The DC-DC converter of claim 18 wherein the adjustment block is further configured to: responsive to the output voltage being below the predetermined voltage range when using the second switching frequency, switch to the first switching frequency and
responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, increase the duty cycle by increasing an on time of the DC-DC converter until a maximum duty cycle for the first switching frequency is reached.
26. The DC-DC converter of claim 25 wherein the adjustment block is further configured to:
responsive to the output voltage remaining below the predetermined voltage range when using the first switching frequency, switch to a third switching frequency wherein the third frequency is greater than the first frequency.
27. The DC-DC converter of claim 15 wherein the adjustment block is configured to alternate between the first switching frequency and the second switching frequency, wherein the first switching frequency is higher than the predefined disallowed frequency band and the second switching frequency is lower than the predefined disallowed frequency band.
28. The DC-DC converter of claim 15 wherein the at least one predefined disallowed frequency band comprises two or more separate predefined disallowed frequency bands.
PCT/GB2018/051383 2017-05-22 2018-05-22 A dc-dc converter Ceased WO2018215754A1 (en)

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GB2578379A (en) 2020-05-06
GB201918612D0 (en) 2020-01-29
CN110945767A (en) 2020-03-31
CN110945767B (en) 2023-08-15
GB2578379B (en) 2022-10-12
GB2578379A8 (en) 2020-06-03
US10958171B2 (en) 2021-03-23
US20180337601A1 (en) 2018-11-22

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