WO2017216894A1 - Transmitter - Google Patents

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Publication number
WO2017216894A1
WO2017216894A1 PCT/JP2016/067756 JP2016067756W WO2017216894A1 WO 2017216894 A1 WO2017216894 A1 WO 2017216894A1 JP 2016067756 W JP2016067756 W JP 2016067756W WO 2017216894 A1 WO2017216894 A1 WO 2017216894A1
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WO
WIPO (PCT)
Prior art keywords
amplifier
signal
digital
characteristic
transmission signal
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PCT/JP2016/067756
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French (fr)
Japanese (ja)
Inventor
一二三 能登
秀樹 森重
檜枝　護重
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三菱電機株式会社
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Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to PCT/JP2016/067756 priority Critical patent/WO2017216894A1/en
Publication of WO2017216894A1 publication Critical patent/WO2017216894A1/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion

Definitions

  • the present invention relates to a transmitter used for satellite communication, terrestrial microwave communication, and mobile communication.
  • Patent Document 1 discloses a technique for improving the linearity of a communication base station transmitter. Patent Document 1 uses a power series method in order to create predistortion in the digital predistortion method, and considers up to the seventh-order distortion component.
  • a predistortion is created in the digital part, and the DAC
  • the data is output to the analog unit via (Digital to Analog Converter).
  • the non-linear characteristic of the analog unit may be any characteristic.
  • the DAC requires a band three times the signal band, so the DAC band is less than the 100 MHz band.
  • the present invention has been made to solve the above-described problems, and an object thereof is to suppress quantization noise generated in a DAC without increasing the number of bits of the DAC.
  • the transmitter of the present invention includes a digital circuit that generates a digital transmission signal, a digital-to-analog converter that converts the predistorted transmission signal into an analog signal, and amplifies the analog transmission signal output by the digital-analog converter.
  • An amplifier is provided between the amplifier and the digital-analog converter and the amplifier, and the amplitude of the analog transmission signal output from the digital-analog converter is changed so that the AM-AM characteristic combined with the amplifier becomes a gain expansion characteristic.
  • An expansion generation circuit for outputting to the digital signal, and a distortion compensation circuit that is provided between the digital circuit and the digital-analog converter and predistorts the distortion of the transmission signal amplified by the amplifier.
  • quantization noise generated in the DAC can be suppressed without increasing the number of DAC bits.
  • FIG. 1 is a block diagram showing a configuration example of a transmitter according to Embodiment 1 of the present invention.
  • the transmitter includes a modem 1 and an RF module 9.
  • the modem 1 includes a DSP (Digital-Signal-Processor) 2, a modulation unit 3, a signal comparison unit 4, a PD (Pre-Distortion) signal generation unit 5, a PD unit 6, a DAC 7, and an ADC (Analog to Digital Converter) 8.
  • DSP Digital-Signal-Processor
  • modulation unit 3 a modulation unit 3
  • signal comparison unit 4 a PD (Pre-Distortion) signal generation unit 5
  • a PD unit 6 a DAC 7, and an ADC (Analog to Digital Converter) 8.
  • ADC Analog to Digital Converter
  • the DSP 2 includes a modulation unit 3, generates a digital signal, converts the digital signal into a baseband signal by the modulation unit 3, and outputs the digital signal.
  • the DSP 2 is connected to the PD 6 and the signal comparison unit 4.
  • the signal comparison unit 4 compares the modulation wave signal output from the modulation unit 3 with the feedback modulation wave signal output from the AD 8, and compares the complex signal vector difference (in the case of the LUT (Look Up Table) method with the amplitude). It is a signal comparison unit that outputs a phase difference) to the PD signal generation unit 5.
  • the signal comparison unit 4 is connected to the DSP 2 and the ADC 8.
  • the PD signal generation unit 5 is a PD signal generation unit that generates an inverse signal that cancels distortion generated from the DAC 7 to the amplifier 15 according to the comparison result output from the signal comparison unit 4.
  • the PD signal generation unit 5 is connected to the PD 6 and the signal comparison unit 4.
  • a method of generating an inverse signal by the PD signal generation unit 5 for example, an LUT method, a polynomial method, a power series method, a memory polynomial method, or a Volterra series method is used.
  • PD 6 is a PD that superimposes the reverse signal generated by the PD signal generation unit 5 and the baseband signal generated by the modulation unit 3 and outputs the superimposed signal.
  • the PD 6 is connected to the DSP 2 and the DAC 7.
  • the DAC 7 is a DAC that converts a digital signal output from the PD 6 into an analog signal.
  • the DAC 7 is connected to the PD 6 and the BPF 10.
  • the ADC 8 is an ADC that changes an analog signal output from the BPF 21 into a digital signal.
  • the ADC 8 is connected to the BPF 21 and the signal comparison unit 4.
  • the DSP 2, the signal comparison unit 4, the PD signal generation unit 5, and the PD 6 are configured by an FPGA (Field-Programmable Gate Array), an ASIC (Application Specific Integrated Circuit), and the like.
  • the DAC 7 and the ADC 8 may be configured to be built in the FPGA.
  • the RF module 9 includes a BPF (Band-Pass Filter) 10, a mixer 11, an LO (Local Signal Source) 12, a BPF 13, an ECC (Expansion / Characteristics / Circuit) 14, an amplifier 15, a coupler 16, an isolator 17, an LPF (Low-Pass Filter). ) 18, an antenna 19, a mixer 20, and a BPF 21.
  • BPF Band-Pass Filter
  • LO Local Signal Source
  • BPF 13 Low-Pass Filter
  • ECC Exposure / Characteristics / Circuit
  • the BPF 10 is a BPF that blocks unnecessary waves from the modulation signal output from the DAC 7 and outputs a modulation signal that blocks unnecessary waves.
  • the BPF 10 is connected to the DAC 7 and the mixer 11.
  • the mixer 11 is a mixer that mixes the modulation signal output from the BPF 11 and the local signal of the LO 12 and converts the frequency of the modulation signal.
  • the mixer 11 has an LO terminal connected to the LO 12, an IF terminal connected to the BPF 11, and an RF terminal connected to the BPF 13.
  • LO 12 is an oscillator that outputs a local signal used for frequency conversion in the mixer 11.
  • the LO 12 is connected to the mixer 11 and the mixer 20.
  • the BPF 13 is a BPF that blocks unnecessary waves outside the desired band from the frequency-converted modulated signal and outputs a modulated signal that blocks unnecessary waves.
  • the BPF 13 is connected to the mixer 11 and the ECC 14.
  • the ECC 14 is an analog circuit that performs gain expansion (gain expansion) on the AM-AM characteristic of the modulation signal.
  • the ECC 14 is connected to the BPF 13 and the amplifier 15.
  • FIG. 2 is a circuit diagram showing a configuration example and characteristics of the ECC 14 according to the first embodiment of the present invention.
  • FIG. 2A shows an example in which an amplifier whose operation class is from class B to class AB is used, and has a characteristic that the gain increases with respect to the output power.
  • Vc is a bias voltage of the amplifier, and Vc is adjusted to set a bias in a range from class B to class AB.
  • FIG. 2B shows an example in which a parallel diode linearizer is used, and has a characteristic of increasing the gain with respect to output power by using the nonlinear characteristic of the diode.
  • FIG. 2C is a modification of FIG. 2B in which the ECC 14 is configured using a diode and a hybrid. The characteristic of increasing the gain with respect to the output power using the nonlinear characteristic of the diode is shown. Give it.
  • the gain characteristic with respect to the output power has been described as an example, but the gain characteristic with respect to the input power is the same as that with the output power.
  • the amplifier 15 is an amplifier that receives a gain-expanded modulation signal and amplifies the modulation signal.
  • the amplifier 15 is connected to the ECC 14 and the coupler 16.
  • the coupler 16 is a coupler that extracts a part of the amplified modulation signal.
  • the coupler 16 is connected to the amplifier 15 and the isolator 17.
  • the isolator 17 is an isolator that allows a modulated signal to pass and absorbs a reflected wave generated by a component connected to a subsequent stage.
  • the isolator 17 is connected to the coupler 16 and the LPF 18.
  • the LPF 18 is an LPF that blocks unnecessary waves outside the desired band from the modulation signal and outputs a modulation signal that blocks unnecessary waves.
  • the LPF 18 is connected to the isolator 17 and the antenna 19.
  • the antenna 19 is an antenna that transmits the modulation signal output from the LPF 18.
  • the antenna 19 is connected to the LPF 18.
  • the mixer 20 is a mixer that mixes the local signal of the LO 12 and the modulation signal amplified by the amplifier 15 and converts the frequency of the modulation signal.
  • the mixer 20 has an RF terminal connected to the coupler 16, an LO terminal connected to the LO 12, and an IF terminal connected to the BPF 21.
  • the BPF 21 is a BPF that blocks unnecessary waves from the modulation signal frequency-converted by the mixer 20 and outputs a modulation signal that blocks unnecessary waves.
  • the BPF 21 is connected to the mixer 20 and the ADC 8.
  • the DSP 2 outputs the modulation wave signal generated by the modulation unit 3 in the DSP 2 to the DAC 7 via the PD 6.
  • the DAC 7 converts the input baseband signal into an analog signal and outputs the analog signal to the BPF 10.
  • the BPF 10 suppresses unnecessary waves outside the passband of the BPF 10 from the analog signal output by the DAC 7, and outputs an analog signal in which unnecessary waves are suppressed to the mixer 11.
  • the mixer 11 up-converts the analog signal output from the BPF 10 using the local signal of the LO 12 and converts it into an RF signal.
  • the mixer 11 outputs an RF signal to the BPF 13.
  • the mixer 11 When the mixer 11 performs frequency conversion, an analog signal and a local signal are mixed, so that an unnecessary wave is output in addition to a desired RF signal.
  • the BPF 11 blocks unnecessary waves from the output signal of the mixer 11 and outputs a desired RF signal to the ECC 14.
  • the ECC 14 has an AM-AM characteristic having a gain expansion characteristic, changes the gain according to the power of the input RF signal, and outputs the gain to the amplifier 15. Details of this point will be described later.
  • the amplifier 15 amplifies the RF signal output from the ECC 14 and outputs the amplified RF signal to the LPF 18 via the coupler 16 and the isolator 17.
  • the LPF 18 suppresses unnecessary waves in the RF signal amplified by the amplifier 15 and outputs an RF signal in which unnecessary waves are suppressed to the antenna 19.
  • the antenna 19 transmits the RF signal output from the LPF 18.
  • the coupler 16 takes out a part of the RF signal and outputs it to the mixer 20 in order to feed back the RF signal amplified by the amplifier 15.
  • the mixer 20 uses the local signal of the LO 12 to down-convert the RF signal input from the coupler 16 and convert it into a baseband signal.
  • the BPF 21 suppresses unnecessary waves in the output signal of the mixer 20 and outputs a baseband signal in which unnecessary waves are suppressed to the ADC 8.
  • the ADC 8 converts the baseband signal into a digital signal and outputs it to the signal comparison unit 4.
  • the signal comparison unit 4 compares the modulation signal generated by the DSP 2 with the modulation signal fed back through the ECC 14 and the amplifier 15, and outputs the difference to the PD signal generation unit 5.
  • the PD signal generation unit 5 generates an inverse signal that cancels distortion generated in the ECC 14 and the amplifier 15 in accordance with the output signal of the signal comparison unit 4 and outputs the reverse signal to the PD 6.
  • the PD 6 superimposes an inverse signal that compensates for distortion of the ECC 14 and the amplifier 15 on the modulation signal generated by the DSP 2, and outputs the superimposed modulation signal to the DAC 7.
  • the generated modulation signal is compared with the modulation signal fed back through the ECC 14 and the amplifier 15, and an inverse signal that cancels the nonlinear characteristics of the ECC 14 and the amplifier 15 is superimposed on the modulation signal.
  • the distortion generated in the amplifier 15 is compensated. This distortion compensation is called predistortion.
  • FIG. 3 is a diagram showing AM-AM characteristics and distortion characteristics in a general amplifier.
  • the AM-AM characteristic of an amplifier changes in the order of dotted line ⁇ broken line ⁇ solid line
  • the distortion characteristic deteriorates in the order of broken line ⁇ dotted line ⁇ solid line. This is because the flatter the AM-AM characteristic, the better the distortion characteristic.
  • the AM-AM characteristic and the distortion characteristic are related, and it is desired to flatten the AM-AM characteristic in order to improve the distortion.
  • FIG. 4 shows AM-AM characteristics when the ECC 14 and the amplifier 15 according to the first embodiment of the present invention are combined.
  • the broken line is the conventional characteristic (characteristic when there is no ECC 14)
  • the dotted line is the characteristic when the distortion characteristic is good
  • an analog circuit corresponding to the ECC 14 is provided at the front stage of the amplifier as shown by the dotted line in FIG. 4, but the ECC 14 is such that the AM-AM characteristic of the ECC 14 and the amplifier 15 is the solid line in FIG.
  • the gain is expanded according to the power of the input signal. That is, the ECC 14 does not compensate for the distortion of the amplifier 15 but causes the AM-AM characteristic of the ECC 14 and the amplifier 15 to be the gain expansion characteristic even if the distortion of the amplifier 15 is deteriorated. Thereby, although the distortion of the signal is deteriorated, the maximum value of the power output from the DAC 7 can be reduced as will be described later. The deteriorated distortion is compensated by the PD 6.
  • FIG. 5 is a diagram showing a cumulative distribution of instantaneous power with respect to the average power of the modulated wave signal.
  • the vertical axis is CCDF (Complementary Cumulative Distribution Function), and the horizontal axis is instantaneous power.
  • PAPR peak to average power ratio
  • the DAC 7 requires a dynamic range of 10 dB for PAPR and Ad dB for distortion compensation of PD6 when there is no ECC14. .
  • the gain of the amplifier decreases as the input power increases. Therefore, it is necessary to increase the gain of AdB in order to compensate for the decrease in the gain.
  • the wider the band of the modulation signal and the larger the PAPR the more the amplifier operates non-linearly and the amount of distortion compensation increases, so the dynamic range required for the DAC 7 increases.
  • the operation speed of the DAC is decreased.
  • FIG. 6 is a conceptual diagram showing signal quantization in the DAC 7 according to the first embodiment of the present invention.
  • a dotted line is an input signal waveform when the ECC 14 is not present, and a solid line is an input signal waveform when the ECC 14 is present.
  • the gain is expanded by the ECC 14, so that the amplitude of the signal output from the DAC 7 is smaller than when the ECC 14 is not present. Therefore, when the number of bits of the DAC 7 is X bits, the resolution per bit (corresponding to the step width in FIG.
  • FIG. 7 is a diagram showing a frequency spectrum before and after distortion compensation of the transmitter according to Embodiment 1 of the present invention.
  • the noise floor after distortion compensation is increased because the quantization noise is increased by the amount of gain expansion by the DAC 7.
  • the quantization noise is lowered, the distortion noise is reduced.
  • the noise floor after compensation can be lowered.
  • FIG. 8 is a characteristic diagram showing distortion characteristics of the RF module 9 when the number of bits of the DAC 7 according to Embodiment 1 of the present invention is 10.
  • the vertical axis is ACPR (Adjacent Channel Power Ratio), and the horizontal axis is Pout.
  • ACPR Adjacent Channel Power Ratio
  • Pout the horizontal axis
  • the ECC 14 is used to change the AM-AM characteristic of the amplifier 15 to the gain expansion characteristic, so that an increase in quantization noise of the DAC 7 can be suppressed.
  • the RF module 9 of the first embodiment does not have a reception function, but may have a reception function.
  • FIG. 9 is a block diagram showing a configuration example of a transmitter according to Embodiment 2 of the present invention. 9, the same reference numerals as those in FIG. 1 denote the same or corresponding parts. 1 is different from FIG. 1 in that a VGA (Variable Gain Amplifier) 22, a VGA 23, a TS 24a, and a control unit 25a are provided.
  • VGA Very Gain Amplifier
  • BPF 10 and BPF 10a mixer 11 and mixer 11a, LO 12 and LO 12a, BPF 13 and BPF 13a, ECC 14 and ECC 14a, amplifier 15 and amplifier 15a, coupler 16 and coupler 16a, isolator 17 and isolator 17a, LPF 18 and LPF 18a, antenna 19 and antenna 19a, the mixer 20 and the mixer 20a, and the BPF 21 and the BPF 21a correspond to each other.
  • the VGA 22 is a VGA that is provided in front of the ECC 14 and adjusts the power level of the signal input to the ECC 14 by changing the gain.
  • the VGA 23 is a VGA that is provided at the subsequent stage of the ECC 14 and adjusts the power level of the signal output from the ECC 14 by changing the gain.
  • the TS 24 a is a temperature sensor that monitors the temperature of the RF module 9.
  • the TS 24a is connected to the control unit 25a and transmits temperature information to the control unit 25a.
  • the control unit 25 a is a control unit that controls the bias voltages of the VGAs 22 and 23, the ECC 14, and the amplifier 15.
  • the control unit 25a receives temperature information from the TS 24a, and controls the gains of the VGAs 22 and 23, the ECC 14, and the amplifier 15 by changing the bias voltage according to the temperature information.
  • the RF module 9 has different characteristics depending on the temperature, as shown in FIG. 9, the modulation signal generated by the signal comparison unit 4 is compared with the modulated signal fed back, and predistortion is performed by the PD 6 so that the two match.
  • the characteristic variation due to temperature is compensated for distortion by feedback control.
  • the characteristic of the RF module 9 is not the gain expansion characteristic, the quantization noise increases in the DAC 7 and the noise floor increases. Therefore, it is necessary to perform control so that the RF module 9 has a gain expansion characteristic even when a temperature change occurs.
  • the current flowing through the ECC 14 becomes small, so that the characteristic that the gain increases with respect to the input power (gain expansion characteristic) becomes small. Further, the gain of the amplifier 15 and the saturated output power are also reduced, and the amplifier 15 has a characteristic that the gain is reduced with respect to the input power.
  • the temperature is detected by the TS 24a, and the bias voltage or bias current of the ECC 14 and the ECC 15 is adjusted so that the AM-AM characteristic of the RF module 9 becomes the gain expansion characteristic according to the temperature. Thereby, even if temperature changes, the RF module 9 can maintain a gain expansion characteristic.
  • FIG. 10 is a characteristic diagram showing characteristics of the RF module 9 with and without temperature compensation according to Embodiment 2 of the present invention.
  • the broken line is the characteristic of the RF module 9 when the ECC 14 and the bias voltage of the amplifier 15 are not controlled with respect to the temperature
  • the solid line is the characteristic of the RF module 9 when the bias voltage of the ECC 14 and the amplifier 15 is controlled with respect to the temperature. It is a characteristic.
  • a dotted line is a characteristic of the RF module 9 when the gain characteristic is flat, and is a characteristic to be compared with a solid line and a broken line.
  • a characteristic in which the gain is higher than this characteristic is a gain expansion characteristic.
  • the control unit 25a also controls the gains of the VGAs 22 and 23 according to the temperature so that the overall gain of the RF module 9 becomes constant regardless of the temperature. As a result, even if the temperature changes, the overall gain of the RF module 9 can be maintained and the gain expansion characteristic can be maintained.
  • the gain of the ECC 14 and the amplifier 15 is controlled so that the RF module 9 maintains the gain expansion characteristic even when the temperature changes. An increase in noise can be suppressed.
  • FIG. 11 is a block diagram showing another configuration example of the transmitter according to the second embodiment of the present invention. 9 and 11 are compared, the modem 1 and the modem 1a, the DSP 2 and the DSP 2a, the modulation unit 3 and the modulation unit 3a, the signal comparison unit 4 and the signal comparison units 4a and 4b, the PD signal generation unit 5 and the PD signal generation unit.
  • FIG. 11 shows a configuration in which a plurality of RF modules 9a shown in FIG. 9 are arranged, the DSP 2a is shared by the plurality of RF modules, and a transmitter configuration of Massive-Input MIMO (Multiple-Input and Multi-Output) is used. Even with this configuration, the same effects as those of the second embodiment are obtained.
  • FIG. 11 shows an example in which two RF modules are provided, but two or more RF modules may be provided.
  • FIG. 12 is a block diagram showing another configuration example of the transmitter according to Embodiment 2 of the present invention. 12, the same reference numerals as those in FIG. 9 indicate the same or corresponding parts, and thus the description thereof is omitted. Further, for example, the components having the same numbers correspond to each other like the amplifier 15a and the amplifier 15b, and thus the description thereof is omitted. Further, the RF module 9a in FIG. 9 corresponds to the RF modules 90a, 90b, 90c, and 90d in FIG. 9 is compared with FIG. 11, in the configuration of the RF module 90a of FIG. 12, the phase shifter 28a is provided in the front stage of the amplifier 15, and the phase shifter 29a is provided in the front stage of the mixer 20a.
  • a DIV 26 and a Com 27 are provided between the modem 1 and the RF modules 90a, 90b, 90c, and 90d.
  • the DIV 26 is a distribution circuit that distributes the output signal of the modem 1 to the RF modules 90a, 90b, 90c, and 90d.
  • Com 27 is a combining circuit that combines the output signals of the RF modules 90 a, 90 b, 90 c, and 90 d and outputs the combined signals to the modem 1.
  • FIG. 12 shows an active phased array transmitter having an analog subarray configuration in which a plurality of RF modules of FIG. 9 are arranged. The signal is distributed to each RF module by the DIV 26, and the signal fed back is synthesized by the Com 27.
  • a phase shifter 28a is inserted into the RF module 90a.
  • the phase shifter 29a is a phase shifter for returning the phase changed by the phase shifter 28a when performing distortion compensation. Even with this configuration, the same effects as those of the second embodiment are obtained.
  • the phase shifters 29a, 29b, 29c, and 29d return the phase even when the phases of the phase shifters 28a, 28b, 28c, and 28d are changed and the transmission beam is shaken.
  • the feedback signals 91a, 91b, 91c, and 91d can be in phase.
  • the feedback signal having the same phase is synthesized by Com27 and predistortion is performed using the synthesized signal. Therefore, even if the direction of the transmission beam is changed, the distortion is low with respect to the front direction of the transmission beam.
  • a signal can be output.
  • the local sources 12a, 12b, 12c, and 12d there are two methods for bringing the local sources 12a, 12b, 12c, and 12d into the same phase in order to make the feedback signal in the same phase.
  • One is to make the local sources 12a, 12b, 12c, and 12d in phase by synchronizing reference signals (reference signals) that drive the local sources.
  • the local sources 12a, 12b are controlled by controlling and correcting the load enable signal for controlling the rising of the local sources 12a, 12b, 12c, 12d.
  • 12c, 12d are in phase.
  • FIG. 13 is a block diagram showing another configuration example of the transmitter according to Embodiment 2 of the present invention.
  • FIG. 13 is a modification of FIG. 12, in which the ECC of each RF module is unified.
  • the same reference numerals as those in FIG. since the components with the same number correspond to each other, the description thereof is omitted.
  • the RF modules 90a, 90b, 90c, and 90d correspond to the RF modules 91a, 91b, 91c, and 91d, respectively.
  • the ECC module 30 is shared by the RF modules 91a, 91b, 91c, and 91d.
  • the ECC module 30 includes a VGA 22, a VGA 23, an ECC 14, a DIV 26, a COM 27, and a control unit 31.
  • the control unit 31 controls the VGA 22 so that the input power entering the ECC 14 becomes the same even when the temperature changes, and the average gain from the ECC 14 to the input ends of the amplifiers 15a, 15b, 15c, and 15d is constant.
  • the VGA 23 is controlled so that Even with this configuration, the same effects as those of the second embodiment are obtained.

Abstract

A conventional transmitter has a problem of increase of quantization noise at a DAC when a transmission signal is wideband. A transmitter according to the present invention is provided with: a digital circuit that generates a digital transmission signal; a digital analog converter that converts a predistorted transmission signal into an analog signal; an amplifier that amplifies the analog transmission signal output by the digital analog converter; an expansion generating circuit that is provided between the digital analog converter and the amplifier, and that changes and outputs, to the amplifier, the amplitude of the analog transmission signal output by the digital analog converter so that an AM-AM characteristic made to match with the amplifier becomes a gain expansion characteristic; and a distortion compensation circuit that is provided between the digital circuit and the digital analog converter, and that predistorts the transmission signal amplified by the amplifier so that the distortion of the transmission signal is compensated, and outputs the predistorted transmission signal to the digital analog converter.

Description

送信機Transmitter
　本発明は、衛星通信、地上波マイクロ波通信、移動体通信に使用される送信機に関するものである。 The present invention relates to a transmitter used for satellite communication, terrestrial microwave communication, and mobile communication.
　近年、無線通信の進展は目覚ましく、無線通信システムには更なる通信容量の拡大及び通信速度の向上が求められている。しかし、無線通信資源には限りがあり、その中で高速かつ大容量な通信を実現する必要がある。そのため、昨今では比較的使いやすい低周波帯から高速かつ大容量な通信が期待できる高周波帯での通信システムが期待されている。高周波帯での通信システムが可能になれば、信号帯域を広くとることができ、高速かつ大容量な通信が実現できる。ところが、無線通信システムで使用する通信用基地局送信機には高い線形性および高い効率が求められる。一般的に低周波帯に比べて高周波帯での送信機の効率は低いため、高い効率を得ようとした場合、線形性が悪化する。この線形性を改善する一つの手法として歪み補償が有る。特許文献１に示すような通信用基地局送信機によく用いられるディジタルプリディストーション方式がその一つである。 In recent years, the progress of wireless communication is remarkable, and the wireless communication system is required to further expand the communication capacity and improve the communication speed. However, wireless communication resources are limited, and it is necessary to realize high-speed and large-capacity communication. For this reason, a communication system in a high frequency band that can be expected to perform high-speed and large-capacity communication from a low frequency band that is relatively easy to use is expected. If a communication system in a high frequency band becomes possible, a wide signal band can be obtained, and high-speed and large-capacity communication can be realized. However, a communication base station transmitter used in a wireless communication system is required to have high linearity and high efficiency. In general, since the efficiency of a transmitter in a high frequency band is lower than that in a low frequency band, linearity deteriorates when trying to obtain high efficiency. One technique for improving this linearity is distortion compensation. One of them is a digital predistortion method often used in a communication base station transmitter as shown in Patent Document 1.
　特許文献１には、通信用基地局送信機の線形性を高める技術が開示されている。特許文献１は、ディジタルプリディストーション方式の中で予歪みを作るために、べき級数方式を用いており、７次の歪み成分まで考慮している。 Patent Document 1 discloses a technique for improving the linearity of a communication base station transmitter. Patent Document 1 uses a power series method in order to create predistortion in the digital predistortion method, and considers up to the seventh-order distortion component.
　特許文献１のような従来の通信用基地局送信機によく用いられるディジタルプリディストータにおいて通信用基地局送信機内のアナログ部の非線形特性を改善する場合、予歪みをディジタル部で作成し、ＤＡＣ（Ｄｉｇｉｔａｌ ｔｏ Ａｎａｌｏｇ Ｃｏｎｖｅｒｔｅｒ）を介して、アナログ部へ出力する。このとき、アナログ部の非線形特性はどのような特性であっても良い。数１０ＭＨｚ帯域の信号を用いる場合、３次歪みまでを考慮するならＤＡＣには信号帯域の３倍の帯域が必要になるので、ＤＡＣの帯域は、１００ＭＨｚ帯域未満となる。 In a digital predistorter often used in a conventional communication base station transmitter such as Patent Document 1, in order to improve the non-linear characteristic of the analog part in the communication base station transmitter, a predistortion is created in the digital part, and the DAC The data is output to the analog unit via (Digital to Analog Converter). At this time, the non-linear characteristic of the analog unit may be any characteristic. When a signal of several tens MHz band is used, if considering up to the third order distortion, the DAC requires a band three times the signal band, so the DAC band is less than the 100 MHz band.
　特許第３９４６１８８号公報 Patent No. 3946188
　今後、さらに高速かつ大容量な通信を実現しようとすると、数１００ＭＨｚ帯域から数ＧＨｚ帯域の信号を用いることになり、ＤＡＣの帯域は非常に広帯域になる。一般的に、動作速度が上がると、ＤＡＣのビット数は、小さくなる傾向があり、量子化雑音が増加するという課題が生じる。量子化雑音を下げるために、動作速度とビット数を両方上げようとすると、ＤＡＣは、高価かつ消費電力が大きくなる。 In the future, if higher-speed and larger-capacity communication is to be realized, signals from several hundred MHz band to several GHz band will be used, and the DAC band becomes very wide. In general, when the operation speed increases, the number of bits of the DAC tends to decrease, resulting in an increase in quantization noise. In order to reduce the quantization noise, if both the operation speed and the number of bits are increased, the DAC is expensive and consumes a large amount of power.
　この発明は、上記のような問題点を解決するためになされたもので、ＤＡＣのビット数を上げずに、ＤＡＣで発生する量子化雑音を抑圧することを目的とする。 The present invention has been made to solve the above-described problems, and an object thereof is to suppress quantization noise generated in a DAC without increasing the number of bits of the DAC.
　本発明の送信機は、ディジタルの送信信号を生成するディジタル回路と、プレディストーションされた送信信号をアナログ信号に変換するディジタルアナログ変換器と、ディジタルアナログ変換器が出力したアナログの送信信号を増幅する増幅器と、ディジタルアナログ変換器と増幅器の間に設けられ、増幅器と合わせたＡＭ－ＡＭ特性が利得伸長特性になるように、ディジタルアナログ変換器が出力したアナログの送信信号の振幅を変化させ、増幅器に出力するエクスパンジョン発生回路と、ディジタル回路とディジタルアナログ変換器との間に設けられ、増幅器が増幅した送信信号の歪みをプレディストーションする歪み補償回路とを備える。 The transmitter of the present invention includes a digital circuit that generates a digital transmission signal, a digital-to-analog converter that converts the predistorted transmission signal into an analog signal, and amplifies the analog transmission signal output by the digital-analog converter. An amplifier is provided between the amplifier and the digital-analog converter and the amplifier, and the amplitude of the analog transmission signal output from the digital-analog converter is changed so that the AM-AM characteristic combined with the amplifier becomes a gain expansion characteristic. An expansion generation circuit for outputting to the digital signal, and a distortion compensation circuit that is provided between the digital circuit and the digital-analog converter and predistorts the distortion of the transmission signal amplified by the amplifier.
　この発明によれば、ＤＡＣのビット数を上げなくても、ＤＡＣで発生する量子化雑音を抑圧できる。 According to the present invention, quantization noise generated in the DAC can be suppressed without increasing the number of DAC bits.
この発明の実施の形態１に係る送信機の一構成例を示す構成図である。It is a block diagram which shows one structural example of the transmitter which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係るＥＣＣ１４の構成例及び特性を示す回路図である。It is a circuit diagram which shows the structural example and characteristic of ECC14 which concern on Embodiment 1 of this invention. 一般的な増幅器におけるＡＭ－ＡＭ特性及び歪み特性を示す図である。It is a figure which shows the AM-AM characteristic and distortion characteristic in a general amplifier. この発明の実施の形態１に係るＥＣＣ１４と増幅器１５とを合わせたときのＡＭ－ＡＭ特性である。This is an AM-AM characteristic when the ECC 14 and the amplifier 15 according to the first embodiment of the present invention are combined. 変調波信号の平均電力に対する瞬時電力の累積分布を示す図である。It is a figure which shows the cumulative distribution of the instantaneous electric power with respect to the average electric power of a modulated wave signal. この発明の実施の形態１に係るＤＡＣ７における信号の量子化を示す概念図である。It is a conceptual diagram which shows the quantization of the signal in DAC7 which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係る送信機の歪み補償前及び歪み補償後の周波数スペクトラムを示す図である。It is a figure which shows the frequency spectrum before and after distortion compensation of the transmitter which concerns on Embodiment 1 of this invention. この発明の実施の形態１に係るＤＡＣ７のビット数が１０である場合におけるＲＦモジュール９の歪み特性を示す特性図である。It is a characteristic view which shows the distortion characteristic of RF module 9 when the bit number of DAC7 which concerns on Embodiment 1 of this invention is 10. この発明の実施の形態２に係る送信機の一構成例を示す構成図である。It is a block diagram which shows one structural example of the transmitter which concerns on Embodiment 2 of this invention. この発明の実施の形態２に係る温度補償の有無に対するＲＦモジュール９の特性を示す特性図である。It is a characteristic view which shows the characteristic of RF module 9 with respect to the presence or absence of temperature compensation which concerns on Embodiment 2 of this invention. この発明の実施の形態２に係る送信機の他の構成例を示す構成図である。It is a block diagram which shows the other structural example of the transmitter which concerns on Embodiment 2 of this invention. この発明の実施の形態２に係る送信機の他の構成例を示す構成図である。It is a block diagram which shows the other structural example of the transmitter which concerns on Embodiment 2 of this invention. この発明の実施の形態２に係る送信機の他の構成例を示す構成図である。It is a block diagram which shows the other structural example of the transmitter which concerns on Embodiment 2 of this invention.
実施の形態１.
　図１は、この発明の実施の形態１に係る送信機の一構成例を示す構成図である。本送信機は、モデム１及びＲＦモジュール９を備える。 Embodiment 1.
FIG. 1 is a block diagram showing a configuration example of a transmitter according to Embodiment 1 of the present invention. The transmitter includes a modem 1 and an RF module 9.
　モデム１は、ＤＳＰ（Ｄｉｇｉｔａｌ Ｓｉｇｎａｌ Ｐｒｏｃｅｓｓｏｒ）２、変調部３、信号比較部４、ＰＤ（Ｐｒｅ-Ｄｉｓｔｏｒｔｉｏｎ）信号生成部５、ＰＤ部６、ＤＡＣ７、ＡＤＣ（Ａｎａｌｏｇ　ｔｏ　Ｄｉｇｉｔａｌ　Ｃｏｎｖｅｒｔｅｒ）８を備える。 The modem 1 includes a DSP (Digital-Signal-Processor) 2, a modulation unit 3, a signal comparison unit 4, a PD (Pre-Distortion) signal generation unit 5, a PD unit 6, a DAC 7, and an ADC (Analog to Digital Converter) 8.
　ＤＳＰ２は、変調部３を備え、ディジタル信号を生成し、変調部３でディジタル信号をベースバンド信号に変換し、出力するＤＳＰである。ＤＳＰ２は、ＰＤ６及び信号比較部４に接続される。 The DSP 2 includes a modulation unit 3, generates a digital signal, converts the digital signal into a baseband signal by the modulation unit 3, and outputs the digital signal. The DSP 2 is connected to the PD 6 and the signal comparison unit 4.
　信号比較部４は、変調部３が出力する変調波信号とＡＤ８が出力するフィードバックされた変調波信号とを比較し、その複素信号のベクトル差（ＬＵＴ（Ｌｏｏｋ Ｕｐ Ｔａｂｌｅ）方式の場合は振幅と位相の差分）をＰＤ信号生成部５に出力する信号比較部である。信号比較部４は、ＤＳＰ２及びＡＤＣ８に接続される。 The signal comparison unit 4 compares the modulation wave signal output from the modulation unit 3 with the feedback modulation wave signal output from the AD 8, and compares the complex signal vector difference (in the case of the LUT (Look Up Table) method with the amplitude). It is a signal comparison unit that outputs a phase difference) to the PD signal generation unit 5. The signal comparison unit 4 is connected to the DSP 2 and the ADC 8.
　ＰＤ信号生成部５は、信号比較部４が出力する比較結果にしたがって、ＤＡＣ７から増幅器１５までで発生した歪みを打ち消す逆信号を生成するＰＤ信号生成部である。ＰＤ信号生成部５は、ＰＤ６及び信号比較部４に接続される。ＰＤ信号生成部５で逆信号を生成する手法として、例えば、ＬＵＴ方式、多項式方式やべき級数方式、メモリポリノミナル方式、もしくはヴォルテラ級数方式が用いられる。 The PD signal generation unit 5 is a PD signal generation unit that generates an inverse signal that cancels distortion generated from the DAC 7 to the amplifier 15 according to the comparison result output from the signal comparison unit 4. The PD signal generation unit 5 is connected to the PD 6 and the signal comparison unit 4. As a method of generating an inverse signal by the PD signal generation unit 5, for example, an LUT method, a polynomial method, a power series method, a memory polynomial method, or a Volterra series method is used.
　ＰＤ６は、ＰＤ信号生成部５が生成した逆信号と変調部３が生成したベースバンド信号とを重畳し、重畳した信号を出力するＰＤである。ＰＤ６は、ＤＳＰ２及びＤＡＣ７に接続される。 PD 6 is a PD that superimposes the reverse signal generated by the PD signal generation unit 5 and the baseband signal generated by the modulation unit 3 and outputs the superimposed signal. The PD 6 is connected to the DSP 2 and the DAC 7.
　ＤＡＣ７は、ＰＤ６が出力するディジタル信号をアナログ信号に変換するＤＡＣである。ＤＡＣ７は、ＰＤ６及びＢＰＦ１０に接続される。 The DAC 7 is a DAC that converts a digital signal output from the PD 6 into an analog signal. The DAC 7 is connected to the PD 6 and the BPF 10.
　ＡＤＣ８は、ＢＰＦ２１が出力するアナログ信号をディジタル信号に変化するＡＤＣである。ＡＤＣ８は、ＢＰＦ２１及び信号比較部４に接続される。 The ADC 8 is an ADC that changes an analog signal output from the BPF 21 into a digital signal. The ADC 8 is connected to the BPF 21 and the signal comparison unit 4.
　例えば、ＤＳＰ２、信号比較部４、ＰＤ信号生成部５、ＰＤ６は、ＦＰＧＡ（Ｆｉｅｌｄ－Ｐｒｏｇｒａｍｍａｂｌｅ Ｇａｔｅ Ａｒｒａｙ）、ＡＳＩＣ（Ａａｐｐｌｉｃａｔｉｏｎ　Ｓｐｅｃｉｆｉｃ　Ｉｎｔｅｇｒａｔｅｄ　Ｃｉｒｃｕｉｔ）などで構成される。また、ＤＡＣ７及びＡＤＣ８は、ＦＰＧＡに内蔵される構成であっても良い。 For example, the DSP 2, the signal comparison unit 4, the PD signal generation unit 5, and the PD 6 are configured by an FPGA (Field-Programmable Gate Array), an ASIC (Application Specific Integrated Circuit), and the like. Further, the DAC 7 and the ADC 8 may be configured to be built in the FPGA.
　ＲＦモジュール９は、ＢＰＦ（Ｂａｎｄ‐Ｐａｓｓ　Ｆｉｌｔｅｒ）１０、ミクサ１１、ＬＯ（ローカル信号源）１２、ＢＰＦ１３、ＥＣＣ（Ｅｘｐａｎｓｉｏｎ Ｃｈａｒａｃｔｅｒｉｓｔｉｃｓ Ｃｉｒｃｕｉｔ）１４、増幅器１５、カップラ１６、アイソレータ１７、ＬＰＦ（Ｌｏｗ－Ｐａｓｓ　Ｆｉｌｔｅｒ）１８、アンテナ１９、ミクサ２０、ＢＰＦ２１を備える。 The RF module 9 includes a BPF (Band-Pass Filter) 10, a mixer 11, an LO (Local Signal Source) 12, a BPF 13, an ECC (Expansion / Characteristics / Circuit) 14, an amplifier 15, a coupler 16, an isolator 17, an LPF (Low-Pass Filter). ) 18, an antenna 19, a mixer 20, and a BPF 21.
　ＢＰＦ１０は、ＤＡＣ７が出力する変調信号から不要波を遮断し、不要波を遮断した変調信号を出力するＢＰＦである。ＢＰＦ１０は、ＤＡＣ７及びミクサ１１に接続される。 The BPF 10 is a BPF that blocks unnecessary waves from the modulation signal output from the DAC 7 and outputs a modulation signal that blocks unnecessary waves. The BPF 10 is connected to the DAC 7 and the mixer 11.
　ミクサ１１は、ＢＰＦ１１が出力する変調信号とＬＯ１２のローカル信号とを混合し、変調信号の周波数を変換する混合器である。ミクサ１１は、ＬＯ端子がＬＯ１２に接続され、ＩＦ端子がＢＰＦ１１に接続され、ＲＦ端子がＢＰＦ１３に接続される。 The mixer 11 is a mixer that mixes the modulation signal output from the BPF 11 and the local signal of the LO 12 and converts the frequency of the modulation signal. The mixer 11 has an LO terminal connected to the LO 12, an IF terminal connected to the BPF 11, and an RF terminal connected to the BPF 13.
　ＬＯ１２は、ミクサ１１において周波数を変換するのに用いられるローカル信号を出力する発振器である。ＬＯ１２は、ミクサ１１及びミクサ２０に接続される。 LO 12 is an oscillator that outputs a local signal used for frequency conversion in the mixer 11. The LO 12 is connected to the mixer 11 and the mixer 20.
　ＢＰＦ１３は、周波数変換された変調信号から所望帯域外の不要波を遮断し、不要波を遮断した変調信号を出力するＢＰＦである。ＢＰＦ１３は、ミクサ１１及びＥＣＣ１４に接続される。 The BPF 13 is a BPF that blocks unnecessary waves outside the desired band from the frequency-converted modulated signal and outputs a modulated signal that blocks unnecessary waves. The BPF 13 is connected to the mixer 11 and the ECC 14.
　ＥＣＣ１４は、変調信号のＡＭ－ＡＭ特性を利得伸長（ゲインエクスパンジョン）させるアナログ回路である。ＥＣＣ１４は、ＢＰＦ１３及び増幅器１５に接続される。
　図２は、この発明の実施の形態１に係るＥＣＣ１４の構成例及び特性を示す回路図である。
　図２の（ａ）は、動作級がＢ級からＡＢ級の増幅器を用いた場合の例であり、出力電力に対して利得が増加する特性をもつ。Ｖｃは、増幅器のバイアス電圧であり、Ｖｃを調整してＢ級からＡＢ級の範囲にバイアスを設定する。
　図２の（ｂ）は、並列型ダイオードリニアライザーを用いた例であり、ダイオードの非線形特性を利用して、出力電力に対して利得を増加させる特性をもたせる。
　図２の（ｃ）は、ダイオード及びハイブリッドを用いてＥＣＣ１４を構成した図２の（ｂ）の変形例であり、ダイオードの非線形特性を利用して、出力電力に対して利得を増加させる特性をもたせる。
　ここでは、出力電力に対する利得特性を例に説明したが、入力電力に対する利得特性の場合も出力電力の場合と同様である。 The ECC 14 is an analog circuit that performs gain expansion (gain expansion) on the AM-AM characteristic of the modulation signal. The ECC 14 is connected to the BPF 13 and the amplifier 15.
FIG. 2 is a circuit diagram showing a configuration example and characteristics of the ECC 14 according to the first embodiment of the present invention.
FIG. 2A shows an example in which an amplifier whose operation class is from class B to class AB is used, and has a characteristic that the gain increases with respect to the output power. Vc is a bias voltage of the amplifier, and Vc is adjusted to set a bias in a range from class B to class AB.
FIG. 2B shows an example in which a parallel diode linearizer is used, and has a characteristic of increasing the gain with respect to output power by using the nonlinear characteristic of the diode.
FIG. 2C is a modification of FIG. 2B in which the ECC 14 is configured using a diode and a hybrid. The characteristic of increasing the gain with respect to the output power using the nonlinear characteristic of the diode is shown. Give it.
Here, the gain characteristic with respect to the output power has been described as an example, but the gain characteristic with respect to the input power is the same as that with the output power.
　増幅器１５は、利得伸長された変調信号が入力され、その変調信号を増幅する増幅器である。増幅器１５は、ＥＣＣ１４及びカップラ１６に接続される。 The amplifier 15 is an amplifier that receives a gain-expanded modulation signal and amplifies the modulation signal. The amplifier 15 is connected to the ECC 14 and the coupler 16.
　カップラ１６は、増幅された変調信号の一部を取り出すカップラである。カップラ１６は、増幅器１５及びアイソレータ１７に接続される。 The coupler 16 is a coupler that extracts a part of the amplified modulation signal. The coupler 16 is connected to the amplifier 15 and the isolator 17.
　アイソレータ１７は、変調信号を通過させ、後段に接続されるコンポーネントにより生じる反射波を吸収するアイソレータである。アイソレータ１７は、カップラ１６及びＬＰＦ１８に接続される。 The isolator 17 is an isolator that allows a modulated signal to pass and absorbs a reflected wave generated by a component connected to a subsequent stage. The isolator 17 is connected to the coupler 16 and the LPF 18.
　ＬＰＦ１８は、変調信号から所望帯域外の不要波を遮断し、不要波を遮断した変調信号を出力するＬＰＦである。ＬＰＦ１８は、アイソレータ１７及びアンテナ１９に接続される。 The LPF 18 is an LPF that blocks unnecessary waves outside the desired band from the modulation signal and outputs a modulation signal that blocks unnecessary waves. The LPF 18 is connected to the isolator 17 and the antenna 19.
　アンテナ１９は、ＬＰＦ１８が出力した変調信号を送信するアンテナである。アンテナ１９は、ＬＰＦ１８に接続される。 The antenna 19 is an antenna that transmits the modulation signal output from the LPF 18. The antenna 19 is connected to the LPF 18.
　ミクサ２０は、ＬＯ１２のローカル信号と増幅器１５が増幅した変調信号とを混合し、変調信号を周波数変換する混合器である。ミクサ２０は、ＲＦ端子がカップラ１６に接続され、ＬＯ端子がＬＯ１２に接続され、ＩＦ端子がＢＰＦ２１に接続される。 The mixer 20 is a mixer that mixes the local signal of the LO 12 and the modulation signal amplified by the amplifier 15 and converts the frequency of the modulation signal. The mixer 20 has an RF terminal connected to the coupler 16, an LO terminal connected to the LO 12, and an IF terminal connected to the BPF 21.
　ＢＰＦ２１は、ミクサ２０が周波数変換した変調信号から不要波を遮断し、不要波を遮断した変調信号を出力するＢＰＦである。ＢＰＦ２１は、ミクサ２０及びＡＤＣ８に接続される。 The BPF 21 is a BPF that blocks unnecessary waves from the modulation signal frequency-converted by the mixer 20 and outputs a modulation signal that blocks unnecessary waves. The BPF 21 is connected to the mixer 20 and the ADC 8.
　次に、この発明の実施の形態１に係る送信機の動作について説明する。
　モデム１において、ＤＳＰ２は、ＤＳＰ２内の変調部３で生成された変調波信号を、ＰＤ６を介してＤＡＣ７に出力する。ＤＡＣ７は、入力されたベースバンド信号をアナログ信号に変換し、ＢＰＦ１０に出力する。 Next, the operation of the transmitter according to Embodiment 1 of the present invention will be described.
In the modem 1, the DSP 2 outputs the modulation wave signal generated by the modulation unit 3 in the DSP 2 to the DAC 7 via the PD 6. The DAC 7 converts the input baseband signal into an analog signal and outputs the analog signal to the BPF 10.
　ＲＦモジュール９において、ＢＰＦ１０は、ＤＡＣ７が出力するアナログ信号からＢＰＦ１０の通過帯域外の不要波を抑圧し、不要波を抑圧したアナログ信号をミクサ１１に出力する。 In the RF module 9, the BPF 10 suppresses unnecessary waves outside the passband of the BPF 10 from the analog signal output by the DAC 7, and outputs an analog signal in which unnecessary waves are suppressed to the mixer 11.
　ミクサ１１は、ＬＯ１２のローカル信号を用いて、ＢＰＦ１０が出力するアナログ信号をアップコンバージョンし、ＲＦ信号に変換する。ミクサ１１は、ＲＦ信号をＢＰＦ１３に出力する。 The mixer 11 up-converts the analog signal output from the BPF 10 using the local signal of the LO 12 and converts it into an RF signal. The mixer 11 outputs an RF signal to the BPF 13.
　ミクサ１１が周波数変換を行なう際、アナログ信号とローカル信号とを混合するため、所望のＲＦ信号の他に、不要波も出力される。ＢＰＦ１１は、ミクサ１１の出力信号から不要波を遮断し、所望のＲＦ信号をＥＣＣ１４に出力する。 When the mixer 11 performs frequency conversion, an analog signal and a local signal are mixed, so that an unnecessary wave is output in addition to a desired RF signal. The BPF 11 blocks unnecessary waves from the output signal of the mixer 11 and outputs a desired RF signal to the ECC 14.
　ＥＣＣ１４は、ＡＭ―ＡＭ特性が利得伸長の特性をもち、入力されるＲＦ信号の電力に応じて利得を変化させ、増幅器１５に出力する。この点についての詳細は後述する。 The ECC 14 has an AM-AM characteristic having a gain expansion characteristic, changes the gain according to the power of the input RF signal, and outputs the gain to the amplifier 15. Details of this point will be described later.
　増幅器１５は、ＥＣＣ１４が出力したＲＦ信号を増幅し、増幅したＲＦ信号をカップラ１６及びアイソレータ１７を介してＬＰＦ１８に出力する。
　ＬＰＦ１８は、増幅器１５が増幅したＲＦ信号のうち、不要波を抑圧し、不要波を抑圧したＲＦ信号をアンテナ１９に出力する。
　アンテナ１９は、ＬＰＦ１８が出力したＲＦ信号を送信する。 The amplifier 15 amplifies the RF signal output from the ECC 14 and outputs the amplified RF signal to the LPF 18 via the coupler 16 and the isolator 17.
The LPF 18 suppresses unnecessary waves in the RF signal amplified by the amplifier 15 and outputs an RF signal in which unnecessary waves are suppressed to the antenna 19.
The antenna 19 transmits the RF signal output from the LPF 18.
　一方、カップラ１６は、増幅器１５が増幅したＲＦ信号をフィードバックさせるために、ＲＦ信号の一部を取り出し、ミクサ２０に出力する。
　ミクサ２０は、ＬＯ１２のローカル信号を用いて、カップラ１６から入力されたＲＦ信号をダウンコンバーションし、ベースバンド信号に変換する。
　ＢＰＦ２１は、ミクサ２０の出力信号のうち、不要波を抑圧し、不要波を抑圧したベースバンド信号をＡＤＣ８に出力する。 On the other hand, the coupler 16 takes out a part of the RF signal and outputs it to the mixer 20 in order to feed back the RF signal amplified by the amplifier 15.
The mixer 20 uses the local signal of the LO 12 to down-convert the RF signal input from the coupler 16 and convert it into a baseband signal.
The BPF 21 suppresses unnecessary waves in the output signal of the mixer 20 and outputs a baseband signal in which unnecessary waves are suppressed to the ADC 8.
　ＡＤＣ８は、ベースバンド信号をディジタル信号に変換し、信号比較部４に出力する。
　信号比較部４は、ＤＳＰ２が生成した変調信号と、ＥＣＣ１４及び増幅器１５を通過してフィードバックされた変調信号とを比較し、その差分をＰＤ信号生成部５に出力する。
　ＰＤ信号生成部５は、信号比較部４の出力信号にしたがって、ＥＣＣ１４及び増幅器１５で生じた歪みを打ち消す逆信号を生成し、ＰＤ６に出力する。
　ＰＤ６は、ＤＳＰ２が生成した変調信号にＥＣＣ１４及び増幅器１５の歪みを補償する逆信号を重畳し、重畳した変調信号をＤＡＣ７に出力する。
　このように、生成した変調信号と、ＥＣＣ１４及び増幅器１５を通過してフィードバックされた変調信号とを比較し、ＥＣＣ１４及び増幅器１５の非線形特性を打ち消す逆信号を変調信号に重畳することで、ＥＣＣ１４及び増幅器１５で生じる歪みを補償している。このように歪み補償を行うことをプリディストーションと言う。 The ADC 8 converts the baseband signal into a digital signal and outputs it to the signal comparison unit 4.
The signal comparison unit 4 compares the modulation signal generated by the DSP 2 with the modulation signal fed back through the ECC 14 and the amplifier 15, and outputs the difference to the PD signal generation unit 5.
The PD signal generation unit 5 generates an inverse signal that cancels distortion generated in the ECC 14 and the amplifier 15 in accordance with the output signal of the signal comparison unit 4 and outputs the reverse signal to the PD 6.
The PD 6 superimposes an inverse signal that compensates for distortion of the ECC 14 and the amplifier 15 on the modulation signal generated by the DSP 2, and outputs the superimposed modulation signal to the DAC 7.
In this way, the generated modulation signal is compared with the modulation signal fed back through the ECC 14 and the amplifier 15, and an inverse signal that cancels the nonlinear characteristics of the ECC 14 and the amplifier 15 is superimposed on the modulation signal. The distortion generated in the amplifier 15 is compensated. This distortion compensation is called predistortion.
　ＥＣＣ１４の動作について詳しく説明する。
　図３は、一般的な増幅器におけるＡＭ－ＡＭ特性及び歪み特性を示す図である。一般的に、増幅器のＡＭ－ＡＭ特性が、点線→破線→実線のように変化すると、歪み特性は、破線→点線→実線の順に劣化する。ＡＭ－ＡＭ特性が平坦であるほど、歪み特性が良くなるからである。このように、ＡＭ－ＡＭ特性と歪み特性とは関係しており、歪みを改善するためには、ＡＭ－ＡＭ特性を平坦にすることが望まれる。
　図４は、この発明の実施の形態１に係るＥＣＣ１４と増幅器１５とを合わせたときのＡＭ－ＡＭ特性である。破線が従来の特性（ＥＣＣ１４がない場合の特性）であり、点線が歪み特性が良い場合の特性であり、本発明の特性（ＥＣＣ１４がある場合の特性）である。 The operation of the ECC 14 will be described in detail.
FIG. 3 is a diagram showing AM-AM characteristics and distortion characteristics in a general amplifier. Generally, when the AM-AM characteristic of an amplifier changes in the order of dotted line → broken line → solid line, the distortion characteristic deteriorates in the order of broken line → dotted line → solid line. This is because the flatter the AM-AM characteristic, the better the distortion characteristic. As described above, the AM-AM characteristic and the distortion characteristic are related, and it is desired to flatten the AM-AM characteristic in order to improve the distortion.
FIG. 4 shows AM-AM characteristics when the ECC 14 and the amplifier 15 according to the first embodiment of the present invention are combined. The broken line is the conventional characteristic (characteristic when there is no ECC 14), the dotted line is the characteristic when the distortion characteristic is good, and is the characteristic of the present invention (characteristic when the ECC 14 is present).
　通常、図４の点線になるように、増幅器の前段にＥＣＣ１４に相当するアナログ回路を設けるが、ＥＣＣ１４は、ＥＣＣ１４と増幅器１５とを合わせたＡＭ－ＡＭ特性が、図４の実線になるように、入力信号の電力に応じで利得伸長を行う。つまり、ＥＣＣ１４は、増幅器１５の歪みを補償するのではなく、増幅器１５の歪みを劣化させても、ＥＣＣ１４と増幅器１５とを合わせたＡＭ－ＡＭ特性が利得伸長特性になるようにする。これにより、信号の歪みは劣化するが、後述するようにＤＡＣ７が出力する電力の最大値を低減できる。なお、劣化した歪みは、ＰＤ６により補償する。 Normally, an analog circuit corresponding to the ECC 14 is provided at the front stage of the amplifier as shown by the dotted line in FIG. 4, but the ECC 14 is such that the AM-AM characteristic of the ECC 14 and the amplifier 15 is the solid line in FIG. The gain is expanded according to the power of the input signal. That is, the ECC 14 does not compensate for the distortion of the amplifier 15 but causes the AM-AM characteristic of the ECC 14 and the amplifier 15 to be the gain expansion characteristic even if the distortion of the amplifier 15 is deteriorated. Thereby, although the distortion of the signal is deteriorated, the maximum value of the power output from the DAC 7 can be reduced as will be described later. The deteriorated distortion is compensated by the PD 6.
　利得伸長とビット数との関係について説明する。
　図５は、変調波信号の平均電力に対する瞬時電力の累積分布を示す図である。縦軸がＣＣＤＦ（Ｃｏｍｐｌｅｍｅｎｔａｒｙ Ｃｕｍｕｌａｔｉｖｅ Ｄｉｓｔｒｉｂｕｔｉｏｎ Ｆｕｎｃｔｉｏｎ）であり、横軸が瞬時電力である。図５に示す変調信号は、ＰＡＰＲ（Ｐｅａｋ ｔｏ Ａｖｅｒａｇｅ Ｐｏｗｅｒ Ｒａｔｉｏ）が約１０ｄＢであるため、ＤＡＣ７は、ＥＣＣ１４がない場合、ＰＡＰＲ分の１０ｄＢ及びＰＤ６の歪み補償分のＡｄＢのダイナミックレンジが必要となる。増幅器は、一般的に、入力電力が大きくなるほど利得が低下するので、その利得の低下を補償するためにＡｄＢの利得伸長が必要になる。
　したがって、変調信号の帯域が広くなり、ＰＡＰＲが大きくなるほど、増幅器が非線形動作し、歪み補償量が大きくなるので、ＤＡＣ７に必要なダイナミックレンジは大きくなる。一般的に、ダイナミックレンジとビット数を上げると、ＤＡＣの動作速度は遅くなる。 The relationship between gain expansion and the number of bits will be described.
FIG. 5 is a diagram showing a cumulative distribution of instantaneous power with respect to the average power of the modulated wave signal. The vertical axis is CCDF (Complementary Cumulative Distribution Function), and the horizontal axis is instantaneous power. Since the modulated signal shown in FIG. 5 has a peak to average power ratio (PAPR) of about 10 dB, the DAC 7 requires a dynamic range of 10 dB for PAPR and Ad dB for distortion compensation of PD6 when there is no ECC14. . In general, the gain of the amplifier decreases as the input power increases. Therefore, it is necessary to increase the gain of AdB in order to compensate for the decrease in the gain.
Therefore, the wider the band of the modulation signal and the larger the PAPR, the more the amplifier operates non-linearly and the amount of distortion compensation increases, so the dynamic range required for the DAC 7 increases. In general, when the dynamic range and the number of bits are increased, the operation speed of the DAC is decreased.
　一方、ＥＣＣ１４がある場合、ＥＣＣ１４で利得伸長を行うので、その分、ＤＡＣ７に掛かる負担は小さくなる。
　図６は、この発明の実施の形態１に係るＤＡＣ７における信号の量子化を示す概念図である。
　点線が、ＥＣＣ１４がない場合の入力信号波形であり、実線が、ＥＣＣ１４がある場合の入力信号波形である。
　図６に示すように、ＥＣＣ１４がある場合、ＥＣＣ１４で利得伸長するので、ＥＣＣ１４がない場合に比べて、ＤＡＣ７が出力する信号の振幅は小さくなる。したがって、ＤＡＣ７のビット数をＸビットとすると、ＥＣＣ１４がある場合の方が、同じビット数でも、ＥＣＣ１４がない場合に比べて、１ビット当たりの分解能（図６中のステップ幅に相当する）は、高くなる。１ビット当たりの分解能が高い方（図６中のステップ幅が小さい方）が、量子化ノイズは小さくなるので、ＥＣＣ１４を設けた方が、ＤＡＣ７における量子化ノイズを小さくできる。 On the other hand, when the ECC 14 is present, the gain is extended by the ECC 14, so that the burden on the DAC 7 is reduced accordingly.
FIG. 6 is a conceptual diagram showing signal quantization in the DAC 7 according to the first embodiment of the present invention.
A dotted line is an input signal waveform when the ECC 14 is not present, and a solid line is an input signal waveform when the ECC 14 is present.
As shown in FIG. 6, when the ECC 14 is present, the gain is expanded by the ECC 14, so that the amplitude of the signal output from the DAC 7 is smaller than when the ECC 14 is not present. Therefore, when the number of bits of the DAC 7 is X bits, the resolution per bit (corresponding to the step width in FIG. 6) in the case where the ECC 14 is present is equal to that in the case where the ECC 14 is not present even when the ECC 14 is present. , Get higher. The higher the resolution per bit (the smaller the step width in FIG. 6), the smaller the quantization noise. Therefore, if the ECC 14 is provided, the quantization noise in the DAC 7 can be reduced.
　もし、ＥＣＣ１４がない場合において、ＥＣＣ１４がある場合と同じ量子化ノイズを実現するためには、Ｘビットより大きいビット数のＤＡＣ７を用いる必要がある。その場合、ＤＡＣ７は、高価かつ消費電力が大きくなる。 If there is no ECC 14, it is necessary to use a DAC 7 having a number of bits larger than X bits in order to realize the same quantization noise as when the ECC 14 is present. In that case, the DAC 7 is expensive and consumes a large amount of power.
　図７は、この発明の実施の形態１に係る送信機の歪み補償前及び歪み補償後の周波数スペクトラムを示す図である。
　従来の場合（ＥＣＣ１４がない場合）、ＤＡＣ７で利得伸長を行う分、量子化ノイズが増加するため、歪み補償後のノイズフロアが上昇するが、本発明では、量子化ノイズを下げられるため、歪み補償後のノイズフロアを下げることができる。 FIG. 7 is a diagram showing a frequency spectrum before and after distortion compensation of the transmitter according to Embodiment 1 of the present invention.
In the conventional case (when there is no ECC 14), the noise floor after distortion compensation is increased because the quantization noise is increased by the amount of gain expansion by the DAC 7. However, in the present invention, since the quantization noise is lowered, the distortion noise is reduced. The noise floor after compensation can be lowered.
　図８は、この発明の実施の形態１に係るＤＡＣ７のビット数が１０である場合におけるＲＦモジュール９の歪み特性を示す特性図である。
　縦軸がＡＣＰＲ（Ａｄｊａｃｅｎｔ Ｃｈａｎｎｅｌ Ｐｏｗｅｒ Ｒａｔｉｏ）であり、横軸がＰｏｕｔである。本構成では、ＥＣＣ１４を設けて、利得伸長を行うことにより、ＤＡＣ７が出力する信号の最大電力値を下げられるので、１ビット当たりの分解能を高めることができ、従来構成（ＥＣＣ１４がない構成）に比べて、歪み特性を改善できる。従来構成で、同じ歪み特性を得るためには、ＤＡＣ７は１２ビット程度必要であるが、本構成では、１０ビットで、１２ビット程度の歪み特性を得ることができる。 FIG. 8 is a characteristic diagram showing distortion characteristics of the RF module 9 when the number of bits of the DAC 7 according to Embodiment 1 of the present invention is 10. In FIG.
The vertical axis is ACPR (Adjacent Channel Power Ratio), and the horizontal axis is Pout. In this configuration, by providing the ECC 14 and performing gain expansion, the maximum power value of the signal output from the DAC 7 can be reduced, so that the resolution per bit can be increased and the conventional configuration (configuration without the ECC 14) can be achieved. Compared to the distortion characteristics. In order to obtain the same distortion characteristics in the conventional configuration, the DAC 7 needs about 12 bits. In this configuration, the distortion characteristics of about 12 bits can be obtained with 10 bits.
　以上の通り、この発明の実施の形態１によれば、ＥＣＣ１４を用いて、増幅器１５のＡＭ－ＡＭ特性を利得伸長特性にするので、ＤＡＣ７の量子化ノイズの増加を抑制することができる。 As described above, according to the first embodiment of the present invention, the ECC 14 is used to change the AM-AM characteristic of the amplifier 15 to the gain expansion characteristic, so that an increase in quantization noise of the DAC 7 can be suppressed.
　なお、実施の形態１のＲＦモジュール９は、受信機能を有していないが、受信機能を有するようにしても良い。 Note that the RF module 9 of the first embodiment does not have a reception function, but may have a reception function.
実施の形態２.
　図９は、この発明の実施の形態２に係る送信機の一構成例を示す構成図である。
　図９において、図１と同一符号は、同一または相当部分を示している。ＶＧＡ（Ｖａｒｉａｂｌｅ　Ｇａｉｎ　Ａｍｐｌｉｆｉｅｒ）２２及びＶＧＡ２３、ＴＳ２４ａ、制御部２５ａを設けている点が、図１と異なる。なお、ＢＰＦ１０とＢＰＦ１０ａ、ミクサ１１とミクサ１１ａ、ＬＯ１２とＬＯ１２ａ、ＢＰＦ１３とＢＰＦ１３ａ、ＥＣＣ１４とＥＣＣ１４ａ、増幅器１５と増幅器１５ａ、カップラ１６とカップラ１６ａ、アイソレータ１７とアイソレータ１７ａ、ＬＰＦ１８とＬＰＦ１８ａ、アンテナ１９とアンテナ１９ａ、ミクサ２０とミクサ２０ａ、ＢＰＦ２１とＢＰＦ２１ａは対応する。 Embodiment 2.
FIG. 9 is a block diagram showing a configuration example of a transmitter according to Embodiment 2 of the present invention.
9, the same reference numerals as those in FIG. 1 denote the same or corresponding parts. 1 is different from FIG. 1 in that a VGA (Variable Gain Amplifier) 22, a VGA 23, a TS 24a, and a control unit 25a are provided. BPF 10 and BPF 10a, mixer 11 and mixer 11a, LO 12 and LO 12a, BPF 13 and BPF 13a, ECC 14 and ECC 14a, amplifier 15 and amplifier 15a, coupler 16 and coupler 16a, isolator 17 and isolator 17a, LPF 18 and LPF 18a, antenna 19 and antenna 19a, the mixer 20 and the mixer 20a, and the BPF 21 and the BPF 21a correspond to each other.
　ＶＧＡ２２は、ＥＣＣ１４の前段に設けられ、利得を変化させることにより、ＥＣＣ１４に入力される信号の電力レベルを調整するＶＧＡである。
　ＶＧＡ２３は、ＥＣＣ１４の後段に設けられ、利得を変化させることにより、ＥＣＣ１４が出力する信号の電力レベルを調整するＶＧＡである。
　ＴＳ２４ａは、ＲＦモジュール９の温度をモニタする温度センサである。ＴＳ２４ａは、制御部２５ａに接続され、温度情報を制御部２５ａに送信する。
　制御部２５ａは、ＶＧＡ２２及び２３、ＥＣＣ１４、増幅器１５のバイアス電圧を制御する制御部である。制御部２５ａは、ＴＳ２４ａから温度情報を受信し、その温度情報にしたがって、ＶＧＡ２２及び２３、ＥＣＣ１４、増幅器１５のバイアス電圧を変化させることにより、それらの利得を制御する。 The VGA 22 is a VGA that is provided in front of the ECC 14 and adjusts the power level of the signal input to the ECC 14 by changing the gain.
The VGA 23 is a VGA that is provided at the subsequent stage of the ECC 14 and adjusts the power level of the signal output from the ECC 14 by changing the gain.
The TS 24 a is a temperature sensor that monitors the temperature of the RF module 9. The TS 24a is connected to the control unit 25a and transmits temperature information to the control unit 25a.
The control unit 25 a is a control unit that controls the bias voltages of the VGAs 22 and 23, the ECC 14, and the amplifier 15. The control unit 25a receives temperature information from the TS 24a, and controls the gains of the VGAs 22 and 23, the ECC 14, and the amplifier 15 by changing the bias voltage according to the temperature information.
　次に、この発明の実施の形態２に係る送信機の動作について説明する。実施の形態１と同様の動作については説明を省略する。
　ＲＦモジュール９は、温度により特性が異なるが、図９に示したように、信号比較部４で生成した変調信号とフィードバックした変調信号とを比較し、両者が一致するようにＰＤ６でプリディストーションを行う場合、温度による特性変動は、フィードバック制御により歪み補償される。
　しかし、ＲＦモジュール９の特性は利得伸長特性でないと、ＤＡＣ７において量子化ノイズが増加し、ノイズフロアが上昇してしまう。したがって、温度変化が生じてもＲＦモジュール９が利得伸長特性になるように、制御を行なう必要がある。 Next, the operation of the transmitter according to Embodiment 2 of the present invention will be described. The description of the same operation as that in Embodiment 1 is omitted.
Although the RF module 9 has different characteristics depending on the temperature, as shown in FIG. 9, the modulation signal generated by the signal comparison unit 4 is compared with the modulated signal fed back, and predistortion is performed by the PD 6 so that the two match. When performing, the characteristic variation due to temperature is compensated for distortion by feedback control.
However, if the characteristic of the RF module 9 is not the gain expansion characteristic, the quantization noise increases in the DAC 7 and the noise floor increases. Therefore, it is necessary to perform control so that the RF module 9 has a gain expansion characteristic even when a temperature change occurs.
　温度が上昇した場合、ＥＣＣ１４に流れる電流は小さくなるので、入力電力に対して利得が増加する特性（利得伸長特性）は小さくなる。また、増幅器１５の利得及び飽和出力電力も減少し、増幅器１５は、入力電力に対して利得が低下する特性になる。
　本送信機では、ＴＳ２４ａで温度を検出し、その温度に応じてＲＦモジュール９のＡＭ－ＡＭ特性が利得伸長特性になるように、ＥＣＣ１４及びＥＣＣ１５のバイアス電圧またはバイアス電流を調整する。これにより、温度が変化してもＲＦモジュール９は、利得伸長特性を維持できる。 When the temperature rises, the current flowing through the ECC 14 becomes small, so that the characteristic that the gain increases with respect to the input power (gain expansion characteristic) becomes small. Further, the gain of the amplifier 15 and the saturated output power are also reduced, and the amplifier 15 has a characteristic that the gain is reduced with respect to the input power.
In this transmitter, the temperature is detected by the TS 24a, and the bias voltage or bias current of the ECC 14 and the ECC 15 is adjusted so that the AM-AM characteristic of the RF module 9 becomes the gain expansion characteristic according to the temperature. Thereby, even if temperature changes, the RF module 9 can maintain a gain expansion characteristic.
　図１０は、この発明の実施の形態２に係る温度補償の有無に対するＲＦモジュール９の特性を示す特性図である。
　破線が、温度に対してＥＣＣ１４及び増幅器１５のバイアス電圧を制御しない場合のＲＦモジュール９の特性であり、実線が、温度に対してＥＣＣ１４及び増幅器１５のバイアス電圧を制御する場合のＲＦモジュール９の特性である。点線は、利得特性が平坦の場合のＲＦモジュール９の特性であり、実線及び破線の比較対象となる特性である。この特性より利得が上に行く特性が、利得伸長特性である。 FIG. 10 is a characteristic diagram showing characteristics of the RF module 9 with and without temperature compensation according to Embodiment 2 of the present invention.
The broken line is the characteristic of the RF module 9 when the ECC 14 and the bias voltage of the amplifier 15 are not controlled with respect to the temperature, and the solid line is the characteristic of the RF module 9 when the bias voltage of the ECC 14 and the amplifier 15 is controlled with respect to the temperature. It is a characteristic. A dotted line is a characteristic of the RF module 9 when the gain characteristic is flat, and is a characteristic to be compared with a solid line and a broken line. A characteristic in which the gain is higher than this characteristic is a gain expansion characteristic.
　ＥＣＣ１４及び増幅器１５のバイアス電圧を変化させると、利得が変化するため、利得伸長特性は維持できても、ＲＦモジュール９の全体の利得は変化してしまう。温度により、ＲＦモジュール９の全体の利得が変化すると、アンテナ１９ａから送信される電波の電力が温度によって異なることになるため、特性として好ましくない。
　そこで、制御部２５ａは、温度によりＶＧＡ２２及び２３の利得も制御することにより、ＲＦモジュール９の全体の利得が、温度によらず一定になるようにする。これにより、温度が変化しても、ＲＦモジュール９の全体の動作利得は、一定のまま、利得伸長特性を維持できる。 When the bias voltage of the ECC 14 and the amplifier 15 is changed, the gain changes. Therefore, even if the gain expansion characteristic can be maintained, the overall gain of the RF module 9 changes. If the overall gain of the RF module 9 changes depending on the temperature, the power of the radio wave transmitted from the antenna 19a varies depending on the temperature.
Therefore, the control unit 25a also controls the gains of the VGAs 22 and 23 according to the temperature so that the overall gain of the RF module 9 becomes constant regardless of the temperature. As a result, even if the temperature changes, the overall gain of the RF module 9 can be maintained and the gain expansion characteristic can be maintained.
　以上の通り、この発明の実施の形態２によれば、温度が変化してもＲＦモジュール９が利得伸長特性を維持するように、ＥＣＣ１４及び増幅器１５の利得を制御するので、温度変化による量子化雑音の増加を抑制できる。 As described above, according to the second embodiment of the present invention, the gain of the ECC 14 and the amplifier 15 is controlled so that the RF module 9 maintains the gain expansion characteristic even when the temperature changes. An increase in noise can be suppressed.
　図１１は、この発明の実施の形態２による送信機の他の構成例を示す構成図である。図９と図１１とを比較すると、モデム１とモデム１ａ、ＤＳＰ２とＤＳＰ２ａ、変調部３と変調部３ａ、信号比較部４と信号比較部４ａ及び４ｂ、ＰＤ信号生成部５とＰＤ信号生成部５ａ及び５ｂ、ＰＤ６とＰＤ６ａ及び６ｂ、ＤＡＣ７とＤＡＣ７ａ及び７ｂ、ＡＤＣ８とＡＤＣ８ａ及び８ｂ、ＲＦモジュール９とＲＦモジュール９ａ及び９ｂ、ＢＰＦ１０とＢＰＦ１０ａ及び１０ｂ、ミクサ１１とミクサ１１ａ及び１１ｂ、ＬＯ１２とＬＯ１２ａ及び１２ｂ、ＢＰＦ１３とＢＰＦ１３ａ及び１３ｂ、ＥＣＣ１４とＥＣＣ１４ａ及び１４ｂ、増幅器１５と増幅器１５ａ及び１５ｂ、カップラ１６とカップラ１６ａ及び１６ｂ、アイソレータ１７とアイソレータ１７ａ及びｂ、ＬＰＦ１８とＬＰＦ１７ａ及び１７ｂ、アンテナ１９とアンテナ１９ａ及び１９ｂ、ミクサ２０とミクサ２０ａ及び２０ｂ、ＢＰＦ２１とＢＰＦ２１ａ及び２１ｂ、図９のＶＧＡ２２ａと図１１のＶＧＡ２２ａ及び２２ｂ、図９のＶＧＡ２３ａと図１１のＶＧＡ２３ａ及び２３ｂ、図９のＴＳ２４ａと図１１のＴＳ２４ａ及び２４ｂ、図９の制御部２５ａと図１１の制御部２５ａ及び２５ｂは、対応している。 FIG. 11 is a block diagram showing another configuration example of the transmitter according to the second embodiment of the present invention. 9 and 11 are compared, the modem 1 and the modem 1a, the DSP 2 and the DSP 2a, the modulation unit 3 and the modulation unit 3a, the signal comparison unit 4 and the signal comparison units 4a and 4b, the PD signal generation unit 5 and the PD signal generation unit. 5a and 5b, PD6 and PD6a and 6b, DAC7 and DAC7a and 7b, ADC8 and ADC8a and 8b, RF module 9 and RF modules 9a and 9b, BPF10 and BPF10a and 10b, mixer 11 and mixers 11a and 11b, LO12 and LO12a and 12b, BPF 13 and BPF 13a and 13b, ECC 14 and ECC 14a and 14b, amplifier 15 and amplifiers 15a and 15b, coupler 16 and couplers 16a and 16b, isolator 17 and isolators 17a and 17b, LPF 18 and LPF 17a and 17b, antenna 19 and antenna 1 a and 19b, mixer 20 and mixers 20a and 20b, BPF 21 and BPF 21a and 21b, VGA 22a in FIG. 9 and VGA 22a and 22b in FIG. 11, VGA 23a in FIG. 9 and VGA 23a and 23b in FIG. 11, TS24a in FIG. TS24a and 24b, the control part 25a of FIG. 9, and the control parts 25a and 25b of FIG. 11 correspond.
　図１１は、図９のＲＦモジュール９ａを複数個並べ、複数のＲＦモジュールに対してＤＳＰ２ａを共通化し、Ｍａｓｓｉｖｅ ＭＩＭＯ（Ｍｕｌｔｉｐｌｅ－Ｉｎｐｕｔ ａｎｄ Ｍｕｌｔｉｐｌｅ－Ｏｕｔｐｕｔ）の送信機構成としたものである。このような構成であっても、実施の形態２と同様の効果を奏する。なお、図１１は、ＲＦモジュールを２個設けた例であるが、２個以上設けても良い。 FIG. 11 shows a configuration in which a plurality of RF modules 9a shown in FIG. 9 are arranged, the DSP 2a is shared by the plurality of RF modules, and a transmitter configuration of Massive-Input MIMO (Multiple-Input and Multi-Output) is used. Even with this configuration, the same effects as those of the second embodiment are obtained. FIG. 11 shows an example in which two RF modules are provided, but two or more RF modules may be provided.
　図１２は、この発明の実施の形態２に係る送信機の他の構成例を示す構成図である。
　図１２において、図９と同一符号は、同一または相当部分を示しているため、説明を省略する。また、例えば、増幅器１５ａと増幅器１５ｂとのように、同じ番号の構成要素同士は対応しているため、説明を省略する。また、図９のＲＦモジュール９ａと図１２のＲＦモジュール９０ａ、９０ｂ、９０ｃ、及び９０ｄとは対応している。図９と図１１とを比較すると、図１２のＲＦモジュール９０ａの構成において、増幅器１５の前段に移相器２８ａを設け、ミクサ２０ａの前段に移相器２９ａを設けている点が、図９のＲＦモジュール９ａと異なる。また、図９と比較して、モデム１と、ＲＦモジュール９０ａ、９０ｂ、９０ｃ及び９０ｄとの間に、ＤＩＶ２６とＣｏｍ２７とを設けている。
　ＤＩＶ２６は、モデム１の出力信号をＲＦモジュール９０ａ、９０ｂ、９０ｃ及び９０ｄに分配する分配回路である。 FIG. 12 is a block diagram showing another configuration example of the transmitter according to Embodiment 2 of the present invention.
12, the same reference numerals as those in FIG. 9 indicate the same or corresponding parts, and thus the description thereof is omitted. Further, for example, the components having the same numbers correspond to each other like the amplifier 15a and the amplifier 15b, and thus the description thereof is omitted. Further, the RF module 9a in FIG. 9 corresponds to the RF modules 90a, 90b, 90c, and 90d in FIG. 9 is compared with FIG. 11, in the configuration of the RF module 90a of FIG. 12, the phase shifter 28a is provided in the front stage of the amplifier 15, and the phase shifter 29a is provided in the front stage of the mixer 20a. Different from the RF module 9a. Compared with FIG. 9, a DIV 26 and a Com 27 are provided between the modem 1 and the RF modules 90a, 90b, 90c, and 90d.
The DIV 26 is a distribution circuit that distributes the output signal of the modem 1 to the RF modules 90a, 90b, 90c, and 90d.
　Ｃｏｍ２７は、ＲＦモジュール９０ａ、９０ｂ、９０ｃ及び９０ｄの出力信号を合成して、モデム１に出力する合成回路である。
　図１２は、図９のＲＦモジュールを複数個並べ、アナログサブアレイ構成のアクティブフェーズドアレイ送信機としたものである。ＤＩＶ２６で信号を各ＲＦモジュールへ分配し、フィードバックしてきた信号はＣｏｍ２７で合成される。アクティブフェーズドアレイ送信機として動作させるため、移相器２８ａがＲＦモジュール９０ａに挿入されている。移相器２９ａは、歪み補償を行う際に移相器２８ａで変化させた位相を戻すための移相器である。このような構成であっても、実施の形態２と同様の効果を奏する。
また、本構成では、移相器２８ａ、２８ｂ、２８ｃ、及び２８ｄの位相をそれぞれ変化させ、送信ビームを振っても、移相器２９ａ、２９ｂ、２９ｃ、及び２９ｄにより位相を戻すため、ＲＦモジュール９１ａ、９１ｂ、９１ｃ、及び９１ｄのフィードバック信号を同相にできる。そして、本構成は、同相にしたフィードバック信号をＣｏｍ２７で合成し、その合成した信号を用いてプリディストーションするので、送信ビームの方向を変化させても、送信ビームの正面方向に対して低歪みな信号を出力できる。 Com 27 is a combining circuit that combines the output signals of the RF modules 90 a, 90 b, 90 c, and 90 d and outputs the combined signals to the modem 1.
FIG. 12 shows an active phased array transmitter having an analog subarray configuration in which a plurality of RF modules of FIG. 9 are arranged. The signal is distributed to each RF module by the DIV 26, and the signal fed back is synthesized by the Com 27. In order to operate as an active phased array transmitter, a phase shifter 28a is inserted into the RF module 90a. The phase shifter 29a is a phase shifter for returning the phase changed by the phase shifter 28a when performing distortion compensation. Even with this configuration, the same effects as those of the second embodiment are obtained.
Further, in this configuration, the phase shifters 29a, 29b, 29c, and 29d return the phase even when the phases of the phase shifters 28a, 28b, 28c, and 28d are changed and the transmission beam is shaken. The feedback signals 91a, 91b, 91c, and 91d can be in phase. In this configuration, the feedback signal having the same phase is synthesized by Com27 and predistortion is performed using the synthesized signal. Therefore, even if the direction of the transmission beam is changed, the distortion is low with respect to the front direction of the transmission beam. A signal can be output.
なお、図１２のアナログサブアレイ構成の場合、フィードバック信号を同相にするためにローカル源１２a、１２ｂ、１２ｃ、１２ｄを同相にする方法は２つある。１つは、ローカル源を駆動するリファレンス信号（参照信号）を同期させることで、ローカル源１２a、１２ｂ、１２ｃ、１２ｄを同相にすることである。２つ目は、ローカル源を駆動するリファレンス信号が同期していない場合でも、ローカル源１２a、１２ｂ、１２ｃ、１２ｄの立ち上がりを制御するロードイネーブル信号を制御し、補正することでローカル源１２a、１２ｂ、１２ｃ、１２ｄを同相にすることである。このような方法により、ダウンコンバージョンに用いるローカル源１２a、１２ｂ、１２ｃ、１２ｄの出力信号を同相にできるので、ミクサ１０a、１０ｂ、１０ｃ、１０ｄでフィードバック信号をダウンコンバーションしても、同相にできる。 In the case of the analog subarray configuration of FIG. 12, there are two methods for bringing the local sources 12a, 12b, 12c, and 12d into the same phase in order to make the feedback signal in the same phase. One is to make the local sources 12a, 12b, 12c, and 12d in phase by synchronizing reference signals (reference signals) that drive the local sources. Second, even when the reference signals for driving the local sources are not synchronized, the local sources 12a, 12b are controlled by controlling and correcting the load enable signal for controlling the rising of the local sources 12a, 12b, 12c, 12d. , 12c, 12d are in phase. By such a method, since the output signals of the local sources 12a, 12b, 12c, and 12d used for down conversion can be made in phase, even if the feedback signals are down-converted by the mixers 10a, 10b, 10c, and 10d, they can be made in phase. .
　図１３は、この発明の実施の形態２に係る送信機の他の構成例を示す構成図である。
　図１３は、図１２の変形で、各ＲＦモジュールのＥＣＣを１つに統一したものである。図１３において、図１２と同一符号は、同一または相当部分を示しているため、説明を省略する。また、同じ番号の構成要素同士は対応しているため、説明を省略する。図１２と図１３とを比較すると、ＲＦモジュール９０ａ、９０ｂ、９０ｃ、及び９０ｄと、ＲＦモジュール９１ａ、９１ｂ、９１ｃ、及び９１ｄとは、それぞれ対応している。ただし、ＥＣＣモジュール３０は、ＲＦモジュール９１ａ、９１ｂ、９１ｃ、及び９１ｄに対して共通化されている。ＥＣＣモジュール３０は、ＶＧＡ２２、ＶＧＡ２３、及びＥＣＣ１４、ＤＩＶ２６、ＣＯＭ２７、及び制御部３１を備える。
　制御部３１は、温度が変化しても、ＥＣＣ１４に入る入力電力が同じ電力となるようにＶＧＡ２２を制御し、また、ＥＣＣ１４から増幅器１５ａ、１５ｂ、１５ｃ、１５ｄの入力端までの平均利得が一定になるようにＶＧＡ２３を制御する。
　このような構成であっても、実施の形態２と同様の効果を奏する。 FIG. 13 is a block diagram showing another configuration example of the transmitter according to Embodiment 2 of the present invention.
FIG. 13 is a modification of FIG. 12, in which the ECC of each RF module is unified. In FIG. 13, the same reference numerals as those in FIG. In addition, since the components with the same number correspond to each other, the description thereof is omitted. Comparing FIG. 12 and FIG. 13, the RF modules 90a, 90b, 90c, and 90d correspond to the RF modules 91a, 91b, 91c, and 91d, respectively. However, the ECC module 30 is shared by the RF modules 91a, 91b, 91c, and 91d. The ECC module 30 includes a VGA 22, a VGA 23, an ECC 14, a DIV 26, a COM 27, and a control unit 31.
The control unit 31 controls the VGA 22 so that the input power entering the ECC 14 becomes the same even when the temperature changes, and the average gain from the ECC 14 to the input ends of the amplifiers 15a, 15b, 15c, and 15d is constant. The VGA 23 is controlled so that
Even with this configuration, the same effects as those of the second embodiment are obtained.
１ １ａ　モデム、２　２ａ　ＤＳＰ、３　３ａ　３ｂ　３ｃ　３ｄ　変調部、４　４ａ　４ｂ　４ｃ　４ｄ　信号比較部、５　５ａ　５ｂ　５ｃ　５ｄ　ＰＤ信号生成部、６　６ａ　６ｂ　６ｃ　６ｄ　ＰＤ、７　７ａ　７ｂ　７ｃ　７ｄ　ＤＡＣ、８　８ａ　８ｂ　８ｃ　８ｄ　ＡＤＣ、９　９ａ　９ｂ　９ｃ　９ｄ　９０ａ　９０ｂ　９０ｃ　９０ｄ　９１ａ　９１ｂ　９１ｃ　９１ｄ　ＲＦモジュール、１０　１０ａ　１０ｂ　１０ｃ　１０ｄ　ＢＰＦ、１１　１１ａ　１１ｂ　１１ｃ　１１ｄ　ミクサ、１２　１２ａ　１２ｂ　１２ｃ　１２ｄ　ＬＯ、１３　１３ａ　１３ｂ　１３ｃ　１３ｄ　ＢＰＦ、１４　１４ａ　１４ｂ　１４ｃ　１４ｄ　ＥＣＣ、１５　１５ａ　１５ｂ　１５ｃ　１５ｄ　増幅器、１６　１６ａ　１６ｂ　１６ｃ　１６ｄ　カップラ、１７　１７ａ　１７ｂ　１７ｃ　１７ｄ　アイソレータ、１８　１８ａ　１８ｂ　１８ｃ　１８ｄ　ＬＰＦ、１９　１９ａ　１９ｂ　１９ｃ　１９ｄ　アンテナ、２０　２０ａ　２０ｂ　２０ｃ　２０ｄ　ミクサ、２１　２１ａ　２１ｂ　２１ｃ　２１ｄ　ＢＰＦ、２２　２２ａ　２２ｂ　２２ｃ　２２ｄ　２３　２３ａ　２３ｂ　２３ｃ　２３ｄ　ＶＧＡ、２４　２４ａ　２４ｂ　２４ｃ　２４ｄ　ＴＳ、２５　２５ａ　２５ｂ　２５ｃ　２５ｄ　制御部、２６　ＤＩＶ、２７　Ｃｏｍ、２８　２９　移相器、３０　ＥＣＣモジュール、３１　制御部。 1 1a modem, 2.2a DSP, 3a 3b 3c 3d modulation unit, 4a 4b 4c 4d signal comparison unit, 5a 5b 5c 5d PD signal generation unit, 6a 6b 6c 6d PD, 7a 7b 7c 8dD 8a 8b 8c 8d ADC, 9 9a 9b 9c 9d 90a 90b 90c 90d 91a 91b 91c 91d RF module, 1010a 10b 10c 10d BPF, 11 11a 11b 11c 11d mixer 13b 12c 13b 12b 13c 12d 14 14a 14b 14c 14d ECC, 15 15a 15b 15c 15d amplifier, 16 16a 16b 6c 16d coupler, 17 17a 17b 17c 17d isolator, 18 18a 18b 18c 18d LPF, 19 19a 19b 19c 19d antenna, 20 20a 20b 20c 20d mixer, 21 21a 21b 21c 22d 23b 23d 23b 22d 23b 22d 23 23B 24 24a 24b 24c 24d TS, 25 25a 25b 25c 25d control unit, 26 DIV, 27 Com, 28 29 phase shifter, 30 ECC module, 31 control unit.

Claims (8)

  1. 　ディジタルの送信信号を生成するディジタル回路と、
    　プレディストーションされた前記送信信号をアナログ信号に変換するディジタルアナログ変換器と、
    　前記ディジタルアナログ変換器が出力したアナログの前記送信信号を増幅する増幅器と、
    　前記ディジタルアナログ変換器と前記増幅器の間に設けられ、前記増幅器と合わせたＡＭ－ＡＭ特性が利得伸長特性になるように、前記ディジタルアナログ変換器が出力したアナログの前記送信信号の振幅を変化させ、前記増幅器に出力するエクスパンジョン発生回路と、
    　前記ディジタル回路と前記ディジタルアナログ変換器との間に設けられ、前記増幅器が増幅した前記送信信号の歪みを補償するように前記送信信号をプレディストーションし、プレディストーションした前記送信信号を前記ディジタルアナログ変換器に出力する歪み補償回路と、
    　を備えた送信機。     A digital circuit for generating a digital transmission signal;
    A digital-to-analog converter for converting the predistorted transmission signal into an analog signal;
    An amplifier that amplifies the analog transmission signal output by the digital-analog converter;
    Provided between the digital-analog converter and the amplifier, the amplitude of the analog transmission signal output from the digital-analog converter is changed so that an AM-AM characteristic combined with the amplifier becomes a gain expansion characteristic. An expansion generation circuit for outputting to the amplifier;
    The transmission circuit is provided between the digital circuit and the digital-analog converter, predistorts the transmission signal so as to compensate for distortion of the transmission signal amplified by the amplifier, and converts the predistorted transmission signal to the digital-analog conversion. Distortion compensation circuit to output to the
    With transmitter.
  2. 　前記増幅器及び前記エクスパンジョン発生回路を合わせた前記ＡＭ－ＡＭ特性が、利得伸長特性になるように、温度変化に対して、前記増幅器及び前記エクスパンジョン発生回路のバイアス電圧を制御する制御回路と、
    　を備えたことを特徴とする請求項１に記載の送信機。     A control circuit for controlling a bias voltage of the amplifier and the expansion generation circuit with respect to a temperature change so that the AM-AM characteristic of the amplifier and the expansion generation circuit is a gain expansion characteristic. When,
    The transmitter according to claim 1, further comprising:
  3. 　前記エクスパンジョン発生回路の前段及び後段に設けられ、温度変化に対して、前記ＡＭ－ＡＭ特性の利得が一定になるように利得を可変する可変利得増幅器と、
    　を備えたことを特徴とする請求項２に記載の送信機。     A variable gain amplifier which is provided in a front stage and a rear stage of the expansion generation circuit and which varies the gain so that the gain of the AM-AM characteristic is constant with respect to a temperature change;
    The transmitter according to claim 2, further comprising:
  4. 　前記増幅器を複数設け、複数の前記増幅器に対して前記ディジタルアナログ変換器を共通化したことを特徴とする請求項１に記載の送信機。 The transmitter according to claim 1, wherein a plurality of the amplifiers are provided, and the digital-analog converter is shared by a plurality of the amplifiers.
  5. 　複数の前記増幅器に対して前記エクスパンジョン発生回路を共通化したことを特徴とする請求項４に記載の送信機。 5. The transmitter according to claim 4, wherein the expansion generation circuit is shared by a plurality of the amplifiers.
  6. 　前記エクスパンジョン発生回路は、Ｂ級もしくはＡＢ級にバイアスされた増幅器であることを特徴とする請求項１に記載の送信機。 The transmitter according to claim 1, wherein the expansion generation circuit is an amplifier biased to a class B or class AB.
  7. 　前記エクスパンジョン発生回路は、ダイオードリニアライザーであることを特徴とする請求項１に記載の送信機。 The transmitter according to claim 1, wherein the expansion generation circuit is a diode linearizer.
  8. 　前記エクスパンジョン発生回路は、ダイオードを用いたハイブリッド回路であることを特徴とする請求項１に記載の送信機。 The transmitter according to claim 1, wherein the expansion generation circuit is a hybrid circuit using a diode.
PCT/JP2016/067756 2016-06-15 2016-06-15 Transmitter WO2017216894A1 (en)

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Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2002009555A (en) * 2000-06-16 2002-01-11 Toshiba Corp Amplifier and pre-distorter
JP2003037451A (en) * 2001-06-08 2003-02-07 Trw Inc Application of doherty amplifier as predistortion circuit for linearizing microwave amplifier
JP2008277908A (en) * 2007-04-25 2008-11-13 Mitsubishi Electric Corp Digital predistorter
JP2012044297A (en) * 2010-08-16 2012-03-01 Nec Corp Am-am distortion generator, am-pm distortion generator, distortion generator, am-am distortion generation method and distortion generation method
JP2015061204A (en) * 2013-09-19 2015-03-30 三菱電機株式会社 Distortion compensation circuit and distortion compensation method
JP2015099972A (en) * 2013-11-18 2015-05-28 三菱電機株式会社 Transmitter module

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2002009555A (en) * 2000-06-16 2002-01-11 Toshiba Corp Amplifier and pre-distorter
JP2003037451A (en) * 2001-06-08 2003-02-07 Trw Inc Application of doherty amplifier as predistortion circuit for linearizing microwave amplifier
JP2008277908A (en) * 2007-04-25 2008-11-13 Mitsubishi Electric Corp Digital predistorter
JP2012044297A (en) * 2010-08-16 2012-03-01 Nec Corp Am-am distortion generator, am-pm distortion generator, distortion generator, am-am distortion generation method and distortion generation method
JP2015061204A (en) * 2013-09-19 2015-03-30 三菱電機株式会社 Distortion compensation circuit and distortion compensation method
JP2015099972A (en) * 2013-11-18 2015-05-28 三菱電機株式会社 Transmitter module

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