WO2015198409A1 - Power conversion device and actuator using same - Google Patents

Power conversion device and actuator using same Download PDF

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Publication number
WO2015198409A1
WO2015198409A1 PCT/JP2014/066765 JP2014066765W WO2015198409A1 WO 2015198409 A1 WO2015198409 A1 WO 2015198409A1 JP 2014066765 W JP2014066765 W JP 2014066765W WO 2015198409 A1 WO2015198409 A1 WO 2015198409A1
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WO
WIPO (PCT)
Prior art keywords
resonance
pulse
circuit
control means
conversion device
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PCT/JP2014/066765
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French (fr)
Japanese (ja)
Inventor
宮崎　泰三
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株式会社日立製作所
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Priority to PCT/JP2014/066765 priority Critical patent/WO2015198409A1/en
Publication of WO2015198409A1 publication Critical patent/WO2015198409A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode

Definitions

  • the present invention relates to a power converter that converts a DC power source into AC power.
  • the power converter according to the present invention is particularly suitable for driving an actuator that is required to have low noise.
  • an actuator for nursing care assumed to be used in combination with a medical precision instrument such as an examination instrument, a wearable actuator used in close proximity to a high precision sensor such as an myoelectric sensor, etc. are driven It assumes use.
  • Switching power supplies are widely used in the field of power electronics. Switching power supplies are excellent in controllability and high in efficiency, but high frequency noise is generated with switching. Further, from the recent demand for further energy saving, a high efficiency, low noise switching system is required.
  • a resonant switching power supply is known as one of the techniques to meet this demand. This is to perform ZVS (zero voltage switching) or ZCS (zero current switching) using resonant operation of an inductor and a capacitor. By switching at zero voltage or zero current, switching loss can theoretically be made zero. In addition, since the current and voltage waveforms are also gradually changed with time, generation of surge voltage particularly in the case of inductive load is reduced, and generation of noise is also suppressed.
  • composite resonance As a resonant switching power supply, one that regulates a generated voltage by using a circuit having two or more resonance points is known. For the sake of simplicity, having two or more resonance points is hereinafter referred to as "composite resonance".
  • the complex resonant circuit is configured by combining a plurality of inductors and capacitors.
  • the generated voltage can be adjusted according to the pulse width. This realizes a high efficiency, low noise variable voltage switching power supply.
  • High efficiency and low noise are required not only for switching power supplies, but also for inverters that supply AC power to motors and actuators.
  • An example of noise reduction means utilizing the concept of resonance also in an inverter is disclosed, for example, in [Patent Document 2].
  • the load power is controlled by controlling the power pulse density by first generating a power pulse by resonance and selecting whether the power pulse is passed to the load side by the switching element or not. It is a thing.
  • this method is referred to as PDM (Pulse Density Modulation) for the sake of simplicity.
  • ZVS can also be realized by the method described in [Patent Document 2], and high efficiency and low noise effects can be expected.
  • the present invention is directed to high efficiency, low noise inverters that provide power to motors and actuators.
  • the inverter is required to be able to operate continuously from zero voltage to the maximum voltage.
  • the complex resonant circuit system used in switching power supplies has a problem that the adjustment range of the voltage is narrow.
  • the adjustment range of the voltage is wide, but the resolution depends on the resonance period. Although it is necessary to shorten the cycle to increase the resolution, it is necessary to increase the switching frequency accordingly, and it has been difficult to improve the resolution using PDM.
  • the power converter according to the present invention is characterized by having two or more complex resonance units provided with a pulse generation circuit at the front stage of a complex resonance circuit having a plurality of resonance points.
  • the power conversion device is characterized in that the pulse control means for controlling the pulse generation circuit selectively operates the complex resonance unit in accordance with the positive / negative of the command signal.
  • the complex resonance unit has a coupling portion in which magnetic coupling or electric field coupling is performed, the power converter is separated into the primary side and the secondary side by the coupling portion, and the power conversion is performed.
  • the apparatus is characterized by including a coupler for attaching and detaching the primary side and the secondary side.
  • the present invention is characterized by using the power converter as an actuator that generates an assisting force for assisting the movement of a living body.
  • the primary side is disposed on the normal environment side of the isolated space maintenance bench, and the secondary side is a management environment of the isolated space maintained bench An actuator is disposed on the side.
  • the present invention relates to a power conversion device having a complex resonance unit provided with a pulse generation circuit at a front stage of a complex resonance circuit having a plurality of resonance points.
  • a pulse control means for changing both the circuit cycle and the pulse density is provided, and the ratio of the output adjustment lower limit to the upper limit by the pulse control means of the complex resonant circuit is 2 or more.
  • the power conversion device of the present invention is characterized in that the pulse control means comprises sampling means for sampling a command signal, and the pulse control means extends the execution cycle of the sampling means when the command signal is small.
  • the power conversion device of the present invention is characterized in that the pulse control means has monitoring means for detecting a sudden change of the command signal, and the monitoring means causes the sampling means to be re-executed when the command signal is suddenly changed.
  • power control can be performed on a plurality of phases by providing two or more complex resonance units provided with a pulse generation circuit in the front stage of the complex resonance circuit having a plurality of resonance points in the power converter.
  • actuators are often driven in two or three phases, and stepping motors and SR motors have an even greater number of phases.
  • the advantages of the complex resonance switching power source of high efficiency and low noise can be applied to the actuator driving inverter.
  • control means for controlling the pulse generation circuit causes the compound resonant unit to operate in a fixed polarity (monopolar) operation by selectively operating the compound resonance unit in accordance with the positive or negative of the command signal. ) Can be run on applications that require it. As a result, the current direction of the load can be switched to positive or negative. In the motor, the applied voltage can be increased by bipolar operation, which is advantageous when realizing a high torque motor.
  • the composite resonance unit has a coupling portion which is magnetically or electrically coupled, and the power converter is separated into the primary side and the secondary side by the coupling portion, and the power converter is the primary
  • the primary side having a power supply and the secondary side having a load can be configured to be detachable.
  • the coupler portion can be covered with a resin or the like, the possibility of an electric shock or the like due to direct contact with a contactor or the like is reduced even when the primary side is directly touched.
  • the power conversion device as an actuator that generates an assisting force for assisting the movement of a living body
  • the low noise effect of the present invention can be maximally exhibited.
  • Some equipment for rehabilitation, actuators for walking assistance, etc. acquire minute muscle current with a sensor and perform feedback.
  • the present invention is applied to an actuator for an isolated space maintenance bench having the power conversion device, the composite resonance unit is disposed on the normal environment side of the isolated space maintenance bench, and the load is the isolated space.
  • the isolated space maintenance bench is assumed to be a biological safety cabinet or a clean room for semiconductor manufacturing.
  • the load side can be realized by only passive components such as an inductor, a capacitor, and a diode. Therefore, there is no taking in of the power line from the outside, and it is easy to cover the entire load. Therefore, by providing a mechanical seal or magnetic fluid seal on the bearing portion, the amount of contamination released to the control environment can be reduced.
  • the complex resonant unit requires an active element such as a switching circuit and sealing is difficult, the load and the complex resonant unit can be separated. Placing the load on the management environment side and placing the complex resonance unit on the normal environment side has the effect of reducing the adverse effect on the management environment.
  • the power conversion device has a complex resonance unit including a pulse generation circuit at the front stage of a complex resonance circuit having a plurality of resonance points, wherein the power conversion device has a cycle and a pulse of the pulse generation circuit.
  • the pulse control means has sampling means for sampling the command signal, and the pulse control means extends the execution cycle of the sampling means when the command signal is small, even if the command signal is small. Resolution can be secured.
  • the pulse control means has a monitoring means for detecting a sudden change of the command signal, and the monitoring means ensures responsiveness by re-executing the sampling means when the command signal is suddenly changed. it can.
  • FIG. 1 shows a block diagram of a power converter according to the present invention.
  • FIG. 2 shows an operation explanatory view of the composite resonance circuit.
  • FIG. 3 shows a graphical representation of the transfer function.
  • FIG. 4 shows an example of quantization of the command signal.
  • FIG. 5 shows the gate signal and the output voltage.
  • FIG. 6 shows an embodiment of pulse width correction.
  • FIG. 7 shows an operation flowchart of the pulse control means.
  • FIG. 8 shows an embodiment of a power converter main circuit performing bipolar operation.
  • FIG. 9 shows an embodiment of a power converter main circuit performing unipolar operation.
  • FIG. 10 shows an embodiment in which the main circuit is constituted by an LCC resonant circuit.
  • FIG. 11 shows an LCC equivalent circuit.
  • FIG. 12 illustrates one embodiment of an actuator within a biological safety cabinet.
  • FIG. 13 shows an embodiment of the end effector of the robot.
  • FIG. 14 shows an embodiment of a power assist device.
  • FIG. This figure shows an example of using the power conversion device according to the present invention as an inverter for driving a three-phase motor.
  • the power converter 10 is composed of a control unit 11 and a main circuit 12.
  • the three-phase motor is realized by three windings of U-phase, V-phase and W-phase, but this figure shows only U-phase for simplicity.
  • the command signal creation means 13 creates a voltage command signal to be applied to the U-phase, and creates a voltage command according to a desired torque or speed.
  • the command signal generating means 13 and the like output the command signal 14, and the image thereof is shown in the figure.
  • the command signal 14 is sent to the positive side signal extraction means 15 and the negative side signal extraction means 16.
  • the positive side signal extraction means 15 generates a signal obtained by cutting off the negative part of the command signal 14.
  • the positive side signal extraction means 15 outputs the generated positive side signal 17.
  • An image of the positive side signal 17 is shown in the figure.
  • the negative signal extraction means 16 cuts off the positive part of the command signal 14 and generates a signal whose absolute value is taken.
  • the negative signal extraction means 16 outputs the generated negative signal 18.
  • An image of the negative side signal 18 is shown in the figure.
  • the positive side signal 17 and the negative side signal 18 are sent to separate pulse control means 19 independent of each other.
  • the pulse control means 19 for inputting the positive side signal is numbered 19a
  • the pulse control means 19 for inputting the negative side signal is numbered 19b.
  • the pulse control means 19 generates a gate signal 20 for operating the main circuit 12 from the positive side signal 17 or the negative side signal 18 inputted. The operation of the pulse control means 19 will be described later with reference to the drawings.
  • gate signals 20 are generated per pulse control means 19. Since there are two pulse control means 19 in this figure, four gate signals are generated.
  • a label 21 is added to the generated gate signal 20 and is electrically connected to a label in the main circuit 12 having the same name.
  • the gate signals 21a, 21b, 21c and 21d are respectively assigned to the four gate signals.
  • the main circuit 12 is composed of a plurality of circuit blocks.
  • a pulse generation circuit 22, a complex resonance circuit 23, and a full wave rectification converter 24 are provided.
  • the full wave rectification converter 24 is used here, the effect of the present invention is not lost even with the half wave rectification converter.
  • the present invention is characterized by having a plurality of combinations of the pulse generation circuit 22 and the complex resonance circuit 23.
  • a combination of the pulse generation circuit 22 and the complex resonance circuit 23 is referred to as a complex resonance unit 25.
  • the output of the full wave rectification converter 24 is connected to one side of the motor coil 26.
  • One side of motor coil 26 is neutral point 27, and neutral point 27 is normally connected to one of V-phase and W-phase motor coils (not shown) when a three-phase equilibrium voltage is applied. .
  • a configuration example in which the neutral point 27 is not connected for each phase is also possible.
  • the direct current component is not transmitted by the full wave rectification converter 24, even if the pulse generation circuit 22 of a certain phase has a short circuit failure, the motor coil 26 of the defective phase is not affected. Therefore, there is an advantage that the degenerate operation using only two phases at the time of failure is also possible.
  • the power converter according to this figure converts the power of the DC power supply 28 into three-phase AC power for driving a motor (not shown).
  • the pulse generation circuit 22 is realized by a bridge by the switching element 31.
  • the switching element 31 is realized by a transistor, a MOS-FET, an IGBT or the like. Since the MOS-FET has relatively high-speed body diode in the device and can perform zero voltage switching (ZVS) and zero current switching (ZCS) using parasitic capacitance between drain and source, here -FET is illustrated. The operations of ZVS and ZCS are omitted because they are well known.
  • the complex resonant circuit 23 is composed of a resonant capacitance 32, a resonant inductance 33, and a transformer magnetization inductance 34.
  • the resonant inductance 33 may be realized by the leakage inductance of the transformer 35. The operation of the complex resonant circuit will be described later.
  • the full wave rectification converter is realized by a transformer 35 and a diode 36.
  • An AC voltage generated by resonance is applied to the primary side of the transformer, and is sent to the secondary side by the transformer.
  • the AC voltage sent to the secondary side is converted to a time-varying DC voltage by using two diodes 36. Since the full wave rectification converter 24 generates only a positive voltage, two sets of the complex resonance unit 25 and the full wave rectification converter 24 attached thereto are used to perform bipolar driving. One set is connected to one side of the motor coil 26 and the other set is connected to the opposite side of the motor coil 26. This makes it possible to make the potential difference between both ends of the motor coil 26 positive and negative.
  • a general resonant circuit is configured by combining one resonant capacitor and one resonant inductor.
  • the resonance frequency at this time is calculated by [Equation 1].
  • L is an inductance and C is a capacitance value.
  • the complex resonant circuit 23 shown in FIG. 2 is configured by connecting one resonant capacitor and two resonant inductors in series.
  • the resonant inductance 33 and the transformer magnetizing inductance 34 in FIG. 1 correspond to the two resonant inductors in FIG.
  • Such a circuit configuration is called an LLC composite resonant circuit.
  • a resonant frequency fr in the transmission line resonant circuit 37 and a resonant frequency fr2 as viewed in the entire composite resonant circuit 23 exist, and the frequency characteristics of the general resonant circuit shown in equation 1 change.
  • Equations 2 and 3 show equations for calculating the resonant frequency fr of the transmission line resonant circuit 37 and the resonant frequency fr2 of the complex resonant circuit 23.
  • Equation 4 a transfer function defined by the absolute value of the input voltage Vin in FIG. 2 and the absolute value of the voltage Vout across the transformer magnetization inductance 34 is M (f).
  • f represents the frequency of Vin.
  • Equation 4 The definition of M (f) is shown in Equation 4.
  • f is defined by normalizing f at the first resonance frequency.
  • Formula 5 defines the definition of fn.
  • Equation 6 the transfer function M (f) is expressed by the normalized frequency fn on the assumption that it is linear.
  • ⁇ is defined by Equation 7.
  • Q is a so-called Q value (dissipation energy ratio), which is a value determined by the impedance of the load and the characteristic impedance of the input path.
  • FIG. 3 is a graphical representation of the transfer function represented by [Equation 6].
  • the composite resonance circuit changes the voltage peak value of the transformer magnetization inductance 34 in accordance with the input frequency.
  • the frequency 41 and the switching element limit frequency 42 at which the transfer function maximizes M (f) are shown. Since it is difficult to obtain the noise reduction effect by ZVS and ZCS at a frequency of 41 or less at which the transfer function maximizes M (f), the LLC composite resonant circuit has a frequency 41 at which the transfer function maximizes M (f). Use at frequencies above.
  • the LLC complex resonant circuit can theoretically operate at a frequency between the frequency 41 which maximizes the transfer function M (f) and the switching element limit frequency 42.
  • the stability is degraded near the frequency 41 where the transfer function maximizes M (f), and when the normalized frequency fn is larger than 1, the sensitivity of the transfer function transfer function M (f) to the increase of the switching frequency Is small. Therefore, the operation region 43 shown in the following description will be driven.
  • the operating region 43 is determined such that the transfer function M (f) is in the range of 1 to 2, and the circuit constant is determined such that the resonant operation is stable within this range.
  • the pulse generation circuit 22 has the same effect by changing the duty ratio of the pulse by the pulse control means 19 instead of the frequency itself.
  • a detailed description of the LLC resonant circuit is described in, for example, application note AN1336 published by Microchip Technology, Inc. shown in [Non-patent Document 1].
  • the pulse control means 19 quantizes the input command signal first.
  • FIG. 5 shows gate signals 20 (Up1 and Up2) generated and voltage Uout appearing in the motor coil 26 with respect to the quantized signal 52 surrounded by a broken line in FIG.
  • the number of pulses in the sampling period 51 is determined first.
  • quantization is performed to four levels, and when the signal level is maximum (4/4), four sets of pulses are generated within the sampling period.
  • the signal level is 3/4, 3 sets, 2/4, 2 sets, and 1/4, 1 set.
  • PDM Pulse Density Modulation
  • time slot 53 The time when this pulse enters is referred to as a time slot, and is shown by a time slot 53 in FIG.
  • a combination of two time slots is represented as a time slot pair 54.
  • Each time slot is numbered from 1a to 4b.
  • the first number is the serial number of the time slot pair 54
  • "a” means the time slot for generating the gate signal of the switching element 31 on the upper arm side in the pulse generation circuit 22.
  • b means a time slot for generating a gate signal on the lower arm side.
  • the quantization result is 3/4, three sets of pulses may be generated, and since a and b of time slot 53 having the same first number operate as a pair, three of four time slot pairs 54 are generated. Operate, one pauses.
  • time slot pair 2 is paused in the figure.
  • known techniques such as ⁇ conversion can be diverted.
  • the time slot interval 55 is described as ts in FIG.
  • the length 56 of one pulse is described as ton.
  • the frequency characteristic shown in FIG. 3 is used to correct this error. That is, the pulse width generated by the pulse generation circuit 22 is changed to finely adjust the magnitude of the output.
  • FIG. 6 is an example of pulse width correction.
  • f 1 / t0
  • t0 2 ⁇ / fr.
  • the frequency fr is expressed by [Equation 2].
  • the pulse width As the pulse width is increased from t0, the pulse period decreases, and the transfer function M (f) increases in the operation region shown in FIG. As a result, the pulse peak value of Uout is higher than in the case where the pulse width is t0. Along with that, the area per pulse increases. By this, the power applied to the motor coil 26 can be finely corrected.
  • the width of the fine correction needs to be larger as the pulse has a smaller signal level.
  • the maximum value of the transfer function M (f) is 2 and the minimum value is 1, the ratio of the maximum value to the minimum value is 2.
  • the pulse width can be changed continuously, an area between signal levels 1/4 and 2/4 can be realized without error. Since an area of 2/4 or more may be corrected with a smaller amount, an error due to PDM can be eliminated as a result in an area of 1 ⁇ 4 or more. This is not limited to the case where the number of time slot pairs is four, and the same holds for an arbitrary two or more natural numbers N, and it is possible to adjust so as to eliminate errors in the area of signal level 1 / N or more.
  • This characteristic is not limited to the assumption based on FIG. 3 where the output adjustment lower limit is 1 and the output adjustment upper limit is 2, but similar effects can be obtained if the ratio of the output adjustment lower limit and the upper limit is 2 or more.
  • the error ⁇ can not be made 0 by the above method, but this range is a region where the output torque is very small and the time change is small in the motor. Therefore, there is no practical problem even if the sampling period 51 is sufficiently wide. For example, if the sampling period 51 is quadrupled and the number of pulse slots is 16, errors due to PDM can be eliminated in a region of 1/16 or more. In the case of a motor inverter, it is possible to reduce the output by setting the sampling cycle as required. When a large output is suddenly required in a state where the sampling cycle is increased, it is possible to cope with it by resetting the sampling cycle by an interrupt.
  • FIG. 7 shows a flowchart illustrating the operation of the pulse control means 19 described above.
  • Step 701 (hereinafter referred to as S701) is a start node, from which the program is started.
  • step 702 initialization is performed for a standard sampling period: T0, a standard number of time slots: Ns0, a maximum number of time slots: Nmax, and an input signal maximum value: vmax.
  • the sampling period 51 and the number (Ns) of time slots 53 are set to initial values. After that, operate the two operations in parallel.
  • One is main control processing and the other is input monitoring processing.
  • the main control process generates a gate signal 20, and the input monitoring process monitors whether or not the input has suddenly changed.
  • the input signal is sampled and held to perform quantization. If the quantization result is other than 0 or 1, the quantization result correction process is performed in S705. This is an operation of subtracting 1 from the quantization result if the quantization signal 52 is larger than the original signal (here, the positive side signal 17). If the quantization result is 0 (S706), the operation of doubling the sampling time 51 (T) and the number of time slots 53 (Ns) is repeated until the quantization result is 1 or more (S708, S709, S711) .
  • the on-time 56 (ton) is set to 0 in S710 and the pulse generation is paused during the sampling period 51 (T).
  • the on-time ton at which the quantization error ⁇ becomes 0 is obtained in S715, and the gate signal 20 is generated and output in S716.
  • the sampling time 51 (T) waits (S717). Note that this standby state is canceled by the input monitoring process when the input suddenly changes, and is canceled.
  • the input signal is monitored in S712, and it is determined in S713 whether or not the predetermined voltage change rate is exceeded. As a result, if it exceeds the base value, an interrupt is generated in S714, and the standby state in S717 is released.
  • S 718 is an annotation. If the voltage change rate is within the base value, the process returns to the monitoring state of S712. This makes it possible to ensure high-speed response even when there is a sudden change in voltage.
  • FIG. 8 is an example of a main circuit 12 of a power converter according to the present invention for driving a motor in bipolar operation.
  • it is a separable line 60, and the main circuit 12 can be separated into the primary side 61 and the secondary side 62 on the primary side and the secondary side of the transformer 35.
  • magnetic coupling is used to energetically connect the primary side 61 and the secondary side 62, and it is characterized in that no electrical contactor is required. For this reason, it is easy to completely seal the secondary side 62, and the secondary side 62 can cope with an electrically severe environment such as water or a corrosive atmosphere.
  • the primary side 61 is normally installed in an environment, and it is suitable for applications in which the secondary side 62 is installed in an isolated environment using a partition wall.
  • the secondary side 62 side is configured only with passive components such as an inductor and a diode, which is robust and has an advantage that the number of maintenance can be reduced.
  • This configuration uses two switching elements for one composite resonance unit 25.
  • bipolar operation two composite resonance units are combined to drive a one-phase motor coil 26.
  • FIG. 9 shows an embodiment of the main circuit 12 in which the motor is driven in unipolar operation and the switching elements are reduced to half.
  • the neutral point 27 is taken as the reference potential 63.
  • one composite resonance unit is sufficient for one phase, and the total number of switching elements is six.
  • FIG. 10 shows an example in which the main circuit 12 is configured by an LCC resonant circuit.
  • the electric field coupling 64 is formed by arranging two conductive plates close to each other, and is equivalent to a capacitor in electric circuit.
  • the neutral point realization capacitor 65 is for creating a neutral point voltage in the primary side 61.
  • the reference potential 63 is a neutral point 27.
  • the separable wire 60 exists between the electric field couplings, and can be separated into the primary side 61 and the secondary side 62 on both sides of the separable wire 60.
  • FIG. 11 is an equivalent circuit of FIG. An inductor 66 and a capacitor 67 are provided.
  • the transmission line resonant unit 37 is composed of an inductor 66 and a capacitor 67, and the complex resonant circuit 23 is composed of two inductors 66 and capacitors 67. Even with this configuration, the power converter can be configured in the same way as in the case of using the LLC composite resonant circuit.
  • FIG. 12 shows an embodiment in which the power converter according to the present invention is used as an actuator in a biological safety cabinet.
  • the biological safety cabinet 71 is a device used for manipulating biological samples such as infectious bacteria and genetically modified cells which should not leak to the outside.
  • the intake filter 72 and the exhaust filter 73 are provided to completely isolate and manage the sample and the normal environment. The operation performed from the outside, such as culture and observation, is performed using the glove 74 so that the operator does not directly contact the biological sample.
  • the biological safety cabinet 71 needs to be exchanged with the outside when introducing or discarding a biological sample, but at that time, the inside of the biological safety cabinet 71 needs to be sterilized.
  • the medicine 76 provided with a medicine input port 75 for sterilization, formaldehyde, hydrogen peroxide water or the like is used. Formaldehyde is oxidized to form formic acid, which is likely to corrode metal materials. Also, since hydrogen peroxide solution is a strong oxidizing agent and causes rusting of metal materials, when installing a mechanical device in biological safety cabinet 71, measures against acid and rust are necessary.
  • an actuator 77 using the power converter according to the present invention is provided.
  • Such an actuator 77 is used, for example, to handle a culture vessel installed on the component shelf 78.
  • the actuator 77 is characterized in that the primary side is disposed in a normal environment and the secondary side is disposed in an isolated space (management environment). Since the secondary side of the power converter according to the present invention is easy to seal, by separately sealing the joint part, it is possible to minimize the influence of corrosion and oxidation due to the dispersion of the drug 76.
  • FIG. 13 shows an embodiment in which a power converter according to the present invention is used as a robot end effector.
  • a hand effect device 80 is shown, which is attached to the hand of the robot to perform operations such as gripping and rotation.
  • 80a and 80b are illustrated as hand effect devices.
  • the end effectors 80a and 80b are attached to the primary side 61 by a coupling mechanism 81 and are interchangeable.
  • a linear actuator 82, a gripping mechanism 83, a rotary actuator 84, and a winch 85 are shown. Since the power converter according to the present invention has less noise generated by the power converter itself and there is no electrical contactor on the primary side 61 and the secondary side 62, the hand effect device 80 can be replaced as shown in the figure. There is also an advantage that noise shielding is easy as a structure.
  • FIG. 14 shows a power assist device using the power converter according to the present invention as an actuator.
  • the power assist device 90 is mounted in contact with the human body 91.
  • the surface myoelectric potential sensor 92 measures a minute electric potential that appears on the surface of the human body 91 when the myoelectric potential changes due to contraction and extension of the muscle.
  • the measurement result of the surface myoelectric potential sensor 92 is sent to the upper control system to determine the assist amount.
  • the surface myoelectric potential sensor can measure noninvasively, but since the voltage to be measured is very small, the signal is easily degraded by noise. Since the power converter according to the present invention has less generation noise, more accurate myoelectric potentials can be measured, and an assist that accurately captures the user's intention is possible.

Abstract

A power conversion device is equipped with two or more multi-resonance units each equipped with a pulse generation circuit at the previous stage of a multi-resonance circuit having a plurality of resonance points. A pulse control means for controlling said pulse generation circuit selects and operates the multi-resonance units in accordance with the positive or negative of a voltage command signal of the power conversion device and enables the multi-resonance units to be operated on an application requiring both polarities, thereby enabling an applied voltage to be increased. Alternatively, the power conversion device is provided with a multi-resonance unit equipped with a pulse generation circuit at the previous stage of a multi-resonance circuit having a plurality of resonance points and has a pulse control means for changing both the frequency and pulse density of said pulse generation circuit, wherein the operation region is set so that the ratio of the upper and lower limits of the transfer function of the multi-resonance circuit becomes equal to 2 or more and circuit constants are determined so that the resonance operation is stable in the operation region, thereby widening the voltage adjustment range and increasing the resolution.

Description

電力変換装置およびそれを用いたアクチュエータPower converter and actuator using the same
　本発明は、直流電力源から交流電力に変換する電力変換装置に係る。本発明による電力変換装置は、特に低ノイズであることを求められるアクチュエータを駆動するのに好適である。応用例としては，検査機器などの医療用精密機器と組み合わせて利用されることが想定される介護用のアクチュエータや，筋電センサ等の高精度センサと近接して用いられるウェアラブルアクチュエータなどを駆動する用途を想定している。 The present invention relates to a power converter that converts a DC power source into AC power. The power converter according to the present invention is particularly suitable for driving an actuator that is required to have low noise. As an application example, an actuator for nursing care assumed to be used in combination with a medical precision instrument such as an examination instrument, a wearable actuator used in close proximity to a high precision sensor such as an myoelectric sensor, etc. are driven It assumes use.
　パワーエレクトロニクス分野においてスイッチング電源が広く用いられている。スイッチング電源は制御性に優れ，高効率であるが，スイッチングに伴う高周波ノイズが発生する。また，昨今のさらなる省エネルギー化の要求から，高効率，低ノイズのスイッチング方式が求められている。 Switching power supplies are widely used in the field of power electronics. Switching power supplies are excellent in controllability and high in efficiency, but high frequency noise is generated with switching. Further, from the recent demand for further energy saving, a high efficiency, low noise switching system is required.
　この要求に応える技術の一つとして，共振型スイッチング電源が知られている。これはインダクタとコンデンサの共振動作を利用してZVS(ゼロ電圧スイッチング)またはZCS（ゼロ電流スイッチング）を行わせるものである。電圧がゼロ，または電流がゼロの時点でスイッチングを行うことにより理論上はスイッチング損失をゼロに出来る。また，電流，電圧波形も時間とともに緩やかに変化させるため，特に誘導性負荷の場合のサージ電圧発生が少なくなり，ノイズの発生も抑えられるという特徴がある。 A resonant switching power supply is known as one of the techniques to meet this demand. This is to perform ZVS (zero voltage switching) or ZCS (zero current switching) using resonant operation of an inductor and a capacitor. By switching at zero voltage or zero current, switching loss can theoretically be made zero. In addition, since the current and voltage waveforms are also gradually changed with time, generation of surge voltage particularly in the case of inductive load is reduced, and generation of noise is also suppressed.
　共振型スイッチング電源には，２つ以上の共振点を有する回路を用いて発生電圧を調整するものが知られている。なお，簡単のため，２つ以上の共振点を有することを，以降，「複合共振」と記載することとする。 As a resonant switching power supply, one that regulates a generated voltage by using a circuit having two or more resonance points is known. For the sake of simplicity, having two or more resonance points is hereinafter referred to as "composite resonance".
　複合共振回路を用いたスイッチング電源としては，例えば〔特許文献１〕に開示された方法がある。この方法では複合共振回路は複数のインダクタとコンデンサを組み合わせて構成される。複合共振回路にパルスを印加すると，そのパルス幅に応じて発生電圧を調整することが可能となる。これによって高効率，低ノイズの可変電圧スイッチング電源を実現するものである。 As a switching power supply using a composite resonant circuit, for example, there is a method disclosed in [Patent Document 1]. In this method, the complex resonant circuit is configured by combining a plurality of inductors and capacitors. When a pulse is applied to the complex resonant circuit, the generated voltage can be adjusted according to the pulse width. This realizes a high efficiency, low noise variable voltage switching power supply.
　高効率，低ノイズはスイッチング電源のみならず，モータやアクチュエータに交流電力を与えるインバータでも求められている。インバータにおいても共振の考え方を利用したノイズ低減手段としては，例えば〔特許文献２〕に開示されたものがある。この方法では，まず共振により電力パルスを発生させ，前記電力パルスをスイッチング素子により負荷側へ通過させるか，通過させないかを選択することで，電力パルス密度を制御することで，負荷電力を制御するものである。以降，簡単のためこの方式をPDM(Pulse Density Modulation)と記載する。〔特許文献２〕に記載された方法によってもZVSを実現することが可能であり，高効率，低ノイズ効果が期待できる。 High efficiency and low noise are required not only for switching power supplies, but also for inverters that supply AC power to motors and actuators. An example of noise reduction means utilizing the concept of resonance also in an inverter is disclosed, for example, in [Patent Document 2]. In this method, the load power is controlled by controlling the power pulse density by first generating a power pulse by resonance and selecting whether the power pulse is passed to the load side by the switching element or not. It is a thing. Hereinafter, this method is referred to as PDM (Pulse Density Modulation) for the sake of simplicity. ZVS can also be realized by the method described in [Patent Document 2], and high efficiency and low noise effects can be expected.
国際公開番号WO2012/101906International Publication Number WO 2012/101906 特開平09-168208号JP 09-168208 A
　本発明は，モータやアクチュエータに電力を供給する，高効率，低ノイズのインバータを対象としている。インバータはゼロ電圧から最大電圧まで，連続的に動作出来ることが求められる。しかし，スイッチング電源で利用されている複合共振回路方式では，電圧の調整範囲が狭いという課題がある。 The present invention is directed to high efficiency, low noise inverters that provide power to motors and actuators. The inverter is required to be able to operate continuously from zero voltage to the maximum voltage. However, the complex resonant circuit system used in switching power supplies has a problem that the adjustment range of the voltage is narrow.
　また，PDM方式を用いた場合，電圧の調整範囲は広いが，分解能が共振周期に依存する。分解能を上げるためには周期を短くする必要があるが，それに伴いスイッチング周波数を高くすることが必要となるため，PDMを用いて分解能を高めることは困難であった。 When the PDM method is used, the adjustment range of the voltage is wide, but the resolution depends on the resonance period. Although it is necessary to shorten the cycle to increase the resolution, it is necessary to increase the switching frequency accordingly, and it has been difficult to improve the resolution using PDM.
　上記課題を解決するために本発明の電力変換器では、共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを２個以上有することを特徴とする。 In order to solve the above problems, the power converter according to the present invention is characterized by having two or more complex resonance units provided with a pulse generation circuit at the front stage of a complex resonance circuit having a plurality of resonance points.
　さらに，本発明の電力変換装置は、前記パルス発生回路を制御するパルス制御手段を指令信号の正負に応じて複合共振ユニットを選択動作させることを特徴とする。 Further, the power conversion device according to the present invention is characterized in that the pulse control means for controlling the pulse generation circuit selectively operates the complex resonance unit in accordance with the positive / negative of the command signal.
　さらに，本発明の電力変換装置は、前記複合共振ユニットは磁界結合または電界結合される結合部分を有し，前記電力変換器は前記結合部分によって一次側と二次側に分離され，前記電力変換装置は前記一次側と二次側とを着脱する結合器を備えることを特徴とする。 Furthermore, in the power conversion device according to the present invention, the complex resonance unit has a coupling portion in which magnetic coupling or electric field coupling is performed, the power converter is separated into the primary side and the secondary side by the coupling portion, and the power conversion is performed. The apparatus is characterized by including a coupler for attaching and detaching the primary side and the secondary side.
　また，本発明は生体の運動を補助する補助力を発生するアクチュエータに前記電力変換装置を用いることを特徴とする。 Further, the present invention is characterized by using the power converter as an actuator that generates an assisting force for assisting the movement of a living body.
　また，本発明は前記電力変換装置を有する隔離空間維持ベンチ用アクチュエータにおいて，前記一次側は前記隔離空間維持ベンチの通常環境側に配置され，かつ前記二次側は前記隔離空間維持ベンチの管理環境側にアクチュエータを配置することを特徴とする。 In the actuator for an isolated space maintenance bench having the power conversion device according to the present invention, the primary side is disposed on the normal environment side of the isolated space maintenance bench, and the secondary side is a management environment of the isolated space maintained bench An actuator is disposed on the side.
　また，上記課題を解決するために本発明は、共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを有した電力変換装置において，前記電力変換装置は前記パルス発生回路の周期およびパルス密度の双方を変更するパルス制御手段を備え，前記複合共振回路のパルス制御手段による出力調整下限と上限の比が２以上であることを特徴とする。 Further, in order to solve the above-mentioned problems, the present invention relates to a power conversion device having a complex resonance unit provided with a pulse generation circuit at a front stage of a complex resonance circuit having a plurality of resonance points. A pulse control means for changing both the circuit cycle and the pulse density is provided, and the ratio of the output adjustment lower limit to the upper limit by the pulse control means of the complex resonant circuit is 2 or more.
　さらに，本発明の電力変換装置は、前記パルス制御手段は指令信号をサンプルするサンプリング手段を備え，前記パルス制御手段は指令信号が小さい時に前記サンプリング手段の実行周期を長くすることを特徴とする。 Further, the power conversion device of the present invention is characterized in that the pulse control means comprises sampling means for sampling a command signal, and the pulse control means extends the execution cycle of the sampling means when the command signal is small.
　さらに，本発明の電力変換装置は、前記パルス制御手段は指令信号の急変を検知する監視手段を有し，前記監視手段は指令信号の急変時に前記サンプリング手段を再実行させることを特徴とする。 Furthermore, the power conversion device of the present invention is characterized in that the pulse control means has monitoring means for detecting a sudden change of the command signal, and the monitoring means causes the sampling means to be re-executed when the command signal is suddenly changed.
　本発明では、共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを２個以上電力変換装置に備えることにより，複数相に対して電力制御を行うことが可能になる。一般的に，アクチュエータは２相または３相で駆動される場合が多く，またステッピングモータやＳＲモータではさらに多い相数を有している。それぞれの相に対して複合共振ユニットを備えることで，高効率，低ノイズという複合共振型スイッチング電源の利点をアクチュエータ駆動用インバータに適用することができる。 In the present invention, power control can be performed on a plurality of phases by providing two or more complex resonance units provided with a pulse generation circuit in the front stage of the complex resonance circuit having a plurality of resonance points in the power converter. Become. In general, actuators are often driven in two or three phases, and stepping motors and SR motors have an even greater number of phases. By providing the complex resonance unit for each phase, the advantages of the complex resonance switching power source of high efficiency and low noise can be applied to the actuator driving inverter.
　さらに，本発明では、前記パルス発生回路を制御する制御手段が指令信号の正負に応じて複合共振ユニットを選択動作させることで，一定極性（モノポーラ）動作の複合共振型スイッチング電源を両極性（バイポーラ）が必要なアプリケーションで動作させることが可能になる。これにより，負荷の電流方向を正負に切り替えることができるようになる。モータにおいてはバイポーラ動作させることで印加電圧を高く取ることができるようになり，高トルクモータを実現する場合に有利となる。 Furthermore, in the present invention, the control means for controlling the pulse generation circuit causes the compound resonant unit to operate in a fixed polarity (monopolar) operation by selectively operating the compound resonance unit in accordance with the positive or negative of the command signal. ) Can be run on applications that require it. As a result, the current direction of the load can be switched to positive or negative. In the motor, the applied voltage can be increased by bipolar operation, which is advantageous when realizing a high torque motor.
　さらに，本発明では、前記複合共振ユニットは磁界結合または電界結合される結合部分を有し，前記電力変換器は前記結合部分によって一次側と二次側に分離され，前記電力変換装置は前記一次側と二次側とを着脱する結合器を有することによって，電源を有する一次側と負荷を有する二次側とを着脱自在に構成することができる。この構成の場合，一次側と二次側を電気的に結合するコンタクタを必ずしも必要としない。そのため，樹脂などで結合器部分を覆うことができるため，一次側に直接触れた場合でも，コンタクタなどに直接接触することによる感電等の可能性が減少する。また，負荷交換時にコンタクタの露出がないために，水中環境で負荷（エンドエフェクタ）の交換を行う際に漏電等のおそれが少ないという効果がある。また，コンタクタの露出がないためにアーク放電も起こりづらく，引火性ガスの存在下で作業するアクチュエータに好適である。 Furthermore, in the present invention, the composite resonance unit has a coupling portion which is magnetically or electrically coupled, and the power converter is separated into the primary side and the secondary side by the coupling portion, and the power converter is the primary By having a coupler for attaching and detaching the side and the secondary side, the primary side having a power supply and the secondary side having a load can be configured to be detachable. In this configuration, it is not necessary to have a contactor for electrically coupling the primary side and the secondary side. Therefore, since the coupler portion can be covered with a resin or the like, the possibility of an electric shock or the like due to direct contact with a contactor or the like is reduced even when the primary side is directly touched. In addition, since there is no exposure of the contactor at the time of load exchange, there is an effect that there is little risk of electric leakage or the like when the load (end effector) is exchanged in the underwater environment. In addition, since there is no exposure of the contactors, arcing is less likely to occur, which is suitable for an actuator that operates in the presence of a flammable gas.
　また，本発明では、生体の運動を補助する補助力を発生するアクチュエータに前記電力変換装置を用いることで，本発明の持つ低ノイズの効果が最大限発揮できる。リハビリ用機材や歩行補助用アクチュエータ等では，微小な筋電流をセンサで取得し，フィードバックを行うものがある。本発明によればセンサの誤動作につながるノイズを減らせるため，より信頼性の高いアクチュエータを実現できるという効果がある。また，周辺の医療機器など精密機器への影響も少ないという効果がある。 Further, according to the present invention, by using the power conversion device as an actuator that generates an assisting force for assisting the movement of a living body, the low noise effect of the present invention can be maximally exhibited. Some equipment for rehabilitation, actuators for walking assistance, etc. acquire minute muscle current with a sensor and perform feedback. According to the present invention, it is possible to reduce the noise leading to the malfunction of the sensor, so that it is possible to realize an actuator with higher reliability. In addition, there is an effect that the influence on precision equipment such as peripheral medical equipment is small.
　また，本発明では、前記電力変換装置を有する隔離空間維持ベンチ用アクチュエータに本発明を適用し，前記複合共振ユニットは前記隔離空間維持ベンチの通常環境側に配置し，かつ前記負荷は前記隔離空間維持ベンチの管理環境側に配置することにより，メンテナンスが容易なアクチュエータを実現できる。ここで，隔離空間維持ベンチとは，生物学的安全キャビネットや半導体製造用クリーンルームなどを想定している。本発明によれば，負荷側はインダクタ，コンデンサ，ダイオードといった受動部品のみで実現することが可能であるため，外部からの電力線の取り込みがなく，負荷全体を被覆することが容易である。したがって，軸受部分にメカニカルシールや磁性流体シールを設けることで，管理環境に放出する汚染は少なくできる。複合共振ユニットはスイッチング回路などの能動素子が必要であり，密閉は難しいが，負荷と複合共振ユニットは分離することができる。負荷を管理環境側に配置し，複合共振ユニットを通常環境側に配置することで，管理環境への悪影響が少なくなるという効果がある。 Furthermore, in the present invention, the present invention is applied to an actuator for an isolated space maintenance bench having the power conversion device, the composite resonance unit is disposed on the normal environment side of the isolated space maintenance bench, and the load is the isolated space. By placing it on the maintenance environment side of the maintenance bench, an actuator that can be easily maintained can be realized. Here, the isolated space maintenance bench is assumed to be a biological safety cabinet or a clean room for semiconductor manufacturing. According to the present invention, the load side can be realized by only passive components such as an inductor, a capacitor, and a diode. Therefore, there is no taking in of the power line from the outside, and it is easy to cover the entire load. Therefore, by providing a mechanical seal or magnetic fluid seal on the bearing portion, the amount of contamination released to the control environment can be reduced. Although the complex resonant unit requires an active element such as a switching circuit and sealing is difficult, the load and the complex resonant unit can be separated. Placing the load on the management environment side and placing the complex resonance unit on the normal environment side has the effect of reducing the adverse effect on the management environment.
　また，本発明では、共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを有した電力変換装置であって，前記電力変換装置は前記パルス発生回路の周期およびパルス密度の双方を変更するパルス制御手段を有し，前記複合共振回路のパルス制御手段による出力調整下限と上限の比を２以上とすることによって，電圧の調整範囲を広く，かつ分解能を高くすることができるという効果がある。 Further, in the present invention, the power conversion device has a complex resonance unit including a pulse generation circuit at the front stage of a complex resonance circuit having a plurality of resonance points, wherein the power conversion device has a cycle and a pulse of the pulse generation circuit. The pulse control means for changing both of the density and the ratio of the output adjustment lower limit and the upper limit by the pulse control means of the complex resonant circuit to be 2 or more, thereby widening the voltage adjustment range and increasing the resolution Has the effect of
　さらに，本発明では、前記パルス制御手段は指令信号をサンプルするサンプリング手段を有し，前記パルス制御手段は指令信号が小さい時に前記サンプリング手段の実行周期を長くすることによって，指令信号が小さい場合でも分解能を確保することができる。 Furthermore, in the present invention, the pulse control means has sampling means for sampling the command signal, and the pulse control means extends the execution cycle of the sampling means when the command signal is small, even if the command signal is small. Resolution can be secured.
　さらに，本発明では、前記パルス制御手段は指令信号の急変を検知する監視手段を有し，前記監視手段は指令信号の急変時に前記サンプリング手段を再実行させることによって，応答性を確保することができる。 Furthermore, in the present invention, the pulse control means has a monitoring means for detecting a sudden change of the command signal, and the monitoring means ensures responsiveness by re-executing the sampling means when the command signal is suddenly changed. it can.
図１は本発明による電力変換装置の構成図を示す。FIG. 1 shows a block diagram of a power converter according to the present invention. 図2は複合共振回路の動作説明図を示す。FIG. 2 shows an operation explanatory view of the composite resonance circuit. 図３は伝達関数のグラフ表示を示す。FIG. 3 shows a graphical representation of the transfer function. 図４は指令信号の量子化例を示す。FIG. 4 shows an example of quantization of the command signal. 図５はゲート信号と出力電圧を示す。FIG. 5 shows the gate signal and the output voltage. 図6はパルス幅修正の一実施例を示す。FIG. 6 shows an embodiment of pulse width correction. 図7はパルス制御手段の動作フローチャートを示す。FIG. 7 shows an operation flowchart of the pulse control means. 図８はバイポーラ動作を行う電力変換器主回路の一実施例を示す。FIG. 8 shows an embodiment of a power converter main circuit performing bipolar operation. 図９はユニポーラ動作を行う電力変換器主回路の一実施例を示す。FIG. 9 shows an embodiment of a power converter main circuit performing unipolar operation. 図１０は主回路をLCC共振回路で構成した一実施例を示す。FIG. 10 shows an embodiment in which the main circuit is constituted by an LCC resonant circuit. 図１１はLCC等価回路を示す。FIG. 11 shows an LCC equivalent circuit. 図１２は生物学的安全キャビネット内部のアクチュエータの一実施例を示す。FIG. 12 illustrates one embodiment of an actuator within a biological safety cabinet. 図１３はロボットの手先効果器の実施例を示す。FIG. 13 shows an embodiment of the end effector of the robot. 図１４はパワーアシスト機器の一実施例を示す。FIG. 14 shows an embodiment of a power assist device.
　以下、実施例を、図面を用いて説明する。 Hereinafter, examples will be described using the drawings.
　図1に本発明による電力変換装置の構成図を示す。本図では三相モータ駆動用のインバータとして本発明による電力変換装置を用いる例について示している。電力変換装置10は，電力変換装置は制御手段11と主回路12から構成される。三相モータはＵ相，Ｖ相，Ｗ相の３つの巻線により実現されるが，本図は簡単のためＵ相のみを示している。 The block diagram of the power converter device by this invention is shown in FIG. This figure shows an example of using the power conversion device according to the present invention as an inverter for driving a three-phase motor. The power converter 10 is composed of a control unit 11 and a main circuit 12. The three-phase motor is realized by three windings of U-phase, V-phase and W-phase, but this figure shows only U-phase for simplicity.
　まず，制御手段11の構成について説明する。指令信号作成手段13はＵ相に与える電圧指令信号を作成し、所望のトルクや速度に応じて電圧指令を作成する。指令信号作成手段13らは指令信号１４が出力され，図にはそのイメージを示している。指令信号14は正側信号抽出手段15および負側信号抽出手段16に送られる。正側信号抽出手段15は指令信号14の負の部分を切り取った信号を生成する。正側信号抽出手段15からは生成された正側信号１７が出力される。図には正側信号17のイメージを示している。負側信号抽出手段16は指令信号14の正の部分を切り取って，その絶対値を取った信号を生成する。負側信号抽出手段16からは生成された負側信号１８が出力される。図には負側信号18のイメージを示している。 First, the configuration of the control means 11 will be described. The command signal creation means 13 creates a voltage command signal to be applied to the U-phase, and creates a voltage command according to a desired torque or speed. The command signal generating means 13 and the like output the command signal 14, and the image thereof is shown in the figure. The command signal 14 is sent to the positive side signal extraction means 15 and the negative side signal extraction means 16. The positive side signal extraction means 15 generates a signal obtained by cutting off the negative part of the command signal 14. The positive side signal extraction means 15 outputs the generated positive side signal 17. An image of the positive side signal 17 is shown in the figure. The negative signal extraction means 16 cuts off the positive part of the command signal 14 and generates a signal whose absolute value is taken. The negative signal extraction means 16 outputs the generated negative signal 18. An image of the negative side signal 18 is shown in the figure.
　正側信号17，負側信号18は，互いに独立した個別のパルス制御手段19に送られる。正側信号を入力するパルス制御手段19には19a，負側信号を入力するパルス制御手段19には19bと付番した。パルス制御手段19は入力された正側信号17，または負側信号18から主回路12を動作させるゲート信号20を発生する。パルス制御手段19の動作については図を改めて後述する。 The positive side signal 17 and the negative side signal 18 are sent to separate pulse control means 19 independent of each other. The pulse control means 19 for inputting the positive side signal is numbered 19a, and the pulse control means 19 for inputting the negative side signal is numbered 19b. The pulse control means 19 generates a gate signal 20 for operating the main circuit 12 from the positive side signal 17 or the negative side signal 18 inputted. The operation of the pulse control means 19 will be described later with reference to the drawings.
　本図においてはパルス制御手段19一つ当たり，ゲート信号20を２つ発生する。本図ではパルス制御手段19が２つあるため，ゲート信号は４つ発生する。発生したゲート信号20にはラベル２１を付加するであり，同じ名前を持つ主回路12中のラベルと電気的に接続される。図中では４つのゲート信号にそれぞれゲート信号21a，21b，21c，21dと付番した。 In the figure, two gate signals 20 are generated per pulse control means 19. Since there are two pulse control means 19 in this figure, four gate signals are generated. A label 21 is added to the generated gate signal 20 and is electrically connected to a label in the main circuit 12 having the same name. In the figure, the gate signals 21a, 21b, 21c and 21d are respectively assigned to the four gate signals.
　次に，主回路12の構成について説明する。主回路12は複数の回路ブロックから構成される。パルス発生回路２２，複合共振回路２３，全波整流コンバータ２４を備える。なお，ここでは全波整流コンバータ24を用いたが，半波整流コンバータでも本発明の効果は失われない。 Next, the configuration of the main circuit 12 will be described. The main circuit 12 is composed of a plurality of circuit blocks. A pulse generation circuit 22, a complex resonance circuit 23, and a full wave rectification converter 24 are provided. Although the full wave rectification converter 24 is used here, the effect of the present invention is not lost even with the half wave rectification converter.
　本発明は，パルス発生回路22と複合共振回路23の組み合わせを複数個有することを特徴の一つとする。ここで，パルス発生回路22と複合共振回路23を合わせたものを複合共振ユニット25と記載する。全波整流コンバータ24の出力はモータコイル26の片側に接続される。モータコイル26の片側は中性点27であり，３相平衡電圧を印加する場合には，通常，中性点27はＶ相，Ｗ相のモータコイル（図示せず）の一方に接続される。 The present invention is characterized by having a plurality of combinations of the pulse generation circuit 22 and the complex resonance circuit 23. Here, a combination of the pulse generation circuit 22 and the complex resonance circuit 23 is referred to as a complex resonance unit 25. The output of the full wave rectification converter 24 is connected to one side of the motor coil 26. One side of motor coil 26 is neutral point 27, and neutral point 27 is normally connected to one of V-phase and W-phase motor coils (not shown) when a three-phase equilibrium voltage is applied. .
　なお，中性点27を各相ごとに接続しない構成例も可能である。本構成例では全波整流コンバータ24によって直流成分が透過しないため，ある相のパルス発生回路22が短絡故障した場合でも故障相のモータコイル26には影響が出てこない。したがって，故障時に二相のみを用いた縮退運転も可能であるという利点がある。 A configuration example in which the neutral point 27 is not connected for each phase is also possible. In this configuration example, since the direct current component is not transmitted by the full wave rectification converter 24, even if the pulse generation circuit 22 of a certain phase has a short circuit failure, the motor coil 26 of the defective phase is not affected. Therefore, there is an advantage that the degenerate operation using only two phases at the time of failure is also possible.
　本図による電力変換装置は直流電源28の有する電力をモータ(図示せず)駆動用の３相交流電力に変換するものである。 The power converter according to this figure converts the power of the DC power supply 28 into three-phase AC power for driving a motor (not shown).
　パルス発生回路22はスイッチング素子31によるブリッジにより実現される。スイッチング素子31はトランジスタ，ＭＯＳ－ＦＥＴ，ＩＧＢＴ等によって実現される。ＭＯＳ－ＦＥＴは比較的高速なボディダイオードを素子内に有し，またドレイン－ソース間寄生容量を利用したゼロ電圧スイッチング（ＺＶＳ）やゼロ電流スイッチング（ＺＣＳ）を行なうことができるため，ここではＭＯＳ－ＦＥＴを図示している。なお，ZVSやZCSの動作については公知であるため省略する。 The pulse generation circuit 22 is realized by a bridge by the switching element 31. The switching element 31 is realized by a transistor, a MOS-FET, an IGBT or the like. Since the MOS-FET has relatively high-speed body diode in the device and can perform zero voltage switching (ZVS) and zero current switching (ZCS) using parasitic capacitance between drain and source, here -FET is illustrated. The operations of ZVS and ZCS are omitted because they are well known.
　複合共振回路23は共振容量32，共振インダクタンス33，トランス磁化インダクタンス34から構成される。共振インダクタンス33はトランス35の漏れインダクタンスによって実現してもよい。複合共振回路の動作については後述する。 The complex resonant circuit 23 is composed of a resonant capacitance 32, a resonant inductance 33, and a transformer magnetization inductance 34. The resonant inductance 33 may be realized by the leakage inductance of the transformer 35. The operation of the complex resonant circuit will be described later.
　全波整流コンバータはトランス35とダイオード36によって実現される。トランスの１次側には共振によって生成された交流電圧がかかり，トランスによって２次側に送られる。２次側に送られた交流電圧はダイオード36を２つ用いることによって，時間的に変化する直流電圧に変換される。なお，全波整流コンバータ24は正の電圧のみ発生するため，バイポーラ駆動を行うために，複合共振ユニット25およびそれに取り付けられる全波整流コンバータ24を２セット用いる。１セットはモータコイル26の片側に，またもう1セットはモータコイル26の反対側に接続する。このことにより，モータコイル26両端の電位差を正負両方にすることが可能になる。 The full wave rectification converter is realized by a transformer 35 and a diode 36. An AC voltage generated by resonance is applied to the primary side of the transformer, and is sent to the secondary side by the transformer. The AC voltage sent to the secondary side is converted to a time-varying DC voltage by using two diodes 36. Since the full wave rectification converter 24 generates only a positive voltage, two sets of the complex resonance unit 25 and the full wave rectification converter 24 attached thereto are used to perform bipolar driving. One set is connected to one side of the motor coil 26 and the other set is connected to the opposite side of the motor coil 26. This makes it possible to make the potential difference between both ends of the motor coil 26 positive and negative.
　次に，図２を用いて複合共振回路23の動作について説明する。 Next, the operation of the complex resonant circuit 23 will be described with reference to FIG.
　一般的な共振回路は，共振容量１個と共振インダクタ１個を結合して構成する。この時の共振周波数は〔数1〕で計算される。ここでＬはインダクタンス，Ｃはキャパシタンスの値である。 A general resonant circuit is configured by combining one resonant capacitor and one resonant inductor. The resonance frequency at this time is calculated by [Equation 1]. Here, L is an inductance and C is a capacitance value.
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
　図２で示す複合共振回路23は共振容量１個と共振インダクタ２個を直列に接続することによって構成されている。図１の共振インダクタンス33とトランス磁化インダクタンス34は図2における２個の共振インダクタに対応する。このような回路構成はＬＬＣ複合共振回路と呼ばれる。この回路は，伝達線路共振回路３７における共振周波数frと，複合共振回路23全体で見たときの共振周波数fr2とが存在し，数1で示した一般的な共振回路とは周波数特性が変化する。伝達線路共振回路37の共振周波数frと複合共振回路23の共振周波数fr2の計算式を数２，数３に示す。 The complex resonant circuit 23 shown in FIG. 2 is configured by connecting one resonant capacitor and two resonant inductors in series. The resonant inductance 33 and the transformer magnetizing inductance 34 in FIG. 1 correspond to the two resonant inductors in FIG. Such a circuit configuration is called an LLC composite resonant circuit. In this circuit, a resonant frequency fr in the transmission line resonant circuit 37 and a resonant frequency fr2 as viewed in the entire composite resonant circuit 23 exist, and the frequency characteristics of the general resonant circuit shown in equation 1 change. . Equations 2 and 3 show equations for calculating the resonant frequency fr of the transmission line resonant circuit 37 and the resonant frequency fr2 of the complex resonant circuit 23.
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000002
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000003
　この時，図２中の入力電圧Vinの絶対値とトランス磁化インダクタンス34の両端電圧Voutの絶対値によって定義される伝達関数をＭ(f)とする。なお，ここでｆはVinの周波数を表す。Ｍ(f)の定義を数４に示す。 At this time, a transfer function defined by the absolute value of the input voltage Vin in FIG. 2 and the absolute value of the voltage Vout across the transformer magnetization inductance 34 is M (f). Here, f represents the frequency of Vin. The definition of M (f) is shown in Equation 4.
Figure JPOXMLDOC01-appb-M000004
Figure JPOXMLDOC01-appb-M000004
　以降の説明の便宜のため，ｆを第一の共振周波数で正規化したfnを定義する。fnの定義式を数５に示す。 For convenience of the following description, f is defined by normalizing f at the first resonance frequency. Formula 5 defines the definition of fn.
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000005
　今，線形を仮定して伝達関数Ｍ(f)を正規化周波数fnで表したＭ(fn)を計算すると，数６のようになる。 Now, assuming that the transfer function M (f) is expressed by the normalized frequency fn on the assumption that it is linear, the result is as shown in Equation 6.
Figure JPOXMLDOC01-appb-M000006
Figure JPOXMLDOC01-appb-M000006
　ここで，λは数７で定義される。また，ＱはいわゆるＱ値（散逸エネルギー比）であり，負荷のインピーダンスと入力経路の特性インピーダンスによって決まる値である。 Here, λ is defined by Equation 7. Also, Q is a so-called Q value (dissipation energy ratio), which is a value determined by the impedance of the load and the characteristic impedance of the input path.
Figure JPOXMLDOC01-appb-M000007
Figure JPOXMLDOC01-appb-M000007
　図３は〔数６〕であらわされる伝達関数をグラフ表示したものである。図３のように，複合共振回路は入力周波数に応じてトランス磁化インダクタンス34の両端電圧波高値を変化させる。なお，図３中で，伝達関数をＭ(f)を最大にする周波数41，スイッチング素子限界周波数42 である。伝達関数をＭ(f)を最大にする周波数41以下の周波数では，ＺＶＳ，ＺＣＳによるノイズ低減効果を得ることが難しいため，ＬＬＣ複合共振回路は伝達関数をＭ(f)を最大にする周波数41以上周波数で用いる。ＬＬＣ複合共振回路は理論上伝達関数をＭ(f)を最大にする周波数41とスイッチング素子限界周波数42の間の周波数で動作できる。なお，伝達関数をＭ(f)を最大にする周波数41付近は安定性が悪くなり，また正規化周波数fnが１より大きいところではスイッチング周波数の増加に対する伝達関数伝達関数をＭ(f)の感度が小さい。したがって，今後の説明では示された動作領域43 駆動することにする。動作領域43は伝達関数Ｍ(f)が１から２の範囲内となるように決めたものであり，この範囲内で共振動作は安定であるように回路定数を決定する。このように設定すると，パルス発生回路22から複合共振回路23に与える周波数を変えれば，トランス磁化インダクタンス34にかかる電圧が変化し，トランス35の二次側に現れる電圧もそれに伴って変化する。したがって，モータコイル26にかかる電圧を変化させることができることになる。制御の容易さから，パルス発生回路22は周波数そのものではなく，パルスのデューティ比をパルス制御手段19によって変化させることで同様の効果を持たせる。ＬＬＣ共振回路についての詳細な説明は，例えば〔非特許文献１〕に示すマイクロチップ・テクノロジー社の発行しているアプリケーションノートAN1336に記載されている。 FIG. 3 is a graphical representation of the transfer function represented by [Equation 6]. As shown in FIG. 3, the composite resonance circuit changes the voltage peak value of the transformer magnetization inductance 34 in accordance with the input frequency. In FIG. 3, the frequency 41 and the switching element limit frequency 42 at which the transfer function maximizes M (f) are shown. Since it is difficult to obtain the noise reduction effect by ZVS and ZCS at a frequency of 41 or less at which the transfer function maximizes M (f), the LLC composite resonant circuit has a frequency 41 at which the transfer function maximizes M (f). Use at frequencies above. The LLC complex resonant circuit can theoretically operate at a frequency between the frequency 41 which maximizes the transfer function M (f) and the switching element limit frequency 42. The stability is degraded near the frequency 41 where the transfer function maximizes M (f), and when the normalized frequency fn is larger than 1, the sensitivity of the transfer function transfer function M (f) to the increase of the switching frequency Is small. Therefore, the operation region 43 shown in the following description will be driven. The operating region 43 is determined such that the transfer function M (f) is in the range of 1 to 2, and the circuit constant is determined such that the resonant operation is stable within this range. With this setting, if the frequency applied from the pulse generation circuit 22 to the complex resonance circuit 23 is changed, the voltage applied to the transformer magnetization inductance 34 changes, and the voltage appearing on the secondary side of the transformer 35 also changes accordingly. Therefore, the voltage applied to the motor coil 26 can be changed. From the easiness of control, the pulse generation circuit 22 has the same effect by changing the duty ratio of the pulse by the pulse control means 19 instead of the frequency itself. A detailed description of the LLC resonant circuit is described in, for example, application note AN1336 published by Microchip Technology, Inc. shown in [Non-patent Document 1].
　次に，パルス制御手段19の動作について説明する。 Next, the operation of the pulse control means 19 will be described.
　パルス制御手段19では，まず入力した指令信号の量子化を行う。図４は正側信号17を４つのレベルに量子化した例を示している。図においてサンプリング周期５１，量子化後の量子化信号５２を示している。一般的な量子化では，誤差が最小となるように量子化を行うが，本図では量子化信号52が原信号（＝正側信号17）を超えない最大値となるように決定する。量子化信号52が0となった場合の処理については後述する。 The pulse control means 19 quantizes the input command signal first. FIG. 4 shows an example in which the positive side signal 17 is quantized to four levels. In the figure, a sampling period 51 and a quantized signal 52 after quantization are shown. In general quantization, quantization is performed so as to minimize the error, but in this figure, the quantization signal 52 is determined so as to have a maximum value that does not exceed the original signal (= positive signal 17). The process when the quantized signal 52 becomes 0 will be described later.
　図５を用いて、図４の破線で囲んだ量子化信号52に対して，作成されるゲート信号20（Up1及びUp2）及びモータコイル26に現れる電圧Ｕoutを示す。本例ではまずサンプリング周期51内のパルス数をまず決定する。ここでは４つのレベルに量子化しており，信号レベルが最大(4/4)のとき，サンプリング周期内で４セット，パルスを発生させる。信号レベルが3/4の時は３セット，2/4の時は２セット，1/4の時は１セットとする。これは，サンプリング時間におけるパルス数の密度を変化させることを意味し，一般的にはＰＤＭ（Ｐｕｌse Ｄｅｎｓｉｔｙ Ｍｏｄｕｌａｔｉｏｎ）と呼ばれる技術である。なお，パルス発生回路22内部には２個のスイッチング素子31があり，これらはペアで動作するため，サンプリング周期内に４×２＝８個のパルスが入る時間を設ける。このパルスが入る時間のことをタイムスロットと称し，図５ではタイムスロット53で示した。タイムスロット２個まとめたものをタイムスロットペア54と表す。各タイムスロットには，1aから4bまで付番している。ここで最初の数字がタイムスロットペア54の通し番号であり，「a」はパルス発生回路22において，上アーム側のスイッチング素子31のゲート信号を発生させるタイムスロットを意味する。また「b」は下アーム側のゲート信号を発生させるタイムスロットを意味する。量子化結果が3/4の時，パルスは３セット発生させればよく，最初の数字が等しいタイムスロット53のaとｂはペアで動作するため，４つのタイムスロットペア54のうち３つは動作させ，１つは休止する。どのスロットペアを休止させるかは任意性があるが，図ではタイムスロットペア２を休止させている。ＰＤＭ信号を作成する方法としては，ΔΣ変換などの公知技術を流用可能である。図５においてタイムスロット間隔５５についてはtsと記載する。1パルスの長さ５６についてはtonと記載する。 FIG. 5 shows gate signals 20 (Up1 and Up2) generated and voltage Uout appearing in the motor coil 26 with respect to the quantized signal 52 surrounded by a broken line in FIG. In this example, first, the number of pulses in the sampling period 51 is determined first. Here, quantization is performed to four levels, and when the signal level is maximum (4/4), four sets of pulses are generated within the sampling period. When the signal level is 3/4, 3 sets, 2/4, 2 sets, and 1/4, 1 set. This means changing the density of the number of pulses in the sampling time, which is a technique generally called PDM (Pulse Density Modulation). Note that there are two switching elements 31 in the pulse generation circuit 22 and these operate in pairs, so that 4 × 2 = 8 pulses are provided in the sampling period. The time when this pulse enters is referred to as a time slot, and is shown by a time slot 53 in FIG. A combination of two time slots is represented as a time slot pair 54. Each time slot is numbered from 1a to 4b. Here, the first number is the serial number of the time slot pair 54, and "a" means the time slot for generating the gate signal of the switching element 31 on the upper arm side in the pulse generation circuit 22. Also, “b” means a time slot for generating a gate signal on the lower arm side. When the quantization result is 3/4, three sets of pulses may be generated, and since a and b of time slot 53 having the same first number operate as a pair, three of four time slot pairs 54 are generated. Operate, one pauses. Although it is optional which slot pair is paused, time slot pair 2 is paused in the figure. As a method of generating a PDM signal, known techniques such as ΔΣ conversion can be diverted. The time slot interval 55 is described as ts in FIG. The length 56 of one pulse is described as ton.
　ＰＤＭのみでは原信号と量子化信号52との間に誤差εが生じるため，この誤差を補正するために図３で示した周波数特性を用いる。すなわち，パルス発生回路22が発生するパルス幅を変化させて，出力の大きさを微調整する。 Since an error ε occurs between the original signal and the quantization signal 52 only with PDM, the frequency characteristic shown in FIG. 3 is used to correct this error. That is, the pulse width generated by the pulse generation circuit 22 is changed to finely adjust the magnitude of the output.
　図６はパルス幅修正の一実施例である。ここで，1パルスの長さ56は２πfn＝１を満足するパルス幅である。〔数５〕とf=1/t0の関係式を用いると，t0＝２π／frで算出することができる。なお，周波数frは〔数２〕で表される。 FIG. 6 is an example of pulse width correction. Here, the length 56 of one pulse is a pulse width satisfying 2πfn = 1. Using the relational expression of [Equation 5] and f = 1 / t0, it can be calculated by t0 = 2π / fr. The frequency fr is expressed by [Equation 2].
　パルス幅をt0から大きくして行くと，パルスの周期が小さくなり，図３で示した動作領域では伝達関数Ｍ(f)が大きくなる。その結果Ｕoutのパルス波高値はパルス幅がt0の場合と比較して高くなる。それに伴い，パルス一つ当たりの面積が増える。このことによってモータコイル26に印加する電力を微修正することができる。 As the pulse width is increased from t0, the pulse period decreases, and the transfer function M (f) increases in the operation region shown in FIG. As a result, the pulse peak value of Uout is higher than in the case where the pulse width is t0. Along with that, the area per pulse increases. By this, the power applied to the motor coil 26 can be finely corrected.
　上記のように複合共振回路の特性を利用して微修正を行う場合，その微修正の幅はパルスが信号レベルが小さいほど大きく必要となる。今，伝達関数Ｍ(f)の最大値を２，最小値を１としたため，最大値と最小値の比は２である。この時，パルス幅は連続的に変化させることができるとすると，信号レベル1/4と2/4の間の領域は誤差なく実現することができる。2/4以上の領域はさらに小さい修正で良いため，結果として1/4以上の領域でＰＤＭに起因する誤差をなくすことができる。これはタイムスロットペア数が４の場合に限らず，任意の２以上の自然数Ｎに関して成立し，信号レベル１／Ｎ以上の領域で誤差をなくすように調整することができる。 As described above, when the fine correction is performed using the characteristics of the complex resonant circuit, the width of the fine correction needs to be larger as the pulse has a smaller signal level. Now, assuming that the maximum value of the transfer function M (f) is 2 and the minimum value is 1, the ratio of the maximum value to the minimum value is 2. At this time, assuming that the pulse width can be changed continuously, an area between signal levels 1/4 and 2/4 can be realized without error. Since an area of 2/4 or more may be corrected with a smaller amount, an error due to PDM can be eliminated as a result in an area of 1⁄4 or more. This is not limited to the case where the number of time slot pairs is four, and the same holds for an arbitrary two or more natural numbers N, and it is possible to adjust so as to eliminate errors in the area of signal level 1 / N or more.
　この特性は，特に出力調整下限が１で，出力調整上限を２とした図３による仮定の場合に限らず，出力調整下限と上限の比が２以上であれば同様の効果が得られる。 This characteristic is not limited to the assumption based on FIG. 3 where the output adjustment lower limit is 1 and the output adjustment upper limit is 2, but similar effects can be obtained if the ratio of the output adjustment lower limit and the upper limit is 2 or more.
　0から1/4の範囲では，上記の方法によっては誤差εを0にすることは出来ないが，この範囲はモータでは出力トルクが非常に小さく，時間変化が小さい領域である。したがって，サンプリング周期51を充分広くとっても実用上問題ない。例えば，サンプリング周期51を４倍にして，パルススロット数を16とすると，1/16以上の領域においてＰＤＭに起因する誤差をなくすことができる。モータ用インバータの場合，必要に応じて出力をサンプリング周期の設定によって小さくしていくことが可能である。なお，サンプリング周期を大きくしている状態で，急に大出力が必要になった場合には，割り込みによりサンプリング周期を再設定することで対処可能である。 In the range of 0 to 1/4, the error ε can not be made 0 by the above method, but this range is a region where the output torque is very small and the time change is small in the motor. Therefore, there is no practical problem even if the sampling period 51 is sufficiently wide. For example, if the sampling period 51 is quadrupled and the number of pulse slots is 16, errors due to PDM can be eliminated in a region of 1/16 or more. In the case of a motor inverter, it is possible to reduce the output by setting the sampling cycle as required. When a large output is suddenly required in a state where the sampling cycle is increased, it is possible to cope with it by resetting the sampling cycle by an interrupt.
　図７に以上述べたパルス制御手段19の動作を説明したフローチャートを示す。 FIG. 7 shows a flowchart illustrating the operation of the pulse control means 19 described above.
　ステップ７０１(以下，S701と称す)は開始ノードであり，ここからプログラムが開始される。 Step 701 (hereinafter referred to as S701) is a start node, from which the program is started.
　ステップ７０２では、標準サンプリング周期：T0、標準タイムスロット数：Ns0、最大タイムスロット数：Nmax、入力信号最大値：vmaxについて初期設定を行う。 In step 702, initialization is performed for a standard sampling period: T0, a standard number of time slots: Ns0, a maximum number of time slots: Nmax, and an input signal maximum value: vmax.
　S703において、サンプリング周期51とタイムスロット53の数(Ns)を初期値に設定する。その後，２つの動作を並行動作させる。一方は主制御処理で，もう一方は入力監視処理である。主制御処理はゲート信号20を作成し，入力監視処理では入力が急変していないかどうかを監視する。 In S703, the sampling period 51 and the number (Ns) of time slots 53 are set to initial values. After that, operate the two operations in parallel. One is main control processing and the other is input monitoring processing. The main control process generates a gate signal 20, and the input monitoring process monitors whether or not the input has suddenly changed.
　主制御処理ではまずS704において，入力信号をサンプルホールドし，量子化を行う。量子化結果が０，１以外の場合には，S705において量子化結果補正処理を行う。これは原信号（ここでは正側信号17）より量子化信号52が大きければ，量子化結果から１を引く操作である。量子化結果が０ならば(S706)，サンプリング時間51(T)とタイムスロット53の数(Ns)を二倍にする操作を量子化結果が１以上になるまで繰り返す（S708，S709，S711）。一定回数(Nmax)以上試みても量子化結果が１以上にならない場合には，S710においてオン時間56(ton)を０とし，パルス発生をサンプリング周期51(T)の間休止する。 In the main control process, first, in S704, the input signal is sampled and held to perform quantization. If the quantization result is other than 0 or 1, the quantization result correction process is performed in S705. This is an operation of subtracting 1 from the quantization result if the quantization signal 52 is larger than the original signal (here, the positive side signal 17). If the quantization result is 0 (S706), the operation of doubling the sampling time 51 (T) and the number of time slots 53 (Ns) is repeated until the quantization result is 1 or more (S708, S709, S711) . If the quantization result does not become 1 or more even if it is tried a fixed number of times (Nmax) or more, the on-time 56 (ton) is set to 0 in S710 and the pulse generation is paused during the sampling period 51 (T).
　この後，S715において量子化誤差εが０となるオン時間tonを求め，S716でゲート信号20を作成・出力する。ゲート信号20の出力後，サンプリング時間51(T)待機する(S717)。なお，この待機状態は入力が急変した際に入力監視処理によって割り込みが行われ，キャンセルされる。入力監視処理ではS712において入力信号を監視し，S713において所定の電圧変化率を超えているかどうかを判定する。その結果，基底値を超えていればS714で割り込みを発生し，S717における待機状態を解除する。S718は注釈である。電圧変化率が基底値内の場合にはS712の監視状態に戻る。このことによって電圧急変時にあっても高速応答性が確保できる。 After that, the on-time ton at which the quantization error ε becomes 0 is obtained in S715, and the gate signal 20 is generated and output in S716. After the gate signal 20 is output, the sampling time 51 (T) waits (S717). Note that this standby state is canceled by the input monitoring process when the input suddenly changes, and is canceled. In the input monitoring process, the input signal is monitored in S712, and it is determined in S713 whether or not the predetermined voltage change rate is exceeded. As a result, if it exceeds the base value, an interrupt is generated in S714, and the standby state in S717 is released. S 718 is an annotation. If the voltage change rate is within the base value, the process returns to the monitoring state of S712. This makes it possible to ensure high-speed response even when there is a sudden change in voltage.
　図８はバイポーラ動作でモータを駆動させる本発明による電力変換器の主回路12の一例である。本図において，分離可能線６０であり，主回路12はトランス35の一次側と二次側とで一次側61と二次側62とに分離できる。一次側61と二次側62とをエネルギー的に接続するには本例の場合磁気結合を用いており，電気的なコンタクタを必要としないという特徴がある。このため，二次側62を完全密閉することが容易であり，二次側62は水中や，腐食雰囲気といった電気的に過酷な環境であっても対応が可能となる。一次側61は通常環境に設置し，隔壁を用いて二次側62を隔離環境に設置するアプリケーションに好適となる。また，二次側62側はインダクタ，ダイオードといった受動部品のみで構成されており堅牢で，メンテナンス回数も低減できるという利点を有する。 FIG. 8 is an example of a main circuit 12 of a power converter according to the present invention for driving a motor in bipolar operation. In the figure, it is a separable line 60, and the main circuit 12 can be separated into the primary side 61 and the secondary side 62 on the primary side and the secondary side of the transformer 35. In the present embodiment, magnetic coupling is used to energetically connect the primary side 61 and the secondary side 62, and it is characterized in that no electrical contactor is required. For this reason, it is easy to completely seal the secondary side 62, and the secondary side 62 can cope with an electrically severe environment such as water or a corrosive atmosphere. The primary side 61 is normally installed in an environment, and it is suitable for applications in which the secondary side 62 is installed in an isolated environment using a partition wall. In addition, the secondary side 62 side is configured only with passive components such as an inductor and a diode, which is robust and has an advantage that the number of maintenance can be reduced.
　この構成は，複合共振ユニット25一個について2個のスイッチング素子を用いる。バイポーラ動作では二個の複合共振ユニットを組み合わせて一相のモータコイル26を駆動する。本構成例ではU，V，Wの三相あるため，スイッチング素子総数は２×２×３＝12個となる。 This configuration uses two switching elements for one composite resonance unit 25. In bipolar operation, two composite resonance units are combined to drive a one-phase motor coil 26. In this configuration example, since there are three phases of U, V, and W, the total number of switching elements is 2 × 2 × 3 = 12.
　図９にユニポーラ動作でモータを駆動させ，スイッチング素子を半減した主回路12の一実施例を示す。本図において中性点27を基準電位63とした。この構成では一相について複合共振ユニット一個でよく，スイッチング素子総数は６個で構成される。 FIG. 9 shows an embodiment of the main circuit 12 in which the motor is driven in unipolar operation and the switching elements are reduced to half. In the figure, the neutral point 27 is taken as the reference potential 63. In this configuration, one composite resonance unit is sufficient for one phase, and the total number of switching elements is six.
　以上LLC共振回路を用いた本発明による電力変換装置の説明を行ったが，本発明の主旨を変えない複合共振回路であれば同様の効果が得られる。 Although the power converter according to the present invention using the LLC resonant circuit has been described above, the same effect can be obtained as long as the composite resonant circuit does not change the gist of the present invention.
　図10は主回路12をLCC共振回路で構成した例を示す。 FIG. 10 shows an example in which the main circuit 12 is configured by an LCC resonant circuit.
　図10において，電界カップリング64は二枚の導電板を近接配置したものであり，電気回路的にはコンデンサと等価である。中性点実現コンデンサ65は一次側61内で中性点電圧を作成するものである。ここで基準電位63は中性点27としている。この構成でも電界カップリング間に分離可能線60が存在し，分離可能線60の両側で一次側61と二次側62とに分離できる。 In FIG. 10, the electric field coupling 64 is formed by arranging two conductive plates close to each other, and is equivalent to a capacitor in electric circuit. The neutral point realization capacitor 65 is for creating a neutral point voltage in the primary side 61. Here, the reference potential 63 is a neutral point 27. Also in this configuration, the separable wire 60 exists between the electric field couplings, and can be separated into the primary side 61 and the secondary side 62 on both sides of the separable wire 60.
　図11は図10の等価回路である。インダクタ６６と，コンデンサ６７を備えている。伝送線路共振部37はインダクタ66とコンデンサ67で構成され，複合共振回路23はインダクタ66とコンデンサ67二個で構成される。この構成であってもLLC複合共振回路を用いた場合と同様の考え方で電力変換装置を構成することができる。 FIG. 11 is an equivalent circuit of FIG. An inductor 66 and a capacitor 67 are provided. The transmission line resonant unit 37 is composed of an inductor 66 and a capacitor 67, and the complex resonant circuit 23 is composed of two inductors 66 and capacitors 67. Even with this configuration, the power converter can be configured in the same way as in the case of using the LLC composite resonant circuit.
　次に，本電力変換器を用いたアクチュエータの実施例について説明する。 Next, an embodiment of an actuator using the power converter will be described.
　図12は生物学的安全キャビネット内部のアクチュエータに本発明による電力変換装置を用いた一実施例を示す。生物学的安全キャビネット７１は伝染性細菌や遺伝子組換細胞など外部に漏れてはならない生物試料の操作に用いられる機器である。吸気フィルタ７２，排気フィルタ７３は，試料と通常環境とを完全に隔離，管理するために設けられる。培養，観察など，外部から行う操作はグローブ74を用いて直接操作者が生物試料に接触しないように行われている。 FIG. 12 shows an embodiment in which the power converter according to the present invention is used as an actuator in a biological safety cabinet. The biological safety cabinet 71 is a device used for manipulating biological samples such as infectious bacteria and genetically modified cells which should not leak to the outside. The intake filter 72 and the exhaust filter 73 are provided to completely isolate and manage the sample and the normal environment. The operation performed from the outside, such as culture and observation, is performed using the glove 74 so that the operator does not directly contact the biological sample.
　生物学的安全キャビネット71は生物試料の導入，廃棄などの際，外部とのやり取りが必要となるが，その際，生物学的安全キャビネット71の内部の滅菌操作が必要となる。また、滅菌のための薬剤投入ポート75を備えている，薬剤76としては，ホルムアルデヒドや過酸化水素水などが用いられる。ホルムアルデヒドは酸化することでギ酸となり，金属材料を腐食しやすい。また，過酸化水素水は強酸化剤であり，金属材料に錆を発生させる原因となるため，生物学的安全キャビネット71内に機械装置を設置する場合には酸，錆に対する対策が必要であった。 The biological safety cabinet 71 needs to be exchanged with the outside when introducing or discarding a biological sample, but at that time, the inside of the biological safety cabinet 71 needs to be sterilized. In addition, as the medicine 76 provided with a medicine input port 75 for sterilization, formaldehyde, hydrogen peroxide water or the like is used. Formaldehyde is oxidized to form formic acid, which is likely to corrode metal materials. Also, since hydrogen peroxide solution is a strong oxidizing agent and causes rusting of metal materials, when installing a mechanical device in biological safety cabinet 71, measures against acid and rust are necessary. The
　この実施例において本発明による電力変換器を用いたアクチュエータ７７を備える。このようなアクチュエータ77は，例えば部品棚78に設置された培養容器のハンドリングに用いられる。アクチュエータ77は1次側を通常環境に，また２次側を隔離された空間（管理環境）内に配置したことが特徴である。本発明による電力変換器の二次側は密閉が容易であるため，関節部のシールを別途行うことで，薬剤76の散布による腐食，酸化の影響を極力小さくすることが可能である。 In this embodiment, an actuator 77 using the power converter according to the present invention is provided. Such an actuator 77 is used, for example, to handle a culture vessel installed on the component shelf 78. The actuator 77 is characterized in that the primary side is disposed in a normal environment and the secondary side is disposed in an isolated space (management environment). Since the secondary side of the power converter according to the present invention is easy to seal, by separately sealing the joint part, it is possible to minimize the influence of corrosion and oxidation due to the dispersion of the drug 76.
　図13はロボットの手先効果器に本発明による電力変換装置を用いた一実施例を示す。ここでは手先効果器80を示し，ロボットの手先に取り付けられて把持，回転などの操作を行うものを指す。図13には手先効果器として80a，80bを図示している。手先効果器80a，80bは結合機構81によって一次側61に取り付けられ，互いに交換可能である。また、直動アクチュエータ８２，把持機構８３，回転アクチュエータ８４，ウィンチ８５を示している。本発明による電力変換装置は電力変換装置そのものの発生ノイズが少なく，さらに1次側61と２次側62とに電気的なコンタクタがないために，図のように手先効果器80を交換可能な構造としてもノイズシールドが容易であるという利点がある。 FIG. 13 shows an embodiment in which a power converter according to the present invention is used as a robot end effector. Here, a hand effect device 80 is shown, which is attached to the hand of the robot to perform operations such as gripping and rotation. In FIG. 13, 80a and 80b are illustrated as hand effect devices. The end effectors 80a and 80b are attached to the primary side 61 by a coupling mechanism 81 and are interchangeable. Also, a linear actuator 82, a gripping mechanism 83, a rotary actuator 84, and a winch 85 are shown. Since the power converter according to the present invention has less noise generated by the power converter itself and there is no electrical contactor on the primary side 61 and the secondary side 62, the hand effect device 80 can be replaced as shown in the figure. There is also an advantage that noise shielding is easy as a structure.
　図14は本発明による電力変換器をアクチュエータとして用いたパワーアシスト機器である。ここで，パワーアシスト機器90が人体９１に接触して装着されている。表面筋電位センサ９２は筋肉の収縮，伸展によって筋電位が変化した際に人体91表面に現れる微小電位を測定する。表面筋電位センサ92の測定結果は上位制御系に送られ，アシスト量が決定される。表面筋電位センサは非侵襲による測定が可能であるが，測定する電圧が微小であるため，ノイズによって信号が劣化しやすい。本発明による電力変換器は発生ノイズが少ないため，より正確な筋電位が測定でき，使用者の意図を正確に汲み取ったアシストが可能になる。 FIG. 14 shows a power assist device using the power converter according to the present invention as an actuator. Here, the power assist device 90 is mounted in contact with the human body 91. The surface myoelectric potential sensor 92 measures a minute electric potential that appears on the surface of the human body 91 when the myoelectric potential changes due to contraction and extension of the muscle. The measurement result of the surface myoelectric potential sensor 92 is sent to the upper control system to determine the assist amount. The surface myoelectric potential sensor can measure noninvasively, but since the voltage to be measured is very small, the signal is easily degraded by noise. Since the power converter according to the present invention has less generation noise, more accurate myoelectric potentials can be measured, and an assist that accurately captures the user's intention is possible.

Claims (8)

  1. 　共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを２個以上有することを特徴とする電力変換装置。 What is claimed is: 1. A power conversion device comprising: two or more complex resonance units including a pulse generation circuit at a front stage of a complex resonance circuit having a plurality of resonance points.
  2. 　請求項１において，
    前記パルス発生回路を制御するパルス制御手段は指令信号の正負に応じて前記複合共振ユニットを選択動作させることを特徴とする電力変換装置。     In claim 1,
    A power converter characterized in that pulse control means for controlling the pulse generation circuit selectively operates the composite resonance unit in accordance with the positive / negative of a command signal.
  3. 　請求項１において，
    前記複合共振ユニットは磁界結合または電界結合される結合部分を備え，前記電力変換器は前記結合部分によって一次側と二次側に分離され，前記電力変換装置は前記一次側と前記二次側とを着脱する結合器を備えることを特徴とする電力変換装置。     In claim 1,
    The composite resonant unit includes a coupling portion which is magnetically or electrically coupled, and the power converter is separated into a primary side and a secondary side by the coupling portion, and the power converter includes the primary side and the secondary side. A power converter comprising: a coupler for attaching and detaching.
  4. 　請求項３記載の電力変換装置を備えて，生体の運動を補助する補助力を発生するアクチュエータ。 An actuator comprising the power conversion device according to claim 3 to generate an assisting force for assisting the movement of a living body.
  5. 　請求項３記載の電力変換装置を備えた有する隔離空間維持ベンチ用アクチュエータにおいて，
    前記一次側は前記隔離空間維持ベンチの通常環境側に配置され，かつ前記二次側は前記隔離空間維持ベンチの管理環境側に配置されたことを特徴とするアクチュエータ。     An actuator for an isolated space maintenance bench, comprising the power conversion device according to claim 3;
    An actuator, wherein the primary side is disposed on the normal environment side of the isolated space maintenance bench, and the secondary side is disposed on the management environment side of the isolated space maintenance bench.
  6. 　共振点を複数個有する複合共振回路の前段にパルス発生回路を備えた複合共振ユニットを有した電力変換装置において，
    前記電力変換装置は前記パルス発生回路の周期およびパルス密度の双方を変更するパルス制御手段を備え，
    前記複合共振回路のパルス制御手段による出力調整下限と上限の比が２以上であることを特徴とする電力変換装置。     In a power converter having a complex resonance unit provided with a pulse generation circuit at the front stage of a complex resonance circuit having a plurality of resonance points,
    The power converter includes pulse control means for changing both the period and pulse density of the pulse generation circuit,
    The power converter characterized in that the ratio of the output adjustment lower limit and the upper limit by the pulse control means of the complex resonance circuit is 2 or more.
  7. 　請求項６において，
    前記パルス制御手段は指令信号をサンプルするサンプリング手段を備え，
    前記パルス制御手段は指令信号が小さい時に前記サンプリング手段の実行周期を長くすることを特徴とする電力変換装置。     In claim 6,
    The pulse control means comprises sampling means for sampling a command signal,
    The power conversion device, wherein the pulse control means lengthens the execution period of the sampling means when the command signal is small.
  8. 　請求項７において，
    前記パルス制御手段は指令信号の急変を検知する監視手段を有し，
    前記監視手段は指令信号の急変時に前記サンプリング手段を再実行させることを特徴とする電力変換装置。     In claim 7,
    The pulse control means has a monitoring means for detecting a sudden change of the command signal,
    The power conversion device according to claim 1, wherein the monitoring means re-executes the sampling means when the command signal is suddenly changed.
PCT/JP2014/066765 2014-06-25 2014-06-25 Power conversion device and actuator using same WO2015198409A1 (en)

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH0965589A (en) * 1995-08-25 1997-03-07 Matsushita Electric Works Ltd Power supply equipment
JP2008515379A (en) * 2004-10-01 2008-05-08 コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ LED wide area light source lamp power converter
WO2012101906A1 (en) * 2011-01-26 2012-08-02 株式会社村田製作所 Switching power supply device

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH0965589A (en) * 1995-08-25 1997-03-07 Matsushita Electric Works Ltd Power supply equipment
JP2008515379A (en) * 2004-10-01 2008-05-08 コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ LED wide area light source lamp power converter
WO2012101906A1 (en) * 2011-01-26 2012-08-02 株式会社村田製作所 Switching power supply device

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