WO2014192373A1 - 電力変換装置、電力変換装置の制御方法、回転センサレス制御装置及び回転センサレス制御装置の制御方法 - Google Patents
電力変換装置、電力変換装置の制御方法、回転センサレス制御装置及び回転センサレス制御装置の制御方法 Download PDFInfo
- Publication number
- WO2014192373A1 WO2014192373A1 PCT/JP2014/057010 JP2014057010W WO2014192373A1 WO 2014192373 A1 WO2014192373 A1 WO 2014192373A1 JP 2014057010 W JP2014057010 W JP 2014057010W WO 2014192373 A1 WO2014192373 A1 WO 2014192373A1
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- frequency
- superimposed high
- carrier
- superimposed
- voltage
- Prior art date
Links
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
- H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
- H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
- H02P6/16—Circuit arrangements for detecting position
- H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
- H02P6/183—Circuit arrangements for detecting position without separate position detecting elements using an injected high frequency signal
Definitions
- Embodiments of the present invention relate to a power conversion device, a control method for the power conversion device, a rotation sensorless control device, and a control method for the rotation sensorless control device.
- Patent Document 1 discloses a PWM power converter that temporally changes the frequency of a carrier wave.
- Patent Document 2 discloses a technique for reducing electromagnetic noise caused by PWM harmonics by dispersing the harmonic components of PWM by changing the carrier frequency according to a random number.
- Patent Document 3 discloses a technique for determining the probability of occurrence of each carrier frequency so that the frequency distribution characteristics of harmonics are flat when changing a plurality of carrier frequencies.
- PMSM rotation sensorless control for controlling a permanent magnet synchronous motor (PMSM) without using a rotation sensor
- the rotation angle using the motor core saliency is generated in a low speed region where the induced voltage is small.
- a method of superimposing a high-frequency voltage and detecting a current response is widely used.
- electromagnetic noise is generated by the superimposed high-frequency voltage, and a technique for switching (changing) the frequency of the superimposed high-frequency voltage with time is proposed in order to reduce the generated electromagnetic noise. (For example, see Patent Document 4).
- the range in which the superimposed high-frequency frequency can be selected is generally limited, and the component of the average frequency increases due to the fact that only a narrow range can be selected, which may generate electromagnetic noise.
- the present invention has been made in view of the above, and is a power conversion device capable of reducing electromagnetic noise during PWM control, a control method for the power conversion device, a rotation sensorless control device, a control method for the rotation sensorless control device, and control. The purpose is to provide a program.
- the setting unit of the power conversion device sets the duration time at random and sets any one of a plurality of different carrier frequencies as the set carrier frequency.
- the carrier wave generation unit generates a carrier wave having a predetermined set carrier frequency for a predetermined duration.
- the PWM signal generation unit generates a PWM signal based on the carrier wave generated by the carrier wave generation unit, and the power conversion unit performs power conversion based on the generated PWM signal and supplies it to the load.
- the setting unit of the rotation sensorless control device sets any one of a plurality of superposed high frequency frequencies that are higher than the fundamental frequency in PWM control, and sets the superposed high frequency.
- the duration of a voltage having a high frequency or a current having a set superimposed high frequency is set at random.
- the generator generates a voltage or current having a superimposed high frequency frequency for a duration
- the estimation unit applies the generated voltage having the superimposed high frequency frequency to the permanent magnet synchronous motor or generates the generated superimposed frequency.
- a current having a high frequency is supplied to the permanent magnet synchronous motor to estimate the rotor magnetic pole position and rotation speed of the permanent magnet synchronous motor.
- FIG. 1 is a schematic configuration block diagram of a PWM power converter according to the first embodiment.
- FIG. 2 is a schematic configuration block diagram of the continuation determination unit.
- FIG. 3 is a timing chart of the first embodiment.
- FIG. 4 is an explanatory diagram of the switching state of the carrier frequency.
- FIG. 5 is an explanatory diagram of the relationship between the carrier frequency and the harmonic component.
- FIG. 6 is an explanatory diagram of a modification of the first embodiment.
- FIG. 7 is a schematic configuration block diagram of the PWM power conversion apparatus according to the second embodiment.
- FIG. 8 is a timing chart of the second embodiment.
- FIG. 9 is an operation explanatory diagram of the second embodiment.
- FIG. 10 is an explanatory diagram of the effect of the second embodiment.
- FIG. 11 is a schematic configuration block diagram of the PWM power conversion apparatus according to the third embodiment.
- FIG. 12 is an explanatory diagram of an example of a current flowing through the load in the mechanical characteristic determination.
- FIG. 13 is an explanatory diagram of the noise characteristic of the load in the mechanical characteristic determination.
- FIG. 14 is an explanatory diagram of the relationship between the mechanical characteristics and the set carrier frequency.
- FIG. 15 is an explanatory diagram of another relationship between mechanical characteristics and a set carrier frequency.
- FIG. 16 is an explanatory diagram of a schematic configuration of the first modification.
- FIG. 17 is a schematic configuration block diagram of the PMSM rotation sensorless control system of the fourth embodiment.
- FIG. 18 is a schematic configuration block diagram of the continuation determination unit.
- FIG. 19 is a timing chart of the fourth embodiment.
- FIG. 12 is an explanatory diagram of an example of a current flowing through the load in the mechanical characteristic determination.
- FIG. 13 is an explanatory diagram of the noise characteristic of the load in the mechanical characteristic determination
- FIG. 20 is an explanatory diagram of the switching state of the superimposed high frequency frequency.
- FIG. 21 is an explanatory diagram of the relationship between the superimposed high-frequency frequency and the harmonic component.
- FIG. 22 is a schematic configuration block diagram of the high-frequency voltage command generation unit.
- FIG. 23 is an operation explanation timing chart of the modified example.
- FIG. 24 is a schematic configuration block diagram of another high-frequency voltage command generation unit.
- FIG. 25 is a schematic configuration block diagram of the PMSM rotation sensorless control system of the fifth embodiment.
- FIG. 26 is an operation explanatory diagram of the fifth embodiment.
- FIG. 27 is an explanatory diagram of the effect of the fifth embodiment.
- FIG. 28 is a schematic configuration block diagram of the PMSM rotation sensorless control system of the sixth embodiment.
- FIG. 29 is a timing chart of the sixth embodiment.
- FIG. 30 is an explanatory diagram of a modification of the sixth embodiment.
- FIG. 1 is a schematic configuration block diagram of a PWM power converter according to a first embodiment.
- the PWM power conversion device 10 is roughly divided into a step-down chopper 13 that functions as a power conversion unit that steps down the input DC voltage from the DC power supply 11 and outputs it as a drive voltage to the load 12, and calculates the carrier frequency to calculate the carrier frequency.
- a carrier frequency calculation unit 14 that outputs a signal
- a carrier signal generation unit 15 that generates a carrier signal having a carrier frequency corresponding to the carrier frequency signal
- an output voltage command generation unit 16 that generates and outputs an output voltage command signal
- a PWM signal generator 17 for outputting a PWM signal to the step-down chopper 13 based on the inputted carrier wave signal and output voltage command signal.
- the carrier frequency calculation unit 14, the carrier signal generation unit 15, the output voltage command generation unit 16, and the PWM signal generation unit 17 constitute a control unit 18 that controls PWM power conversion.
- the carrier frequency calculator 14 determines a duration (duration) based on the random number generator 21 that generates a random number and the input random number value, and outputs a duration data (duration data).
- the continuation determination unit 23 that performs continuation determination based on the input duration data and outputs the frequency selection signal, and the highest frequency among the frequencies that can be set as the carrier frequency based on the frequency selection signal
- a frequency selection unit 24 that outputs, as a carrier frequency signal, either the maximum carrier frequency fmax or the lowest carrier frequency fmin that can be set as the carrier frequency.
- FIG. 2 is a schematic configuration block diagram of the continuation determination unit.
- the continuation determination unit 23 receives the continuation period data and the count data, determines whether the value of the continuation period data and the value of the count data match, and outputs a comparison result signal.
- the counter 32 is reset when the comparison result signal is inconsistent and the count value is incremented and output as count data, and the comparison result signal and frequency selection signal are input, and the comparison result signal and frequency selection are input.
- an inversion processor 33 for inverting the frequency selection signal when the signals do not match.
- the random number generation unit 21 of the carrier frequency calculation unit 14 generates a random number value and outputs it to the duration determination unit 22.
- the random number generator 21 calculates a pseudo-random number and outputs it as a random value, or refers to a random number table and outputs a random value.
- the continuation determination unit 23 performs continuation determination based on the input duration data and outputs a frequency selection signal.
- the frequency selection signal is a binary signal having a value of “0” or “1”. It is data. Therefore, a frequency selection signal corresponding to either the maximum carrier frequency fmax or the minimum carrier frequency fmin is output to the frequency selection unit 24 according to the value of the duration data.
- the frequency selection unit 24 outputs the carrier frequency exclusively from the maximum carrier frequency fmax or the minimum carrier frequency fmin as the carrier frequency signal based on the frequency selection signal.
- the carrier signal generation unit 15 generates a carrier signal having a frequency corresponding to the carrier frequency signal (in the first embodiment, either the maximum carrier frequency fmax or the minimum carrier frequency fmin), and the PWM signal generation unit 17 Output to.
- the PWM signal generator 17 outputs a PWM signal to the step-down chopper 13 based on the input carrier wave signal and output voltage command signal.
- the step-down chopper 13 steps down the input DC voltage from the DC power source 11 based on the PWM signal and outputs it as a drive voltage to the load 12 so that the load 12 is driven.
- FIG. 3 is a timing chart of the first embodiment.
- the carrier signal SC generated by the carrier signal generator 15 is 0 [V] and the DC power supply voltage Vdc [V]. ]
- a triangular wave that transitions between the two voltage levels.
- the voltage of the output voltage command signal SB is constant.
- the PWM signal SP generated by the PWM signal generator 17 is at the “H” level when the carrier signal SC ⁇ the output voltage command signal SB, and when the carrier signal SC ⁇ the output voltage command signal SB. Becomes “L” level.
- FIG. 4 is an explanatory diagram of the switching state of the carrier frequency.
- the frequency of the carrier signal SC (set carrier frequency) is the maximum carrier frequency fmax or the minimum carrier frequency fmin. The duration of either of them is made to change randomly.
- the average frequency component in the carrier frequency selection range can be reduced. Also, in the first embodiment, by changing the duration according to the random number value, the change in the duration of the carrier wave of the same frequency is not regular, so there is a sense of incongruity in hearing with the change in duration. It never happens.
- the random number generator 21 generates a random number, and the duration is changed according to the random number. For example, even if the duration is changed according to the sine wave, the average frequency of the carrier frequency selection range is changed. Components can be reduced. In the frequency spectrum distribution of the harmonics generated by the carrier wave, the peak of the harmonic component generated by each carrier frequency appears within the carrier frequency selection range.
- the selection range of the carrier frequency is limited by the control calculation processing time of the microcomputer at the upper limit frequency (fmax) and by the deterioration of controllability due to control delay. Therefore, the frequency band from the upper limit frequency (fmax) to the lower limit frequency (fmin) cannot be wide.
- FIG. 5 is an explanatory diagram of the relationship between the carrier frequency and the harmonic component. That is, as shown in FIG. 5, the peak 71 of the harmonic component due to the minimum carrier frequency fmin that is the lower limit carrier frequency, the peak 72 of the harmonic component due to the maximum carrier frequency fmax that is the upper limit carrier frequency, and the frequency change The peaks 73 of the harmonic components overlap each other. Therefore, in order to expand the range of dispersion as much as possible in the carrier frequency selection range, two frequencies of the minimum carrier frequency fmin at the lower limit of the carrier frequency selection range and the maximum carrier frequency fmax at the upper limit are selected as the carrier frequencies. Thus, it is more preferable to disperse by changing the duration.
- the duration is an integer multiple of half the carrier period when the carrier wave is a triangular wave and the carrier wave is a sawtooth wave so that the carrier frequency update timing is a peak or valley of the carrier wave.
- FIG. 6 is an explanatory diagram of a modification of the first embodiment.
- the control of the load 12 by the step-down chopper 13 has been described.
- FIG. A dispersion effect of harmonic components can be obtained, and the occurrence of mechanical resonance or the like can be suppressed.
- FIG. 7 is a schematic configuration block diagram of a PWM power converter according to a second embodiment.
- the second embodiment is different from the first embodiment in that the first embodiment simply sets a duration for continuously using a carrier wave of the same frequency to generate a PWM signal by a random value.
- the second embodiment is provided with a carrier frequency calculation unit that uses a random number value and a transition probability value, and determines whether or not the random value satisfies a condition for transition based on the transition probability value. In this case, by using a plurality of transition probability values, it is determined whether or not to switch from a certain carrier frequency to another carrier frequency, so that the carrier frequency can be switched more uniformly and the resulting harmonic component Can be uniformly distributed.
- the carrier frequency calculation unit 14A of the PWM power conversion apparatus 10 can be broadly divided into a random number generation unit 21 that generates a random number and a plurality of transition probability values set in advance based on a frequency selection signal, as shown in FIG.
- a transition probability selection unit 41 that selects and outputs one of them, and a transition that determines whether or not to perform frequency transition based on the input random number value and the input transition probability value, and outputs a transition command signal
- the determination unit 42, the frequency selection instruction unit 43 that outputs a frequency selection signal based on the input shift command signal, and the carrier frequency is either the maximum carrier frequency fmax or the minimum carrier frequency fmin based on the frequency selection signal.
- a frequency selection unit 24 that exclusively outputs the signal as a carrier frequency signal.
- the transition probability value is a probability value for shifting the carrier frequency from the maximum carrier frequency fmax to the minimum carrier frequency fmin.
- a transition probability value P1h which is a probability value for shifting the carrier frequency from the minimum carrier frequency fmin to the maximum carrier frequency fmax, is used.
- the random number generation unit 21 of the carrier frequency calculation unit 14 ⁇ / b> A generates a random value and outputs it to the transition determination unit 42.
- the random number generator 21 calculates a pseudo-random number and outputs it as a random value, or refers to a random number table and outputs the random value to the transition determination unit 42.
- the transition probability selection unit 41 selects one of a plurality of transition probability values Ph1, Plh set in advance based on the frequency selection signal output from the frequency selection instruction unit 43, and performs transition determination. To the unit 42.
- the frequency selection signal output from the frequency selection instruction unit 43 corresponds to the maximum carrier frequency fmax
- a transition probability value Phl is output to the transition determination unit 42.
- the probability value Plh is output to the transition determination unit 42.
- the frequency selection instruction unit 43 outputs a frequency selection signal to the frequency selection unit 24 based on the input shift command signal.
- the frequency selection unit 24 outputs the carrier frequency to the carrier signal generation unit 15 using either the maximum carrier frequency fmax or the minimum carrier frequency fmin as the carrier frequency signal.
- the carrier signal generator 15 generates a carrier signal having a frequency corresponding to the carrier frequency signal (in the second embodiment, either the maximum carrier frequency fmax or the minimum carrier frequency fmin) and outputs the carrier signal to the PWM signal generator 17. .
- the output voltage command generation unit 16 generates an output voltage command signal (corresponding to a fundamental wave in PWM control) corresponding to the output voltage of the step-down chopper 13 and outputs it to the PWM signal generation unit 17.
- the PWM signal generator 17 outputs a PWM signal to the step-down chopper 13 based on the input carrier wave signal and output voltage command signal.
- the step-down chopper 13 steps down the input DC voltage from the DC power source 11 based on the PWM signal and outputs it as a drive voltage to the load 12 so that the load 12 is driven.
- FIG. 8 is a timing chart of the second embodiment.
- the carrier frequency calculation unit 14A outputs the carrier signal SC1 in which the duration of the maximum carrier frequency fmax or the minimum carrier frequency fmin changes at random as in the first embodiment, and the PWM signal generation unit 17 As a result, a PWM signal SP1 as shown in FIG. 8 is output.
- the shift state of the carrier frequency can be adjusted by using an appropriate shift probability value, so that the frequency spectrum distribution of the desired harmonics can be adjusted. Is easily generated.
- FIG. 9 is an operation explanatory diagram of the second embodiment.
- the frequency spectrum distribution of the generated harmonics is three types of frequency spectrum distributions as shown in FIG. (3) Since the transition from the minimum carrier frequency fmin to the maximum carrier frequency fmax and (4) the transition from the maximum carrier frequency fmax to the minimum carrier frequency fmin are both 73, they are handled in common.
- the amplitude of the harmonic component at the peak of each frequency spectrum distribution can be expressed by equations (2) to (4).
- the constant C varies depending on the modulation rate, the dispersion range, etc., but is a constant common to the three dispersions. Based on the above, in the second embodiment, the frequency spectrum distribution of harmonics is adjusted.
- the transition probability value Ph1 for changing the carrier frequency from the maximum carrier frequency fmax to the minimum carrier frequency fmin is increased, or the carrier frequency is changed to the minimum carrier frequency fmin.
- the transition probability value Plh for shifting from the maximum carrier frequency fmax to the maximum carrier frequency may be reduced.
- the transition probability value Ph1 for changing the carrier frequency from the maximum carrier frequency fmax to the minimum carrier frequency fmin is reduced, or the carrier frequency is changed from the minimum carrier frequency fmin. It is only necessary to increase the transition probability value Plh for shifting to the maximum carrier frequency fmax.
- the carrier frequency is decreased from the maximum carrier frequency fmax to the minimum.
- the transition probability value Phl for shifting to the carrier frequency fmin or the transition probability value Plh for shifting the carrier frequency from the minimum carrier frequency fmin to the maximum carrier frequency fmax may be increased. This makes it possible to generate an arbitrary frequency spectrum distribution.
- the selection range of the carrier frequency cannot usually be wide. Therefore, as shown in FIG. 5, the peak 71 of the harmonic component due to the minimum carrier frequency fmin which is the lower carrier frequency of the carrier frequency selection range and the harmonic due to the maximum carrier frequency fmax which is the upper carrier frequency of the carrier frequency selection range.
- the peak 72 of the wave component and the peak 73 of the harmonic component resulting from the shift of the carrier frequency overlap each other.
- FIG. 10 is an explanatory diagram of the effect of the second embodiment. Therefore, the harmonic peak 73 caused by the carrier frequency shift is made smaller than the harmonic component peak 71 due to the minimum carrier frequency fmin and the harmonic component peak 72 due to the maximum carrier frequency fmax, so that FIG. As shown, it can be as flat as possible within the selected range of carrier frequencies.
- the component of the average frequency in the carrier frequency selection range can be reduced even if the shift determination is performed according to, for example, a sine wave instead of a random number.
- the control of the load 12 by the step-down chopper 13 is shown as in the first embodiment.
- the PWM power converter is used, the dispersion effect of the harmonic components can be obtained similarly.
- the carrier frequency is changed only at the peak position or the bottom position so that the update timing of the carrier frequency is at the peak (peak) or bottom (valley) of the carrier.
- the PWM premise that the average value of the output voltage between the peak and the bottom of the carrier wave becomes the output voltage command is maintained, and the occurrence of an error in the output voltage can be prevented by changing the carrier frequency.
- FIG. 11 is a schematic configuration block diagram of the PWM power conversion apparatus according to the third embodiment.
- the same parts as those of the second embodiment of FIG. 7 are denoted by the same reference numerals.
- the carrier wave frequency calculation unit 14B of the PWM power conversion apparatus 10 is roughly classified as shown in FIG. 11, based on a random number generation unit 21 that generates random numbers and a transition probability value Ph1, Based on the frequency selection signal, a transition probability carrier frequency determination unit 51 that determines Plh, maximum carrier frequency fmax, and minimum carrier frequency fmin, and a plurality of transition probability values set by the transition probability carrier frequency determination unit 51
- a transition probability selection unit 41 that outputs a transition command
- a transition determination unit 42 that performs a transition determination as to whether or not to perform a frequency shift based on the input random number value and the input transition probability value, and outputs a transition command signal
- a frequency selection instruction unit 43 Based on the input shift command signal, a frequency selection instruction unit 43 that outputs a frequency selection signal and a shift probability carrier frequency determination unit 51 set Maximum carrier frequency fmax or the minimum carrier frequency either fmin which includes a frequency selector 24 for outputting a carrier frequency signal based on the frequency selection signal.
- FIG. 12 is an explanatory diagram of an example of a current flowing through the load in the mechanical characteristic determination.
- FIG. 13 is an explanatory diagram of the noise characteristic of the load in the mechanical characteristic determination.
- the mechanical characteristics are, for example, the load noise shown in FIG. 13 when a current close to white noise as shown in FIG. These are the characteristics and the frequency of the mechanical resonance point of the load (for example, 3.4 kHz and 3.9 kHz in FIG. 13).
- FIG. 14 is an explanatory diagram of the relationship between the mechanical characteristics and the set carrier frequency. Then, according to the input mechanical characteristics, for example, as shown in FIG. 14, the transition probability carrier frequency determination unit 51 sets the mechanical resonance point in the mechanical characteristics 82 to the average frequency of the maximum carrier frequency fmax and the minimum carrier frequency fmin.
- FIG. 15 is an explanatory diagram of another relationship between mechanical characteristics and a set carrier frequency.
- the carrier frequency in the case of the mechanical characteristic 84 in which the noise at the maximum carrier frequency fmax is large, the carrier frequency is higher than the transition probability value Phl that changes the carrier frequency from the maximum carrier frequency fmax to the minimum carrier frequency fmin.
- the mechanical resonance is reduced so that the transition probability value Plh that shifts the minimum carrier frequency fmin from the minimum carrier frequency fmin to a smaller value and the harmonic peak of the minimum carrier frequency fmin becomes larger than the harmonic peak of the maximum carrier frequency fmax.
- the avoided frequency spectrum distribution 83 is set.
- the third embodiment configured as described above, mechanical resonance can be avoided and electromagnetic noise can be reduced.
- the component of the average frequency in the carrier frequency selection range is reduced even if the transition determination is performed according to, for example, a sine wave instead of a random number. it can.
- control of the load 12 by the step-down chopper 13 is shown as in the first and second embodiments.
- the dispersion effect of the harmonic components is similarly shown. Is obtained.
- the control of the load 12 by the step-down chopper 13 has been shown as in the first and second embodiments.
- an inverter that controls the AC motor 12A that functions as the load 12 is used. Even if it exists, if this invention is used according to the mechanical characteristic of 12 A of AC motors, the electromagnetic noise reduction effect will be acquired similarly.
- FIG. 16 is a schematic configuration explanatory diagram of a first modified example of the first to third embodiments.
- a configuration in which the mechanical characteristics of the load are separately input (set) is adopted.
- a noise detecting microphone 91 is provided in the vicinity of the load, and the output of the microphone 91 is obtained.
- FFT analyzer 92 that performs analysis by fast Fourier transform (FFT) and to determine the maximum carrier frequency fmax, the minimum carrier frequency fmin, and the carrier duration based on the FFT analysis result.
- FFT fast Fourier transform
- the configuration having the transition probability carrier frequency determination unit 51 is adopted.
- the transition probability is previously avoided so as to avoid mechanical resonance. It is also possible to calculate the values Phl and Plh and the carrier wave frequencies fmax and fmin and set them.
- a control unit of the PWM power conversion device includes a control device such as a CPU, a storage device such as a ROM (Read Only Memory) and a RAM, and an HDD.
- a control device such as a CPU
- a storage device such as a ROM (Read Only Memory) and a RAM
- an HDD high-density digital versatile disk
- an external storage device such as a CD drive device
- a display device such as a display device
- an input device such as a keyboard and a mouse
- a hardware configuration using a normal computer can also be configured.
- the control program executed by the control unit of the PWM power conversion device is a CD-ROM in an installable or executable file.
- the program may be recorded on a computer-readable recording medium such as a flexible disk (FD), a CD-R, a DVD (Digital Versatile Disk), or the like.
- a control program executed by the control unit of the PWM power converter according to the first to third embodiments is stored on a computer connected to a network such as the Internet, You may comprise so that it may provide by downloading via a network.
- the control program executed by the PWM power conversion device of the present embodiment may be provided or distributed via a network such as the Internet.
- control program of the control unit of the PWM power converter according to the first to third embodiments may be provided by being incorporated in advance in a ROM or the like.
- FIG. 17 is a schematic configuration block diagram of a PMSM rotation sensorless control system of a fourth embodiment.
- the PMSM rotation sensorless control system 110 is broadly divided into an inverter 111 that performs power conversion, a PMSM 112 that is rotationally driven by the inverter 111, and a high frequency that is superimposed to estimate the rotation angle of the PMSM 112 using the motor core saliency.
- a superposed high frequency frequency computing unit 13 for computing and setting the voltage frequency
- an inverter control unit for controlling the inverter 111 while superposing a high frequency signal having a frequency set by the superposed high frequency frequency computing unit 113 on the control signal.
- the superimposed high-frequency calculation unit 113 can be provided inside or outside the inverter control unit 114.
- the superposition high-frequency calculation unit 113 determines a duration based on a random number generation unit 121 that generates a random number, based on the value of the input random number, and outputs a duration period determination unit 122 that outputs the duration data.
- a continuation determination unit 123 that performs continuation determination, which will be described later, based on the period data, and outputs a frequency selection signal, and a maximum superimposed high-frequency frequency f having the highest frequency among frequencies that can be set as the superimposed high-frequency frequency based on the frequency selection signal and a frequency selection unit 124 that outputs one of the lowest superposed high frequency f fmin that can be set as “max” or a superposed high frequency as a superposed high frequency signal.
- FIG. 18 is a schematic configuration block diagram of the continuation determination unit.
- the continuation determination unit 123 includes a comparator 161, a counter 162, and an inversion processor 163.
- the comparator 161 receives the duration data and the count data, determines whether or not the values of the duration data and the count data match, and outputs a comparison result signal.
- the counter 162 has a memory inside, and the count value is stored in the memory.
- the initial value of the count value is 0, for example.
- a constant value (for example, 1) is added to the count value every control cycle to update the count value in the memory.
- the counter 162 receives the comparison result signal, and the count value is reset to the initial value when the comparison result signal corresponds to coincidence. In addition, when the comparison result signal corresponds to a mismatch, the counter 162 continuously adds a constant value for each control cycle and outputs it as a count value.
- the inversion processor 163 inverts the frequency selection signal when the comparison result signal and the frequency selection signal are input and the comparison result signal does not match the frequency selection signal.
- the inverter control unit 114 outputs a superimposed voltage command signal vdch * for instructing the frequency of the high frequency voltage to be superimposed based on the superimposed high frequency signal output from the frequency selection unit 124. It has. Further, the inverter control unit 114 includes a rotation phase angle estimation unit 132 that estimates the rotation phase angle of the PMSM 12 based on the input superimposed voltage command signal vdch * and the q-axis current detection signal iqc and outputs the estimated phase angle ⁇ est. I have.
- the inverter control unit 114 also outputs a d-axis current detection signal idc calculated from a d-axis current command signal idc * , a q-axis current command signal iqc * , and a detection value of the current detection unit 137, which are input from the outside of the cab or the like.
- Current controller 133 for generating and outputting fundamental wave voltage command signals vdcf * and vqcf * for current control based on q-axis current detection signal iqc, fundamental wave voltage command signals vdcf * and vqcf *, and superimposed voltage
- a high-frequency voltage superimposing unit 134 that outputs a d-axis voltage command signal vdc * and a q-axis voltage command signal vqc * based on the command signal vdch * .
- the inverter control unit 114 performs coordinate conversion of the input d-axis voltage command signal vdc * and q-axis voltage command signal vqc * to perform U-phase voltage command signal vu * , V-phase voltage command signal vv * , W-phase.
- the first coordinate conversion unit 135 that outputs the voltage command signal vw * , the input voltage command signals vu * , vv * , and vw * and the triangular wave or sawtooth wave that is a carrier wave are compared, and PWM modulation is performed.
- a PWM modulator 136 that outputs a gate signal that is an on / off command for each phase switching element of the inverter 11.
- the inverter control unit 114 detects a current response value of a plurality of phases (two phases of U phase and W phase in the example of FIG. 17) out of the three-phase alternating current flowing through the PMSM 12, and a current detection signal (of FIG. 17).
- a current detection unit 137 that outputs a U-phase current detection signal iu and a W-phase current detection signal iw), and a current detection signal output by the current detection unit 137 (in the example of FIG. 17, the U-phase current detection signal iu).
- a second coordinate conversion unit 138 that performs coordinate conversion (UVW / dcqc conversion) of the W-phase current detection signal iw) and outputs a d-axis current detection signal idc and a q-axis current detection signal iqc.
- the random number generation unit 121 of the superimposed high-frequency calculation unit 113 generates a random value and outputs it to the duration determination unit 122.
- the random number generator 121 calculates a pseudo-random number and outputs it as a random value, or refers to a random number table and outputs a random value.
- the continuation determination unit 123 performs continuation determination based on the input duration data and outputs a frequency selection signal.
- the frequency selection signal is “0” or “1”. Is binary data having any of the following values. Therefore, a frequency selection signal corresponding to either the maximum superposition high frequency f'max or the minimum superposition high frequency f'min is output to the frequency selection unit 124 according to the value of the duration data.
- the frequency selection unit 124 outputs either the maximum superimposed high frequency f'max or the minimum superimposed high frequency f'min to the high frequency voltage command generation unit 131 as a superimposed high frequency signal based on the frequency selection signal.
- the high-frequency voltage command generation unit 131 of the inverter control unit 114 generates a superimposed voltage command signal vdch * for instructing the frequency of the high-frequency voltage to be superimposed based on the input superimposed high-frequency signal and the rotational phase angle estimation unit 132 and Output to the high-frequency voltage superimposing unit 134.
- the current control unit 133 performs basic control based on the input d-axis current command signal idc * , q-axis current command signal iqc * , d-axis current detection signal idc, and q-axis current detection signal iqc.
- the wave voltage command signals vdcf * and vqcf * are generated and output to the high frequency voltage superimposing unit 134.
- the high-frequency voltage superimposing unit 134 the fundamental wave voltage instruction signal Vdcf * and superimposed voltage command signal VDCH * d-axis voltage command signal vdc based on *, generates a fundamental wave voltage command signal Vqcf * and superimposed voltage
- a q-axis voltage command signal vqc * is generated based on the command signal vdch *
- the generated d-axis voltage command signal vdc * and q-axis voltage command signal vqc * are output to the first coordinate conversion unit 135.
- the first coordinate conversion unit 135 performs coordinate conversion of the input d-axis voltage command signal vdc * and q-axis voltage command signal vqc * to perform U-phase voltage command signal vu * , V-phase voltage command signal vv *, and W-phase.
- the voltage command signal vw * is output to the PWM modulation unit 136.
- the PWM modulation unit 136 compares the input voltage command signals vu * , vv * , vw * with a triangular wave or sawtooth wave as a carrier wave, performs PWM modulation, and turns on / off each phase switching element of the inverter 111.
- a gate signal that is an OFF command is output to the inverter 11.
- U-phase current, V-phase current, and W-phase current flow from the inverter 111 to the PMSM 112 in a synchronized state, and a rotor (not shown) of the PMSM 112 rotates.
- the current detection unit 137 detects a current response value of a plurality of phases (two phases of the U phase and the W phase in the example of FIG. 17) out of the three-phase alternating current flowing in the PMSM 112, and a current detection signal (In the example of FIG. 17, the U-phase current detection signal iu and the W-phase current detection signal iw) are output to the second coordinate conversion unit 138.
- the second coordinate conversion unit 138 performs coordinate conversion (UVW / dcqc conversion) of the current detection signal output by the current detection unit 137 (in the example of FIG. 17, the U-phase current detection signal iu and the W-phase current detection signal iw).
- the d-axis current detection signal idc is output to the current control unit 133 and the q-axis current detection signal iqc is output to the rotational phase angle estimation unit 132 and the current control unit 133.
- the rotational phase angle estimation unit 132 estimates the rotational phase angle of the PMSM 112 based on the input superimposed voltage command signal vdch * and the q-axis current detection signal iqc, and calculates the estimated phase angle ⁇ est as the first coordinate conversion unit 135. And output to the second coordinate converter 138.
- the first coordinate conversion unit 135 outputs to the PWM modulation unit 136 voltage command signals vu * , vv * , vw * that are optimal for the rotation state of the PMSM 112 corresponding to the estimated phase angle ⁇ est. Therefore, the inverter 111 performs rotation driving according to the rotation state of the PMSM 112 while suppressing noise.
- FIG. 19 is a timing chart of the fourth embodiment. As shown in FIG. 19, when the DC power supply voltage of the PWM modulation unit 136 is Vdc [V], the carrier wave signal SC generated by the carrier wave generation unit is 0 [V] and the DC power supply voltage Vdc [V]. A triangular wave transitions between the two voltage levels.
- the cycle of the superimposed voltage command signal vdch * output from the high-frequency voltage command generation unit 131 is set to the superimposed high-frequency frequency (maximum superimposed high-frequency frequency f′max or minimum superimposed high-frequency frequency f′min).
- FIG. 20 is an explanatory diagram of the switching state of the superimposed high frequency frequency.
- FIG. 20 is a diagram expressed on the long-term time axis of FIG.
- the superposition high frequency is the maximum superposition high frequency f'max or the minimum superposition high frequency f'min. The duration of either of them is made to change randomly.
- the average frequency component in the superimposed high frequency selection range can be reduced. Further, in the fourth embodiment, by changing the duration according to the random number value, the change in the duration of the superimposed high frequency of the same frequency is not regular, so that the sense of incongruity caused by the change in duration Will not occur.
- the average frequency component of the superimposed high frequency selection range can be reduced. Further, since the duration is changed according to the random number value, the same superimposed high-frequency frequency is maintained, the regularity of the continuous duration change is lost, and the duration change It is less likely to cause a sense of incongruity in hearing.
- the selection range of the carrier frequency is such that the upper limit frequency (f′max) is limited by the microcomputer's control calculation processing time, and the lower limit frequency (f′min) is limited by the deterioration of controllability due to control delay. receive. Therefore, the frequency band from the upper limit frequency (f′max) to the lower limit frequency (f′min) cannot be wide.
- FIG. 21 is an explanatory diagram of the relationship between the superimposed high-frequency frequency and the harmonic component. That is, as shown in FIG. 21, the peak 171 of the harmonic component by the minimum superposition high frequency f'min that is the lower limit frequency, the peak 172 of the harmonic component by the maximum superposition high frequency f'max that is the upper limit frequency, and the frequency The peaks 173 of the harmonic components that accompany each shift overlap. Therefore, in order to expand the dispersion range as much as possible in the superposition high frequency frequency selection range, the minimum superposition high frequency f'min that is the lower limit frequency of the superposition high frequency frequency selection range and the maximum superposition high frequency that is the upper limit frequency. It can be seen that it is more preferable to disperse by selecting two frequencies f′max as superposed high-frequency frequencies and changing the duration.
- the duration of the superimposed high-frequency frequency is a peak or valley of the carrier, and the duration is an integral multiple of a half cycle of the carrier cycle when the carrier is a triangular wave, and the carrier is a sawtooth wave.
- the duration period in which the superimposed high frequency is maintained is an integral multiple of a half period of the superimposed high frequency, It is also possible to configure so that the frequency of the superimposed high frequency is switched at a timing when the high frequency current becomes zero.
- FIG. 22 is a schematic configuration block diagram of a high-frequency voltage command generation unit.
- the high-frequency voltage command generation unit 131 uses the superimposed high-frequency frequency (f′max or f′min) input from the superimposed high-frequency frequency setting unit 13X as the center superimposed high-frequency frequency fh_c expressed by the equation (6) or (7). Divided by the first superimposed high-frequency voltage amplitude command value Vdh * and output as a second superimposed high-frequency voltage amplitude command value Vdh ** , and a superimposed high-frequency voltage amplitude command generation unit 301 that is input.
- a rectangular wave generating unit 302 that uses the superimposed high-frequency voltage amplitude of the frequency (f′max or f′min) as a superimposed voltage command signal vdch * as a value corresponding to the second superimposed high-frequency voltage amplitude command value Vdh **. ing.
- FIG. 23 is an operation explanation timing chart of the third modification.
- the superposed high frequency frequency f′min ( ⁇ f′max) in the period from time t1 to time t2 and the period from time t3 to time t4.
- the amplitude vhf′min ⁇ vh of the superimposed voltage command signal vdch * .
- the amplitude Ih of the high frequency current idch can be approximated by the following equation using the amplitude vh of the superimposed voltage command signal vdch * , the superimposed high frequency fh, and the inductance L. Ih ⁇ vh / (4fh ⁇ L) Therefore, as a result of the above configuration, vh / fh becomes constant, and as shown in FIG. 23, the amplitude of the high-frequency current idch generated by the superimposed high-frequency voltage is the value of the superimposed high-frequency frequency (f′min or f′max).
- the signal-to-noise ratio is constant regardless of whether the signal-to-noise ratio is constant.
- FIG. 24 is a schematic configuration block diagram of another high-frequency voltage command generation unit.
- a frequency selection signal is sent from the continuation determination unit 123 of FIG. 17 to the high-frequency voltage command generation unit 131.
- the high-frequency voltage command generation unit 131 superimposes the superimposed high-frequency voltage amplitude vhf′max corresponding to the superimposed high-frequency frequency f′max or the superimposed high-frequency frequency f′min.
- a voltage amplitude selection unit 303 that outputs one of the high frequency voltage amplitudes vhf′min, an input superimposed high frequency frequency (f′max or f′min), and an input superimposed high frequency voltage amplitude (vhf′max or vhf′min) ), And a rectangular wave generation unit 302 that uses the superimposed voltage command signal vdch * . Also with this configuration, the amplitude of the high-frequency current idch generated by the superimposed high-frequency voltage is constant regardless of the value of the superimposed high-frequency voltage frequency, and the SN ratio can be kept constant.
- voltage amplitude selection is performed using the frequency selection signal generated by the superimposed high-frequency frequency calculation unit 113 as shown in FIG.
- the superposed high-frequency voltage amplitude command Vdh * can be selected only by switching with a switch as in the section 303, and the program can be simplified.
- FIG. 25 is a schematic configuration block diagram of a PMSM rotation sensorless control system of a fifth embodiment.
- the same parts as those in the fourth embodiment in FIG. 17 are denoted by the same reference numerals.
- the fifth embodiment differs from the fourth embodiment in that the fourth embodiment simply sets the duration of the same superimposed high-frequency frequency that is continuously used by a random value, whereas the second embodiment
- the embodiment is provided with a superposition high-frequency calculation unit that uses a random number value and a transition probability value and determines whether or not the random value satisfies a condition for transition based on the transition probability value.
- a plurality of transition probability values are used to determine whether or not to switch from one superimposed high frequency to another superimposed high frequency, it is possible to more uniformly superimpose the high frequency by controlling the generated high frequency component.
- the frequency can be switched, and the obtained harmonic components can be uniformly dispersed.
- the superimposed high-frequency calculation unit 113A of the PMSM rotation sensorless control system 110A is roughly divided into a random number generator 121 that generates random numbers and a plurality of transition probabilities set in advance based on a frequency selection signal.
- a transition probability selection unit 141 that selects and outputs one of the values, and determines whether or not frequency displacement should be performed based on the input random number value and the input transition probability value, and outputs a transition command signal
- the shift determining unit 142, the frequency selection instruction unit 143 that outputs a frequency selection signal based on the input shift command signal, and the maximum superimposed high-frequency frequency f′max or the minimum based on the frequency selection signal.
- a frequency selection unit 124 that exclusively outputs any one of the superimposed high-frequency frequencies f′min as a superimposed high-frequency signal. Eteiru.
- the transition probability value includes a transition probability value P′hl, which is a probability value for shifting the superimposed high frequency frequency from the maximum superimposed high frequency frequency f′max to the minimum superimposed high frequency frequency f′min, and the superimposed high frequency frequency as the minimum superimposed high frequency frequency f′min.
- a transition probability value P′lh which is a probability value for transition from 1 to the maximum superimposed high-frequency frequency f′max, is used.
- the random number generation unit 121 of the superposition high-frequency calculation unit 113A generates a random value and outputs it to the transition determination unit 142.
- the random number generation unit 121 calculates a pseudo-random number and outputs it as a random value or refers to a random number table and outputs the random value to the transition determination unit 142.
- the transition probability selection unit 141 selects one of a plurality of transition probability values P′hl and P′lh set in advance based on the frequency selection signal output from the frequency selection instruction unit 143. And output to the transition determination unit 142.
- the transition probability selecting unit 141 when the frequency selection signal corresponding to the maximum superimposed high-frequency frequency f′max is input from the frequency selection instruction unit 143 to the transition probability selecting unit 141, the superimposed high-frequency frequency is changed from the maximum superimposed high-frequency frequency f′max.
- a transition probability value P′hl which is a probability value to be shifted to the minimum superimposed high frequency f′min, is output to the transition determination unit 142.
- the transition probability selection unit 141 When the frequency selection signal corresponding to the minimum superimposed high frequency f'min is input from the peripheral frequency selection instruction unit 143 to the transition probability selection unit 141, the superimposed high frequency is changed from the minimum superimposed high frequency f'min to the maximum superimposed frequency.
- a transition probability value P′lh that is a probability value to be shifted to the high frequency f′max is output to the transition determination unit 142.
- the frequency selection instruction unit 143 outputs a frequency selection signal to the frequency selection unit 124 based on the input shift command signal.
- the frequency selection unit 124 sets the superposed high frequency as a superposed high frequency signal using either the maximum superposed high frequency f'max or the minimum superposed high frequency f'min as a superposed high frequency signal.
- the high-frequency voltage command generator 131 generates a superimposed voltage command signal corresponding to the frequency corresponding to the superimposed high-frequency signal (in this second embodiment, either the maximum superimposed high-frequency frequency f′max or the minimum superimposed high-frequency frequency f′min).
- Generate vdch * is output to the rotational phase angle estimating unit 132 and the high frequency voltage superimposing unit 134.
- the current control unit 133 performs basic control based on the input d-axis current command signal idc * , q-axis current command signal iqc * , d-axis current detection signal idc, and q-axis current detection signal iqc.
- the wave voltage command signals vdcf * and vqcf * are generated and output to the high frequency voltage superimposing unit 134.
- the high-frequency voltage superimposing unit 134 the fundamental wave voltage instruction signal Vdcf * and superimposed voltage command signal VDCH * d-axis voltage command signal vdc based on *, generates a fundamental wave voltage command signal Vqcf * and superimposed voltage
- a q-axis voltage command signal vqc * is generated based on the command signal vqch * .
- the generated d-axis voltage command signal vdc * and q-axis voltage command signal vqc * are output to the first coordinate converter 135.
- the first coordinate conversion unit 135 performs coordinate conversion of the input d-axis voltage command signal vdc * and q-axis voltage command signal vqc * to perform U-phase voltage command signal vu * , V-phase voltage command signal vv *, and W-phase.
- a voltage command signal vw * is generated.
- the generated U-phase voltage command signal vu * , V-phase voltage command signal vv *, and W-phase voltage command signal vw * are output to the PWM modulation unit 136.
- the PWM modulation unit 136 compares the input voltage command signals vu * , vv * , vw * with a triangular wave or sawtooth wave as a carrier wave, performs PWM modulation, and turns on / off each phase switching element of the inverter 111.
- a gate signal that is an off command is output to the PMSM 112.
- U-phase current, V-phase current, and W-phase current flow through PMSM 112 in a synchronized state, and a rotor (not shown) of PMSM 112 rotates.
- the current detection unit 137 detects a current response value of a plurality of phases (two phases of the U phase and the W phase in the example of FIG. 25) out of the three-phase alternating current flowing in the PMSM 112, and the current detection signal (In the example of FIG. 25, the U-phase current detection signal iu and the W-phase current detection signal iw) are output to the second coordinate conversion unit 138.
- the second coordinate conversion unit 138 performs coordinate conversion (UVW / dcqc conversion) of the current detection signal output by the current detection unit 137 (in the example of FIG. 25, the U-phase current detection signal iu and the W-phase current detection signal iw).
- the d-axis current detection signal idc is output to the current control unit 133 and the q-axis current detection signal iqc is output to the rotational phase angle estimation unit 132 and the current control unit 133.
- the rotational phase angle estimation unit 132 estimates the rotational phase angle of the PMSM 12 based on the input superimposed voltage command signal vdch * and the q-axis current detection signal iqc, and calculates the estimated phase angle ⁇ est as the first coordinate conversion unit 135. And output to the second coordinate converter 138.
- the first coordinate conversion unit 135 outputs to the PWM modulation unit 136 voltage command signals vu * , vv * , vw * that are optimal for the rotation state of the PMSM 12 corresponding to the estimated phase angle ⁇ est. Therefore, the inverter 111 performs rotation driving according to the rotation state of the PMSM 112 while suppressing noise.
- the shift state of the superimposed high frequency can be adjusted by using an appropriate shift probability value, so that the frequency spectrum of the desired harmonics can be adjusted. Distribution generation is facilitated.
- FIG. 26 is an operation explanatory diagram of the fifth embodiment.
- the frequency spectrum distribution of the generated harmonics is three types of frequency spectrum distributions as shown in FIG. That is, (1) The frequency spectrum distribution 171 when the minimum superposition high frequency f′min is continued, (2) The frequency spectrum distribution 172 when the maximum superimposed high frequency f′max is continued, (3) Both the frequency spectrum distribution 173 and the transition time from the minimum superposition high frequency f'min to the maximum superposition high frequency f'max and the transition from the maximum superposition high frequency f'max to the minimum superposition high frequency f'min So it will be treated in common. Therefore, it considers using these three types of frequency spectrum distribution.
- the amplitude of the harmonic component at the peak of each frequency spectrum distribution can be expressed by equations (8) to (10).
- the constant C varies depending on the modulation rate, the dispersion range, and the like, but is a constant common to the three dispersions. Based on the above, in the fifth embodiment, the frequency spectrum distribution of harmonics is adjusted.
- the transition probability value P′hl for shifting the superimposed high-frequency frequency from the maximum superimposed high-frequency frequency f′max to the minimum superimposed high-frequency frequency f′min is increased.
- the transition probability value P′lh for shifting the superimposed high frequency frequency from the minimum superimposed high frequency f′min to the maximum superimposed high frequency f′max may be reduced.
- the transition probability value P'hl for shifting the superimposed high frequency from the maximum superimposed high frequency f'max to the minimum superimposed high frequency f'min is reduced.
- the transition probability value P′lh for shifting the superposed high frequency frequency from the minimum superposed high frequency f′min to the maximum superposed high frequency f′max may be increased.
- transition probability value P′hl for shifting the superimposed high-frequency frequency from the maximum superimposed high-frequency frequency f′max to the minimum superimposed high-frequency frequency f′min, or the superimposed high-frequency frequency from the minimum superimposed high-frequency frequency f′min to the maximum superimposed high-frequency frequency f′min.
- the transition probability value P′lh to be shifted to f′max may be increased. This makes it possible to generate an arbitrary frequency spectrum distribution.
- the selection range of the superimposed high-frequency frequency cannot normally be wide. Therefore, as shown in FIG. 21, the frequency spectrum distribution 171 of the harmonic component due to the lower limit superposed high frequency f'min, the frequency spectrum distribution 172 of the harmonic component due to the maximum superposed high frequency f'max, and the transition of the superposed high frequency. Thus, the frequency spectrum distributions 173 of the harmonic components resulting from are overlapped.
- FIG. 27 is an explanatory diagram of the effect of the fifth embodiment.
- the frequency spectrum distribution 173 of harmonics resulting from the transition of the superimposed high-frequency frequency is the frequency spectrum distribution 171 of harmonic components due to the minimum superimposed high-frequency frequency f′min and the harmonic component due to the maximum superimposed high-frequency frequency f′max.
- the sum of the spectra is flat.
- FIG. As shown, it is possible to make it as flat as possible within the selection range of the superposed high frequency.
- the component of the average frequency in the superimposed high-frequency frequency selection range can be reduced even if the shift determination is performed according to, for example, a sine wave instead of a random number.
- FIG. 28 is a schematic configuration block diagram of a PMSM rotation sensorless control system of a sixth embodiment.
- the same parts as those in the fourth embodiment in FIG. 17 are denoted by the same reference numerals.
- the sixth embodiment differs from the fourth embodiment in that the carrier frequency used in the PWM modulator 136 (in the case of FIG. 28, the maximum carrier frequency) in the PMSM rotation sensorless control system 110B instead of the superposed high-frequency calculation unit 113.
- PWM modulation unit by generating a carrier SC having a carrier frequency determined by the carrier frequency calculation unit 151 and a point using the carrier frequency calculation unit 151 that determines either f′max1 or the minimum carrier frequency f′min1)
- a high frequency voltage command generation unit 131A that outputs a superimposed voltage command signal vdch * for instructing the frequency of the high frequency voltage to be superimposed based on the carrier frequency signal; It is a point with.
- the random number generation unit 121 of the carrier frequency calculation unit 151 generates a random value and outputs it to the duration determination unit 122.
- the random number generator 121 calculates a pseudo-random number and outputs it as a random value, or refers to a random number table and outputs a random value.
- the duration determination unit 122 determines the duration of the superimposed high-frequency frequency based on the input random number and outputs it as duration data. More specifically, in the first embodiment, the calculation is performed by the above-described equation (1) based on one period of the superimposed high frequency frequency.
- the continuation determination unit 123 performs continuation determination based on the input duration data and outputs a frequency selection signal.
- the frequency selection signal since there are two types of carrier frequency, the maximum carrier frequency f′max1 or the minimum carrier frequency f′min1, the frequency selection signal has a value of “0” or “1”. Is binary data. Therefore, a frequency selection signal corresponding to either the maximum carrier frequency f′max1 or the minimum carrier frequency f′min1 is output to the frequency selection unit 124 according to the continuation determination.
- the frequency selection unit 124 outputs either the maximum carrier frequency f′max1 or the minimum carrier frequency f′min1 to the high frequency voltage command generation unit 131A and the carrier generation unit 152 as a carrier frequency signal based on the frequency selection signal.
- the high-frequency voltage command generation unit 131A generates the superposed voltage command signal vdch * for instructing the frequency of the high-frequency voltage to be superposed on the basis of the input carrier frequency signal.
- the data is output to the superimposing unit 134.
- the carrier generation unit 152 generates a carrier signal SC having a frequency corresponding to the carrier frequency signal (in this third embodiment, either the maximum carrier frequency f′max1 or the minimum carrier frequency f′min1), and performs PWM modulation. Output to the unit 136.
- FIG. 29 is a timing chart of the sixth embodiment.
- the carrier frequency calculation unit 151 of the third embodiment generates a carrier frequency in which the duration of the maximum carrier frequency f′max1 or the minimum carrier frequency f′min1 changes randomly.
- the carrier frequency is output from the carrier frequency calculation unit 151 to the carrier generation unit 152.
- the carrier wave generator 152 outputs the carrier signal SC to the PWM modulator 136 using the input carrier frequency.
- both the electromagnetic noise due to the superimposed high frequency and the electromagnetic noise due to the superimposed high frequency can be simultaneously reduced by synchronizing the superimposed high frequency with the carrier wave.
- the frequency of the superimposed high-frequency voltage is made equal to the carrier frequency.
- the frequency may be, for example, half or one third of the superimposed high-frequency.
- FIG. 30 is an explanatory diagram of a modification of the sixth embodiment.
- the same calculation method as that of the superimposed high frequency calculation unit 13 of the first embodiment is used as the carrier frequency calculation unit 151.
- the modification of the sixth embodiment is a PMSM rotation sensorless.
- the control system 110C as shown in FIG. 30, it is also possible to adopt a configuration of a carrier frequency calculation unit 151A using the same calculation method as the superimposed high frequency frequency calculation unit 113A of the fifth embodiment.
- the component of the average frequency in the superimposed high-frequency frequency selection range even if the transition determination is performed according to, for example, a sine wave instead of a random value. Can be reduced.
- the same effect can be obtained by the method of superimposing the high-frequency current. The same effect can be obtained even when a voltage is superimposed on both the d-axis and the q-axis, or only on the q-axis, or a sine wave is superimposed.
- the rotation sensorless control device of the fourth to sixth embodiments includes a control device such as a CPU, a ROM (Read It is equipped with a storage device such as an only memory (RAM), a RAM, an external storage device such as an HDD or a CD drive device, a display device such as a display device, and an input device such as a keyboard and a mouse. It is also possible to configure as a hardware configuration.
- a control device such as a CPU, a ROM (Read It is equipped with a storage device such as an only memory (RAM), a RAM, an external storage device such as an HDD or a CD drive device, a display device such as a display device, and an input device such as a keyboard and a mouse. It is also possible to configure as a hardware configuration.
- the control program executed by the rotation sensorless control device of the fourth to sixth embodiments is a CD-ROM, flexible disk in an installable or executable file. (FD), CD-R, DVD (Digital Versatile Disk), etc. may be recorded on a computer-readable recording medium and provided.
- a control program executed by the rotation sensorless control device is stored on a computer connected to a network such as the Internet, and is transmitted via the network. You may comprise so that it may provide by downloading. Further, the control program executed by the control unit of the rotation sensorless control apparatus of the present embodiment may be provided or distributed via a network such as the Internet.
- control program for the rotation sensorless control device may be provided by being incorporated in advance in a ROM or the like.
Abstract
Description
しかしながら、重畳される高周波電圧により電磁騒音が発生することがわかっており、この発生する電磁騒音を低減するために、重畳する高周波電圧の周波数を時間的に切り替える(変更する)技術が提案されている(たとえば、特許文献4参照)。
したがって、PWM電力変換に用いるPWM信号の搬送波周波数の影響を低減するために、複数の搬送波周波数を切り替えて用いた場合には、切替回数が多くなるほど平均周波数の成分が大きくなり、狙った通りの周波数スペクトル分布を実現できず、機械共振を回避するのが困難となり電磁騒音が発生する虞があった。
また、PWM電力変換に用いるPWM信号に重畳される重畳高周波周波数の影響を低減するために、複数の重畳高周波周波数を切り替えて用いた場合には、切替回数が多くなるほど平均周波数の成分が大きくなり、平坦な周波数スペクトル分布の実現が困難となる虞れがある。特に、一般的には重畳高周波周波数を選択できる範囲には制約があり、狭い範囲しか選択できないことに起因して平均周波数の成分が大きくなり、電磁騒音が発生する虞があった。
本発明は、上記に鑑みてなされたものであって、PWM制御時の電磁騒音を低減可能な電力変換装置、電力変換装置の制御方法、回転センサレス制御装置、回転センサレス制御装置の制御方法及び制御プログラムを提供することを目的としている。
これにより、搬送波発生部は、所定の設定搬送波周波数の搬送波を所定の継続時間の間発生する。
そして、PWM信号発生部は、搬送波発生部において発生された搬送波に基づいて、PWM信号を発生し、電力変換部は、発生されたPWM信号に基づいて電力変換を行って負荷に供給する。
[1]第1実施形態
図1は、第1実施形態のPWM電力変換装置の概要構成ブロック図である。
PWM電力変換装置10は、大別すると、直流電源11からの入力直流電圧を降圧して駆動電圧として負荷12に出力する電力変換部として機能する降圧チョッパ13と、搬送波周波数を演算して搬送波周波数信号を出力する搬送波周波数演算部14と、搬送波周波数信号に相当する搬送波周波数を有する搬送波信号を発生する搬送波信号発生部15と、出力電圧指令信号を生成し、出力する出力電圧指令生成部16と、入力された搬送波信号及び出力電圧指令信号に基づいてPWM信号を降圧チョッパ13に出力するPWM信号発生部17と、を備えている。
上記構成において、搬送波周波数演算部14、搬送波信号発生部15、出力電圧指令生成部16及びPWM信号発生部17は、PWM電力変換を制御する制御部18を構成している。
継続判定部23は、継続期間データ及びカウントデータが入力され、継続期間データの値及びカウントデータの値が一致したか否かを判別して、比較結果信号を出力する比較器31と、比較結果信号が一致を表す場合にリセットされ、比較結果信号が不一致の状態でカウント値を増加してカウントデータとして出力するカウンタ32と、比較結果信号及び周波数選択信号が入力され、比較結果信号と周波数選択信号とが不一致の場合に、周波数選択信号を反転する反転処理器33と、を備えている。
まず、搬送波周波数演算部14の乱数発生部21は、乱数値を発生して継続期間決定部22に出力する。ここで、乱数発生部21は、疑似乱数を演算して乱数値として出力したり、乱数テーブルを参照したりして乱数値を出力する。
継続期間=搬送波1周期×乱数値 ……(1)
降圧チョッパ13は、PWM信号に基づいて直流電源11からの入力直流電圧を降圧して駆動電圧として負荷12に出力し、負荷12が駆動されることとなる。
図3は、第1実施形態のタイミングチャートである。
図3に示すように、PWM信号発生部17の直流電源電圧をVdc[V]とした場合に、搬送波信号発生部15が発生する搬送波信号SCは、0[V]と直流電源電圧Vdc[V]との二つの電圧レベルの間で遷移する三角波となる。
一方、出力電圧指令信号SBの電圧は、一定である。
これらの結果、PWM信号発生部17で発生されるPWM信号SPは、搬送波信号SC<出力電圧指令信号SBの場合には、“H”レベルとなり、搬送波信号SC≧出力電圧指令信号SBの場合には、“L”レベルとなる。
図4に示すように、本第1実施形態によれば、乱数発生部20において発生された乱数に基づいて、搬送波信号SCの周波数(設定搬送波周波数)は、最大搬送波周波数fmaxあるいは最小搬送波周波数fminのいずれかの継続期間がランダムに変化するようにされている。
また、本第1実施形態においては、乱数値に応じて継続期間を変更することで、同一周波数の搬送波の継続期間の変化に規則性が無くなるため、継続期間の変化に伴う聴覚上の違和感が生じることが無くなる。
搬送波によって発生する高調波の周波数スペクトル分布において、各搬送波周波数によって発生する高調波成分のピークは搬送波周波数選択範囲内に表れる。
すなわち、図5に示すように、下限の搬送波周波数である最小搬送波周波数fminによる高調波成分のピーク71と上限の搬送波周波数である最大搬送波周波数fmaxによる高調波成分のピーク72、周波数の変移に伴う高調波成分のピーク73はそれぞれ重なることになる。したがって、搬送波周波数選択範囲の中で、分散の範囲を可能な限り拡げるためには、搬送波周波数選択範囲の下限の最小搬送波周波数fminと上限の最大搬送波周波数fmaxの2つの周波数を搬送波周波数として選択して、継続期間を変化させることで分散するのがより好ましいことがわかる。
本第1実施形態では、降圧チョッパ13による負荷12の制御について示したが、例えば、図6に示すように、インバータ13Aによる交流モータ12Aの制御の場合等、PWM電力変換装置であれば同様に高調波成分の分散効果が得られ、機械的共振などの発生を抑制することができる。
図7は、第2実施形態のPWM電力変換装置の概要構成ブロック図である。
図7において、図1の第1実施形態と同様の部分には、同一の符号を付すものとする。
本第2実施形態が第1実施形態と異なるのは、第1実施形態が、単純に乱数値によって同一周波数の搬送波をPWM信号の生成に継続して用いる継続期間を設定していたのに対し、本第2実施形態は、乱数値及び変移確率値を用い、乱数値が変移すべき条件を満たしているか否かを変移確率値に基づいて行う搬送波周波数演算部を備えた点である。この場合に、変移確率値を複数用いることにより、ある搬送波周波数から他の搬送周波数に切り替えるか否かを判別しているため、より一層均一に搬送波周波数を切り替えることができ、得られる高調波成分の周波数を均一に分散させることができる。
まず、搬送波周波数演算部14Aの乱数発生部21は、乱数値を発生して変移判定部42に出力する。ここで、乱数発生部21は、第1実施形態と同様に、疑似乱数を演算して乱数値として出力したり、乱数テーブルを参照したりして乱数値を変移判定部42に出力する。
より具体的には、例えば、乱数値を0~1として、入力された乱数値が変移確率値(例えば、変移確率値Phl=0.45)以下の場合には、現在の搬送波周波数とは異なる搬送波周波数に変移する変移指令信号を周波数選択指示部43に出力する。
この結果、周波数選択部24は、入力された周波数選択信号に基づいて、搬送波周波数を最大搬送波周波数fmaxあるいは最小搬送波周波数fminのうちいずれか一方を搬送波周波数信号として搬送波信号発生部15に出力する。
搬送波信号発生部15は、搬送波周波数信号に対応する周波数(本第2施形態では、最大搬送波周波数fmaxあるいは最小搬送波周波数fminのいずれか)の搬送波信号を生成し、PWM信号発生部17に出力する。
これらの結果、PWM信号発生部17は、入力された搬送波信号及び出力電圧指令信号に基づいてPWM信号を降圧チョッパ13に出力する。
降圧チョッパ13は、PWM信号に基づいて直流電源11からの入力直流電圧を降圧して駆動電圧として負荷12に出力し、負荷12が駆動されることとなる。
以上の構成により、搬送波周波数演算部14Aにおいては、第1実施形態と同様に、最大搬送波周波数fmaxあるいは最小搬送波周波数fminの継続期間がランダムに変化する搬送波信号SC1が出力され、PWM信号発生部17により、図8に示すようなPWM信号SP1が出力される。
本第2実施形態においては、動作モードとしては、以下の4つしか存在しない。
(1)最小搬送波周波数fminの継続
(2)最大搬送波周波数fmaxの継続
(3)最小搬送波周波数fminから最大搬送波周波数fmaxへの変移
(4)最大搬送波周波数fmaxから最小搬送波周波数fminへの変移
この場合において、生じる高調波の周波数スペクトル分布としては、図9に示すように、3種類の周波数スペクトル分布となる。(3)最小搬送波周波数fminから最大搬送波周波数fmaxへの変移と(4)最大搬送波周波数fmaxから最小搬送波周波数fminへの変移はいずれも73となるので、共通で扱う。
各周波数スペクトル分布のピークにおける高調波成分の振幅は、(2)式~(4)式で表せる。
以上に基づいて、本第2実施形態においては、高調波の周波数スペクトル分布を調整している。
したがって、図5に示したように、搬送波周波数選択範囲の下限の搬送波周波数である最小搬送波周波数fminによる高調波成分のピーク71、搬送波周波数選択範囲の上限の搬送波周波数である最大搬送波周波数fmaxによる高調波成分のピーク72及び搬送波周波数の変移に起因する高調波成分のピーク73はそれぞれ重なることになる。
そのため、搬送波周波数の変移に起因する高調波のピーク73を、最小搬送波周波数fminによる高調波成分のピーク71及び最大搬送波周波数fmaxによる高調波成分のピーク72に対して小さくすることで、図10に示すように、搬送波周波数の選択範囲内で可能な限り平坦にすることが可能になる。
本第3実施形態が上記第2実施形態と異なる点は、負荷の機械特性に基づいて変移確率値および搬送波周波数を決定する変移確率搬送波周波数決定部を備えた点である。
図11は、第3実施形態のPWM電力変換装置の概要構成ブロック図である。
図11において、図7の第2実施形態と同一の部分には、同一の符号を付すものとする。
図13は、機械特性判定における負荷の騒音特性の説明図である。
上述した第3実施形態のPWM電力変換装置10の構成において、機械特性とは、例えば、図12のようなホワイトノイズに近いような電流を負荷に流した際における、図13に示す負荷の騒音特性や、単純に負荷の機械共振点の周波数(例えば、図13においては、3.4kHz及び3.9kHz)である。
そして、変移確率搬送波周波数決定部51は、入力された機械特性に応じて、例えば、図14に示すように、機械特性82における機械共振点を、最大搬送波周波数fmax及び最小搬送波周波数fminの平均周波数[=2fmin×fmax/(fmax+fmin)≒(fmax+fmin)/2]と一致させるように最大搬送波周波数fmax及び最小搬送波周波数fminを決定する。さらに、変移確率値Phl、Plhを小さくすることにより、搬送波周波数の変移の回数を抑制し、平均周波数における高調波のピークの発生を抑制することで、機械共振を回避した周波数スペクトル分布81にする。
例えば、図15に示すように、最大搬送波周波数fmaxにおける騒音が大きく出る機械特性84の場合には、搬送波周波数を最大搬送波周波数fmaxから最小搬送波周波数fminに変移させる変移確率値Phlよりも、搬送波周波数を最小搬送波周波数fminから最大搬送波周波数fmaxに変移させる変移確率値Plhを小さくして最大搬送波周波数fmaxの高調波のピークよりも最小搬送波周波数fminの高調波のピークが大きくなるように、機械共振を回避した周波数スペクトル分布83にする。
本第3実施形態においても、第1実施形態及び第2実施形態と同様に、乱数ではなく、例えば正弦波に応じて変移判定を実施しても、搬送波周波数選択範囲の平均周波数の成分を低減できる。
また、第3実施形態においても、第1実施形態及び第2実施形態と同様に、降圧チョッパ13による負荷12の制御について示したが、例えば、負荷12として機能する交流モータ12Aを制御するインバータであっても交流モータ12Aの機械特性に応じて本発明を用いれば同様に電磁騒音低減効果が得られる。
[4.1]第1変形例
図16は、第1実施形態乃至第3実施形態の第1変形例の概要構成説明図である。
以上の各実施形態においては、負荷の機械特性を別途入力(設定)する構成を採っていたが、図16に示すように、負荷の近傍に騒音検出用のマイクロフォン91を設け、マイクロフォン91の出力を高速フーリエ変換(FFT)して分析するFFTアナライザ92を設け、FFT分析結果に基づいて最大搬送波周波数fmax、最小搬送波周波数fmin及び搬送波継続期間を定めるように構成することも可能である。
第1実施形態乃至第3実施形態の第1変形例と同様にFFTアナライザを設け、さらに、第2実施形態で用いた変移確率値Phl及び変移確率値PlhをFFT分析結果に基づいて定めるように構成することも可能である。
上記第3実施形態においては、変移確率搬送波周波数決定部51を持つ構成を採っていたが、第2実施形態の構成において、機械共振を避けるようにあらかじめ変移確率値Phl、Plh及び搬送波周波数fmax、fminを計算しておき、それを設定するように構成することも可能である。
第1実施形態乃至第3実施形態のPWM電力変換装置の制御部は、CPUなどの制御装置と、ROM(Read Only Memory)やRAMなどの記憶装置と、HDD、CDドライブ装置などの外部記憶装置と、ディスプレイ装置などの表示装置と、キーボードやマウスなどの入力装置を備えており、通常のコンピュータを利用したハードウェア構成として構成することも可能である。
また第1実施形態乃至第3実施形態のPWM電力変換装置の制御部で実行される制御プログラムは、インストール可能な形式又は実行可能な形式のファイルでCD-ROM、フレキシブルディスク(FD)、CD-R、DVD(Digital Versatile Disk)等のコンピュータで読み取り可能な記録媒体に記録されて提供されるようにしてもよい。
また、第1実施形態乃至第3実施形態のPWM電力変換装置の制御部で実行される制御プログラムを、インターネット等のネットワークに接続されたコンピュータ上に格納し、ネットワーク経由でダウンロードさせることにより提供するように構成しても良い。また、本実施形態のPWM電力変換装置で実行される制御プログラムをインターネット等のネットワーク経由で提供または配布するように構成してもよい。
また、第1実施形態乃至第3実施形態のPWM電力変換装置の制御部の制御プログラムを、ROM等に予め組み込んで提供するように構成してもよい。
図17は、第4実施形態のPMSM回転センサレス制御システムの概要構成ブロック図である。
PMSM回転センサレス制御システム110は、大別すると、電力変換を行うインバータ111と、インバータ111により回転駆動されるPMSM112と、電動機鉄心突極性を利用してPMSM112の回転角度を推定するために重畳する高周波電圧の周波数を演算し、設定するための重畳高周波周波数演算部13と、重畳高周波周波数演算部113により設定された周波数を有する高周波信号を制御信号に重畳しつつ、インバータ111を制御するインバータ制御部114と、を備えている。このとき、重畳高周波周波数演算部113はインバータ制御部114内部または外部に設けることが可能である。
継続判定部123は、比較器161、カウンタ162、反転処理器163を備えている。
比較器161は、継続期間データ及びカウントデータが入力され、継続期間データの値及びカウントデータの値が一致したか否かを判別して、比較結果信号を出力する。
反転処理器163は、比較結果信号及び周波数選択信号が入力され、比較結果信号と周波数選択信号とが不一致となった場合に、周波数選択信号を反転する。
以上の一連の処理が継続判定となる。
まず、重畳高周波周波数演算部113の乱数発生部121は、乱数値を発生して継続期間決定部122に出力する。ここで、乱数発生部121は、疑似乱数を演算して乱数値として出力したり、乱数テーブルを参照したりして乱数値を出力する。
継続期間=重畳高周波周波数の1周期×乱数値 ……(5)
この結果、インバータ111よりPMSM112には、同期した状態でU相電流、V相電流及びW相電流が流れて、PMSM112の図示しない回転子が回転することとなる。
図19は、第4実施形態のタイミングチャートである。
図19に示すように、PWM変調部136の直流電源電圧をVdc[V]とした場合に、搬送波発生部が発生する搬送波信号SCは、0[V]と直流電源電圧Vdc[V]との2つの電圧レベルの間で遷移する三角波となる。
より詳細には、図19において、時刻t1から時刻t2に至る期間及び時刻t3から時刻t4に至る期間においては、重畳電圧指令信号vdch*は、重畳高周波周波数=最小重畳高周波周波数f’minであるので、比較的低い周波数の矩形波となっている。
ここで図20は、図19(a)の長期的な時間軸で表現した図になっている。
図20に示すように、本第4実施形態によれば、乱数発生部121において発生された乱数値に基づいて、重畳高周波周波数は、最大重畳高周波周波数f’maxあるいは最小重畳高周波周波数f’minのいずれかの継続期間がランダムに変化するようにされている。
また、本第4実施形態においては、乱数値に応じて継続期間を変更することで、同一周波数の重畳高周波の継続期間の変化に規則性が無くなるため、継続期間の変化に伴う聴覚上の違和感を生じることが無くなる。
また、本第4実施形態によれば、乱数値に応じて継続期間を変更しているので、同一の重畳高周波周波数を維持し、継続する継続期間の変化に規則性が無くなり、継続期間の変化に伴う聴覚上の違和感が生じることが少なくなる。
すなわち、図21に示すように、下限の周波数である最小重畳高周波周波数f’minによる高調波成分の山171、上限の周波数である最大重畳高周波周波数f’maxによる高調波成分の山172及び周波数の変移に伴う高調波成分の山173はそれぞれ重なることになる。したがって、重畳高周波周波数選択範囲の中で、分散の範囲を可能な限り広げるためには、重畳高周波周波数選択範囲の下限の周波数である最小重畳高周波周波数f’minと上限の周波数である最大重畳高周波周波数f’maxの2つの周波数を重畳高周波周波数として選択して、継続期間を変化させることで分散するのがより好ましいことがわかる。
[5.1.1]第1変形例
上記構成において、重畳高周波周波数を維持する継続期間期を、重畳高周波の周期の半周期の整数倍とし、高周波電流が零となるタイミングで重畳高周波の周波数を切り替えるように構成することも可能である。
また、さらに、継続期間を、重畳高周波の1周期の整数倍とすることで、高周波1周期のフーリエ級数演算によって回転位相角の推定、すなわち、推定位相角θestを正確に算出することができる。
上記第4実施形態では、高周波電圧を重畳する方法を示したが、高周波電流を重畳する方法でも同様の効果を得ることが可能である。
また、上記第4実施形態では、重畳高周波電圧をd軸に矩形波を重畳している例を示したが、d軸とq軸の両方あるいはq軸だけに電圧を重畳する、あるいは正弦波を重畳する場合でも同様の効果が得られる。
図22は、高周波電圧指令生成部の概要構成ブロック図である。
上記構成の結果、図23に示すように、時刻t1から時刻t2に至る期間及び時刻t3から時刻t4に至る期間は、重畳高周波周波数=f’min(<f’max)となっている。この期間における重畳電圧指令信号vdch*の値は、重畳高周波周波数が、中心重畳高周波周波数fh_cにおける重畳電圧指令信号vdch*の振幅=vh[V]の場合に、重畳高周波周波数=f’minにおける重畳電圧指令信号vdch*の振幅vhf’min<vhとなっている。
Ih≒vh/(4fh×L)
したがって、上記構成の結果、vh/fhが一定となるので、図23に示すように、重畳した高周波電圧によって発生する高周波電流idchの振幅が重畳高周波周波数の値(f’minまたはf’max)にかかわらず一定となり、SN比を一定に保つことができる。
このように構成することで、騒音を低減するために重畳高周波周波数を可変にすることで回転子磁極位置の推定精度が落ちるような場合があっても、重畳高周波周波数の電圧振幅に応じて制御することができるため、重畳高周波周波数が一定のときと同様の推定精度を維持することができる。
図24の構成においては、図17の継続判定部123から高周波電圧指令生成部131へ周波数選択信号が送られる。
高周波電圧指令生成部131は、継続判定部123から入力された周波数選択信号に基づいて、重畳高周波周波数f’maxに対応する重畳高周波電圧振幅vhf’maxあるいは重畳高周波周波数f’minに対応する重畳高周波電圧振幅vhf’minのいずれかを出力する電圧振幅選択部303と、入力された重畳高周波周波数(f’maxまたはf’min)及び入力された重畳高周波電圧振幅(vhf’maxまたはvhf’min)に基づいて、重畳電圧指令信号vdch*とする矩形波生成部302と、を備えている。
本構成によっても、重畳した高周波電圧によって発生する高周波電流idchの振幅が重畳した高周波電圧周波数の値にかかわらず一定となり、SN比を一定に保つことができる。
図24の高周波電圧指令生成部131を第4実施形態の重畳高周波周波数演算部113と組み合わせることで、図24に示すように重畳高周波周波数演算部113で生成した周波数選択信号を用いて電圧振幅選択部303のようにスイッチによる切り替えだけで重畳高周波電圧振幅指令Vdh*を選択でき、プログラムを単純にすることができる。
図25は、第5実施形態のPMSM回転センサレス制御システムの概要構成ブロック図である。
図25において、図17の第4実施形態と同様の部分には、同一の符号を付すものとする。
まず、重畳高周波周波数演算部113Aの乱数発生部121は、乱数値を発生して変移判定部142に出力する。ここで、乱数発生部121は、第1実施形態と同様に、疑似乱数を演算して乱数値として出力したり、乱数テーブルを参照したりして乱数値を変移判定部142に出力する。
より具体的には、例えば、乱数値を0~1として、入力された乱数値が変移確率値(たとえば、変移確率値P’hl=0.45)以下の場合には、現在の重畳高周波周波数とは異なる重畳高周波周波数に変移する変移指令信号を周波数選択指示部143に出力する。
この結果、周波数選択部124は、入力された周波数選択信号に基づいて、重畳高周波周波数を最大重畳高周波周波数f’maxあるいは最小重畳高周波周波数f’minのうちいずれか一方を重畳高周波周波数信号として高周波電圧指令生成部131に出力する。高周波電圧指令生成部131は、重畳高周波周波数信号に対応する周波数(本第2施形態では、最大重畳高周波周波数f’maxあるいは最小重畳高周波周波数f’minのいずれか)に対応する重畳電圧指令信号vdch*を生成する。また、生成した重畳電圧指令信号vdch*を回転位相角推定部132及び高周波電圧重畳部134に出力する。
この結果、PMSM112には、同期した状態でU相電流、V相電流及びW相電流が流れて、PMSM112の図示しない回転子が回転することとなる。
本第5実施形態において、動作モードとしては、以下の4つしか存在しない。
(1)最小重畳高周波周波数f’minの継続
(2)最大重畳高周波周波数f’maxの継続
(3)最小重畳高周波周波数f’minから最大重畳高周波周波数f’maxへの変移
(4)最大重畳高周波周波数f’maxから最小重畳高周波周波数f’minへの変移
この場合において、生じる高調波の周波数スペクトル分布としては、図26に示すように、3種類の周波数スペクトル分布となる。
すなわち、
(1)最小重畳高周波周波数f’minの継続時が周波数スペクトル分布171、
(2)最大重畳高周波周波数f’maxの継続時が周波数スペクトル分布172、
(3)最小重畳高周波周波数f’minから最大重畳高周波周波数f’maxへの変移時及び最大重畳高周波周波数f’maxから最小重畳高周波周波数f’minへの変移時はいずれも周波数スペクトル分布173となるので、共通で扱う。
したがって、これら3種類の周波数スペクトル分布を用いて考察する。
各周波数スペクトル分布のピークにおける高調波成分の振幅は、(8)式~(10)式で表せる。
以上に基づいて、本第5実施形態においては、高調波の周波数スペクトル分布を調整している。
したがって、図21に示したように、下限の重畳高周波周波数f’minによる高調波成分の周波数スペクトル分布171と最大重畳高周波周波数f’maxによる高調波成分の周波数スペクトル分布172、重畳高周波周波数の変移に起因する高調波成分の周波数スペクトル分布173はそれぞれ重なることになる。
図27では、重畳高周波周波数の変移に起因する高調波の周波数スペクトル分布173が、最小重畳高周波周波数f’minによる高調波成分の周波数スペクトル分布171及び最大重畳高周波周波数f’maxによる高調波成分の周波数スペクトル分布172と重なり合った際にスペクトルの和が平坦となっている。スペクトルの和を平坦とするために、周波数スペクトル分布173が、周波数スペクトル分布171及び周波数スペクトル分布172に対して低くなるように変移確率P’hlとP’lhを設定することで、図27に示すように、重畳高周波周波数の選択範囲内で可能な限り平坦にすることが可能になる。
図28は、第6実施形態のPMSM回転センサレス制御システムの概要構成ブロック図である。
図28において、図17の第4実施形態と同様の部分には、同一の符号を付すものとする。
まず、搬送波周波数演算部151の乱数発生部121は、乱数値を発生して継続期間決定部122に出力する。ここで、乱数発生部121は、疑似乱数を演算して乱数値として出力したり、乱数テーブルを参照したりして乱数値を出力する。
以上の構成により、第3実施形態の搬送波周波数演算部151においては、最大搬送波周波数f’max1あるいは最小搬送波周波数f’min1の継続期間がランダムに変化する搬送波周波数が生成される。搬送波周波数は搬送波周波数演算部151より搬送波発生部152へ出力される。搬送波発生部152において、入力された搬送波周波数を用いて搬送波信号SCをPWM変調部136に出力する。
以上の説明においては、重畳高周波電圧の周波数を搬送波周波数と等しくしたが、同期していれば、例えば重畳高周波周波数の半分、3分の1などにしてもよい。
また、本第6実施形態では、搬送波周波数演算部151として、第1実施形態の重畳高周波周波数演算部13と同様の演算方法を用いたが、本第6実施形態の変形例は、PMSM回転センサレス制御システム110Cにおいて、図30に示すように、第5実施形態の重畳高周波周波数演算部113Aと同様の演算方法を用いた搬送波周波数演算部151Aの構成を採ることも可能である。
また、本第6実施形態においても、第4実施形態及び第5実施形態と同様に、高周波電流を重畳する方法でも同様の効果が得られる。また、d軸とq軸の両方あるいはq軸だけに電圧を重畳する、あるいは正弦波を重畳する場合でも同様の効果が得られる。
[8.1]第1変形例
第4実施形態乃至第6実施形態の回転センサレス制御装置は、CPUなどの制御装置と、ROM(Read Only Memory)やRAMなどの記憶装置と、HDD、CDドライブ装置などの外部記憶装置と、ディスプレイ装置などの表示装置と、キーボードやマウスなどの入力装置を備えており、通常のコンピュータを利用したハードウェア構成として構成することも可能である。
また第4実施形態乃至第6実施形態の回転センサレス制御装置で実行される制御プログラムは、インストール可能な形式又は実行可能な形式のファイルでCD-ROM、フレキシブルディスク(FD)、CD-R、DVD(Digital Versatile Disk)等のコンピュータで読み取り可能な記録媒体に記録されて提供されるようにしてもよい。
また、第4実施形態乃至第6実施形態の回転センサレス制御装置で実行される制御プログラムを、インターネット等のネットワークに接続されたコンピュータ上に格納し、ネットワーク経由でダウンロードさせることにより提供するように構成しても良い。また、本実施形態の回転センサレス制御装置の制御部で実行される制御プログラムをインターネット等のネットワーク経由で提供または配布するように構成しても良い。
また、第4実施形態乃至第6実施形態の回転センサレス制御装置の制御プログラムを、ROM等に予め組み込んで提供するように構成してもよい。
Claims (15)
- 所定の設定搬送波周波数の搬送波を所定の継続時間の間発生する搬送波発生部と、
前記継続時間をランダムに設定するとともに、互いに異なる複数の搬送波周波数のうちいずれか一の搬送波周波数を前記設定搬送波周波数として設定する設定部と、
前記搬送波発生部において発生された前記搬送波に基づいて、PWM信号を発生するPWM信号発生部と、
前記PWM信号に基づいて電力変換を行って負荷に供給する電力変換部と、
を備えた電力変換装置。 - 前記設定部は、乱数を発生する乱数発生部と、
発生された前記乱数に基づいて、前記継続時間を決定する継続時間決定部と、
を備えた請求項1記載の電力変換装置。 - 前記設定部は、乱数を発生する乱数発生部と、
発生された前記乱数及び所定の設定変移確率値に基づいて、現在の設定搬送波周波数を他の搬送波周波数に変移させるか否かを判定する変移判定部と、を備え、
前記設定部は、前記変移判定部の判定結果が変移させるものであった場合に、前記設定搬送波周波数を他の搬送波周波数に設定するとともに、互いに異なる複数の変移確率値のうちいずれか一の変移確率値を前記設定変移確率値として設定する、
請求項1記載の電力変換装置。 - 前記負荷の機械的共振特性が入力され、当該入力された機械的共振特性に基づいて、前記複数の搬送波周波数及び前記複数の変移確率値のうち少なくともいずれか一方をあらかじめ決定する値決定部を備えた、
請求項3記載の電力変換装置。 - PWM信号に基づいて電力変換を行って負荷に供給する電力変換部を備えた電力変換装置において実行される電力変換装置の制御方法であって、
所定の設定搬送波周波数の搬送波を所定の継続時間の間発生する搬送波発生過程と、
前記継続時間をランダムに設定するとともに、互いに異なる複数の搬送波周波数のうちいずれか一の搬送波周波数を前記設定搬送波周波数として設定する設定過程と、
前記搬送波発生過程において発生された前記搬送波に基づいて、PWM信号を発生する信号発生過程と、
を備えた電力変換装置の制御方法。 - PWM制御における基本波周波数よりも高い周波数である互いに異なる複数の重畳高周波周波数のうちいずれか一の重畳高周波周波数を設定するとともに、設定した重畳高周波周波数を有する電圧あるいは設定した重畳高周波周波数を有する電流の継続時間をランダムに設定する設定部と、
前記重畳高周波周波数の電圧あるいは電流を前記継続時間の間発生する発生部と、
発生された前記重畳高周波周波数を有する電圧を永久磁石同期電動機に印加し、あるいは、発生された前記重畳高周波周波数を有する電流を前記永久磁石同期電動機に供給して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する推定部と、
を備えた回転センサレス制御装置。 - 前記設定部は、乱数を発生する乱数発生部と、
発生された前記乱数に基づいて、前記継続時間を決定する継続時間決定部と、
を備えた請求項6記載の回転センサレス制御装置。 - 前記設定部は、乱数を発生する乱数発生部と、
発生された前記乱数及び所定の設定変移確率値に基づいて、現在の設定重畳高周波周波数を他の重畳高周波周波数に変移させるか否かを判定する変移判定部と、を備え、
前記設定部は、前記変移判定部の判定結果が変移させるものであった場合に、前記設定重畳高周波周波数を他の重畳高周波周波数に設定するとともに、互いに異なる複数の変移確率値のうちいずれか一の変移確率値を前記設定変移確率値として設定する、
請求項6記載の回転センサレス制御装置。 - PWM制御における基本波周波数よりも高い周波数である互いに異なる複数の重畳高周波周波数のうちいずれか一の重畳高周波周波数を設定するとともに、設定した重畳高周波周波数を有する電圧の継続時間をランダムに設定する設定部と、
前記重畳高周波周波数の電圧を前記継続時間の間発生する発生部と、
発生された前記重畳高周波周波数を有する電圧を永久磁石同期電動機に印加して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する推定部と、
前記重畳高周波周波数に応じて重畳高周波周波数の電圧の振幅を決定する高周波電圧指令生成部と、
を備えた回転センサレス制御装置。 - 前記設定部は、二つの重畳高周波周波数を交互に設定する、
請求項7または請求項9記載の回転センサレス制御装置。 - PWM制御における基本波周波数よりも高い周波数である重畳高周波周波数を可変設定可能な第1設定部と、
設定した重畳高周波周波数を有する電圧あるいは設定した重畳高周波周波数を有する電流の継続時間をランダムに設定する第2設定部と、
前記設定部において設定された重畳高周波周波数に応じて振幅を決定する振幅決定部と、
設定された前記重畳高周波周波数及び前記振幅に基づいて重畳高周波電圧指令を生成する指令生成部と、
前記重畳高周波周波数及び前記振幅を有する電圧を永久磁石同期電動機に印加し、あるいは、発生された前記重畳高周波周波数を有する電流を前記永久磁石同期電動機に供給して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する推定部と、
を備えた回転センサレス制御装置。 - 前記設定部は、前記PWM制御における搬送波周波数と、前記重畳高周波周波数を同期させる、
請求項6、請求項9又は請求項11のいずれかに記載の回転センサレス制御装置。 - 永久磁石同期電動機をPWM制御するに際し、回転センサレス制御を行う回転センサレス制御装置において実行される回転センサレス制御装置の制御方法であって、
PWM制御における基本波周波数よりも高い周波数である互いに異なる複数の重畳高周波周波数のうちいずれか一の重畳高周波周波数を設定するとともに、設定した重畳高周波周波数を有する電圧あるいは設定した重畳高周波周波数を有する電流の継続時間をランダムに設定する過程と、
前記重畳高周波周波数の電圧あるいは電流を前記継続時間の間発生する過程と、
発生された前記重畳高周波周波数を有する電圧を永久磁石同期電動機に印加し、あるいは、発生された前記重畳高周波周波数を有する電流を前記永久磁石同期電動機に供給して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する過程と、
を備えた回転センサレス制御装置の制御方法。 - 永久磁石同期電動機をPWM制御するに際し、回転センサレス制御を行う回転センサレス制御装置において実行される回転センサレス制御装置の制御方法であって、
PWM制御における基本波周波数よりも高い周波数である互いに異なる複数の重畳高周波周波数のうちいずれか一の重畳高周波周波数を設定するとともに、設定した重畳高周波周波数を有する電圧の継続時間をランダムに設定する過程と、
前記重畳高周波周波数の電圧を前記継続時間の間発生する過程と、
発生された前記重畳高周波周波数を有する電圧を前記永久磁石同期電動機に印加して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する過程と、
前記重畳高周波周波数の電圧によって発生する高周波電流の振幅が一定となるように制御する過程と、
を備えた回転センサレス制御装置の制御方法。 - PWM制御における基本周波数よりも高い周波数である重畳高周波周波数を可変に設定する重畳高周波周波数設定と、
前記重畳高周波周波数に応じて振幅を決定する重畳高周波周波数振幅決定部と、
前記重畳高周波周波数と前記重畳高周波周波数振幅より重畳高周波電圧指令を生成する高周波電圧指令生成部と、
発生された前記重畳高周波周波数を有する電圧を永久磁石同期電動機に印加して、前記永久磁石同期電動機の回転子磁極位置及び回転速度を推定する推定部と、
を備えた回転センサレス制御装置。
Priority Applications (6)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
KR1020157033042A KR101765407B1 (ko) | 2013-05-27 | 2014-03-14 | 전력 변환 장치 및 전력 변환 장치의 제어 방법 |
KR1020177003963A KR101812458B1 (ko) | 2013-05-27 | 2014-03-14 | 회전 센서리스 제어 장치 및 회전 센서리스 제어 장치의 제어 방법 |
EP14803567.8A EP3007345B1 (en) | 2013-05-27 | 2014-03-14 | Power conversion device |
CN201480030250.8A CN105432010B (zh) | 2013-05-27 | 2014-03-14 | 电力变换装置、电力变换装置的控制方法、无旋转传感器控制装置以及无旋转传感器控制装置的控制方法 |
SG11201509697QA SG11201509697QA (en) | 2013-05-27 | 2014-03-14 | Power conversion device, control method thereof, rotation sensorless control device, and control method thereof |
US14/894,469 US9923447B2 (en) | 2013-05-27 | 2014-03-14 | Power conversion device having improved noise characteristics, and control method thereof |
Applications Claiming Priority (4)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP2013-111321 | 2013-05-27 | ||
JP2013111321A JP6304942B2 (ja) | 2013-05-27 | 2013-05-27 | 電力変換装置、電力変換装置の制御方法及び制御プログラム |
JP2013-132093 | 2013-06-24 | ||
JP2013132093A JP6261889B2 (ja) | 2013-06-24 | 2013-06-24 | 回転センサレス制御装置、回転センサレス制御装置の制御方法及び制御プログラム |
Publications (1)
Publication Number | Publication Date |
---|---|
WO2014192373A1 true WO2014192373A1 (ja) | 2014-12-04 |
Family
ID=51988417
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/JP2014/057010 WO2014192373A1 (ja) | 2013-05-27 | 2014-03-14 | 電力変換装置、電力変換装置の制御方法、回転センサレス制御装置及び回転センサレス制御装置の制御方法 |
Country Status (6)
Country | Link |
---|---|
US (1) | US9923447B2 (ja) |
EP (1) | EP3007345B1 (ja) |
KR (2) | KR101812458B1 (ja) |
CN (1) | CN105432010B (ja) |
SG (1) | SG11201509697QA (ja) |
WO (1) | WO2014192373A1 (ja) |
Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP3051691A1 (en) * | 2015-02-02 | 2016-08-03 | LG Electronics Inc. | Motor driving device and laundry treatment apparatus including the same |
WO2018038111A1 (ja) * | 2016-08-22 | 2018-03-01 | 株式会社 東芝 | インバータ制御装置およびドライブシステム |
CN113098337A (zh) * | 2021-04-09 | 2021-07-09 | 哈尔滨理工大学 | 一种伪随机等差注入pmsm驱动器噪声抑制方法 |
Families Citing this family (14)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP6750364B2 (ja) | 2016-07-22 | 2020-09-02 | 株式会社デンソー | 回転電機の回転角推定装置 |
JP6766538B2 (ja) * | 2016-09-09 | 2020-10-14 | 株式会社デンソー | 駆動装置 |
JP2018046712A (ja) * | 2016-09-16 | 2018-03-22 | 株式会社ジェイテクト | モータ制御装置 |
CN106452028B (zh) * | 2016-09-26 | 2019-03-26 | 中国人民解放军海军工程大学 | 一种三角载波斜率随机分布脉宽调制方法 |
JP6838469B2 (ja) | 2017-04-10 | 2021-03-03 | トヨタ自動車株式会社 | 駆動装置 |
JP6812896B2 (ja) * | 2017-04-28 | 2021-01-13 | 株式会社デンソー | 駆動装置および自動車 |
CN110582929B (zh) * | 2017-05-09 | 2021-03-12 | 三菱电机株式会社 | 电力变换装置 |
US10731907B2 (en) | 2017-06-12 | 2020-08-04 | Lennox Industries, Inc. | Controlling systems with motor drives using pulse width modulation |
DE102017114526A1 (de) * | 2017-06-29 | 2019-01-03 | Hanon Systems | Verfahren zur Ansteuerung von Leistungshalbleitern in einem Inverter |
JP6937708B2 (ja) | 2018-02-21 | 2021-09-22 | 日立Astemo株式会社 | モータ制御装置およびそれを用いる電動車両システム |
DE102018206596A1 (de) * | 2018-04-27 | 2019-12-24 | Siemens Mobility GmbH | Verfahren zum Ansteuern eines Pulswechselrichters, Verwendung, Steuereinheit und stationäres oder mobiles System |
CN111049453A (zh) * | 2018-10-15 | 2020-04-21 | 广东威灵电机制造有限公司 | 转子角速度和转子位置检测方法及设备 |
DE102019200992A1 (de) * | 2019-01-28 | 2020-07-30 | Robert Bosch Gmbh | Verfahren zur Steuerung eines Inverters |
JP6685452B1 (ja) * | 2019-05-16 | 2020-04-22 | 三菱電機株式会社 | 回転電機の制御装置 |
Citations (8)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPH0379959A (ja) | 1989-08-22 | 1991-04-04 | Daikin Ind Ltd | 冷凍装置 |
JPH0614557A (ja) * | 1992-06-23 | 1994-01-21 | Toyo Electric Mfg Co Ltd | Pwmインバータの変調方式 |
JPH11220895A (ja) * | 1998-01-30 | 1999-08-10 | Toshiba Corp | インバータ制御装置 |
JP2002252970A (ja) * | 2001-02-26 | 2002-09-06 | Hitachi Ltd | 電力変換装置 |
JP2004343833A (ja) | 2003-05-13 | 2004-12-02 | Toshiba Corp | モータ制御装置 |
JP2007325406A (ja) * | 2006-05-31 | 2007-12-13 | Toshiba Corp | 鉄道車両用モータ制御装置 |
JP2009303288A (ja) | 2008-06-10 | 2009-12-24 | Toyota Motor Corp | インバータ制御装置 |
JP2012228058A (ja) * | 2011-04-19 | 2012-11-15 | Hitachi Industrial Equipment Systems Co Ltd | 電力変換装置、電動機駆動システム |
Family Cites Families (24)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPH03154965A (ja) | 1989-11-13 | 1991-07-02 | Kawasaki Steel Corp | 搬送装置のスケジューリング方法 |
JP2003324944A (ja) * | 2002-05-08 | 2003-11-14 | Fuji Electric Co Ltd | 電源回路 |
JP4085976B2 (ja) * | 2003-12-25 | 2008-05-14 | 日産自動車株式会社 | インバータの制御装置及び制御方法 |
JP4537802B2 (ja) | 2004-08-24 | 2010-09-08 | 株式会社東芝 | 電力変換装置 |
JP5050395B2 (ja) * | 2006-04-24 | 2012-10-17 | 日産自動車株式会社 | 電力制御装置及び電力制御方法 |
US7639518B2 (en) * | 2006-04-26 | 2009-12-29 | Nissan Motor Co., Ltd. | Device and method for controlling power converting device |
JP5239235B2 (ja) * | 2006-10-13 | 2013-07-17 | 日産自動車株式会社 | 電力変換装置および電力変換方法 |
US7733674B2 (en) * | 2006-11-22 | 2010-06-08 | Nissan Motor Co., Ltd. | Power conversion apparatus for converting direct current to polyphase alternating current |
JP4978429B2 (ja) * | 2007-11-01 | 2012-07-18 | アイシン・エィ・ダブリュ株式会社 | 電動機制御装置,電気自動車およびハイブリッド電気自動車 |
US8537580B2 (en) * | 2008-01-18 | 2013-09-17 | Mitsubishi Electric Corporation | Controller of power converter |
US8736220B2 (en) * | 2008-04-28 | 2014-05-27 | Daikin Industries, Ltd. | Inverter control device and power conversion device |
JP4497235B2 (ja) * | 2008-08-08 | 2010-07-07 | トヨタ自動車株式会社 | 交流電動機の制御装置および制御方法 |
JP5472327B2 (ja) * | 2010-02-03 | 2014-04-16 | トヨタ自動車株式会社 | 回転電機の制御装置および回転電機の制御方法 |
EP2566040A1 (en) * | 2010-04-28 | 2013-03-06 | Toyota Jidosha Kabushiki Kaisha | Device for controlling electric motor |
JP5059163B2 (ja) | 2010-05-10 | 2012-10-24 | 株式会社東芝 | 電力変換装置 |
EP2579451B1 (en) * | 2010-05-27 | 2016-03-30 | Toyota Jidosha Kabushiki Kaisha | Control apparatus and control method for motor |
JP5549384B2 (ja) * | 2010-06-03 | 2014-07-16 | 日産自動車株式会社 | 電動機の制御装置および電動機制御システム |
JP5398873B2 (ja) | 2012-04-27 | 2014-01-29 | 株式会社東芝 | 電力変換装置 |
JP6158115B2 (ja) * | 2013-02-21 | 2017-07-05 | 株式会社東芝 | 磁石磁束量推定装置、異常減磁判定装置、同期電動機駆動装置および電動車両 |
US9270168B2 (en) * | 2013-03-15 | 2016-02-23 | Hamilton Sundstrand Corporation | Electromagnetic interference (EMI) reduction in multi-level power converter |
US20140268948A1 (en) * | 2013-03-15 | 2014-09-18 | Hamilton Sundstrand Corporation | Electromagnetic interference (emi) reduction in interleaved power converter |
JP6184753B2 (ja) * | 2013-05-30 | 2017-08-23 | コベルコ建機株式会社 | 電動機駆動用インバータ装置 |
JP6086085B2 (ja) * | 2014-03-18 | 2017-03-01 | 株式会社安川電機 | 電力変換装置、発電システム、電力変換装置の制御装置および電力変換装置の制御方法 |
JP6295809B2 (ja) * | 2014-04-28 | 2018-03-20 | 株式会社安川電機 | 電力変換装置、制御装置および電力変換装置の制御方法 |
-
2014
- 2014-03-14 US US14/894,469 patent/US9923447B2/en active Active
- 2014-03-14 KR KR1020177003963A patent/KR101812458B1/ko active IP Right Grant
- 2014-03-14 SG SG11201509697QA patent/SG11201509697QA/en unknown
- 2014-03-14 EP EP14803567.8A patent/EP3007345B1/en active Active
- 2014-03-14 KR KR1020157033042A patent/KR101765407B1/ko active IP Right Grant
- 2014-03-14 WO PCT/JP2014/057010 patent/WO2014192373A1/ja active Application Filing
- 2014-03-14 CN CN201480030250.8A patent/CN105432010B/zh active Active
Patent Citations (9)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPH0379959A (ja) | 1989-08-22 | 1991-04-04 | Daikin Ind Ltd | 冷凍装置 |
JPH0614557A (ja) * | 1992-06-23 | 1994-01-21 | Toyo Electric Mfg Co Ltd | Pwmインバータの変調方式 |
JPH11220895A (ja) * | 1998-01-30 | 1999-08-10 | Toshiba Corp | インバータ制御装置 |
JP3154965B2 (ja) | 1998-01-30 | 2001-04-09 | 株式会社東芝 | インバータ制御装置 |
JP2002252970A (ja) * | 2001-02-26 | 2002-09-06 | Hitachi Ltd | 電力変換装置 |
JP2004343833A (ja) | 2003-05-13 | 2004-12-02 | Toshiba Corp | モータ制御装置 |
JP2007325406A (ja) * | 2006-05-31 | 2007-12-13 | Toshiba Corp | 鉄道車両用モータ制御装置 |
JP2009303288A (ja) | 2008-06-10 | 2009-12-24 | Toyota Motor Corp | インバータ制御装置 |
JP2012228058A (ja) * | 2011-04-19 | 2012-11-15 | Hitachi Industrial Equipment Systems Co Ltd | 電力変換装置、電動機駆動システム |
Non-Patent Citations (1)
Title |
---|
See also references of EP3007345A4 |
Cited By (9)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP3051691A1 (en) * | 2015-02-02 | 2016-08-03 | LG Electronics Inc. | Motor driving device and laundry treatment apparatus including the same |
US20160226426A1 (en) * | 2015-02-02 | 2016-08-04 | Lg Electronics Inc. | Motor driving device and laundry treatment apparatus including the same |
CN105846742A (zh) * | 2015-02-02 | 2016-08-10 | Lg电子株式会社 | 电机驱动装置及具备该电机驱动装置的洗涤物处理设备 |
US9973131B2 (en) | 2015-02-02 | 2018-05-15 | Lg Electronics Inc. | Motor driving device and laundry treatment apparatus including the same |
CN105846742B (zh) * | 2015-02-02 | 2018-08-24 | Lg电子株式会社 | 电机驱动装置及具备该电机驱动装置的洗涤物处理设备 |
WO2018038111A1 (ja) * | 2016-08-22 | 2018-03-01 | 株式会社 東芝 | インバータ制御装置およびドライブシステム |
TWI668953B (zh) * | 2016-08-22 | 2019-08-11 | 日商東芝股份有限公司 | Inverter control device and drive system |
US10637381B2 (en) | 2016-08-22 | 2020-04-28 | Kabushiki Kaisha Toshiba | Inverter control device and drive system |
CN113098337A (zh) * | 2021-04-09 | 2021-07-09 | 哈尔滨理工大学 | 一种伪随机等差注入pmsm驱动器噪声抑制方法 |
Also Published As
Publication number | Publication date |
---|---|
KR101765407B1 (ko) | 2017-08-07 |
KR101812458B1 (ko) | 2017-12-26 |
KR20170019490A (ko) | 2017-02-21 |
CN105432010B (zh) | 2018-01-09 |
EP3007345A4 (en) | 2016-08-17 |
SG11201509697QA (en) | 2015-12-30 |
US20160111951A1 (en) | 2016-04-21 |
US9923447B2 (en) | 2018-03-20 |
EP3007345B1 (en) | 2022-10-05 |
CN105432010A (zh) | 2016-03-23 |
EP3007345A1 (en) | 2016-04-13 |
KR20150143847A (ko) | 2015-12-23 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
WO2014192373A1 (ja) | 電力変換装置、電力変換装置の制御方法、回転センサレス制御装置及び回転センサレス制御装置の制御方法 | |
KR101928437B1 (ko) | 전동기를 구동하는 인버터의 출력전압을 제어하는 방법 및 장치. | |
WO2011040159A1 (ja) | 電動機駆動装置の制御装置 | |
TWI654827B (zh) | 換流器控制裝置及馬達驅動系統 | |
US9431951B2 (en) | Direct torque control motor controller with transient current limiter | |
JP5914777B2 (ja) | 圧縮機のトルクの自動補正方法、その装置及び圧縮機並びにその制御方法 | |
JP2004343833A (ja) | モータ制御装置 | |
WO2013128799A1 (ja) | 電動モータの制御装置 | |
JP2013223352A (ja) | モータ制御装置及びモータ制御システム | |
WO2016017304A1 (ja) | 電力変換装置 | |
KR20200124787A (ko) | 모터 구동을 위한 인버터 제어 장치 및 방법 | |
JP2009100613A (ja) | Pwmインバータの制御装置 | |
JP6351652B2 (ja) | 電力変換器制御装置 | |
JP6261889B2 (ja) | 回転センサレス制御装置、回転センサレス制御装置の制御方法及び制御プログラム | |
US10778134B2 (en) | Apparatus and method for controlling inverter for driving motor | |
Tang et al. | Compensation of dead-time effects based on revised repetitive controller for PMSM drives | |
JP2010063221A (ja) | モータ制御装置 | |
JP6304942B2 (ja) | 電力変換装置、電力変換装置の制御方法及び制御プログラム | |
JP5851662B1 (ja) | 交流回転機の制御装置 | |
CN112567620B (zh) | 逆变装置 | |
JP2011066974A (ja) | 回転機の制御装置 | |
KR100933393B1 (ko) | 유도 전동기의 직접 토크 제어 장치 및 방법 | |
WO2023067798A1 (ja) | モータ制御装置、およびモータ制御方法 | |
JP7077038B2 (ja) | 同期モータの駆動装置 | |
JP2018082544A (ja) | モータの制御方法、及び、モータ制御装置 |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
WWE | Wipo information: entry into national phase |
Ref document number: 201480030250.8 Country of ref document: CN |
|
121 | Ep: the epo has been informed by wipo that ep was designated in this application |
Ref document number: 14803567 Country of ref document: EP Kind code of ref document: A1 |
|
ENP | Entry into the national phase |
Ref document number: 20157033042 Country of ref document: KR Kind code of ref document: A |
|
WWE | Wipo information: entry into national phase |
Ref document number: 2014803567 Country of ref document: EP |
|
NENP | Non-entry into the national phase |
Ref country code: DE |
|
WWE | Wipo information: entry into national phase |
Ref document number: 14894469 Country of ref document: US |