WO2003021826A1  Method for measuring and compensating gain and phase imbalances in quadrature modulators  Google Patents
Method for measuring and compensating gain and phase imbalances in quadrature modulators Download PDFInfo
 Publication number
 WO2003021826A1 WO2003021826A1 PCT/US2002/027192 US0227192W WO03021826A1 WO 2003021826 A1 WO2003021826 A1 WO 2003021826A1 US 0227192 W US0227192 W US 0227192W WO 03021826 A1 WO03021826 A1 WO 03021826A1
 Authority
 WO
 WIPO (PCT)
 Prior art keywords
 phase
 frequency
 gain
 input
 signal
 Prior art date
Links
Classifications

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
 H04L27/00—Modulatedcarrier systems
 H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
 H04L27/34—Amplitude and phasemodulated carrier systems, e.g. quadratureamplitude modulated carrier systems
 H04L27/36—Modulator circuits; Transmitter circuits
 H04L27/362—Modulation using more than one carrier, e.g. with quadrature carriers, separately amplitude modulated
 H04L27/364—Arrangements for overcoming imperfections in the modulator, e.g. quadrature error or unbalanced I and Q levels

 H—ELECTRICITY
 H03—BASIC ELECTRONIC CIRCUITRY
 H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
 H03D7/00—Transference of modulation from one carrier to another, e.g. frequencychanging
 H03D7/16—Multiplefrequencychanging
 H03D7/165—Multiplefrequencychanging at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature
 H03D7/166—Multiplefrequencychanging at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature using two or more quadrature frequency translation stages

 H—ELECTRICITY
 H03—BASIC ELECTRONIC CIRCUITRY
 H03F—AMPLIFIERS
 H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
 H03F1/32—Modifications of amplifiers to reduce nonlinear distortion
 H03F1/3241—Modifications of amplifiers to reduce nonlinear distortion using predistortion circuits

 H—ELECTRICITY
 H03—BASIC ELECTRONIC CIRCUITRY
 H03F—AMPLIFIERS
 H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
 H03F1/32—Modifications of amplifiers to reduce nonlinear distortion
 H03F1/3241—Modifications of amplifiers to reduce nonlinear distortion using predistortion circuits
 H03F1/3247—Modifications of amplifiers to reduce nonlinear distortion using predistortion circuits using feedback acting on predistortion circuits

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04B—TRANSMISSION
 H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00  H04B13/00; Details of transmission systems not characterised by the medium used for transmission
 H04B1/06—Receivers
 H04B1/16—Circuits
 H04B1/30—Circuits for homodyne or synchrodyne receivers

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04B—TRANSMISSION
 H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00  H04B13/00; Details of transmission systems not characterised by the medium used for transmission
 H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
 H04B1/40—Circuits
 H04B1/403—Circuits using the same oscillator for generating both the transmitter frequency and the receiver local oscillator frequency

 H—ELECTRICITY
 H03—BASIC ELECTRONIC CIRCUITRY
 H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
 H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
 H03D2200/0041—Functional aspects of demodulators
 H03D2200/0084—Lowering the supply voltage and saving power

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04B—TRANSMISSION
 H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00  H04B13/00; Details of transmission systems not characterised by the medium used for transmission
 H04B1/02—Transmitters
 H04B1/04—Circuits
 H04B2001/0408—Circuits with power amplifiers
 H04B2001/0425—Circuits with power amplifiers with linearisation using predistortion

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
 H04L27/00—Modulatedcarrier systems
 H04L27/0014—Carrier regulation
 H04L2027/0016—Stabilisation of local oscillators
Abstract
Description
METHOD FOR MEASURING AND COMPENSATING GAIN AND PHASE IMBALANCES IN QUADRATURE MODULATORS
FIELD AND BACKGROUND OP THE INVENTION The present invention relates to measurement and calibration of quadrature modulators as used in transmitters for digital communication. Quadrature modulators (sometimes referred to as IQ modulators), in particular those used in RFIC (Radio Frequency Integrated Circuits) operating at high frequencies in the GHz range, may incur significant gain and phase imbalances in the baseband path, as well as orthogonality phase imbalance in the local oscillator path. The effect of these impairments, generally denoted as gain and phase imbalances (or "IQ" imbalance), is distortion of the transmitted signal, which translates to reduced or even unacceptable performance.
In many cases it is not practical, and sometimes even not feasible, to design and build quadrature modulators with sufficiently low values of gain and phase imbalances. However, if the quadrature modulator gain and phase imbalances can be estimated, there exist known methods to compensate or equivalently predistort the transmitted input signal, thus canceling their effect. Typically, the values of gain and phase imbalances are not fixed, and may change as a function of operating conditions, aging, etc., thus requiring a simple and efficient builtin method to perform these measurements and evaluate the compensation terms, on a timely basis, while the quadrature modulator is installed and operational.
SUMMARY OF THE INVENTION
The method applies a sequence of test signals at the input of the quadrature modulator, with the resulting output coupled to a detector and processed in order to evaluate estimates of modulator gain and phase imbalances. In normal transmit operation these estimates are used to compensate, or equivalently predistort, the transmitted signal, and as such to cancel the effects of the gain and phase imbalances on the transmitted signal.
The method proposed herein uses a sequence of test input signals, which, combined with the operation of the detector circuit, provides a simple and accurate evaluation of the modulator imbalance terms. In a preferred embodiment, the test signals are sine waveforms with specific amplitude and phase, resulting in a specific signal at the quadrature modulator output. The quadrature modulator gain and phase imbalances distort this signal as compared with an ideal quadrature modulator. This distortion is equivalent to the generation of additional spectral components, whose frequency, amplitude, and phase are a function of the modulator imbalance values. A detector coupled to the modulator output performs a nonlinear operation, which generates intermodulation products between the various spectral components. It is shown below that the amplitude of the 2fj component, where fj is the frequency of the input sine waveform, is proportional to the modulator gain and phase imbalance terms, and that by controlling the amplitude and phase of the test signals, it is possible to derive these terms from the amplitude of the 2f component. According to the present invention, there is provided a method for calibrating a quadrature modulator having an I input and a Q input for inputting baseband I(t) and Q(t) signals, the modulator used to transmit quadrature modulated signals, comprising: a) estimating in sequence values of modulator gain imbalances and of modulator phase imbalances while the modulator is operational, the estimating including inputting at least one test signal at a baseband frequency f] to the modulator to generate detected output signals having a term at frequency 2fj, first in the gain imbalance estimation, then in the phase imbalance estimation, and computing the gain and phase imbalances based on the 2fi term of the detected output signals, and b) in normal transmit operation, compensating first for the gain and then for the phase imbalances to obtain an essentially ideal quadrature modulated signal, the compensating including inputting a transmission signal to the modulator, and based on the computed gain and phase imbalances, applying a predistortion transformation on the input transmission signal.
According to one feature in the method of the present invention, the computing of the gain imbalance is based on an iterative operation that includes modifying the test signals and repeating the measurement of the detected output signal terms at frequency 2f; until reaching a reference value of the detected output signal.
According to another feature in the method of the present invention, the method further includes: for the gain imbalance, inputting in a first step a cosine waveform at the I input, and a zero waveform at the Q input, and in a second step a zero waveform at the I input and the same cosine waveform at the Q input, and, for the phase imbalance, inputting in a first step at the I and Q inputs two sine waveforms of equal amplitude and frequency but shifted by 90° + Q_{\} as given by eqn. 17, and, optionally, inputting in a second step two sine waveforms of equal amplitude and frequency but shifted by +90° + θ_{2} as given by eqn. 21.
According to yet another feature in the method of the present invention, the computing of the phase imbalances includes computing separately a baseband phase imbalance Δθ and a local oscillator orthogonality phase imbalance Δφ, using the inputting of test signals and an iterative operation that includes modifying the test signals by varying the phases θι and θ_{2} and repeating the measurement of the detector output signals until effectively cancelling the signal tenns at frequency 2fj .
According to yet another feature in the method of the present invention, the reference value mentioned above is the result of the first measurement of the detected output amplitude at frequency 2fj as generated by a first test signal.
According to yet another feature in the method of the present invention, the inputting of at least one test signal at a baseband frequency f, includes inputting a plurality N of test signals, each at a different baseband frequency fj(N), and the applying of a predistortion transformation on the input transmission signal includes applying a frequencydependent predistortion transformation.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention is herein described, by way of example only, with reference to the accompanying drawings, wherein: FIG. 1 shows a top level block diagram of the system including a quadrature modulator in which the method of the present invention is applied;
FIG. 2 shows a general block diagram illustrating the main steps of the method; FIG. 3 shows the signal flow through the quadrature modulator; FIG. 4 shows a gain imbalance measurement flow diagram; FIG. 5 shows a phase imbalance measurement flow diagram;
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Method overview
In FIG. 1, a quadrature modulator 12 receives as input test signals from a test signal generator 14, specifically sine waveforms with specific amplitude and phase, resulting in a specific output signal 16 at the quadrature modulator output. The quadrature modulator amplitude and phase imbalances distort this signal as compared with an ideal quadrature modulator. This distortion is equivalent to the generation of additional spectral components whose frequency, amplitude, and phase are functions of the modulator imbalance values. A sample of the output signal is coupled through an RF coupler 22 into a detector 18, which performs a nonlinear operation that generates intermodulation products between the various spectral components of the output signal, including a component at 2fj, where fj is the frequency of the input sine waveform. A detector output 20 is processed in a followup bandpass filter 24, which extracts the component at 2fj, while rejecting the unwanted components. A filter output 30 is then input to a processor 32, preferably a digital processor, which computes estimates 34 of the gain and phase imbalance values. In other embodiments, filter 24 can be implemented as part of processor 32.
In normal transmit operation, an input signal 40 is passed through a predistortion device 42 that performs a predistortion transformation on the signal, which compensates the imbalance effects of the quadrature modulator. Computed estimates 34 are used to program predistortion device 42. Methods to design and program compensation devices such as device 42 are well known in the art.
FIG. 2 shows a general block diagram of the method of the present invention. The measurement and calibration phase consists of the following steps, which are performed in sequence in order to evaluate first the gain imbalance terms and then repeated in order to evaluate the phase imbalance terms.
Signal generator 14 in FIG. 1 in a generating step 102 provides a test input signal. The test signals are then transmitted through quadrature modulator 12 to produce an output signal 16 that includes gain and phase distortion in a test signal transmission step 104. Detector 18 then detects signal 16 in a detection step 106 in which the detector output is also measured. Processor 32 then performs a gain and phase imbalance computing step 108, using the detector measurement. First, an estimate of gain imbalance is computed in an iterative way as shown by the arrow 112, by modifying the test input and repeating steps 102, 104 and 106 until the measured output reaches a predetermined (reference) value. Finally, a programming step 110 is carried out in processor 32, which programs the predistortion device to compensate for the gain imbalance effect. A similar sequence is carried out for compensating the phase imbalance effect. Following the measurement and calibration stage, the transmitter is switched to a "normal" transmit operational mode. As shown in FIG. 2, the following steps are then performed: a predistortion transformation 152 is applied on a transmit signal 150 by predistortion device 42. The predistorted signal is then transmitted through quadrature modulator 12 in a transmitting step 154. The predistorted signal incurs existing gain and phase imbalances, however due to the predistortion, the net effect is almost an "ideal" quadrature modulated signal.
Quadrature modulator
To better understand the method, a short discussion of the quadrature modulator operation and the effect of gain and phase imbalances is presented below.
An ideal quadrature modulator implements the following mathematical operation on a pair of input signals 1(f) and Q(t):
Y(t) = I(t)sin(α>Lot) + Q(t)cos(ω_{LO}t) (1) where I(t) and Q(t) are the baseband input signals and GO_{LO}  πf o is the local oscillator (LO) radial frequency (rad/sec). The operation of a "practical" quadrature modulator that experiences gain and phase imbalances is shown in FIG. 3. The modulator has two input ports, one denoted I, or "inphase" port 50, and the other Q or "quadrature" port 60. The corresponding baseband input signals incur gain and phase imbalances. Without any loss of generality, we can represent all imbalance effects as occurring in the path of the I signal. A block 52 represents the relative phase shift Δθ between the I and Q paths, while a block 54 represents the amplitude gain imbalance ratio 1+ε between the I and Q paths. The values ε and Δθ may be frequency dependent, i.e. function of input signal frequency fj.
The I signal (distorted due to gain and phase imbalances) is input to the baseband input of a first mixer 70 while a first local oscillator signal 72 at frequency f o is input to the first mixer LO port. As is well known, a mixer such as mixer 70 effectively performs a multiplication between both input signals, thus frequency upconverting the input baseband signal. The Q signal is input to a second mixer 74, with a second local oscillator signal 76 also at frequency f o at the second mixer LO port, thus frequency upconverting the Q input baseband signal. The phase relation between the two upconverted signals is dependent on the relative phase between the two local oscillator signals. Local oscillator signals 72 and 76 are generated from a common input signal 78 by passing the input signal through a phase splitter device 80 having the property that its two outputs are 90° phase shifted (orthogonal) one with respect to the other. A phase error block 82 represents a phase error Δφ of this shifting, or, equivalently, an orthogonality phase imbalance Δφ between the two local oscillator signals. Here too, without any loss of generality, we can assume the orthogonality phase error is on the I path. A combining network 90 adds the output of the two mixers into quadrature modulator output 16. Below, the following notation is used:
The quadrature modulator output, including gain and phase (both baseband and orthogonality) imbalance is:
Y(t) = (1+ε )I'(t, ΔΘ)sin(ω_{LO}t+ Δφ) + Q(t)cos(ω_{LO}t) (2)
and I'(t, Δθ) represents the phase shifted (by Δθ) transformation of the input I(t). When we deal with sine/cosine waveforms, then for 1(f) = Acos(ωjt + θ) we have F(t, Δθ) = Acos(cθit + θ+ Δθ). So:
Q(t) = Bcos(cθif)
and
Y(t) = (1+ε )Acos(ωjt + 0+ ΔΘ)sin(ω_{LO}t+ Δφ) + Bcos(ωit)cos(ω_{LO}t) (3)
It is well known that this modulated waveform can be put in a form:
Y(t) = S_{L}cos[(oo_{L}oG)i)t+ ψ_{L}) + Sucos[(ω_{LO}+ωi)t+ ψu) (4)
showing that the quadrature modulator output for I and Q sine like inputs consists of two sideband carriers, one below the LO frequency at fLo L and the other above it at fLo+fi The amplitude and phase of these sidebands is related to the input amplitude and phase, as well as to the values of modulator imbalance. As shown below in the detailed method derivation, we preferably select specific values for the I and Q input amplitude and phase, in order to derive a simple dependency on the imbalance terms. Meanwhile, let us look at an interesting example:
Example: 1(f), Q(t) sine/cosine waveforms with equal amplitude and frequency, but shifted by 90 degrees one with respect to the other, e.g.:
I(t) = Acos(<Bit 90°)  Asin(coif) Q(t) = Acos(cdif)
Ignoring the modulator imbalance we get:
Y(t) = Asin(ωjf)sin(cθLot) + Acos(ω;t)cos(ωLot) = Acos[(ωLo ω;)t] (5)
i.e. the output contains a single sideband carrier at frequency fLo  fj = (G>L_{O}  ω;)/2π. Now, for the same sine inputs waveforms, we compute the effect of gain and phase imbalances:
Y(t) = A(l+ε)sin(ωjt + ΔΘ)sin(ωLot + Δφ) + Acos(ωit)cos(ω_{L}ot) (6)
By standard trigonometric manipulation we can show that:
Y(f) = S_{L}cos[(ω_{L}oωi)t+ ψ_{L}) + Sucos[(ω_{L}o+ωi)t+ ψu) (7)
where:
S_{L} = A k _{+ e)∞8}2 £ ._{) + ε}> ,4 _{^} A(\ + εl2) (8)
S_{u} = AMI + ε) sin^{2} ^ ^{+ AΘ}) + _{ε} ^{2} 14 =~ A[ε/2 + (Aφ + AΘ)/2] (9)
and where the approximations hold for sufficiently small values of the imbalance terms. This expression shows that with gain and phase imbalances, we get a small component at the image frequency f o + i with amplitude proportional to the imbalance terms (in addition to the main term at fLo  fi whose amplitude is only slightly changed due to the gain imbalance).
Detector The output signal (16, FIG. 1) of the quadrature modulator is coupled via a coupler (22 in FIG. 1) into a detector (18 in FIG. 1). The detector performs a nonlinear transformation on its input (signal 16), which can be approximated (square law detector) by:
where Vj_{n} is the input, V_{out} is the output and K_{t} is a proportionality factor. It is well known that when input Vj_{n} contains more than one carrier, the output will contain intermodulation terms. Proper filtering (filter 24 of FIG. 1) allows selection of a desired term. For I and Q sine inputs we get:
V_{in} = K_{2}Y(t) = K_{2}[S_{L}cos[(a>_{L}ocoi)t+ ψ_{L}) + Sucos[(ω_{L}o+cθi)t+ _{Ψu})] (10)
The filter extracts the term at 2f; while rejecting the unwanted terms. That is, while the modulator output has components at ^o  fi and fLo + fi, the detected output has components at 2fj due to a nonlinear effect. Then, the only relevant term (after filtering), Z, is a beat component between the two subcarriers at a frequency equal to the frequency difference between the two subcarriers, or equivalently at twice the baseband input frequency, and with an amplitude So proportional to the product of their amplitudes, i.e.:
Z = S_{D}cos(2ωit + ψ) = kSuS cos(2ωit + ψ) (11)
where k is a proportionality factor and ψ is a phase term (of no interest). Substituting in the above equation the values for the example above (eqn. 8 and 9), So is proportional to the imbalance terms, i.e. S_{D} ~= kA^{2}(l+ ε/2 )[ε/2 + (ΔΘ+Δφ)/2] — K[ε/2 + (ΔΘ+Δφ)/2] (12)
and where the approximations hold for sufficiently small imbalance terms. The nonlinear transformation function of a "practical" detector may deviate from the above simple function (square law). As it will be shown in the detailed derivation for the gain and phase imbalance measurements, the proposed method is insensitive to the knowledge of the exact description of the detector transformation function.
Method Implementation The method of the present invention includes two main stages:
Stage I: evaluate the gain imbalance term ε and apply the resulting compensation.
Stage II: evaluate the phase imbalance terms Δθ and Δφ and apply the resulting compensation.
Stage I  Gain Imbalance: FIG. 4 shows the signal flow for the gain imbalance measurement. To evaluate the gain imbalance between the I and Q baseband paths, we transmit in step 1 (as shown in more detail below) a test signal on I input port (henceforth "I input") 50 with a zero signal on a Q input port (henceforth "Q input") 60, and measure the resulting (first) signal amplitude at the filtered detector 18 output (this first amplitude will be used as a reference value in the next step). The experiment is repeated with the inputs interchanged ("step 2", see below), that is a test signal on Q input 60 and zero signal on I input 50, measuring the resulting (second) signal amplitude at the detector output. The amplitude of the Q input test signal (step 2) is varied (up and down in small incremental steps) until the resulting signal amplitude at the detector output equals the reference value of step 1. This method enables the measurement of gain imbalance independent of the amount of phase imbalance. The gain imbalance is corrected by proper scaling of the I or Q inputs.
Stage I  detailed procedure
Step 1:
Transmit the following signals: L(t) = Acos(cθif) (13) Ql(t) = o
Then
Yι(t) = A(l+ε)cos(ωit + ΔΘ)sin(ω_{L}ot + Δφ)
The signal spectrum of Y^t) consists of two subcarriers at fLofi HO and fLo+fj H , with almost equal amplitudes (equal when gain balance). The filtered detector output (i.e. the term at frequency 2fj) equals to:
Zι(t) = kA^{2}(l+ ε)^{2}cos(2ωit+ψ (14)
where k is a proportionality factor and ψj a phase term of no interest. Subscript "1" refers to parameters of step 1.
Step 2: Repeat the experiment while interchanging between I and Q test signals, that is transmit:
l_{2}(t) = 0 (15)
Q_{2}(t) = A(l+Δ)cos(ωit) where Δ is a control variable of the Q input amplitude. Then: Y_{2}(t) = A(l+Δ)cos(ωit)cos(ω_{LO}t)
It is easily shown that the filtered detector output (i.e. the term at frequency 2fi) equals
Z_{2}(t) = kA^{2} (l+Δ)^{2}cos(2ωit+ψ_{2}) (16)
where k is a proportionality factor and ψ_{2} a phase term of no interest. Subscript "2" refers to parameters of step 2. Under the assumption of a squarelaw detector and with Δ=0, the gain imbalance can be easily derived from the ratio of the amplitudes of Z_{\} and Z_{2} (equations 14 and 16) as amp(Z_{1})/amp(Z_{2}) = (1+ ε)^{2}. Instead of computing this ratio, we use the measured amplitude of Z_{\} (of step 1) as a reference value, and iteratively modify the input amplitude in step 2 (up and down in small incremental steps via the control variable Δ) until the resulting amplitude of Z equals the reference value. Let Λ be the value of Δ for which this equality occurs. Then:
ε = Δ
The equalization of the amplitudes via this iterative procedure enables solving for the gain imbalance term without any assumption on the exact form of the nonlinear transformation function of a "practical" detector. Following the measurement and evaluation of the gain imbalance term, the signals transmitted on the I and Q ports are properly scaled to compensate for this effect.
Stage II  Phase Imbalance: FIG. 5 shows the signal flow for the phase imbalance measurement. To evaluate the phase imbalance between the I and Q baseband paths, as well as the orthogonality phase imbalance in the LO path, we transmit in step 1 a sine test signal on I input 50 with a cosine (i.e. 90 phase relation) signal on Q input 60, and measure the resulting (first) signal amplitude at the filtered detector 18 output. The input test signal is such that ideally (with no phase imbalance) we should get a single sideband carrier 210. Due to the phase imbalance, we get also a small sideband component 212 at the image frequency. After detector 18, we get a component at frequency 2f,, whose amplitude is proportional to the phase imbalance. The procedure is repeated in step 2 with new test inputs 50 and 60 selected such that the phase relation between inputs I and Q is (nominally) +90°. In this case we get a a main sideband carrier 222 and a small sideband component 220 at the image frequency, however their frequency positions are reversed with respect to step 1. We measure the resulting (second) signal amplitude at the detector output. Based on the two measurements we could solve for both baseband phase imbalance Δθ, and orthogonality (local oscillator) phase imbalance Δφ. However, this would require apriori calibration of the detector and knowledge of its nonlinear transformation function. In the detailed description below we prefer a variation of the above procedure (suitable for "practical" detectors) where, instead of measuring the 2f; signal amplitude at detector output (requiring calibrated measurement), we shift the Q input with respect to the I input by a (known) control phase θ till we cancel the 2fi term. The θ value for which cancellation occurs is used to solve for Δθ and Δφ. We assume that gain imbalance measurements have been performed, and that the signals transmitted on the I and Q ports are properly scaled. However, phase imbalance measurements can also be performed while there is still a small residual gain imbalance.
Stage II  Detailed procedure
Step 1
Transmit two sine waveforms with equal amplitude and frequency, but phaseshifted with respect to the other by 90°  Θi where θi is a control (tuning) phase variable:
Iι(t) = Acos(ω_{i}t 90^{0}+ θ_{1}) = Asin(ω_{i}t +θ_{1}) (17)
Assuming gain balance (equivalently, gain imbalance has been precompensated) we get
Yι(t) = Asin(oθit + θι + ΔΘ)sin(ω_{L}ot + Δφ) + Acos(ωit)cos(ω_{L}ot) (18)
By standard trigonometric manipulation we can show that:
Yι(t) = S_{L1}cos[(ω_{LO}ωi)t+ ψ_{L1}) + S_{ul}cos[(ω_{L}o+ω_{i})t+ ψ_{πl}) (19)
where:
S_{L}i = Acos[(θ_{!}+ΔΘΔφ)/2] = ~ A
Suι= Asin [(θι+ΔΘ+Δφ) 2] =~ A(θ!+Δθ+Δφ)/2
and the filtered detector output (i.e. the term at frequency 2fj) equals:
Zι(t) = kA^{2} sin [(θ_{1}+ΔΘ+Δφ)/2]cos(2ω_{i}t+ψ_{1}) (20) where k is a proportionality factor and ψ i is a phase term of no interest. Let β be the value of θ_{\} which cancels the beat component Zi.
^ = (Δ# + Δζø)
β is evaluated by varying θι in small steps (within a range) till we cancel the beat component Zi. Note that in this step we found the combined baseband Δθ and local oscillator Δφ phase imbalance. For some quadrature modulators, Δθ is negligible, in which case Aφ =~ (l)θ_{1 .} However when this assumption is not valid, an additional step is required to solve individually for Δθ and Δφ, as explained below in step 2. When the measurement is noisy or there is a residual gain imbalance, β is the value that minimizes (not cancels) Z_{\}.
Step 2:
We repeat the experiment but use the inputs:
I_{2}(t) = Acos(ω_{f}t +90° + θ_{2}) = Asin(ωjt +θ_{2}) (21)
where θ is a control (tuning) phase variable.
Y_{2}(t) = Asin(ωjt + θ_{2} + ΔΘ)sin(α)Lot + Δφ) + Acos(ωit)cos(ω_{L}ot) (22)
By standard trigonometric manipulation we get:
Y_{2}(t) = S_{L2}cos[(co_{L}ocoi)t+ ψ_{L2}) + Su_{2}cos[(ω_{L}o+ωi)t+ ψ_{U2}) (23)
where:
S_{L2} = Asin [(θ_{2}+ΔΘΔφ)/2] =~ A(θ_{2}+ΔΘΔφ)/2 Su_{2}= Acos[(θ_{2}+ΔΘ+Δφ)/2] = ~ A
Ψ _{2} and ψu_{2} are phase terms of no interest. As expected, this signal has its large component at frequency f o +fi while in Step 1 we got it at fLo fi Similar to Step 1, the filtered detector output (i.e. the term at frequency 2fj) equals:
Z_{2}(t) = kA^{2} sin [(θ_{2}+ΔΘΔφ)/2]cos(2ωit+ ψ_{2}) (24)
where k is a proportionality factor and ψ_{2} a phase term of no interest. Let β. be the value of θ_{2} which cancels the beat component Z_{2}. θ_{2} = Δ< + Δζ2>
Solving the two equations results in: Aβ = (β[ + T_{2})l2 Δ^ = (f^ + έ^)/2 Cancellation of the beat components Zi and Z_{2} via this iterative procedure enables solving for the phase imbalance terms without any assumption on the knowledge of the exact form of the nonlinear transformation function of the detector. The resulting phase imbalance terms are used to generate a transformation on the input signals thus compensating for the quadrature modulator baseband and orthogonality phase imbalance. The compensation, equivalently predistortion methods are known and not repeated here.
In summary, the proposed method uses in its preferred embodiment test input signals with controlled amplitude and phase. It is well known that the gain and phase imbalances reflect on the spectral contents of the output, requiring a complex narrow band receiver to extract this information. However, in an innovative way and in contrast with prior art techniques, the proposed method uses a sequence of test signals at baseband frequency f; together with a detector to analyze the amplitude of the detector output at frequency 2£. This output term is proportional to the gain and phase imbalances and, by controlling the test signals it is possible to compute an estimate of the gain and phase imbalances. The test inputs are selected such that the value of the term at frequency 2f; has sufficient immunity from measurement and detector generated noise, thus generating accurate results. In addition, the gain and phase imbalance measurements can be performed at several values N of the input frequency fj, thus mapping their frequency dependence. For single carrier modulation, we typically perform compensation by using an average (or midfrequency) value of the gain and phase imbalances, however, for multicarrier modulation, such as Orthogonal Frequency Division Multiplexing (OFDM), frequency dependent compensation based on the 2fj component (N) can be applied individually for each carrier. For clarity, the sequence of measurement and compensation at each frequency ft is carried out as explained above.
All publications, patents and patent applications mentioned in this specification are herein incorporated in their entirety by reference into the specification, to the same extent as if each individual publication, patent or patent application was specifically and individually indicated to be incorporated herein by reference. In addition, citation or identification of any reference in this application shall not be construed as an admission that such reference is available as prior art to the present invention. While the invention has been described with respect to a limited number of embodiments, it will be appreciated that many variations, modifications and other applications of the invention may be made.
Claims
Priority Applications (2)
Application Number  Priority Date  Filing Date  Title 

US31696201P true  20010905  20010905  
US60/316,962  20010905 
Applications Claiming Priority (1)
Application Number  Priority Date  Filing Date  Title 

US10/479,056 US20040165678A1 (en)  20020827  20020827  Method for measuring and compensating gain and phase imbalances in quadrature modulators 
Publications (1)
Publication Number  Publication Date 

WO2003021826A1 true WO2003021826A1 (en)  20030313 
Family
ID=23231477
Family Applications (2)
Application Number  Title  Priority Date  Filing Date 

PCT/US2002/027191 WO2003021804A1 (en)  20010905  20020827  New rfic transceiver architecture and method for its use 
PCT/US2002/027192 WO2003021826A1 (en)  20010905  20020827  Method for measuring and compensating gain and phase imbalances in quadrature modulators 
Family Applications Before (1)
Application Number  Title  Priority Date  Filing Date 

PCT/US2002/027191 WO2003021804A1 (en)  20010905  20020827  New rfic transceiver architecture and method for its use 
Country Status (2)
Country  Link 

TW (1)  TW578406B (en) 
WO (2)  WO2003021804A1 (en) 
Cited By (2)
Publication number  Priority date  Publication date  Assignee  Title 

KR101258193B1 (en)  20111122  20130425  주식회사 이노와이어리스  I/q imbalance compensation apparatus and method for direct ubconversion system 
EP2725726A1 (en) *  20121026  20140430  Tektronix, Inc.  Method and apparatus for magnitude and phase response calibration of receivers 
Families Citing this family (1)
Publication number  Priority date  Publication date  Assignee  Title 

FR2949631A1 (en)  20090828  20110304  Thomson Licensing  Broadband reception device emission to the emission and reception of signals of a selected channel in a bandwidth extended dynamically 
Citations (5)
Publication number  Priority date  Publication date  Assignee  Title 

US5396656A (en) *  19930902  19950307  Motorola, Inc.  Method for determining desired components of quadrature modulated signals 
US5694433A (en) *  19940914  19971202  Ericsson Inc.  Efficient linear power amplification 
US5930286A (en) *  19951206  19990727  Conexant Systems, Inc.  Gain imbalance compensation for a quadrature receiver in a cordless direct sequence spread spectrum telephone 
US6009317A (en) *  19970117  19991228  Ericsson Inc.  Method and apparatus for compensating for imbalances between quadrature signals 
US6034573A (en) *  19971030  20000307  Uniden San Diego Research & Development Center, Inc.  Method and apparatus for calibrating modulation sensitivity 
Family Cites Families (7)
Publication number  Priority date  Publication date  Assignee  Title 

US3944747A (en) *  19720824  19760316  Zenith Radio Corporation  Multiple channel FM stereo system 
US4631496A (en) *  19810406  19861223  Motorola, Inc.  Battery saving system for a frequency synthesizer 
US5416449A (en) *  19940523  19950516  Synergy Microwave Corporation  Modulator with harmonic mixers 
US6118810A (en) *  19970508  20000912  Ericsson, Inc.  Multichannel base station/terminal design covering complete system frequency range 
US6288618B1 (en) *  19991220  20010911  Agere Systems Guardian Corp.  Logicbased architecture for FSK modulation and demodulation 
US6348830B1 (en) *  20000508  20020219  The Regents Of The University Of Michigan  Subharmonic doublebalanced mixer 
US6373422B1 (en) *  20001026  20020416  Texas Instruments Incorporated  Method and apparatus employing decimation filter for down conversion in a receiver 

2002
 20020827 WO PCT/US2002/027191 patent/WO2003021804A1/en active Application Filing
 20020827 WO PCT/US2002/027192 patent/WO2003021826A1/en active Application Filing
 20020905 TW TW091120347A patent/TW578406B/en not_active IP Right Cessation
Patent Citations (5)
Publication number  Priority date  Publication date  Assignee  Title 

US5396656A (en) *  19930902  19950307  Motorola, Inc.  Method for determining desired components of quadrature modulated signals 
US5694433A (en) *  19940914  19971202  Ericsson Inc.  Efficient linear power amplification 
US5930286A (en) *  19951206  19990727  Conexant Systems, Inc.  Gain imbalance compensation for a quadrature receiver in a cordless direct sequence spread spectrum telephone 
US6009317A (en) *  19970117  19991228  Ericsson Inc.  Method and apparatus for compensating for imbalances between quadrature signals 
US6034573A (en) *  19971030  20000307  Uniden San Diego Research & Development Center, Inc.  Method and apparatus for calibrating modulation sensitivity 
Cited By (5)
Publication number  Priority date  Publication date  Assignee  Title 

KR101258193B1 (en)  20111122  20130425  주식회사 이노와이어리스  I/q imbalance compensation apparatus and method for direct ubconversion system 
WO2013077507A1 (en) *  20111122  20130530  Innowireless Co., Ltd.  Apparatus and method of compensating for i/q imbalance in direct upconversion system 
US9042483B2 (en)  20111122  20150526  Innowireless Co., Ltd.  Apparatus and method of compensating for I/Q imbalance in direct upconversion system 
EP2725726A1 (en) *  20121026  20140430  Tektronix, Inc.  Method and apparatus for magnitude and phase response calibration of receivers 
US8805313B2 (en)  20121026  20140812  Tektronix, Inc.  Magnitude and phase response calibration of receivers 
Also Published As
Publication number  Publication date 

TW578406B (en)  20040301 
WO2003021804A1 (en)  20030313 
Similar Documents
Publication  Publication Date  Title 

US6081698A (en)  Radio apparatus and offset compensating method  
US7529313B2 (en)  Distortion compensation quadrature modulator and radio transmitter  
US8090320B2 (en)  Strong signal tolerant OFDM receiver and receiving methods  
US6987954B2 (en)  Feedback compensation detector for a direct conversion transmitter  
US5793817A (en)  DC offset reduction in a transmitter  
US9001875B1 (en)  Onchip IQ imbalance and LO leakage calibration for transceivers  
US7689194B2 (en)  Balanced circuit arrangement and method for linearizing such an arrangement  
CA2119867C (en)  Apparatus for compensating of phase rotation in a final amplifier stage  
EP1477005B1 (en)  I/q mismatch compensation in an ofdm receiver in presence of frequency offset  
EP1120944A2 (en)  Modulation system with online IQ calibration  
US7336940B2 (en)  Frequency conversion techniques using antiphase mixing  
US7280612B2 (en)  Digital branch calibrator for an RF transmitter  
EP0977351B1 (en)  Method and apparatus for radio communication  
CN100423447C (en)  Transmitter with transmitter chain phase adjustment on the basis of prestored phase information  
US20060009171A1 (en)  LO leakage and sideband image calibration system and method  
JP3805221B2 (en)  Distortion compensation device  
FI117494B (en)  The method of digital quadrature modulator and a quadrature demodulator, a digital quadrature modulator and a quadrature demodulator  
EP0598585B1 (en)  Automatic calibration of the quadrature balance within a cartesian amplifier  
CN1105424C (en)  Compensation for second order intermodulation in homodyne receiver  
US5293406A (en)  Quadrature amplitude modulator with distortion compensation  
CN1154245C (en)  Reception circuit  
US6073001A (en)  Down conversion mixer  
CA2054995C (en)  System and method for compensation of inphase and quadrature phase and gain imbalance  
FI107212B (en)  I / Q modulator output DC voltage offset correction  
US6670900B1 (en)  Quadrature mismatch compensation 
Legal Events
Date  Code  Title  Description 

AL  Designated countries for regional patents 
Kind code of ref document: A1 Designated state(s): GH GM KE LS MW MZ SD SL SZ UG ZM ZW AM AZ BY KG KZ RU TJ TM AT BE BG CH CY CZ DK EE ES FI FR GB GR IE IT LU MC PT SE SK TR BF BJ CF CG CI GA GN GQ GW ML MR NE SN TD TG Kind code of ref document: A1 Designated state(s): GH GM KE LS MW MZ SD SL SZ TZ UG ZM ZW AM AZ BY KG KZ MD RU TJ TM AT BE BG CH CY CZ DE DK EE ES FI FR GB GR IE IT LU MC NL PT SE SK TR BF BJ CF CG CI CM GA GN GQ GW ML MR NE SN TD TG 

AK  Designated states 
Kind code of ref document: A1 Designated state(s): AE AG AL AM AT AU AZ BA BB BG BR BY BZ CA CH CN CO CR CU CZ DE DK DM DZ EC EE ES FI GB GD GE GH GM HR HU ID IL IN IS JP KE KG KP KR KZ LC LK LR LS LT LU LV MA MD MG MK MN MW MX MZ NO NZ OM PH PL PT RO RU SD SE SG SI SK SL TJ TM TN TR TT TZ UA UG US UZ VN YU ZA ZM ZW Kind code of ref document: A1 Designated state(s): AE AG AL AM AT AU AZ BA BB BG BY BZ CA CH CN CO CR CU CZ DE DM DZ EC EE ES FI GB GD GE GH HR HU ID IL IN IS JP KE KG KP KR LC LK LR LS LT LU LV MA MD MG MN MW MX MZ NO NZ OM PH PL PT RU SD SE SG SI SK SL TJ TM TN TR TZ UA UG US UZ VN YU ZA ZM 

121  Ep: the epo has been informed by wipo that ep was designated in this application  
WWE  Wipo information: entry into national phase 
Ref document number: 10479056 Country of ref document: US 

122  Ep: pct application nonentry in european phase  
WWW  Wipo information: withdrawn in national office 
Country of ref document: JP 

NENP  Nonentry into the national phase in: 
Ref country code: JP 