WO2002095962A2 - Quadrature envelope-sampling of intermediate frequency signal in receiver - Google Patents

Quadrature envelope-sampling of intermediate frequency signal in receiver Download PDF

Info

Publication number
WO2002095962A2
WO2002095962A2 PCT/IB2002/001823 IB0201823W WO02095962A2 WO 2002095962 A2 WO2002095962 A2 WO 2002095962A2 IB 0201823 W IB0201823 W IB 0201823W WO 02095962 A2 WO02095962 A2 WO 02095962A2
Authority
WO
WIPO (PCT)
Prior art keywords
sampling
signal
intermediate frequency
analog
receiver according
Prior art date
Application number
PCT/IB2002/001823
Other languages
French (fr)
Other versions
WO2002095962A3 (en
Inventor
Yiping Fan
Original Assignee
Koninklijke Philips Electronics N.V.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Koninklijke Philips Electronics N.V. filed Critical Koninklijke Philips Electronics N.V.
Priority to JP2002592305A priority Critical patent/JP2004527187A/en
Priority to KR10-2003-7001049A priority patent/KR20030017649A/en
Priority to EP02730609A priority patent/EP1396088A2/en
Publication of WO2002095962A2 publication Critical patent/WO2002095962A2/en
Publication of WO2002095962A3 publication Critical patent/WO2002095962A3/en

Links

Classifications

    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B14/00Transmission systems not characterised by the medium used for transmission
    • H04B14/02Transmission systems not characterised by the medium used for transmission characterised by the use of pulse modulation
    • H04B14/04Transmission systems not characterised by the medium used for transmission characterised by the use of pulse modulation using pulse code modulation
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0007Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage
    • H04B1/0014Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage using DSP [Digital Signal Processor] quadrature modulation and demodulation
    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03DDEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D3/00Demodulation of angle-, frequency- or phase- modulated oscillations
    • H03D3/007Demodulation of angle-, frequency- or phase- modulated oscillations by converting the oscillations into two quadrature related signals
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0007Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage
    • H04B1/0025Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage using a sampling rate lower than twice the highest frequency component of the sampled signal
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0028Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage
    • H04B1/0039Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage using DSP [Digital Signal Processor] quadrature modulation and demodulation
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/16Circuits
    • H04B1/26Circuits for superheterodyne receivers
    • H04B1/28Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/32Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38Demodulator circuits; Receiver circuits
    • H04L27/3845Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
    • H04L27/3881Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using sampling and digital processing, not including digital systems which imitate heterodyne or homodyne demodulation
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/0014Carrier regulation
    • H04L2027/0016Stabilisation of local oscillators
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/0014Carrier regulation
    • H04L2027/0024Carrier regulation at the receiver end

Definitions

  • the present invention relates to sampling of intermediate frequency signals in receivers, and, more particularly, to quadrature envelope sampling of intermediate frequency signals in receivers.
  • a radio frequency (RF) signal is converted into an intermediate frequency (IF) signal.
  • IF intermediate frequency
  • One IF stage is typically used.
  • the received signal is converted by IF mixers into in-phase and quadrature (I/Q) baseband signals.
  • I/Q signals are filtered by a pair of lowpass channel filters.
  • the I/Q lowpass filter outputs are sampled simultaneously by a pair of lowpass analog-to-digital converters (ADC).
  • ADC analog-to-digital converters
  • the digitized data produced by the converters are processed by digital signal hardware to recover the desired information, such as voice, image, and other data. Due to the circuit mismatch from the I/Q IF mixers and the I/Q lowpass filters, the gain and phase frequency response between the I channel and the Q channel are often not the same. This is called I/Q imbalance. In addition, the DC offset problem is very common with this approach.
  • Bandpass sampling of an IF signal is another sampling scheme.
  • the received signal is directly sampled at the IF stage by a bandpass sampling ADC.
  • the sampling can take place with either oversampling or subsampling.
  • the scheme eliminates two IF mixers and analog lowpass filters as compared to the conventional I/Q lowpass sampling scheme previously described.
  • the bandpass sampling scheme eliminates the I/Q imbalance and the DC offset.
  • the cost and complexity of designing, fabricating, and implementing a bandpass ADC and a bandpass digital filter as well as the associated power consumption may limit the usefulness of this sampling approach.
  • the present invention provides an apparatus and method for the direct intermediate frequency (IF) sampling of a received signal which is modulated by a two- dimensional signal constellation, such as quadrature phase shift keying (QPSK) and quadrature amplitude modulation (QAM).
  • IF direct intermediate frequency
  • QPSK quadrature phase shift keying
  • QAM quadrature amplitude modulation
  • the IF signal is sampled by a pair of lowpass analog-to-digital converters thereby achieving significant savings in power consumption and fabrication cost as compared to the more complex and expensive bandpass analog-to-digital converters and digital bandpass filters while maintaining comparable performance with previous designs.
  • the present invention in one form thereof, includes a receiver which overcomes the shortcomings of the prior art.
  • the receiver includes a radio frequency (RF) mixer, an IF filter, and an amplifier.
  • RF radio frequency
  • IF filter Directly connected to the amplifier is a first and a second ADC which are operable to directly sample the IF signal using a quadrature envelope sampling scheme.
  • DSP digital signal processor
  • the present invention includes a method for direct IF sampling of a signal which is modulated by a two-dimensional signal constellation in a receiver.
  • the method includes the steps of receiving a signal and converting the signal to an intermediate frequency using an RF mixer.
  • the method further includes filtering and amplifying the resultant IF signal.
  • the amplified IF signal is directly sampled by a pair of lowpass analog- to-digital converters using a quadrature envelope sampling scheme.
  • a DSP is then used to process the sampled information extracted by the lowpass analog-to-digital converters to recover the desired information.
  • An advantage of the present invention is the reduced power consumption as compared to previous sampling schemes while maintaining good results.
  • Another advantage of the present invention is the reduced complexity as compared to previous sampling schemes while maintaining good results.
  • Yet another advantage of the present invention is the shifting of the digital signal processing toward the antenna.
  • Fig. 1 is a prior art super heterodyne receiver architecture implementing a lowpass sampling scheme.
  • Fig. 2 is a prior art super heterodyne receiver architecture implementing a bandpass sampling scheme.
  • Fig. 3 is a super heterodyne receiver architecture implementing a quadrature envelope sampling scheme according to the present invention.
  • Fig. 4 is a representation of the quadrature envelope sampling scheme according to the present invention.
  • Fig. 5 is a plot of the I-channel baseband signal with the quadrature envelope sampling.
  • Fig. 6 is a plot of the Q-channel baseband signal with the quadrature envelope sampling.
  • Fig. 7 is a plot of the Q-channel signal distortion with the quadrature envelope sampling.
  • Fig. 8 is a plot of the power spectrum of the signal and the distortion produced by the quadrature envelope sampling scheme.
  • FIG. 1 a prior art super heterodyne receiver architecture with lowpass sampling is shown.
  • This super heterodyne receiver 100 utilizes lowpass sampling.
  • Antenna 101 receives an incoming transmitted signal.
  • Antenna 101 is connected to duplex 102.
  • Duplex 102 includes two bandpass filters 104 and 106.
  • Receive filter 104 is operable to pass the frequency of the received signal.
  • Transmit filter 106 is operable to pass the frequency of a transmitted signal.
  • the radio frequency output from receive filter 104 is received by low noise amplifier 108.
  • the amplified output is received by surface acoustic wave filter 110.
  • the filtered signal is then communicated to radio frequency mixer 112.
  • Radio frequency mixer 112 uses radio frequency mixer input 114 to convert the input signal to an intermediate frequency signal.
  • the IF output from mixer 112 is input into surface acoustic wave filter 116.
  • the filtered signal is then input into variable gain amplifier 118.
  • Connected to amplifier 118 are a pair of IF mixers 120.
  • IF mixers 120 down-convert the received signal into in-phase and quadrature (I/Q) baseband signals.
  • the I/Q signals are then filtered by a pair of lowpass channel filters 126.
  • Lowpass analog-to-digital converters 128 may be converters using Sigma Delta modulation technique.
  • Fig. 2 is a super heterodyne receiver architecture with bandpass sampling shown generally at 200.
  • Antenna 201 receives an incoming transmitted signal.
  • Antenna 201 is connected to duplex 202.
  • Duplex 202 includes two filters 204 and 206.
  • Receive filter 204 is operable to pass the frequency of the received signal.
  • Transmit filter 206 is operable to pass the frequency of a transmitted signal.
  • the radio frequency output from receive filter 204 is received by low noise amplifier 208.
  • the amplified output is received by surface acoustic wave filter 210.
  • the filtered signal is then communicated to radio frequency mixer 212.
  • Radio frequency mixer 212 uses radio frequency mixer input 214 to convert the input signal to an intermediate frequency signal.
  • the IF output from mixer 212 is input into surface acoustic wave filter 216.
  • the filtered signal is then input into variable gain amplifier 218.
  • the IF amplified output from amplifier 218 is input into bandpass analog-to-digital converter 220 which either oversamples or sub-samples the signal.
  • the output of converter 220 is filtered by digital bandpass filter 222 and then transmitted for further processing by digital signal processor 224.
  • Fig. 3 is a preferred embodiment of a receiver architecture according to the present invention.
  • Receiver 300 uses quadrature envelope sampling.
  • Antenna 301 receives an incoming transmitted signal.
  • Antenna 301 is connected to duplex 302.
  • Duplex 302 includes two filters 304 and 306.
  • Receive filter 304 is operable to pass the frequency of the received signal.
  • Transmit filter 306 is operable to pass the frequency of a transmitted signal.
  • the radio frequency output from receive filter 304 is received by low noise amplifier 308.
  • the amplified output is received by surface acoustic wave filter 310.
  • the filtered signal is then communicated to radio frequency mixer 312.
  • Radio frequency mixer 312 uses radio frequency mixer input 314 to convert the input signal to an intermediate frequency signal.
  • the IF output from mixer 312 is input into surface acoustic wavefilter 316.
  • the filtered signal is then input into variable gain amplifier 318.
  • the IF signal is then directly sampled by a pair of lowpass analog-to-digital converters 320.
  • Direct sampling involves no intervening components between the amplifier and the analog-to-digital converters. Instead of including mixers and filters before the sampling of the IF signal as is present in the prior art, direct sampling permits the sampling of the IF signal without any mixers, analog channel filters, or similar intervening components. The elimination of intervening components reduces cost and complexity by introducing fewer parts into the design and fabrication of the receiver.
  • the output of converters 320 is input into digital signal processor 322 for further processing.
  • the channel filtering with this architecture is performed by the DSP and the I/Q imbalance is minimized.
  • DSP digital signal processor
  • I/Q imbalance is minimized.
  • a significant saving can be achieved in both power consumption and fabrication cost.
  • Such a configuration discloses a direct IF quadrature envelope sampling scheme for an I/Q signal pair by using a pair of lowpass analog-to-digital converters 320.
  • a fast sample-and-hold circuit must be present at the input of lowpass analog-to-digital converter 320 in order for the quadrature envelope sampling approach to function properly.
  • lowpass analog-to-digital converter 320 is a Sigma Delta analog-to-digital converter.
  • lowpass analog- to-digital converter 320 is a flash-type ADC. If lowpass analog-to-digital converter 320 were a flash-type converter, only one converter would be necessary instead of a pair of converters, thereby further reducing cost and complexity. This is because in the quadrature envelope sampling, the I/Q channels are not sampled simultaneously as in the prior art. Therefore, by time multiplexing, both I and Q channels get sampled by one ADC.
  • Fig. 4 is a graphical representation of the inventive quadrature envelope sampling scheme according to the present invention.
  • the I/Q samples from the quadrature envelope sampling scheme are not taken at the same sampling time.
  • the directly sampled IF signal is sampled in a scheme in which the Q-channel ADC takes a sample a quarter of the IF carrier period before or after the I-channel ADC takes a sample.
  • the Q channel sample is taken a quarter of the IF carrier period later.
  • An I-channel sampling point is shown generally at 410.
  • a Q-channel sampling point is shown generally at 420, ninety degrees after I-channel sampling point 410.
  • the IF carrier period is denoted by T IF and the separation of point 410 and point 420 is shown with arrows indicated by TT .F / 4. The distance between these arrows represent a quarter of the IF carrier period.
  • the sampling frequency is the same as the intermediate frequency or sub-harmonic frequencies of the intermediate frequency. Essentially, the sampling frequency is equal to the intermediate frequency divided by the order of the sub-harmonics (an integer). In Fig. 4a, the order of the sub-harmonic is one (1), which yields a sampling frequency equal to the intermediate frequency. In Fig. 4b, the order of the sub-harmonic is two (2), which yields a sampling frequency equal to one-half (1/2) of the intermediate frequency. Since the typical intermediate frequency is much greater than the information bandwidth, the sampling delay (equal to a quarter of the IF carrier period) in the Q-channel (or in the I channel) will not have any practical negative impact as will be shown below.
  • Fig. 5 provides a graphical plot of the directly sampled I-channel baseband signal with the quadrature envelope sampling scheme from Fig. 4.
  • Fig. 5 includes a plot of two curves which, however, are indistinguishable as they are identical.
  • One curve represents the typical I-channel baseband sampling.
  • the second represents the I-channel with quadrature envelope sampling from the IF signal.
  • the two I-channel curves are identical.
  • Fig. 6 provides a graphical plot of the directly sampled Q-channel baseband signal with the quadrature envelope sampling scheme from Fig. 4.
  • Fig. 6 includes a plot of two curves.
  • One curve represents the typical Q-channel baseband sampling.
  • the second represents the Q-channel with quadrature envelope sampling from the IF signal. The difference between the two curves is very small and, therefore, the curves appear to overlap one another.
  • Fig. 7 provides a graphical plot of the distortion calculated by Equation (6) for the Q-channel over the same period as used in Fig. 6.
  • Fig. 7 illustrates the distortion which is the difference between the two curves in Fig. 6.
  • the amount of distortion is very small because the intermediate frequency is much higher than the information bandwidth.
  • a theoretical mathematical analysis of the quadrature envelope sampling scheme is given below.
  • the received signal denoted as S(t) with its amplitude and phase as m(t) and ⁇ (t) and an arbitrary constant initial phase ⁇ , is represented in Equation (1).
  • the Q-channel sampled data with the quadrature envelope sampling scheme is distorted and the amount of distortion is given by Equation (6) and is shown in Fig. 7 over the same period as the signals shown in Figs. 5 and 6.
  • ⁇ (t, ) m(t l ) • sinf ⁇ (t, )] ⁇ m t, + ⁇ ) ⁇ sinf ⁇ (t, + ⁇ )] (6)
  • Fig. 8 is a graphical plot of power spectrum 810 of the signal and distortion spectrum 820 as calculated by Equation (6).
  • the ratio of the desired signal energy over the distortion energy (SDR) averaged over M sampling points is calculated using Equation (7).
  • a calculation over a 1280-chip period for the CDMA communication system gives an SDR of approximately 53 dB.
  • the SDR value can be seen in Fig. 8 by observing the difference between power spectrum 810 and distortion spectrum 820.
  • the spectrum analysis reveals that the spectrum of the distortion signal is also band limited and has the same bandwidth as the signal in Fig. 7.
  • the quadrature envelope sampling scheme uses the aliasing property of digital sampling. Therefore, the noise in the image bands will fall back into the signal band. Due to the filtering protection of the IF surface acoustic wavefilter, the noise from the image band is greatly reduced. Therefore, the aliasing noise effect should not be a concern.
  • the sampling frequency is the third subharmonic frequency of the intermediate frequency, e.g., IF - 183.6 MHz, the image band is already outside of the US cellular receive band.

Landscapes

  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Superheterodyne Receivers (AREA)

Abstract

An apparatus and method for the two-dimensional direct intermediate frequency sampling of a received signal. A receiver is equipped with a circuit for converting a received radio frequency signal to an intermediate frequency signal. The converted intermediate frequency signal is sampled by a pair of lowpass analog-to-digital converters. The sampling scheme involves quadrature envelope sampling of the intermediate frequency signal. The sampling scheme further involves sampling the Q-channel signal at a quarter of the intermediate frequency carrier period after the I-channel signal is sampled.

Description

Quadrature envelope-sampling of intermediate frequency signal in receiver
The present invention relates to sampling of intermediate frequency signals in receivers, and, more particularly, to quadrature envelope sampling of intermediate frequency signals in receivers.
The sampling of analog signals by receivers in wireless devices, such as code division multiple access (CDMA) or time division multiple access (TDMA) devices, is performed in several ways. In receivers, a radio frequency (RF) signal is converted into an intermediate frequency (IF) signal. One IF stage is typically used. After proper amplification and filtering at radio frequency (RF) and IF, the received signal is converted by IF mixers into in-phase and quadrature (I/Q) baseband signals. The I/Q signals are filtered by a pair of lowpass channel filters. The I/Q lowpass filter outputs are sampled simultaneously by a pair of lowpass analog-to-digital converters (ADC). The digitized data produced by the converters are processed by digital signal hardware to recover the desired information, such as voice, image, and other data. Due to the circuit mismatch from the I/Q IF mixers and the I/Q lowpass filters, the gain and phase frequency response between the I channel and the Q channel are often not the same. This is called I/Q imbalance. In addition, the DC offset problem is very common with this approach.
Bandpass sampling of an IF signal is another sampling scheme. In this scheme, the received signal is directly sampled at the IF stage by a bandpass sampling ADC. The sampling can take place with either oversampling or subsampling. The scheme eliminates two IF mixers and analog lowpass filters as compared to the conventional I/Q lowpass sampling scheme previously described. Furthermore, the bandpass sampling scheme eliminates the I/Q imbalance and the DC offset. However, the cost and complexity of designing, fabricating, and implementing a bandpass ADC and a bandpass digital filter as well as the associated power consumption may limit the usefulness of this sampling approach.
What is needed is an apparatus and method for sampling received signals which possess the benefits of the previous designs, and, furthermore, which eliminate the extra cost and complexity associated with such previous designs. The present invention provides an apparatus and method for the direct intermediate frequency (IF) sampling of a received signal which is modulated by a two- dimensional signal constellation, such as quadrature phase shift keying (QPSK) and quadrature amplitude modulation (QAM). The IF signal is sampled by a pair of lowpass analog-to-digital converters thereby achieving significant savings in power consumption and fabrication cost as compared to the more complex and expensive bandpass analog-to-digital converters and digital bandpass filters while maintaining comparable performance with previous designs.
The present invention, in one form thereof, includes a receiver which overcomes the shortcomings of the prior art. The receiver includes a radio frequency (RF) mixer, an IF filter, and an amplifier. Directly connected to the amplifier is a first and a second ADC which are operable to directly sample the IF signal using a quadrature envelope sampling scheme. Furthermore, a digital signal processor (DSP) is connected to the first and second lowpass analog-to-digital converters and is operable to process the sampled data to recover the desired information.
Furthermore, the present invention includes a method for direct IF sampling of a signal which is modulated by a two-dimensional signal constellation in a receiver. The method includes the steps of receiving a signal and converting the signal to an intermediate frequency using an RF mixer. The method further includes filtering and amplifying the resultant IF signal. The amplified IF signal is directly sampled by a pair of lowpass analog- to-digital converters using a quadrature envelope sampling scheme. A DSP is then used to process the sampled information extracted by the lowpass analog-to-digital converters to recover the desired information.
An advantage of the present invention is the reduced power consumption as compared to previous sampling schemes while maintaining good results.
Another advantage of the present invention is the reduced complexity as compared to previous sampling schemes while maintaining good results.
Yet another advantage of the present invention is the shifting of the digital signal processing toward the antenna.
The above-mentioned and other features and advantages of this invention, and the manner of attaining them, will become more apparent and the invention itself will be better understood by reference to the following description of an embodiment of the invention taken in conjunction with the accompanying drawings, wherein:
Fig. 1 is a prior art super heterodyne receiver architecture implementing a lowpass sampling scheme. Fig. 2 is a prior art super heterodyne receiver architecture implementing a bandpass sampling scheme.
Fig. 3 is a super heterodyne receiver architecture implementing a quadrature envelope sampling scheme according to the present invention.
Fig. 4 is a representation of the quadrature envelope sampling scheme according to the present invention.
Fig. 5 is a plot of the I-channel baseband signal with the quadrature envelope sampling.
Fig. 6 is a plot of the Q-channel baseband signal with the quadrature envelope sampling. Fig. 7 is a plot of the Q-channel signal distortion with the quadrature envelope sampling.
Fig. 8 is a plot of the power spectrum of the signal and the distortion produced by the quadrature envelope sampling scheme.
Corresponding reference characters indicate corresponding parts throughout the several views. The exemplification set out herein illustrates one preferred embodiment of the invention, in one form, and such exemplification is not to be construed as limiting the scope of the invention in any manner. Referring now to the drawings and particularly to Fig. 1 , a prior art super heterodyne receiver architecture with lowpass sampling is shown. This super heterodyne receiver 100 utilizes lowpass sampling. Antenna 101 receives an incoming transmitted signal. Antenna 101 is connected to duplex 102. Duplex 102 includes two bandpass filters 104 and 106. Receive filter 104 is operable to pass the frequency of the received signal. Transmit filter 106 is operable to pass the frequency of a transmitted signal. The radio frequency output from receive filter 104 is received by low noise amplifier 108. The amplified output is received by surface acoustic wave filter 110. The filtered signal is then communicated to radio frequency mixer 112. Radio frequency mixer 112 uses radio frequency mixer input 114 to convert the input signal to an intermediate frequency signal. The IF output from mixer 112 is input into surface acoustic wave filter 116. The filtered signal is then input into variable gain amplifier 118. Connected to amplifier 118 are a pair of IF mixers 120. IF mixers 120 down-convert the received signal into in-phase and quadrature (I/Q) baseband signals. The I/Q signals are then filtered by a pair of lowpass channel filters 126. The analog outputs of lowpass filters 126 are sampled by a pair of lowpass analog-to-digital converters 128. The digitized output of converters 128 is input into digital signal processor 130 for further processing to recover the desired information. Lowpass analog-to-digital converters 128 may be converters using Sigma Delta modulation technique.
Fig. 2 is a super heterodyne receiver architecture with bandpass sampling shown generally at 200. Antenna 201 receives an incoming transmitted signal. Antenna 201 is connected to duplex 202. Duplex 202 includes two filters 204 and 206. Receive filter 204 is operable to pass the frequency of the received signal. Transmit filter 206 is operable to pass the frequency of a transmitted signal. The radio frequency output from receive filter 204 is received by low noise amplifier 208. The amplified output is received by surface acoustic wave filter 210. The filtered signal is then communicated to radio frequency mixer 212.
Radio frequency mixer 212 uses radio frequency mixer input 214 to convert the input signal to an intermediate frequency signal. The IF output from mixer 212 is input into surface acoustic wave filter 216. The filtered signal is then input into variable gain amplifier 218. The IF amplified output from amplifier 218 is input into bandpass analog-to-digital converter 220 which either oversamples or sub-samples the signal. The output of converter 220 is filtered by digital bandpass filter 222 and then transmitted for further processing by digital signal processor 224.
Fig. 3 is a preferred embodiment of a receiver architecture according to the present invention. Receiver 300 uses quadrature envelope sampling. Antenna 301 receives an incoming transmitted signal. Antenna 301 is connected to duplex 302. Duplex 302 includes two filters 304 and 306. Receive filter 304 is operable to pass the frequency of the received signal. Transmit filter 306 is operable to pass the frequency of a transmitted signal. The radio frequency output from receive filter 304 is received by low noise amplifier 308. The amplified output is received by surface acoustic wave filter 310. The filtered signal is then communicated to radio frequency mixer 312. Radio frequency mixer 312 uses radio frequency mixer input 314 to convert the input signal to an intermediate frequency signal. The IF output from mixer 312 is input into surface acoustic wavefilter 316. The filtered signal is then input into variable gain amplifier 318. The IF signal is then directly sampled by a pair of lowpass analog-to-digital converters 320. Direct sampling involves no intervening components between the amplifier and the analog-to-digital converters. Instead of including mixers and filters before the sampling of the IF signal as is present in the prior art, direct sampling permits the sampling of the IF signal without any mixers, analog channel filters, or similar intervening components. The elimination of intervening components reduces cost and complexity by introducing fewer parts into the design and fabrication of the receiver. The output of converters 320 is input into digital signal processor 322 for further processing. The channel filtering with this architecture is performed by the DSP and the I/Q imbalance is minimized. By directly sampling with lowpass analog-to-digital converters 320, a significant saving can be achieved in both power consumption and fabrication cost. Such a configuration discloses a direct IF quadrature envelope sampling scheme for an I/Q signal pair by using a pair of lowpass analog-to-digital converters 320. A fast sample-and-hold circuit must be present at the input of lowpass analog-to-digital converter 320 in order for the quadrature envelope sampling approach to function properly.
In an alternative embodiment, lowpass analog-to-digital converter 320 is a Sigma Delta analog-to-digital converter. In another alternative embodiment, lowpass analog- to-digital converter 320 is a flash-type ADC. If lowpass analog-to-digital converter 320 were a flash-type converter, only one converter would be necessary instead of a pair of converters, thereby further reducing cost and complexity. This is because in the quadrature envelope sampling, the I/Q channels are not sampled simultaneously as in the prior art. Therefore, by time multiplexing, both I and Q channels get sampled by one ADC.
Fig. 4 is a graphical representation of the inventive quadrature envelope sampling scheme according to the present invention. In contrast with conventional prior art I/Q sampling schemes, the I/Q samples from the quadrature envelope sampling scheme are not taken at the same sampling time. In the quadrature envelope sampling scheme, the directly sampled IF signal is sampled in a scheme in which the Q-channel ADC takes a sample a quarter of the IF carrier period before or after the I-channel ADC takes a sample. For demonstration purpose, the Q channel sample is taken a quarter of the IF carrier period later. An I-channel sampling point is shown generally at 410. A Q-channel sampling point is shown generally at 420, ninety degrees after I-channel sampling point 410. The IF carrier period is denoted by TIF and the separation of point 410 and point 420 is shown with arrows indicated by TT.F / 4. The distance between these arrows represent a quarter of the IF carrier period. The sampling frequency is the same as the intermediate frequency or sub-harmonic frequencies of the intermediate frequency. Essentially, the sampling frequency is equal to the intermediate frequency divided by the order of the sub-harmonics (an integer). In Fig. 4a, the order of the sub-harmonic is one (1), which yields a sampling frequency equal to the intermediate frequency. In Fig. 4b, the order of the sub-harmonic is two (2), which yields a sampling frequency equal to one-half (1/2) of the intermediate frequency. Since the typical intermediate frequency is much greater than the information bandwidth, the sampling delay (equal to a quarter of the IF carrier period) in the Q-channel (or in the I channel) will not have any practical negative impact as will be shown below.
Fig. 5 provides a graphical plot of the directly sampled I-channel baseband signal with the quadrature envelope sampling scheme from Fig. 4. Fig. 5 includes a plot of two curves which, however, are indistinguishable as they are identical. One curve represents the typical I-channel baseband sampling. The second represents the I-channel with quadrature envelope sampling from the IF signal. As expected, the two I-channel curves are identical. Fig. 6 provides a graphical plot of the directly sampled Q-channel baseband signal with the quadrature envelope sampling scheme from Fig. 4. Fig. 6 includes a plot of two curves. One curve represents the typical Q-channel baseband sampling. The second represents the Q-channel with quadrature envelope sampling from the IF signal. The difference between the two curves is very small and, therefore, the curves appear to overlap one another.
Fig. 7 provides a graphical plot of the distortion calculated by Equation (6) for the Q-channel over the same period as used in Fig. 6. Fig. 7 illustrates the distortion which is the difference between the two curves in Fig. 6. The amount of distortion is very small because the intermediate frequency is much higher than the information bandwidth. A theoretical mathematical analysis of the quadrature envelope sampling scheme is given below.
The received signal, denoted as S(t) with its amplitude and phase as m(t) and θ(t) and an arbitrary constant initial phase θ, is represented in Equation (1).
Figure imgf000007_0001
When the sampling point for the I-channel on this received waveform is aligned to the positive peak of cos(ωIFt) (this assumption can be made because θ is arbitrary), i.e. : cos(ωIFt) = 1 and sin(ω/ t) = 0 , the sampled I-channel data at the i-th instance ( t = tt ) is given in Equation (2).
I(tl) = m(tl) - cos[φ(tl) + θ] (2) The sampled Q channel at the i-th instance is ( t = /, + δ , δ =TIF I , TIF is the IF carrier period) given by Equation (3).
Q(t, ) = -m(t, + δ) ■ sm[φ(t, + δ) + θ] (3)
Due to phase derotation processing in DSP which uses a reference phase information such as in CDMA cellular communication system, the arbitrary phase θ is removed. Therefore, the effective sampled I/Q data are given by Equations (4) and (5).
7(t, ) = m(t, ) - cosfø(t, )] (4)
Q(t, ) = -m(tl + δ) - stn[ φ (tl + δ)l (5)
The Q-channel sampled data with the quadrature envelope sampling scheme is distorted and the amount of distortion is given by Equation (6) and is shown in Fig. 7 over the same period as the signals shown in Figs. 5 and 6.
Δ(t, ) = m(tl ) • sinfø(t, )] ~ m t, + δ) sinfø(t, + δ)] (6)
Fig. 8 is a graphical plot of power spectrum 810 of the signal and distortion spectrum 820 as calculated by Equation (6). The ratio of the desired signal energy over the distortion energy (SDR) averaged over M sampling points is calculated using Equation (7). A calculation over a 1280-chip period for the CDMA communication system gives an SDR of approximately 53 dB. The SDR value can be seen in Fig. 8 by observing the difference between power spectrum 810 and distortion spectrum 820.
M
SZ)i? = ∑[Δ(t()/ (t;)]2 / (7)
The very high value of the SDR theoretically predicts that the quadrature envelope sampling scheme will not have any negative effects.
As shown in Fig. 8, the spectrum analysis reveals that the spectrum of the distortion signal is also band limited and has the same bandwidth as the signal in Fig. 7. In the frequency domain, the quadrature envelope sampling scheme uses the aliasing property of digital sampling. Therefore, the noise in the image bands will fall back into the signal band. Due to the filtering protection of the IF surface acoustic wavefilter, the noise from the image band is greatly reduced. Therefore, the aliasing noise effect should not be a concern. When the sampling frequency is the third subharmonic frequency of the intermediate frequency, e.g., IF - 183.6 MHz, the image band is already outside of the US cellular receive band. While this invention has been described as having a preferred design, the present invention can be further modified within the spirit and scope of this disclosure. This application is therefore intended to cover any variations, uses, or adaptations of the invention using its general principles. Further, this application is intended to cover such departures from the present disclosure as come within known or customary practice in the art to which this invention pertains and which fall within the limits of the appended claims.

Claims

CLAIMS:
1. A receiver comprising :
- a radio frequency mixer (312);
- an intermediate frequency filter (316);
- an amplifier (318); characterized in that: * a first lowpass analog-to-digital converter (320) is directly connected to said amplifier (318);
* a second lowpass analog-to-digital converter (320) is directly connected to said amplifier (318); and
* a digital signal processor (322) connected to said first and second lowpass analog-to-digital converters (320).
2. The receiver according to claim 1 wherein said receiver forms a part of a communications device.
3. The receiver according to claim 2 wherein said communications device comprises a cellular phone.
4. The receiver according to claim 2 wherein said communications device comprises a wireless device.
5. The receiver according to claim 2 wherein said communications device comprises a code division multiple access (CDMA) device.
6. The receiver according to claim 2 wherein said communications device comprises a time division multiple access (TDMA) device.
7. The receiver according to claim 1 further comprising a radio frequency filter (310).
8. The receiver according to claim 7 wherein said radio frequency filter comprises a surface acoustic wave filter (310).
9. The receiver according to claim 1 wherein said intermediate frequency filter comprises a surface acoustic wave filter (316).
10. The receiver according to claim 1 wherein said amplifier comprises a variable gain amplifier (318).
11. The receiver according to claim 1 wherein said first and second lowpass analog-to-digital converters (320) comprise Sigma Delta analog-to-digital converters.
12. The receiver according to claim 1 wherein said first lowpass analog-to-digital converter (320) comprises a flash-type analog-to-digital converter.
13. A method for direct sampling of an intermediate frequency signal in a receiver comprising:
- receiving a signal;
- converting said signal to an intermediate frequency signal; - filtering said intermediate frequency signal;
- amplifying said filtered intermediate frequency signal; characterized in directly sampling said amplified intermediate frequency signal; and processing said directly sampled signal with a digital signal processor (322).
14. The method according to claim 13 wherein said direct sampling comprises:
- sampling a first channel at a predetermined time; and
- sampling a second channel a quarter of the intermediate frequency carrier period after said sampling of said first channel.
15. The method according to claim 13 wherein said direct sampling is accomplished with a pair of lowpass analog-to-digital converters (320).
16. The method according to claim 15 wherein said lowpass analog-to-digital converters (320) comprise Sigma Delta analog-to-digital converters.
17. The method according to claim 13 wherein said direct sampling is accomplished with a single flash-type lowpass analog-to-digital converter.
PCT/IB2002/001823 2001-05-25 2002-05-22 Quadrature envelope-sampling of intermediate frequency signal in receiver WO2002095962A2 (en)

Priority Applications (3)

Application Number Priority Date Filing Date Title
JP2002592305A JP2004527187A (en) 2001-05-25 2002-05-22 Quadrature envelope sampling of intermediate frequency signals at the receiver
KR10-2003-7001049A KR20030017649A (en) 2001-05-25 2002-05-22 Quadrature envelope-sampling of intermediate frequency signal in receiver
EP02730609A EP1396088A2 (en) 2001-05-25 2002-05-22 Quadrature envelope-sampling of intermediate frequency signal in receiver

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US09/865,236 US20020176522A1 (en) 2001-05-25 2001-05-25 Quadrature envelope-sampling of intermediate frequency signal in receiver
US09/865,236 2001-05-25

Publications (2)

Publication Number Publication Date
WO2002095962A2 true WO2002095962A2 (en) 2002-11-28
WO2002095962A3 WO2002095962A3 (en) 2003-02-13

Family

ID=25345016

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/IB2002/001823 WO2002095962A2 (en) 2001-05-25 2002-05-22 Quadrature envelope-sampling of intermediate frequency signal in receiver

Country Status (6)

Country Link
US (1) US20020176522A1 (en)
EP (1) EP1396088A2 (en)
JP (1) JP2004527187A (en)
KR (1) KR20030017649A (en)
CN (1) CN1463501A (en)
WO (1) WO2002095962A2 (en)

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2005002073A1 (en) * 2003-06-30 2005-01-06 Via Technologies, Inc. Radio receiver supporting multiple modulation formats with a single pair of adcs
WO2007099512A1 (en) * 2006-03-03 2007-09-07 Nxp B.V. Method and apparatus for generating clock signals for quadrature sampling
CN101998459A (en) * 2009-08-27 2011-03-30 中兴通讯股份有限公司 Method and device for measuring single-tone field strength

Families Citing this family (47)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7837116B2 (en) 1999-09-07 2010-11-23 American Express Travel Related Services Company, Inc. Transaction card
US7889052B2 (en) 2001-07-10 2011-02-15 Xatra Fund Mx, Llc Authorizing payment subsequent to RF transactions
US7239226B2 (en) 2001-07-10 2007-07-03 American Express Travel Related Services Company, Inc. System and method for payment using radio frequency identification in contact and contactless transactions
US7172112B2 (en) 2000-01-21 2007-02-06 American Express Travel Related Services Company, Inc. Public/private dual card system and method
US8429041B2 (en) 2003-05-09 2013-04-23 American Express Travel Related Services Company, Inc. Systems and methods for managing account information lifecycles
US8543423B2 (en) 2002-07-16 2013-09-24 American Express Travel Related Services Company, Inc. Method and apparatus for enrolling with multiple transaction environments
US7627531B2 (en) 2000-03-07 2009-12-01 American Express Travel Related Services Company, Inc. System for facilitating a transaction
US7725427B2 (en) 2001-05-25 2010-05-25 Fred Bishop Recurrent billing maintenance with radio frequency payment devices
US7650314B1 (en) 2001-05-25 2010-01-19 American Express Travel Related Services Company, Inc. System and method for securing a recurrent billing transaction
US7119659B2 (en) 2001-07-10 2006-10-10 American Express Travel Related Services Company, Inc. Systems and methods for providing a RF transaction device for use in a private label transaction
US7762457B2 (en) 2001-07-10 2010-07-27 American Express Travel Related Services Company, Inc. System and method for dynamic fob synchronization and personalization
US8538863B1 (en) 2001-07-10 2013-09-17 American Express Travel Related Services Company, Inc. System and method for facilitating a transaction using a revolving use account associated with a primary account
US7493288B2 (en) 2001-07-10 2009-02-17 Xatra Fund Mx, Llc RF payment via a mobile device
US7303120B2 (en) 2001-07-10 2007-12-04 American Express Travel Related Services Company, Inc. System for biometric security using a FOB
US9031880B2 (en) 2001-07-10 2015-05-12 Iii Holdings 1, Llc Systems and methods for non-traditional payment using biometric data
US8294552B2 (en) 2001-07-10 2012-10-23 Xatra Fund Mx, Llc Facial scan biometrics on a payment device
US8960535B2 (en) 2001-07-10 2015-02-24 Iii Holdings 1, Llc Method and system for resource management and evaluation
US20040236699A1 (en) 2001-07-10 2004-11-25 American Express Travel Related Services Company, Inc. Method and system for hand geometry recognition biometrics on a fob
US7996324B2 (en) 2001-07-10 2011-08-09 American Express Travel Related Services Company, Inc. Systems and methods for managing multiple accounts on a RF transaction device using secondary identification indicia
US8548927B2 (en) 2001-07-10 2013-10-01 Xatra Fund Mx, Llc Biometric registration for facilitating an RF transaction
US8635131B1 (en) 2001-07-10 2014-01-21 American Express Travel Related Services Company, Inc. System and method for managing a transaction protocol
US8001054B1 (en) 2001-07-10 2011-08-16 American Express Travel Related Services Company, Inc. System and method for generating an unpredictable number using a seeded algorithm
US7668750B2 (en) 2001-07-10 2010-02-23 David S Bonalle Securing RF transactions using a transactions counter
US8284025B2 (en) 2001-07-10 2012-10-09 Xatra Fund Mx, Llc Method and system for auditory recognition biometrics on a FOB
US9024719B1 (en) 2001-07-10 2015-05-05 Xatra Fund Mx, Llc RF transaction system and method for storing user personal data
US7249112B2 (en) 2002-07-09 2007-07-24 American Express Travel Related Services Company, Inc. System and method for assigning a funding source for a radio frequency identification device
US7827106B2 (en) 2001-07-10 2010-11-02 American Express Travel Related Services Company, Inc. System and method for manufacturing a punch-out RFID transaction device
US7805378B2 (en) 2001-07-10 2010-09-28 American Express Travel Related Servicex Company, Inc. System and method for encoding information in magnetic stripe format for use in radio frequency identification transactions
US7925535B2 (en) 2001-07-10 2011-04-12 American Express Travel Related Services Company, Inc. System and method for securing RF transactions using a radio frequency identification device including a random number generator
US7360689B2 (en) 2001-07-10 2008-04-22 American Express Travel Related Services Company, Inc. Method and system for proffering multiple biometrics for use with a FOB
US7746215B1 (en) 2001-07-10 2010-06-29 Fred Bishop RF transactions using a wireless reader grid
US7705732B2 (en) 2001-07-10 2010-04-27 Fred Bishop Authenticating an RF transaction using a transaction counter
US9454752B2 (en) 2001-07-10 2016-09-27 Chartoleaux Kg Limited Liability Company Reload protocol at a transaction processing entity
US7503480B2 (en) 2001-07-10 2009-03-17 American Express Travel Related Services Company, Inc. Method and system for tracking user performance
US7167513B2 (en) * 2001-12-31 2007-01-23 Intel Corporation IQ imbalance correction
US6805287B2 (en) 2002-09-12 2004-10-19 American Express Travel Related Services Company, Inc. System and method for converting a stored value card to a credit card
US6987953B2 (en) * 2003-03-31 2006-01-17 Nortel Networks Limited Digital transmitter and method
US7136430B2 (en) 2003-03-31 2006-11-14 Nortel Networks Limited Digital receiver and method
CN1957535A (en) * 2004-05-28 2007-05-02 艾利森电话股份有限公司 Digit transducer device
US7318550B2 (en) 2004-07-01 2008-01-15 American Express Travel Related Services Company, Inc. Biometric safeguard method for use with a smartcard
JP4492264B2 (en) * 2004-09-13 2010-06-30 株式会社日立製作所 Quadrature detector and quadrature demodulator and sampling quadrature demodulator using the same
CN100471577C (en) * 2006-01-17 2009-03-25 潘尧钊 Outlet water control mechanism for shower nozzle
DE102006006572A1 (en) * 2006-02-13 2007-08-16 Vega Grieshaber Kg Method of measuring fill level in a container with a pulse propagation time level sensor using intermediate frequency sampling
US8284704B2 (en) * 2007-09-28 2012-10-09 Broadcom Corporation Method and system for utilizing undersampling for crystal leakage cancellation
US9112570B2 (en) * 2011-02-03 2015-08-18 Rf Micro Devices, Inc. Femtocell tunable receiver filtering system
US10833711B2 (en) 2018-12-19 2020-11-10 Silicon Laboratories Inc. System, apparatus and method for concurrent reception of multiple channels spaced physically in radio frequency spectrum
US11171674B2 (en) * 2019-09-18 2021-11-09 Texas Instruments Incorporated Low-complexity inverse sinc for RF sampling transmitters

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE3733967A1 (en) * 1987-10-08 1989-04-27 Bermbach Rainer Quadrature superposition method for demodulating the carrier-frequency received signal in radio clock receivers
US5995556A (en) * 1990-06-06 1999-11-30 California Institute Of Technology Front end for GPS receivers

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6301287B1 (en) * 1995-12-06 2001-10-09 Conexant Systems, Inc. Method and apparatus for signal quality estimation in a direct sequence spread spectrum communication system
US6256477B1 (en) * 1998-09-30 2001-07-03 Conexant Systems, Inc. Avoiding interference from a potentially interfering transmitter in a wireless communication system
US6650264B1 (en) * 1999-03-10 2003-11-18 Cirrus Logic, Inc. Quadrature sampling architecture and method for analog-to-digital converters
US6587530B1 (en) * 2000-10-05 2003-07-01 International Business Machines Corporation Method and apparatus for signal integrity verification
US6639946B2 (en) * 2000-12-01 2003-10-28 International Business Machines Corporation Sigma delta modulator with SAW filter

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE3733967A1 (en) * 1987-10-08 1989-04-27 Bermbach Rainer Quadrature superposition method for demodulating the carrier-frequency received signal in radio clock receivers
US5995556A (en) * 1990-06-06 1999-11-30 California Institute Of Technology Front end for GPS receivers

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
HENTSCHEL T ET AL: "THE DIGITAL FRONT-END OF SOFTWARE RADIO TERMINALS" IEEE PERSONAL COMMUNICATIONS, IEEE COMMUNICATIONS SOCIETY, US, vol. 6, no. 4, August 1999 (1999-08), pages 40-46, XP000849382 ISSN: 1070-9916 *

Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2005002073A1 (en) * 2003-06-30 2005-01-06 Via Technologies, Inc. Radio receiver supporting multiple modulation formats with a single pair of adcs
GB2413908A (en) * 2003-06-30 2005-11-09 Via Tech Inc Radio receiver supporting multiple modulation formats with a single pair of ADCs
GB2413908B (en) * 2003-06-30 2006-06-07 Via Tech Inc Radio receiver supporting multiple modulation formats with a single pair of ADCs
WO2007099512A1 (en) * 2006-03-03 2007-09-07 Nxp B.V. Method and apparatus for generating clock signals for quadrature sampling
CN101998459A (en) * 2009-08-27 2011-03-30 中兴通讯股份有限公司 Method and device for measuring single-tone field strength
CN101998459B (en) * 2009-08-27 2013-03-27 中兴通讯股份有限公司 Method and device for measuring single-tone field strength

Also Published As

Publication number Publication date
EP1396088A2 (en) 2004-03-10
CN1463501A (en) 2003-12-24
JP2004527187A (en) 2004-09-02
US20020176522A1 (en) 2002-11-28
WO2002095962A3 (en) 2003-02-13
KR20030017649A (en) 2003-03-03

Similar Documents

Publication Publication Date Title
US20020176522A1 (en) Quadrature envelope-sampling of intermediate frequency signal in receiver
EP1522151B1 (en) System and method for a direct conversion multi-carrier processor
US6104764A (en) Radio receiving apparatus for receiving communication signals of different bandwidths
EP1249944B1 (en) A subsampling RF receiver architecture
US5619536A (en) Digital superheterodyne receiver and baseband filter method used therein
TWI431966B (en) Ofdm receiving circuit having multiple demodulation paths using oversampling analog-to-digital converter
US5661433A (en) Digital FM demodulator
US8224276B2 (en) Method and arrangement for signal processing in a receiver that can be tuned to different carriers
US20070060077A1 (en) Receiver architecture for wireless communication
US20030072393A1 (en) Quadrature transceiver substantially free of adverse circuitry mismatch effects
KR100736057B1 (en) Dual digital low IF complex receiver
JP2003509909A (en) Phase interpolation receiver for angle modulated RF signals
JP4836041B2 (en) Method and apparatus for sampling an RF signal
CN101207392B (en) Radio communication device
US8102943B1 (en) Variable digital very low intermediate frequency receiver system
US7020221B2 (en) System and method for an IF-sampling transceiver
GB2345230A (en) Image rejection filters for quadrature radio receivers
JPH10215200A (en) Receiver
CN114900405A (en) Acars signal demodulation method based on Soc
KR20060044112A (en) Dc offset suppression apparatus and direct conversion receiving system using it
KR20020063051A (en) Accepting apparatus for code division multiple access base station
JP2002314454A (en) Wireless communication unit

Legal Events

Date Code Title Description
AK Designated states

Kind code of ref document: A2

Designated state(s): CN JP KR

AL Designated countries for regional patents

Kind code of ref document: A2

Designated state(s): AT BE CH CY DE DK ES FI FR GB GR IE IT LU MC NL PT SE TR

WWE Wipo information: entry into national phase

Ref document number: 2002730609

Country of ref document: EP

121 Ep: the epo has been informed by wipo that ep was designated in this application
WWE Wipo information: entry into national phase

Ref document number: 1020037001049

Country of ref document: KR

Ref document number: 028018362

Country of ref document: CN

AK Designated states

Kind code of ref document: A3

Designated state(s): CN JP KR

AL Designated countries for regional patents

Kind code of ref document: A3

Designated state(s): AT BE CH CY DE DK ES FI FR GB GR IE IT LU MC NL PT SE TR

WWP Wipo information: published in national office

Ref document number: 1020037001049

Country of ref document: KR

WWE Wipo information: entry into national phase

Ref document number: 2002592305

Country of ref document: JP

WWP Wipo information: published in national office

Ref document number: 2002730609

Country of ref document: EP

WWW Wipo information: withdrawn in national office

Ref document number: 2002730609

Country of ref document: EP