WO2002004977A2

WO2002004977A2 - Geolocation of telecommunications devices by means of space-based signals processed in a networked computer architecture

Info

Publication number: WO2002004977A2
Application number: PCT/US2001/021878
Authority: WO
Inventors: Peter F. Macdoran; Kenn L. Gold; Mark A. Coffey
Original assignee: Cyberlocator, Inc.
Priority date: 2000-07-12
Filing date: 2001-07-12
Publication date: 2002-01-17
Also published as: AU2001273368A1; WO2002004977A3

Abstract

A method and associated system and apparatus for providing satellite positioning of wireless communication devices meets the needs of wireless communication consumers and service providers at lower operating and manufacturing costs and with reduced form factor requirements.

Description

GEOLOCATION OF TELECOMMUNICATIONS DEVICES

BY MEANS OF SPACE-BASED SIGNALS PROCESSED

IN A NETWORKED COMPUTER ARCHITECTURE

Technical Field:

The present invention relates to the passive reception of space-based radio signals utilizing a simplified sensor that performs digital operations to effectively compress the bandwidth of the space- based signals. The compressed band raw observations are then passed to an associated telecommunications device that is linked to a network-based central processor. This central processor node performs the sophisticated computational operations that determine the geolocation of the telecommunications device. The geolocation information is then transferred to an appropriate network node where geolocation sensitive next-action decisions are made.

A current embodiment of the above-described invention would involve the utilization of signals from the Global Positioning System (GPS). The spread spectrum GPS signals are many bit digitized and input to a digital signal processor (DSP) chip that performs GPS codeless non-linear operations that results in an effective 53 dB of bandwidth compression. The DSP compressed band parameters are then transferred via a cellular or personal communications service (PCS) mobile wireless telephone to an Internet node that completes the processing of the remote sensor GPS compressed band signals. The central processor node ingests the compressed band parameters and converts the sensor raw observations into pseudoranges. A simultaneous parameter estimation procedure allows the derivation of the geocentric or differential separation vector between the cellular phone and a reference terrestrial site. Where the cellular phone user has invoked an E911 call, the central processing node transfers the sensor's geolocation information to an emergency dispatch center for appropriate next action. The form of the geolocation information could be presented in a digital map presentation transferred to an in-vehicle display system. In that way, the most up to date road information can be resident at the E911 dispatcher's facility. Thus, the ever changing road and traffic conditions infoπnation need not be carried as a digital data base in the emergency response vehicle. The simplicity of RF to digital to DSP functions allows for the integration with existing radio and digital processing subsystems that are already a part of the remote wireless telecommunications devices. The dual use of RF receiver /DSP functionality achieves a reduction in cost, saving of power and minimizes form-factor growth while bringing about an important enhancement of telecommunications device capability, E911 regulatory compliance. In an even broader sense, this invention creates a favorable environment for the ubiquitous deployment of a wide range of location- sensitive services and regulatory enforcement options. Background Art:

The use of mobile, wireless communication devices is rapidly expanding worldwide. Wireless telephones are a significant example of wireless communication devices, but wireless communication is quickly spreading to all types of electronic devices such as remote computer terminals that are used to remotely access communication networks, such as the Internet. In many cases, the telephones and the computer terminals are integrated into a single, combined function wireless communication device or "smart device." As the use of wireless communication devices increases, manufacturers and consumers alike are demanding reduced size (i.e., a smaller form factor), the inclusion of more functions such as access to the Internet or other communication networks, lower initial and operating costs (including longer battery life), increased security, and enhanced performance (i.e., better geographical coverage and clearer reception).

In general, wireless communication devices are electronic devices that transmit and receive conversations and/or data using radio waves rather than copper wires or fiber optic cables. In this regard, each device typically includes a radio frequency (RF) antenna to receive and transmit signals in the form of electromagnetic waves. These devices use a RF front end chip for processing the received radio waves (e.g., a mixer for reducing the frequency of the received signal and an analog to digital (AID) converter for converting the signal from analog to digital format and vice versa), and a digital signal processor (DSP). The DSP is a special purpose microprocessor typically placed on a silicon substrate that executes programmed or "hardwired" instructions such as compressing voice, packetizing voice, and converting analog voice to digital. Because the design and manufacture of the DSP chip comprise a significant portion of the cost of the wireless communication device, it is desirable to minimize the cost of the DSP chip and to limit the number and complexity of other microprocessor chips included in wireless communication devices to control costs and the size of such devices.

Adding to the difficulty of producing multifunctional but less expensive and more compact wireless communication devices is the fact that there are several, often incompatible wireless standards that are followed by consumer wireless communication device manufacturers and service providers. Analog cellular operates in the 800 to 900 megahertz (MHz) frequency range, and the operating system for analog is called Advanced Mobile Phone Service (AMPS). Digital cellular shares the 800 to 900 MHz frequency band with analog and is usually available where analog service is offered. Several operating system standards or air interfaces are used to implement digital cellular networks including Code Division Multiple Access (CDMA) and Time Division Multiple Access (TDMA). Personal Communications Service (PCS) is an all-digital service that operates in the 1800 and 1900 MHz frequency band, and PCS networks typically operate with CDMA, TDMA or a global system for mobile communications (GSM). Wireless communication devices generally work on just one of the three operating standards, but some more complex and expensive devices work on both analog cellular and digital cellular networks or on both PCS and analog cellular networks. Digital cellular and PCS devices are becoming more prevalent because they enhance the transfer of information in addition to the voice message which allows the inclusion of features such as caller ID, call waiting, alphanumeric paging, and the like as well as increasing the battery life of the device by reducing the time for data transfer, and data processing, with the included DSP chip performing this digital data processing.

During operation, a user turns on the wireless communication device and it seeks out a signal from the nearest cellular antenna. The antennas are called "cellular" because the antennas are arranged in a honeycomb pattern, and as the user moves around with the wireless communication device, a network computer automatically hands off the user's "call" to the nearest antenna. Each cellular antenna is linked to a mobile telephone switching office (MTSO) which connects the user's wireless call to the local "wired" telephone network. This connection to the wired telephone network can also readily provide access for wireless communication devices to data communication networks such as the Internet.

It is increasingly common for wireless communication devices to include a satellite positioning functionality that unfortunately increases the complexity, size, and cost of such wireless communication devices. Satellite positioning refers generally to the positioning of an object, such as a wireless communication device, through the use of signals transmitted between orbiting satellites and the communications device, for example satellites of the Global Positioning System (GPS), low earth orbiting satellite networks (LEOS), and middle earth orbiting satellite networks (MEOS). GPS satellite positioning is generally based on satellite ranging which allows the position of an object receiving signals from the GPS satellites to be determined by calculating the distance between the object and a group of satellites in space which act as precise frame of reference points. To calculate a position, i.e., latitude, longitude and height, it is necessary to obtain a "fix" on a number of the satellites based on received signals (which are transmitted continuously from the satellites on two frequencies: 1575.42 MHz referred to as the LI signal and 1227.60 MHz referred to as the L2 signal). The signals carry a set of data that includes the satellite's position, the satellite's time measured at transmission of the signal, and a digital sequence known as a pseudo-random noise code (PRN). The GPS receiver at the device uses the PRN code to calculate the location of the device by determining the apparent range or distance from the device to the known position of at least four satellites. By using a simultaneous estimation procedure, the location of the device is derived in terms of the three position parameters (latitude, longitude and height) and a fourth parameter that synchronizes the time within the device's GPS receiver.

A third L-band channel, L5, will be added to future GPS satellites and is expected to operate at a lower frequency than the present L2. The signaling methods will again be of the CDMA spread spectrum type and coherently derived from the same fundamental atomic reference oscillators operating at 10.23 MHz on-board the satellites. The L5 channel is expected to contain only a C/A code modulation. L5 will provide redundancy and increased precision of measurements especially because this band may be broadcast with more power than the present LI or L2 channels. With L5 operating at a frequency below the L2 channel, a phase comparison between the pseudo range as derived at LI C/A and L5C/A will provide the opportunity to measure the ionospheric delay along the line of sight to each of the received satellites. In the past, the L2 channel contained only the P(Y) code so that only the U.S. military or allied forces or users the codeless methods described in U.S. Patent No. 4,797,677 of MacDoran, et al., could derive ionospheric measurements. However, with the openly available C/A code on the L5, code correlating receivers will be able to derive ionospheric calibrations. The codeless methodology here discussed will function equally well and will allow a simplified utilization of the L5 spectrum.

Typically wireless communication devices incorporating GPS location capability have are implemented as stand alone autonomous positioning devices as is required to meet the original military positioning architecture. To achieve the stand alone autonomous capability requires explicit knowledge of the PRN code sequences used in the satellites in order to form the pseudo ranging observables and to decode the satellite orbit information that are required to perform the positioning fix.

A typical implementation for inclusion of a GPS capability into other devices is to introduce specific circuit chips with the required functionality. The functions are a RF chip to bring the L band microwave signals to suitable lower intermediate frequency for the purposes of filtering and amplification. The second specialized chip is a cross correlation processor containing the specific digital codes used by the GPS satellites as published in the U.S. Air Force, Interface Control Document 200 declassified in 1978.

For example, where the remote GPS sensor is a Coarse Acquisition (C/A) code cross correlator (see, Understanding GPS ~ Principles and Applications, E.D. Kaplan, Ed.), the receiver performs as a variety of digital compressions by deriving the pseudo ranges between the remote sensor and all the satellites in view.

Typical commercial receivers use the LI C/A (coarse acquisition) codes that repeat every one millisecond using a code chipping rate of 1.023 MHz. Another digital code known as the P(Y) is cryptographic enabled and provides the Precise Positioning Service. The P(Y) code is transmitted on both the LI and L2 bands and is intended for use by the U.S. military and allied forces. Each of the satellites uses a separate seven-day section of this 264 day long digital code sequence that has a code chipping rate of 10.23 MHz. This cross correlation processor time shifts the code sequence relative to an internal receiver time reference until a match is found between the incoming GPS signals and thereby derives an apparent time of flight from the satellite to the receiver. Multiplying the apparent time of flight by the speed of light gives a pseudo range, pseudo because the receiver clock is not initially synchronized with the satellite clocks.

Once the receiver's model of the satellite code is aligned with that of the incoming signals, the original spread spectrum is despread into a carrier that contains phase modulated telemetry that transfers satellite orbit and clock information required by the receiver in order to complete the positioning function. This final step of the GPS operations often occurs in a third chip that is a reduced instruction set computer that performs input and output/display functions.

While providing desired functionality to wireless communication devices, the addition of a full GPS (or other satellite positioning) receiver to the wireless communication device increases the complexity of the device, adds to the form factor of the device, reduces the battery life, and increases the manufacturing and operation costs for the device.

While initially an unusual feature, satellite positioning is expected to be provided in nearly all wireless communication devices in the near future. In the United States, the driving force behind the design change is the Federal Communication Commission's (FCC) mandated enhancements to the information that cellular service providers are required to transmit to public safety access points (i.e., the people and equipment that handle 911 emergency calls). By October 1, 2001, all cellular service providers must be able to provide the longitude and latitude within 125 meters of the location of wireless communication devices 67 percent of the time. This has led cellular service providers and wireless communication device manufacturers to work together to provide wireless communication devices, such as wireless telephones, with the satellite positioning functionality. As discussed above, the typical positioning implementation approach has been to add GPS technology (i.e., a GPS antenna, a GPS receiver, and support circuitry) to each wireless communication device. The device then transmits the latitude and longitude to the nearest cellular antenna site which then relays the information to the appropriate public safety access point. To reduce the processing required within the wireless communication device, some GPS receiver methods have been designed to be assisted by a server at or associated with the cellular antenna site. For example, U.S. Patent No. 5,999,124 by Sheynblat describes a server-aided system in which satellite signal acquisition parameters and other information are stored and/or calculated at a cell site and then transferred to a cellular telephone. Using this acquisition prompting information, the cellular phone more quickly achieves correlation and in weaker than normal signal conditions and more quickly produces the pseudo range observables for the cellular telephone. The calculated pseudo range value is then transferred to the server via the cellular antenna site for processing into latitude and longitude information. The cellular site prompting provides some reduction in the amount of battery power required for completing satellite signals acquisition and positioning computations. The Sheynblat server-aided system method is a GPS code-dependent system that teaches the addition of a GPS receiver with a code correlator and device resident computations, which results in an increase in the foπn factor (i.e., overall size), some addition power requirements and basic manufacturing costs of the cellular telephone. Additionally, the Sheynblat system must be integrated into the cellular communication technology, employing the mobile telephone control channel, possibly requiring numerous servers (i.e., one at each cellular antenna cell site, or at least regional servers) and associated technology infrastructure to be added in order to provide the 911 location service. Further, the Sheynblat and other systems that provide for at least some processing at the cell site must be configured to work with all wireless multiplexing schemes (CDMA, TDMA, GSM, and others) otherwise the system may be unable to position users who travel outside their wireless service area. There is also an electromagnetic incompatibility concern regarding the introduction of high speed (e.g. 50 MHz) massively parallel cross correlation processor chip into the small form factor of cellular phone devices. In general, code-tracking GPS receivers, and particularly, common C/A code-based receivers, lack a large digital dynamic range and are sensitive to interference and unintentional jamming in urban environments induced by atmospheric and diffraction effects that interfere with the line of sight.

Consequently, there remains a need for a method and associated apparatus for providing satellite positioning of wireless communication devices that meets the needs of wireless communication consumers and service providers while satisfying the FCC mandate. Preferably, such an improved satellite positioning system for wireless communication devices can be provided at lower operating and manufacturing costs and with reduced form factor requirements. Disclosure Of The Invention:

The utilization of the Global Positioning System (GPS) within mobile computer applications and cellular/PCS telephones is becoming a high demand feature for new telecommunications products. The evolution of the GPS from its original military utilization as a system for positioning and navigation has been embraced by the entire civilian user community in substantially the same modes as structured by the military where stand-alone autonomy of operation was a critical system attribute. However, within the context of civilian utilization, stand-alone autonomous operation does not necessarily serve the same civilian/commercial task objectives.

For example, where the GPS device is imbedded into a cellular/PCS phone and the caller needs emergency assistance, it is of little value to the caller to have the handset display the latitude /longitude / height values to inform the handset operator where they are (the caller already knows where they are - they are in trouble). From the perspective of the caller with an emergency, the critical issue is to inform some other entity of the caller's location so that appropriate aid can be dispatched to the caller's location in a timely manner. Thus, in the civilian application of E911, the stand-alone autonomy of a self-contained imbedded full GPS receiver patterned after military applications is not the model that best serves the civilian application. Furthermore, this stand-alone autonomy is accompanied with higher than necessary cost arising from: increased battery power consumption/lesser talk-time, complexity of engineering integration for the manufacturer, increased remote terminal size and the potential necessity of installing a regional supporting infrastructure.

Cellular/PCS phones are an ubiquitous example of mobile telecomm devices whose intrinsic utility derives from being part of a computer based network that bridges the gap between free space wireless and the landline public switched telephone network. In the cellular/PCS networked applications, the location of the remote terminal is made possible by the GPS receiver is often critical to the subsequent actions taken, such as the rendering of aid in the mobile telephone E911 situation. Other examples of the network nodal point location based decision making occurs in asset tracking and with location being the principal element as an authenticating attribute for regulatory enforcement (geographic specific privilege granting or e-commerce taxation) or for computer network security methodology (U.S. Patent No. 5,757,916 of MacDoran, et al.).

In the rendering of emergency aid, asset tracking or location-based authentication, the positional information is required to be present at some network node that needs the information in order to proceed into its next action. For example, the next action for the E911 dispatch center might be for the caller's GPS position information to be available for display on an electronic map that will guide the emergency responders, with high precision, to the location of the in-distress caller. Therefore, if the telecommunications device positioning architecture can be made to be of less cost, simpler and have the device's position determined by a central processing nodal point, the overall user benefits are the same or better with the distributed positioning architecture about to be described. It is to be understood that the concept of a network connection is a completely generalized notion that includes satellite linked information back-hauls into a networked communications infrastructure.

The current technical approaches to adding GPS receiver capability into a remote terminal is to emulate the military modes of stand-alone autonomous operations. This capability addition is typically done by the addition of GPS receiver specific chips, with their supporting circuitry into the remote terminal. The consequences of these chip and circuitry inclusions (ref. www.SiRF.com and www.SnapTrack.com) results in increased bill of materials cost, increased power consumption / reduced time of operation.

Other features and advantages of the invention will become clear from the following detailed description and drawings of particular embodiments of the wireless communication device and codeless positioning system and method of the present invention. Brief Description Of The Drawings:

The accompanying drawings, which are incorporated in and form a part of the specification, illustrate the preferred embodiments of the present invention, and together with the descriptions serve to explain the principles of the invention.

Figure 1 is a functional schematic diagram of a generalized system according to the present invention (generalized geolocation architecture). Figure 2 is a functional schematic diagram of a specialized system according to the present invention (specific E911 Implementation).

Figure 3 is a functional block diagram of an exemplary telecommunications device useful in the system of Figure 1 (elements 14, 16, 18 and 20).

Figure 4 is a functional block diagram of one embodiment of a wireless telecommunications device useful in the system of Figure 2 (elements 53, 54, 56 and 58).

Figure 5 is a block diagram of shared RF architectural elements of Figure 4.

Figure 6 illustrates codeless GPS digital signal processing for multiple data-types generation.

Figure 7 is a software processing data flow within the central processing node.

Best Mode for Carrying Out the Invention:

1. Background of the GPS

For background of the Global Positioning System and its relationship to the art of codeless GPS methods, the reader is referred to U.S. Patent No. 4,797,677 of MacDoran, et al. (incorporated herein by this reference) and U.S. Patent No. 5,757,916 of MacDoran, et al. (incorporated herein by this reference).

2. Overview of the Invention

The preferred embodiment as described in this invention is to achieve a high degree of spectral compression by utilizing certain enhancements to the codeless GPS methods as described U.S. Patent No. 4,797,677 of MacDoran, et al. The motivation for this methodology is to achieve the advantageous architectural characteristics of:

(a) Reduced complexity within the remote terminal because the sophisticated data handling is performed at a network node.

(b) Where the telecommunications device is a wireless mobile platform, to achieve shared use of radio frequency (RF) and digital signal processor (DSP) subsystems functions in order to implement satellite signal sensor operations together with the communications functions.

(c) Near instantaneous (<1 second from power-up) remote terminal raw data acquisition without external information input to enable receiver functions.

(d) Lower power requirements to achieve satellite signal reception capability because of shared RF and DSP chips.

(e) Ability to calibrate the ionospheric errors in the position determination because of the GPS codeless acquisition of the two bands broadcast by the GPS satellites (usually only available to U.S. military users).

(f) Design options to implement many bits to digitize the GPS signals in order to achieve a larger dynamic range and tolerance to GPS in-band interference compared to the usual one or two bit signal processing done in conventional GPS receivers.

(g) Centralized sophisticated data processing of raw remote terminal observations enabled through network connections reduces the need for sophistication in the handset. (h) Central processing can perform either single receiver point geocentric positioning with 20 meter accuracy or full differential positioning relative to known locations , with accuracy of a few meters to even millimeter precision.

(i) The remote terminal raw data is difficult to impossible to forge which can be of significant importance when needing to establish, with high confidence, the actual remote terminal's location as in the case of location-based authentication for computer network transactions or the tracking of hazardous cargo shipments (i.e., nuclear materials transport).

(j) Small to moderate data transfer volumes, approximately 200 to 1000 bytes.

(k) The distributed network architecture allows this GPS sensor methodology to function anywhere on Earth. There is no need to have a deployed server-assist capability in place in order to prompt the remote receiver into acquiring either ordinary, unobstructed sky view or weak signals.

The methodology of this codeless GPS technology works by digitally sampling the signals arriving at the GPS antenna and then performing non-linear operations on those GPS spread spectrum signals. This unconventional, non-linear architecture has the effect of reconstructing a signal that contains the Doppler and phase information for all satellites in view, all without any knowledge of the codes used by the GPS satellites (U.S. Patent No. 4,797,677 of MacDoran, et al.). The codeless despreading of the GPS spread spectrum in effect becomes a data compression implementation in order to same channel communication capacity.

All of the sophisticated processing is performed at a central site, perhaps well removed from either the remote terminals or network interconnection paths. The central processing site determines the geocentric location of the remote terminal or the separation between the remote terminal and the cellular service site and then passes that information to the 911-service center for use by the emergency service response team. 3. Detailed Description of the Invention

Figure 1 illustrates the Generalized Geolocation Architecture of the invention. Consider the existence of a space-based constellation of radio signal sources 10. These sources of radio signals may be based upon Earth-orbiting satellites, or perhaps high altitude aircraft or balloon platforms. These radio signal sources transmit into the free space, 12, and are received by at least one codeless sensor 14 and perhaps simultaneously at 28 and other locations. Without benefit of the explicit knowledge of the space-based signal modulation form, the codeless sensor performs a compression of the signal bandwidth and foπns a set of compression parameters (CPs) 16 that are transferred into an associated telecomm device 18. These CPs are transferred via a link 20 into a networked communications infrastructure (NCI) 22 with the data 24 addressed to a Central Processing Node (CPN) 26. In order to facilitate the possibility of high accuracy baseline vector formation, a second simultaneously receiving codeless sensor 28 is used to input data to the CPN, 26. A source of time information 30 and satellite orbit information 32 is also input to the CPN 26. Within the CPN is a processor to convert the telecomm device CPs into conventional signal characteristics 34 suitable for input to the baseline vector estimation processor 36. By processing the telecomm device data alone, the geocentric geolocation of the device can be determined or by knowing the geocentric location of the CPN's sensor, the derived baseline vector can be represented in a geocentric form of latitude, longitude and height 38. The telecomm device geolocation can then be linked 40 back into the NCI 22 which routes that geolocation information to a Location-Sensitive Next-Action Decision Processor 44. The Next- Action processor 44 authenticates, based upon the telecomm device's geolocation, that the operator of the telecomm device is entitled to further privileges, information and/or services that is then authorized via instruction 46. The location-sensitive final action processor 48 uses the instruction 46 for a variety of functions such as the verification of regulatory compliance, location- based authentication network security, the allocation and directing of emergency services or the dispensing of commercial services.

Figure 2 illustrates the Specific E911 Implementation architecture of the invention. Consider the presence of radio signals from the Global Positioning System satellite constellation 50 consisting of 28 satellites as of mid-2000. These satellites are continuously transmitting signals into the free space, 52, that is received by a shared antenna 53 with the GPS signal routed through the shared RF subsystems of the cellular/PCS phone 58. The codeless sensor 54 digitizes the RF signals and inputs the samples into the shared digital signal processor (DSP) of the cellular phone. The codeless processing of the GPS signals proceeds without benefit of the explicit knowledge of the pseudo random noise digital sequence used to modulate the GPS satellite transmitters using methods detailed in U.S. Patent No. 4,797,677 of MacDoran, et al. in order to form the set of bandwidth compression parameters CPs 56 The CPs are transferred into the cellular phone 58 for transfer via antenna 53 into a wireless link 60 that is received by the mobile telephone switching office (MTSO) 61. The MTSO has a direct link into the Internet 62. These CPs of the cellular/PCS phone are Internet Protocol (IP) addressed to the Central Processing Node (CPN) 66. In order to facilitate the possibility of high accuracy baseline vector formation, a second simultaneously receiving codeless sensor 68 is used to input data to the CPN, 66. A second GPS receiver using the commonly available C/A code 70 is used as a source of Universal Time Coordinated (UTC) time information 71 and satellite orbit information 72 is also input to the CPN 66. Within the CPN is a processor to convert the cellular/PCS phone CPs into conventional signal characteristics of amplitude, frequency and phase 74. From conventional signal characteristics are derived the satellite specific observables of Doppler shift, phase ranging that allows the deduction of which satellites are being received. Once the satellites are identified the pseudo ranges 75 can be formed as if the codes to the GPS signals were available. These pseudo ranges are then input to the baseline vector estimation processor 76. By processing the cellular/PCS data alone, the geocentric geolocation of the device can be determined. A full differential GPS (DGPS) processing is possible by knowing the geocentric location of the CPN's codeless sensor 68 and deriving the baseline vector can be represented in a geocentric form of latitude, longitude and height 77. The cellular/PCS phone geolocation can then be linked 80 back into the Internet via 82 which routes that geolocation information to the E911 Emergency Dispatcher 83. The dispatch facility authenticates, based upon the cellular/PCS phone geolocation, that the cellular phone is entitled to further privileges and what jurisdiction has the responsibility. Once the E911 caller is authenticated, the directive for emergency response is issued 86 via the MTSO 61 for relay via a wireless link 86 to the emergency response resources 88. The response directive could be in the form of an in-vehicle digital map display to guide the emergency team to the E911 caller.

In Figure 3, a generalized telecommunications device is illustrated with signals arriving from a space-based environment 12, which are initially intercepted by an antenna that is comiected to a radio frequency to intermediate frequency (RF to IF) stage 100. Element 100 amplifies and may also heterodyne down-convert the space-based signals into a form suitable for analog to digital conversion 102 using a frequency reference oscillator 104 that is common to the frequency conversion operations of the RF to IF stage and the AD/C sampling in 102. Element 102 is possibly an intrinsic part of the telecommunications' device central processing unit (CPU) 106 and has specific digital processing (DSP) functionality integrated into the CPU chip. The CPU/DSP element 106 performs the codeless signal processing that generates spectral compression parameters 108 that are passed to the voice and data input/output subsystem 110. Voice and /or data (compression parameters 108) are then transferred to the connectivity subsystem 112. Element 112 may be either cable connected or wireless (radio, infrared or possibly acoustic) and may have an antenna element 114 that could be shared with element 100.

Figure 4 illustrates a GPS specific wireless telecommunications device embodiment of generalized case illustrated in Figure 3. A GPS antenna 140 receives signals arriving from a space- based environment 12, and is connected to radio frequency to intermediate frequency (RF to IF) stage 140. Element 140 amplifies and heterodyne down-convert the GPS wide band (20 MHz) signals into an IF suitable for analog to digital conversion 154. A frequency reference oscillator 148 is shared in common with the frequency conversion operations of the RF to IF stage, the AD/C sampling in 154 and the wireless RF receiver/transmitter subsystem 146 connected wireless communications antenna 144 which be combined with the GPS antenna 140.

The 8 bit AD/C values formed by 154 are input for codeless data process that is a portion of the combined DSP functionality of element 150 that is shared with the other conventional telecommunications operations of wireless transmitting and reception controller 160, digital audio encoding and decoding 162, The DSP element 156 performs the codeless signal processing that generates spectral compression parameters that are passed to the receive/transmit subsystem 160. The compression parameters 157 are then transferred to the wireless subsystems 162, 160, 146 and 144 for transfer of data to the MTSO and Internet per Figure 2.

Given the availability of space-based radio signal sources (i.e., GPS, GLONASS, GNSS/Galileo, mobile phone communications satellites or commercial broadcast satellites), it is possible by the methods to be described to derive the geolocation of simplified sensors without the explicit knowledge of the signaling methodology employed within the satellites. Codeless Sensor Characteristics:

(a) An antenna (shared with telecomm device, if a wireless system)

(b) Microwave filter, low noise amplifier (LNA), mixer/local oscillator (LO) (shared with telecomm device).

(c) Intermediate frequency (IF) amplifier/automatic again control (AGC) to drive an analog to digital converter (AD/C) with a preferred 8 bit conversion resolution at a 46 mega-sample per second (Msps) rate with sampling epochs derived from the LO reference oscillator.

(d) Output of the AD/C sampling into a digital signal processor (DSP) chip (shared with telecomm device). The DSP performs non-linear operations to form a superposition of sinewaves one for each satellite, along with their associated Doppler shifts. There are no spectral line cross products because the digital PRN code sequences used in a code division multiple access (CDMA) satellite constellation have low cross correlation properties (nearly orthogonal code space occupations, see, Spread Spectrum Systems by R.L. Dixon) by their fundamental design. The array of non-linear operations upon the wide band satellite signals involves (U.S. Patent No. 4,797,677 of MacDoran, et al.):

Figure 5 illustrates shared RF subsystem components with the wireless telecommunications device shown in Figure 4. Of particular interest is the ability to share the reference oscillator signal 148 that for a cellular/PCS phone design is at a frequency of 1750 MHz. The telecommunications device shared antenna is routed to a RF power divider 180 to splits the signal power into two parts for subsequent LI filtering in 184 for a band centered at 1575.42 MHz +/- 10 MHz. An amplifier stage 186 provides adequate power to a mixer stage 188 that results in an IF center frequency of 175.58 MHz that is filtered into bandpass +/- 10 MHz 190 and amplified again in 192 for subsequent processing in the shared DSP. The L2 signal path from the power divider 180 is to a filter 196 with a bandpass +/- 10 MHz centered at 1227.6 MHz. An amplifier stage 198 provides adequate power to a mixer stage 200 that results in an IF center frequency of 522.4 MHz that is filtered into bandpass +/- 10 MHz 202 and amplified again in 204 for subsequent processing in the shared DSP. The coherence between these two IF signals is constrained by the fact that a common local oscillator has been used in the creation of the IF signals. Figure 6 illustrates a block diagram implementation of the digital signal processing that is performed within the telecomm device that will generate a variety of data-types of differing effective wavelength. It is important to recognize that there exists a fixed numerical relationship between these various data-types. The relationship begins with the fundamental base frequency of the GPS frequency architecture that is 10.23 MHz or f₀. Thus, the chipping frequency of the P(Y) modulation is f₀, the chipping frequency of the C/A modulation is f_o/10, the LI suppressed carrier frequency is 154 x f_0ι the L2 suppressed carrier frequency is 120 x f₀, the second harmonic recovery of the LI suppressed carrier (2L1) is 308 x f₀, the cross correlation between the LI and L2 bands (LI x L2) produces sinewaves at the difference frequency (L1-L2 = 347.82) that is exactly 34 x f₀.

In Figure 6, the processing begins with digitizing the 175 MHz centered IF signal by performing a multi-level (i.e., 8 bit) analog to digital conversion (AD/C) in 300 with a sampling signal 304 of 46 MHz (a sample every 21.7 nanosecond). This sampling rate has the effect of an alias down conversion from an IF band of 175.42 +/- 10 MHz to a digital band extending from 0.58 MHz to 20.58 MHz.

(a) The digitized representation of the LI spread spectrum 305 is then bandpass filtered in 306 to optimize the useful signal input to the squaring operation of 308 that has the property of recovering the satellite modulation suppressed carrier second harmonic satellites (for GPS 2L1 a 95 mm wavelength). The output of the squaring stage is bandpass limited to approximately 30 kHz 310 in order to encompass all the physics of the satellites on the line of sight and the possible movement of the codeless sensor (Earth's rotation or possible dynamical effects). For the typical sensor on the Earth at a fix location, the maximum satellite range rate is approximately 800 meter per second. With an effective 2L1 wavelength of 95 mm per cycle, the satellite motion will create a Doppler shift of +/- 8.4 kHz. The reference oscillator used for the analog down conversion and the 46 MHz AD/C of the IF also has an effect upon the location of the spectral lines created in the squaring operation. For a 1 part per million (PPM) frequency accuracy of a TCXO, the effective 2L1 frequency, 3150 MHz, will shift by 3.1 kHz. The combined satellite apparent motion and the reference oscillator is then 11.5 kHz for a total bandwidth effect of 23 kHz which is rounded up to be 30 kHz. Element 312 is a down- sample operation by the factor M that optimizes the digital data stream from the bandpass filter 310. In the time domain, the output of 310 is digital data that represents a superposition of sinewaves, one for each satellite, while the corresponding frequency domain representation is of spectral lines, one line for each satellite. Element 314, Peak Model, can be considered as a model of the spectral lines present in the incoming data when algebraically subtracted in a summing operation in element 316. The criterion for a matched spectrum is that the output of 316 is white noise. The set of filter taps from the element 314 that satisfies the resultant white noise criterion constitutes the phase locked tracking condition of the digital signal processing and is designated as 318, the compression parameters for the second harmonic of LI, CPs2Ll.

(b) The digital data stream 305 is now processed in a delay (ideally ^X chip time, 49 ns, however, two sample times of approximately 44 ns is available) 320 and multiply manner 322 in order to recover the chipping frequency 10.23 MHz of the satellite pseudo random noise sequence generator. The output of the delay/multiply stage is bandpass limited to approximately 100 Hz 324 in order to encompass all the physics of the satellites on the line of sight and the possible movement of the codeless sensor (Earth's rotation or possible dynamical effects). For the typical sensor on the Earth at a fix location, the maximum satellite range rate is approximately 800 meter per second. With an effective P(Y) code chip wavelength of 29.3m per cycle, the satellite motion will create a Doppler shift of +/- 27 Hz. The reference oscillator used for the analog down conversion and the 46 MHz AD/C of the IF also has an effect upon the location of the spectral lines created in the delay/multiply operation. For a 1 part per million (PPM) frequency accuracy of a TCXO, the effective L1P(Y) frequency, 10.23 MHz, will shift by 10 Hz. The combined satellite apparent motion and the reference oscillator is then 37 Hz for a total bandwidth effect of 74 Hz which is rounded up to be 100 Hz. Element 326 is a down-sample operation by the factor N that optimizes the digital data stream from the bandpass filter 324. In the time domain, the output of 324 is digital data that represents a superposition of sinewaves, one for each satellite, while the corresponding frequency domain representation is of spectral lines, one line for each satellite. Element 328, Peak Model, can be considered as a model of the spectral lines present in the incoming data when algebraically subtracted in a summing operation in element 330. The criterion for a matched spectrum is that the output of 330 is white noise. The set of filter taps from the element 328 that satisfies the resultant white noise criterion constitutes the phase locked tracking condition of the digital signal processing and is designated as 332, the compression parameters for L1P(Y).

(c) The digital data stream 305 is now bandpass filtered in 336 to optimize the sideband power for the recovery of the C/A portion of the GPS spectrum. The output of 336 is processed in a delay (by Vi chip time, 490 ns) 338 and multiply manner 3 in order to recover the chipping frequency 1.023 MHz of the satellite pseudo random noise sequence generator for the C/A channel . The output of the delay/multiply stage is bandpass limited to approximately 10 Hz 342 in order to encompass all the physics of the satellites on the line of sight and the possible movement of the codeless sensor (Earth's rotation or possible dynamical effects). For the typical sensor on the Earth at a fix location, the maximum satellite range rate is approximately 800 meter per second. With an effective P(Y) code chip wavelength of 293m per cycle, the satellite motion will create a Doppler shift of +/- 2.7 Hz. The reference oscillator used for the analog down conversion and the 46 MHz AD/C of the IF also has an effect upon the location of the spectral lines created in the delay/multiply operation. For a 1 part per million (PPM) frequency accuracy of a TCXO, the effective L1P(Y) frequency, 1.023 MHz, will shift by 1 Hz. The combined satellite apparent motion and the reference oscillator is then 3.7 Hz for a total bandwidth effect of 7.4 Hz which is rounded up to be 10 Hz. Element 344 is a down-sample operation by the factor P that optimizes the digital data stream from the bandpass filter 342. In the time domain, the output of 342 is digital data that represents a superposition of sinewaves, one for each satellite, while the corresponding frequency domain representation is of spectral lines, one line for each satellite. Element 346, Peak Model, can be considered as a model of the spectral lines present in the incoming data. These modeled data are algebraically subtracted in a summing operation in element 348. The criterion for a matched spectrum is that the output of 348 is white noise. The set of filter taps from the element 346 that satisfies the resultant white noise criterion constitutes the phase locked tracking condition of the digital signal processing and is designated as 350, the compression parameters for L1C/A.

Because the method described is codeless, it becomes possible to implement an architecture that extracts additional signal power for the sensor as compared with a conventional C/A code correlating receiver. For example, in the case of a conventional receiver architecture there is a filter that limits the C/A spread spectrum signal bandwidth is approximately 2 MHz that contains only the central lobe of the spread spectrum resulting from the PRN sequence whose chipping frequency is 1.023 MHz. As illustrated in the GPS background publication by JJ. Spilker, from the Institute of Navigation, the C/A modulation bandwidth is substantially wider than 2 MHz with sidelobes extend +/- 10 MHz, although with ever diminishing power in the sidelobes. The satellite transmitter exciter stage to the final power amplifier band limits the signals to be amplified and transmitted to only the +/- 10.23 MHz central lobe of the P(Y) channel spread spectrum.

By setting the bandwidth to 10 MHz, the majority of the P(Y) channel power of LI is present and also includes the C/A central lobe and +/- five sidelobes of the C/A modulation. Thus, when the squaring operation to recover the 2L1 data type occurs, more signal power by approximately 1.7 dB will be available. Because of the squaring operation also squares the noise, the ability to input more actual satellite signal power will actually benefit the signal to noise ratio amplitude by approximately 3.4 dB available to the digital tracking loops.

(d) The digital data streams from LI and L2 are now processed in a cross-band manner. A device resident in-siru version of this cross correlation method is described in: "Method and Apparatus for Calibrating the Ionosphere and Application to Surveillance of Geophysical Events," MacDoran, 1984, and U.S. Patent No. 4,463,357 of P.F. MacDoran (incorporated by reference herein). The L2 channel IF signal is 8 bit digitized in the AD/C stage 301 and then multiplied with the digitized LI channel 305. The data-type produced by the cross correlation of the LI P(Y) and L2 P(Y) wide band signals for a combined direct measurement of the ionosphere along the line of sight to each of the satellites being received. In addition, a sinewave results from this cross correlation because although the modulating P(Y) channel codes onto the LI and L2 carriers imposed by the satellite transmitter are fully phase coincident, the fact that the sensor cross correlation process is not an exact modeling of the other channel causes signal of LI -L2 (347.82 MHz) to exist with a wavelength of 86 cm. The (L1-L2) sinewave with the associated Doppler shift exists for each satellite received. Because of electromagnetic dispersion effects upon the wide band group velocity, the L1P(Y) signals will arrive before the L2P(Y) and the amplitude of the resultant (L1-L2) sinewave for each satellite will be diminished. However, by delaying the LI signals, in increments of 22 nanoseconds, a maximum amplitude for each satellite sinewave can be determined. The magnitude of the digital delay is a direct measure of the ionospheric delay along the line of sight to the satellite received. For each satellite, the frequency difference between the measured (L1-L2) sinewave and the nominal rest frequency of 347.82 MHz is a direct measurement of the Doppler shift along the particular line of sight combined with the reference oscillator frequency offset. The offset frequency will be determined by a simultaneous estimation procedure to be described. In Figure 6, element 354 is used to introduce a delay into the LI digital data stream. As will be discussed in the section dealing with the ionosphere, the maximum delay difference between LI and L2 is estimated to be equivalent to 750 ns. Given an AD/C sampling frequency, there is a sample every 22 ns. To cover the 750 ns delay span, the delay span in 354 will be from 0 to 34 lags. The delayed LI data stream from 354 is multiplied with the L2 data stream 302 in element 356. This operation is a codeless GPS (LlxL2) cross correlation operation with the LI delay value that yields the maximized amplitude of the 347.82 MHz sinewave being a direct measure of the ionospheric delay for the particular satellite at the time of the measurement.

The output of the multiplication stage 356 is bandpass limited to 3 kHz in element 358 in order to encompass all the physics of the satellites on the line of sight and the possible movement of the codeless sensor (Earth's rotation or possible dynamical effects). For the typical sensor on the Earth at a fix location, the maximum satellite range rate is approximately 800 meter per second. With an effective L1-L2 wavelength of 86 cm per cycle, the satellite motion will create a Doppler shift of +/- 34 times the 27 Hz that would be observed in the P(Y) channel or 918 Hz. The reference oscillator used for the analog down conversion and the 46 MHz AD/C of the IF also has an effect upon the location of the spectral lines created in the codeless cross correlation operation. For a 1 part per million (PPM) frequency accuracy of a TCXO, the effective L1-L2 frequency, 347.82 MHz, will shift by 347 Hz. The combined satellite apparent motion and the reference oscillator is then 1265 Hz for a total bandwidth effect of 2530 Hz which is rounded up to be 3 kHz. Element 360 is a down- sample operation by the factor P that optimizes the digital data stream from the bandpass filter 358. In the time domain, the output of 358 is digital data that represents a superposition of sinewaves, one for each satellite, while the corresponding frequency domain representation is of spectral lines, one line for each satellite. Element 362, Peak Model, can be considered as a model of the spectral lines present in the incoming data when algebraically subtracted in a summing operation in element 364. The criterion for a matched spectrum is that the output of 364 is white noise. The set of filter taps from the element 362 that satisfies the resultant white noise criterion constitutes the phase locked tracking condition of the digital signal processing and is designated as 366, the compression parameters for (L1-L2) and 368 for the LI ionospheric delay values that maximum the amplitude of the LI x L2 function.

Another approach the measurement of the ionosphere is determine the phase shift between the L1P(Y) and L2P(Y) chipping frequencies. Thus, the L2P(Y) data-type must be created. The digital data stream 302 is now processed in a delay (ideally ^lA chip time, 49 ns, however, two sample times of approximately 44 ns is available) 370 and multiply manner 372 in order to recover the chipping frequency 10.23 MHz of the satellite pseudo random noise sequence generator. The output of the delay/multiply stage is bandpass limited to approximately 100 Hz 374 in order to encompass all the physics of the satellites on the line of sight and the possible movement of the codeless sensor (Earth's rotation or possible dynamical effects). For the typical sensor on the Earth at a fix location, the maximum satellite range rate is approximately 800 meter per second. With an effective P(Y) code chip wavelength of 29.3m per cycle, the satellite motion will create a Doppler shift of +/- 27 Hz. The reference oscillator used for the analog down conversion and the 46 MHz AD/C of the IF also has an effect upon the location of the spectral lines created in the delay/multiply operation. For a 1 part per million (PPM) frequency accuracy of a TCXO, the effective L1P(Y) frequency, 10.23 MHz, will shift by 10 Hz. The combined satellite apparent motion and the reference oscillator is then 37 Hz for a total bandwidth effect of 74 Hz which is rounded up to be 100 Hz. Element 376 is a down-sample operation by the factor R that optimizes the digital data stream from the bandpass filter 374. In the time domain, the output of 374 is digital data that represents a superposition of sinewaves, one for each satellite, while the corresponding frequency domain representation is of spectral lines, one line for each satellite. Element 378, Peak Model, can be considered as a model of the spectral lines present in the incoming data when algebraically subtracted in a summing operation in element 380. The criterion for a matched spectrum is that the output of 380 is white noise. The set of filter taps from the element 378 that satisfies the resultant white noise criterion constitutes the phase locked tracking condition of the digital signal processing and is designated as 382, the compression parameters for L2P(Y).

DSP performs closed loop tracking of each sinewave and outputs phase locked tracking parameters (filter taps) that effectively makes this operation a method of bandwidth compression. The estimated maximum (12 satellites) digital volume required to specify the filter taps is 200 bits per data-type times six data-type [2L1, LI x L2, LI - L2, L1C/A, L1P(Y), L2P(Y)] equals 1200 bits per update. Assuming an update rate of once per second (higher rates such as 10 Hz are also possible, however, the statistical independence of rapidly acquired measurements must be considered), the maximum data rate to be transferred to the central processing node (CPN) will be 1200 bps. Assuming the reception of 12 satellites, each transmitting a bandwidth of 20 MHz, the effective bandwidth compression is given as the ratio of 12 x 20 MHz = 240 MHz compressed into 1200 bps is equivalent to 53 dB of compression. Compression parameters can be buffered over an interval (seconds to minutes to perhaps hours) and are then available for transfer via the telecommunications device to the networked communications infrastructure for routing to a central processing node. The Determination of the Geolocation of Telecommunication Devices Central Processing Node (CPN)

Figure 7 describes the software processing flow at the Central Processing Node.

(a) The telecomm device compression parameters are received by the CPN at a nominal rate of 1200 bps for an update rate of once per second.

(b) CPN time tags the initial set of arriving compression parameters from both the telecomm device 401 and its own locally resident codeless radio signal sensor 403 with subsequent compression parameter sets being of known time offsets because of the tracking loops are interrogated at regular intervals (i.e., once per second).

It is preferred that the filter tap compression parameters remain in their original raw form as computed by the telecomm device DSP for two reasons. Firstly, in keeping with the original system design approach, keeping the remote device as simple as possible form and secondly to further raise the barrier to any adversary attempting a spoofing attack in a security application of this technology. The CPN processes these compression parameters into typical signal characterizations of amplitude, frequency and phase 405, 407(constant, and higher order derivatives), a data quality factor (analogous to the static phase error in closed-loop tracking). The implementation of the transformation of the compression parameters into signal characteristics is described in a text such as Introduction to Digital Signal Processing, by John G. Proakis and Dimitris G. Manolakis, Macmillan Publishing Company, 1988, chapter 11.3.

The array of time varying sinewaves can then be transformed into a power spectrum that has multiple spectral lines, one for each satellite. It then remains to deduce a correspondence between the spectral lines and the individual satellites.

(c) A conventional code correlating receiver (typically a GPS C/A code receiver) provides time information (Universal Time Coordinated, UTC) as well as orbit elements for the space-based radio signal sources illuminating the code correlating receiver and codeless sensors.

(d) Based upon the UTC time tags on the compression parameters and a crude knowledge of the region of the Earth at which the telecomm device may be situated, a model of spectral lines expected from the DSP non-linear operations can be generated (one spectral line per satellite for each data type employed).

(e) The spectral lines from the telecomm device codeless sensor are compared with the modeled spectrum in order to identify which spectral line is to be associated with individual satellites expected to be available to the sensors 409.

The satellite identification search strategy may involve a cross-spectral matching in a least squares sense that would also yield an initial estimate of the reference oscillator frequency offset.

Identification of satellites could also be done by forming an array of differences between the spectral lines (this approach eliminates the effect of the codeless oscillator having a frequency offset).

Where only a few satellite signals may be available, a span of data can be utilized to form the time derivative of the available data types, especially the second harmonic of the suppressed carrier, in order to distinguish between possible satellites. This approach eliminates the reference oscillator offset but leaves sensitivity to the higher order terms in the ID process. Once each of the spectral lines has been associated with the appropriate satellite, the simultaneous estimation procedure can begin in order to derive the geolocation of the telecomm device. The geolocation can be performed either in a geocentric coordinate frame (this could have a security concern because spoofing by means of a satellite signal simulator is a possibility). A more accurate method of location is the full differential satellite signals processing that produces a differential baseline vector relative to the CPN or regional reference sites available to the CPN (this allows for world-wide service coverage). The GPS solutions in the processing software make use of a data type formed by differencing ranges observed at the Central Processing Node, with those observed at the wireless communications device. Common mode differences 413 (Orbit Error, atmospheric effects, etc) are either completely removed, or mitigated based on actual baseline separation. The mathematical manipulations used to invert range measurements into position are formulated to use these differential range measurements, and to produce an offset between the CPN and device locations.

The software filter formulation used at the CPN is sequential (Bierman, 1976). In a sequential filter, data from many epochs are used, however the data is applied to the filter in time sequence, and model updates are performed on a sequential and on-going basis. It is possible to deal with time varying parameters that may or may not change according to known dynamics. These include relative clock offset between the two receivers.

(f) The motion of a satellite and other parameters affecting measured range are governed by a complicated set of non-linear second order differential equations. If a reference model is available which remains close enough to the true range throughout the time of interest, the filtering problem can be linearized thus allowing the use of least squares methods (Tapley, et al., 1991). The set of observed ranges is differenced with a set of ranges based on the mathematical models of all of the parameters affecting range. These residuals should fall within the linear regime, assuming that the model accurately describes what is happening. The orbit trajectory can also be expanded in a Taylor series about each point

(g) The processing proceeds by deriving the distance between the sensor and each of the satellites received at the sensor. The phase locked tracking of the suppressed carrier second harmonic produces a relatively sensitive and unambiguous set observables. The effective frequency of this observable is 2L1 = 2 x 1575.42 = 3150.84 MHz with a wavelength of 95 mm. The maximum partial derivative sensitivity of this observable is 308 times the sensitivity of the P(Y) channel chipping frequency, 308 x 5 micro Hertz per meter (uHz) = 1540 uHz/m = 1.5 mHz/m. Given four satellites reasonably well spaced across the sky (under certain conditions, the number of satellites simultaneously to the sensors could be as many as 12), the geometric dilution of precision (GDOP) may be a factor of four, results in an effect sensitivity of approximately 0.38 mHz/m.

Consider now the ability to phase connect the 2L1 data-type over an interval of five seconds and with a signal strength sufficient to meet the minimal criterion for phase-locked tracking, one radian RMS phase tracking noise. The tracking loop time constant is a few seconds and the tracking loop compression parameters are acquired every one second so as to assure connected phase conditions. The effective precision of the frequency measurement is approximated by the phase noise (1 radian = 57.3 degrees = 0.159 cycle) divided by the time interval (5 seconds) = 0.0318 Hz =31.8 mHz. Scaling by the sensitivity of 0.38 mHz/m yields a positioning precision of approximately 83.6 m. The intervening measured phase values at epochs 2, 3, 4 seconds provides additional precision by a factor of the square root of the degree of over-determinedness or the square root of three, a factor of 1.7 making the effective positioning precision of 49 meters.

The C/A chipping frequency with a wavelength of 293 meters can now be exploited as an unambiguous observable 415 because the 2L1 data-type discussed in (b.) allows the ambiguity resolution to within one-sixth of a C/A wavelength. Consider the DSP closed loop tracking of the C/A chipping frequency to be at a nominal signal to noise ratio of 30 dB-Hz resulting in a 0.03 radian RMS phase noise that is equivalent to 0.005 cycle x 293 m/cy = 1.5 meters. Assuming a GDOP of four, the positioning accuracy is expected to be 6 meters.

The P(Y) chipping frequency with a wavelength of 29.3 meters can now be exploited as an unambiguous observable 417 because the C/A chipping frequency data-type discussed in (c.) allows the ambiguity resolution to within one-fifth of a P(Y) wavelength. Consider the DSP closed loop tracking of the P(Y) chipping frequency to be at a nominal signal to noise ratio of 30 dB-Hz resulting in a 0.03 radian RMS phase noise that is equivalent to 0.005 cycle x 29.3 m/cy = 15 cm. Assuming a GDOP of four, the positioning accuracy is expected to be 60 centimeters. Processing of additional P(Y) channel data that are statistically independent can improve the measurement precision. Consider ten samples over a 100 second interval. The position solution 419 precision will improve by a factor of the square root of 10 or 3.3 for an overall positioning precision of 18 cm.

The data-type produced by the cross correlation of the LI P(Y) and L2 P(Y) wide band signals for a combined direct measurement of the ionosphere along the line of sight to each of the satellites being received. In addition, a sinewave results from this cross correlation because although the modulating P(Y) channeLcodes onto the LI and L2 carriers imposed by the satellite transmitter are full phase coincident, the fact that the sensor cross correlation process is not an exact modeling of the other channel causes signal of LI -L2 (347.82 MHz) to exist with a wavelength of 86 cm. This (L1-L2) data-type can now be exploited as an unambiguous observable because the P(Y) chipping frequency data-type discussed in (d.) allows the ambiguity resolution to within one-fifth of a L1-L2 wavelength. Consider the DSP closed loop tracking of the L1-L2 frequency to be at a nominal signal to noise ratio of 30 dB-Hz resulting in a 0.03 radian RMS phase noise that is equivalent to 0.005 cycle x 86 cm/cy = 4.3 mm. Assuming a GDOP of four, the positioning accuracy is expected to be 1.7 centimeters.

Ionospheric Effects

The total electron columnar content of the ionosphere can be a major limiting error source depending on the nature of the location-sensitive transaction to be performed. The methodology already described offers three methods for the direct measurement of this ionospheric effect.

(a) The cross correlation of the LI x L2 processing where the multiplication operations are incremented in steps of 22 nanoseconds (equivalent to a path of 6.6 m) as determined by the 46 Msps analog to digital conversion sampling performed on the wide band IF output. The maximum zenith ionospheric total electron columnar content to be expected could be as high as 5 x 10¹⁸ electrons per square meter. That columnar content would result in a LI path error of 81 m at the zenith. For a satellite at a low elevation angle relative to the sensor (i.e., 5 degree elevation angle), the error can be a factor of 2.3 larger for a total path error of 115 meters. For the lower L2 frequency, the 5 x 10¹⁸ electrons per square meter ionospheric column will cause a path error of 134 meters at the zenith. For a satellite at a 5 degree elevation angle, the L2 total path error becomes 308 meters. Thus, the L2 modulated signals have a path error that is (308 m - 115 m) = 193 m different than the LI signals. The cross correlation processor will thus have a peak amplitude output when the LI sampled signals are delayed by 193 m / 6.6 m / sample = 29 samples. In order to cover the rising and falling of this correlation peak, it is advisable to examine the correlation amplitude function over a span of 34 lags of the LI signals because the LI signal always arrives before the L2 signal. Having both the rising and falling of the correlation peak will allow the maximum of the cross correlation function to be estimated to within approximately 1% of lag or approximately 5 x 10¹⁶ electrons per square meters or 5 TEC units.

(b) Another method for determining the ionospheric total electron content is to measure the phase difference between the chipping frequencies derived by the sensor from the LI P(Y) and L2 P(Y) channels. The GPS satellite transmitters are configured so that the PRN codes are in-phase and coherent at the LI and L2 transmitters and thus it is the inverse frequency squared dispersion of the ionospheric free electrons that is the principal cause for a phase shift between the LI and L2 channels when received by the sensors. Consider the DSP closed loop tracking of the P(Y) chipping frequency from either LI or L2 to be at a nominal signal to noise ratio of 30 dB-Hz resulting in a 0.03 radian RMS phase noise that is equivalent to 0.005 cycle x 29.3 m/cy = 15 cm of path equivalent noise. Forming the difference between LI P(Y) and L2 P(Y) will create a square root of two larger path uncertainty of approximately 21 cm. The scaling of this LI / L2 channel phase differential is 10 TEC units per meter of differential path implies that the precision of a single measurement is 2 TEC units. With an additional 10 statistically independent measurements, the precision of the ionospheric columnar content measurement is 0.6 TEC.

(c) Where even higher precision is needed, for perhaps scintillation measurements, it is possible to form the 2L1 and 2L2 data-types in order to form the differential carrier phase. The 2L1 channel is ambiguous at every 9.5 cm while the 2L2 is ambiguous at 12 cm. A change of 0.1 TEC in the ionospheric column will cause a 1.67 cm path change at LI and a 2.66 cm change at L2. By forming an accumulated phase from the 2L1 and 2L2 channels, it will be possible to directly measure the changes in the ionospheric columnar electron content to a precision of better than 0.1 TEC. Satellite Orbital Errors

The accuracy of the GPS broadcast orbit elements is expected to be 10 meters, now that selective availability (SA) degradation has been removed from the GPS signals. With a 10 m accuracy in orbit elements, the GPS C/A receiver positioning accuracy of 20 to 30 meters is expected. From step (c.) above, a geocentric positioning of the sensor to within a precision of 6 m would not be supported by an orbit accuracy of 10 m. Therefore, a full differential processing involving a reference site needs to be employed in order to exploit the higher accuracy offered by the codeless sensor methodology. Specific Shared RF and Digital Resources

In Figure 5, the 1^st LO function of the RF front-end could be supplied by the PCS cell phone receiver LO at 1750 MHz. This LO could serve very well as a high side LO for a down conversion of the GPS LI signal at 1575.42 MHz to an IF at 174.58 MHz and the 1227.6 MHz band is converted to an IF at 522.4 MHz. One or both of these IF outputs can then be digitized at 40 mega-samples per second to capture the central lobe of the P(Y) channels broadcast by the GPS satellites at 1575.42 and 1227.6 MHz.

There is an immediate advantage here because the phase locked synthesis of the 1^st LO of the PCS phone receiver is a relatively power intensive operation of the phone. Therefore, being able to make use of it to support a GPS receiver implementation is clearly an advantage. Also, the codeless GPS implementation does not require that the 1^st LO be programmable, as is often the case for conventional GPS receivers that use a numerically controlled 1^st LO. The codeless receiver does have an accuracy requirement of keeping the GPS spectrum within the IF bandpass within perhaps 1% of the intrinsic spectral wide of 2 MHz or the C/A channel and 20 MHz for the P(Y) channel. The PCS phones make use of a temperature compensated crystal oscillator (TCXO) with what is assumed here to be a frequency accuracy and stability of one part per million. The desire to hold the spectrum within 1% is then easily achieved with an accuracy of 20 Hz or better. The 174.58 MHz IF is then input to an analog device filter of 20 MHz bandwidth and then to the digital signal processor (DSP) chip that is already a standard feature of remote terminals. The DSP will have an analog to digital conversion stage using an intentional under sampled technique. Consider an analog to digital converter operating at a rate of 40.92 Msps (mega-samples per second or a sample every 24.5 nanoseconds). The choice of this sampling frequency is made because the later process of delay and multiply in order to recover the P(Y) channel PRN sequence chipping frequency will need a one-half chip time (49 ns or 2 samples) delayed signal. Sampling at 40.92 Msps would imply a Nyquist bandwidth of 20.46 MHz, that captures the entire central lobe of the precision channel and places it at the 1^st IF frequency of 174.58 MHz containing both the C/A and P(Y) channels (S.J. Spilker, Institute of Navigation, Vol. I, The Global Positioning System).

The number of bits in the AD/C is a parameter that can be varied depending upon the particular characteristics of the cellular system and can be between 1 and 12 with the larger number of bits also giving a greater dynamic range for tolerance to in-band GPS signal interference.

The digitized IF data stream is then multiplied by a two sample delayed version of itself. The resultant digital stream is a 40.92 Msps that contains a superposition of 10.23 MHz sinewaves, one for each of the satellites in view. All of the spectral lines will be confined in a relatively narrow band of +/- 27 Hz resulting from the Doppler shifts on the line of sight between each of the satellites for a fixed Earth based GPS sensor. The clustering of these spectral lines is centered at 10.23 MHz +/- 10 Hz for a TCXO reference oscillator of one part per million (lppm) frequency accuracy.

By exploiting the carrier phase data type, it is possible to derive an approximate location determination set of observables. Within the GPS satellites, there is a carrier frequency of 1575.42 MHz that is designated as the LI channel. That LI sinewave is then spread in frequency space by the effects of phase keyed modulation using a pseudo random noise sequence binary code. As described in 4,797,677, by the non-linear process of squaring this spread signal, the second harmonic of the original satellite suppressed carrier can be recovered. The effective frequency is then 3150.84 MHz +/- Doppler shift +/- the reference frequency error and with an effective wavelength of 95 mm. For a lppm frequency error, the spectral lines associated with each of the GPS satellites will be displaced by 3.2 kHz. The maximum Doppler shifts on the lines of sights will be 2 x 154 the P(Y) channel Doppler of 27 Hz = 8.3 kHz. Thus, the band to be searched is +/- 11.5 kHz or a total bandwidth of 23 kHz.

Accordingly, resort may be made to all suitable modifications and equivalents that fall within the scope of the invention as defined by the claims which follows. The words "comprise," "comprises," "comprising," "include(s)," and "including" when used in this specification and in the following claims are intended to specify the presence of stated features or steps, but they do not preclude the presence or addition of one or more other features, steps, or groups thereof.

Claims

Claims:

1. A system for passive reception of space-based radio signals utilizing a simplified sensor that performs digital operations to effectively compress the bandwidth of the space-based radio signals, comprising: a first sensor locationally associated with a telecommunications device for sensing transmissions from two or more signal sources that contain information sufficient to derive a position and velocity vector that identifies a geolocation where the telecommunications device is positioned, said first sensor including a converter for converting the sensed transmissions into first compressed band raw observations for communication to a central server accessed via a computer network connection, and circuitry for communicating the first compressed band raw observations to the centralized server; and wherein the centralized server, comprises: a processor for receiving and processing the first compressed band raw observations and for comparing one or more attributes of the position and velocity vector contained in the first compressed band raw observations to predetermined privilege criteria, and circuitry for developing a geolocation sensitive user privilege signal when the one or more attributes of the first compressed band raw observations satisfy the predetermined privilege criteria.

2. A system as recited in claim 1, further comprising: a second sensor at a predefined different geolocation than the first sensor for sensing transmissions from the same two or more signal sources sensed by the first sensor that produce transmissions containing information sufficient to derive a position and velocity for the predefined different geolocation; a converter for converting the sensed transmissions to second compressed band raw observations for communication to the centralized server; circuitry for communicating the second compressed band raw observations to the centralized server, said centralized server having a processor for receiving and processing the second compressed band raw observations and for preparing a differential position and velocity vector from the first and second compressed band raw observations and comparing one or more attributes of the differential position and velocity vector to predefined privilege criteria; and circuitry for developing a user privilege signal when the one or more attributes of the differential position and velocity vector satisfy the predefined privilege criteria.

3. A system as recited in claim 2, wherein the second sensor is located at the central processor.

4. A system as recited in claim 2, wherein the second sensor is located at a site different from the central processor.

5. A system as recited in claim 1, wherein the first sensor uses codeless techniques to convert the sensed transmissions into first position and velocity observations.

6. A system as recited in claim 2, wherein the second sensor uses codeless techniques to convert the sensed transmissions into second position and velocity observations.

7. A system as recited in claim 2, wherein the first sensor and the second sensor use codeless techniques to convert the sensed transmissions into position and velocity observations.

8. A system as recited in claim 1, wherein the privilege granting criteria comprise geodetic location of the telecommunications device.

9. A system as recited in claim 1, wherein the privilege granting criteria comprise latitude and longitude of the telecommunications device.

10. A system as recited in claim 1, wherein the privilege granting criteria comprise latitude, longitude, and height of the telecommunications device.

11. A system as recited in claim 1, wherein the privilege granting criteria comprise latitude, longitude, and height of the telecommunications device and a non-zero velocity of the telecommunications device.

12. A system as recited in claim 1, wherein the privilege granting criteria comprise geodetic location of the telecommunications device with an offset from a standard geodetic reference system.

13. A system as recited in claim 1, wherein the privilege granting criteria comprise geodetic location of the telecommunications device and the privilege granting criteria are stored in encrypted form.

14. A system as recited in claim 1, wherein the first sensor observations are compressed relative to the transmissions actually sensed by the first sensor.

15. A system as recited in claim 1, wherein the first sensor observations are compressed relative to the transmissions actually sensed by the first sensor and the circuitry for communicating the first compressed band raw observations to the privilege granting server communicates the transmissions sensed for a real time interval of X duration in a transmission to the privilege granting server having a real time duration less than X.

16. A system as recited in claim 1, wherein the first sensor observations are compressed relative to the transmissions actually sensed by the first sensor in a ratio of at least 100,000 to 1.

17. A system as recited in claim 1, wherein the first sensor uses codeless techniques to convert the sensed transmissions into first position and velocity observations and establishes an epoch time and frequency synchronization of the telecommunications device traceable to an external reference system.

18. A system as recited in claim 2, wherein the second sensor uses codeless techniques to convert the sensed transmissions into second position and velocity observations and establishes an epoch time and frequency synchronization of the telecommunications device traceable to an external reference system.

19. A system as recited in claim 1, wherein the first sensor uses codeless techniques to convert the sensed transmissions into first position and velocity observations and determines ionospheric spatial and time characteristics of columnar electron content.

20. A system as recited in claim 2, wherein the second sensor uses codeless techniques to convert the sensed transmissions into second position and velocity observations and determines ionospheric spatial and time characteristics of columnar electron content.

21. A system for applying to an electronic message, which exists at a telecommunications device location and is intended for a destination, information for determining privileges allowed to that electronic message, comprising: a sensor locationally associated with the telecommunications device for sensing transmissions from two or more signal sources that produce transmissions containing information sufficient to derive a position and velocity vector that identifies a location of the telecommunications device, said sensor including a converter for converting the sensed , transmissions into position and velocity observations for communication to a destination; circuitry for associating sensed compressed band raw observations from the sensor with the electronic message; and circuitry for sending the electronic message toward the destination with the sensed compressed band raw observations.

22. A system as recited in claim 21, wherein the message with associated compressed band raw observations is encrypted.

23. A system as recited in claim 21, wherein the message with associated compressed band raw observations is digitally signed and encrypted.

24. A system as recited in claim 21, wherein the message requests a commercial transaction.

25. A system as recited in claim 21, wherein the commercial transaction is a transfer from one account to another.

26. A system as recited in claim 21, wherein the message is sent by a payor and requests a debit of an account of the payor.

27. A system as recited in claim 21, wherein a function of the message is used in the production of the compressed band raw observations as these are associated with the message, whereby changes to the message cause the compressed band raw observations to become invalid.

28. A method for determining the privileges of a telecommunications device seeking access to a location-sensitive decision processor, comprising: at a telecommunications device privilege requesting circuitry, which is remote from a centralized server and is associated with a remote user's geolocation: at a first sensor locationally associated with the telecommunications device, sensing transmissions from two or more signal sources that produce constantly changing transmissions containing information sufficient to derive a position and velocity, which identifies the remote user's geolocation; converting the sensed transmissions into first compressed band raw observations for communication to a centralized serve; and communicating the first compressed band raw observations to the centralized server; at the centralized server: receiving and processing the first compressed band raw observations; comparing one or more attributes of the position and velocity contained in the first compressed band raw observations to predetermined privilege criteria; and developing a telecommunications device privilege signal when the one or more attributes of the first compressed band raw observations satisfy the predetermined privilege determining criteria.

29. A method as recited in claim 28, further comprising: at a predefined different location than the first sensor: sensing transmissions from the same two or more signal sources sensed by the first sensor that produce transmissions containing information sufficient to derive a position and velocity for the predefined different location; converting the sensed transmissions to second compressed band raw observations for communication to the centralized server; and communicating the second compressed band raw observations to the centralized server; at the centralized server: receiving and processing the second compressed band raw observations; preparing a differential position and velocity vector from the first and second compressed band raw observations; comparing one or more attributes of the differential position and velocity vector to predefined privilege criteria; and developing a telecommunications device privilege signal when one or more attributes of the differential position and velocity satisfy the predefined privilege criteria.

30. A method for applying to an electronic message, which exists at a telecommunications device geolocation and is intended for a destination, information for determining a privileged next-action of that message, comprising: sensing at a sensor, which is locationally associated with telecommunications device transmissions from two or more signal sources that produce transmissions containing information sufficient to derive a position and velocity vector that identifies the telecommunications device geolocation; converting the sensed transmissions into compressed band raw observations for communication to the destination; associating the sensed observations from the sensor with the electronic message; and sending the electronic message toward the destination with the sensed compressed band raw observations.

31. A system for passive reception of space-based signals comprising:

(a) a sensor which receives said signal and forms a set of compression parameters;

(b) a central processing node which is in communication with said sensor, capable of receiving the set of compression parameters;

(c) a processor within the central processing node, capable of converting the set compression parameters into a set of conventional signal characteristics suitable for input to a baseline vector estimation processor.