US3136848A - Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio - Google Patents

Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio Download PDF

Info

Publication number
US3136848A
US3136848A US42729A US4272960A US3136848A US 3136848 A US3136848 A US 3136848A US 42729 A US42729 A US 42729A US 4272960 A US4272960 A US 4272960A US 3136848 A US3136848 A US 3136848A
Authority
US
United States
Prior art keywords
signal
frequency
transistor
vidicon
noise ratio
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Lifetime
Application number
US42729A
Inventor
William H Woodworth
Original Assignee
William H Woodworth
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by William H Woodworth filed Critical William H Woodworth
Priority to US42729A priority Critical patent/US3136848A/en
Application granted granted Critical
Publication of US3136848A publication Critical patent/US3136848A/en
Anticipated expiration legal-status Critical
Expired - Lifetime legal-status Critical Current

Links

Images

Classifications

    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04NPICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N5/00Details of television systems
    • H04N5/222Studio circuitry; Studio devices; Studio equipment ; Cameras comprising an electronic image sensor, e.g. digital cameras, video cameras, TV cameras, video cameras, camcorders, webcams, camera modules for embedding in other devices, e.g. mobile phones, computers or vehicles
    • H04N5/225Television cameras ; Cameras comprising an electronic image sensor, e.g. digital cameras, video cameras, camcorders, webcams, camera modules specially adapted for being embedded in other devices, e.g. mobile phones, computers or vehicles
    • H04N5/228Circuit details for pick-up tubes

Description

June 9, 1964 w WOQDWORTH 3,136,848
VIDICON WITH LOW IMPEDANCE AMPLIFIER FOR EXTENDED HIGH FREQUENCY RESPONSE AND IMPROVED SIGNAL TO NoIsE RATIO Filed July 13, 1960 I. (PRIOR ART) FIG.
FIG. 2.
16m: 160 MC IOKC IOOKC liuc IKC (PRIOR ART) FIG. 3.
FIG. 4.
INVENTOR. WILLIAM H. WOODWORTH ATTO'RNEYSE United States Patent 3,136,848 VEDICGN WITH LOW HVIPEDANCE AMPLIFIER FUR EXTENDED HIGH FREQUENCY RESPONSE AND HMPROVED SIGNAL TO NOISE RATIO William H. Woodworth, China Lake, Calif., assiguor to the United States of America as represented by the Secretary of the Navy Filed July 13, 1960, Ser. No. 42,729 2 Claims. (Cl. 178-72) (Granted under Title 35, US. Code (1952), sec. 266) The invention herein described may be manufactured and used by or for the Government of the United States of America for governmental purposes without the payment of any royalties thereon or therefor.
The present invention relates to a vidicon and amplifier circuit combination for operation with high frequency low current signals and particularly to such a vidicon and amplifier circuit combination which provides minimum attenuation at high frequencies.
The current output of vidicons, photo-multipliers, photo-cells and other similar devices is of a relatively low order of magnitude of approximately 1X10 amperes. Prior methods of amplifying signals of this order of magnitude have been by means of vacuum tubes wherein a resistor in series with the signal source was included in shunt with the grid-cathode impedance, yielding suitable voltage division to provide an adequate grid signal. Transistor circuitries which have been used for amplification of signals of this magnitude have been developed by one-to-one substitution for the vacuum tubes wherein this substitution has resulted in the use of a load resistor in series with the emitter (equivalent to the grid-cathode impedance) and the use of a shunting resistor in series with the signal source.
These prior methods are satisfactory so long as the frequency of the output signal is of a relatively low order of magnitude since there will be negligible attenuation. However, when it is desirable to amplify signals of a higher order of frequency magnitude there is attenuation and a resultant reduction of the signal to noise ratio. This higher-frequency attenuation results from the presence of the shunting resistor in parallel with the small current signal to be amplified wherein the resistor is unavoidably shunted by the output capacitance of the signal source, the input capacitance of the amplifier and the lead wire capacitance. In these prior devices, to obtain an acceptable signal to noise ratio by having a sufficiently large voltage at the control element of the tube or transistor, the effective load resistance in series with the small current signal source must be of a relatively large order of magnitude. By employing such resistance of relatively large magnitude, which is shunted by the above mentioned unavoidable capacitance, the frequency at which attenuation occurs is of comparatively low value. Since it is frequently necessary to operate at higher frequencies it has been found necessary to employ high peaking circuits. By using high peaking circuits it is possible to obtain a predetermined voltage throughout the entire frequency range by providing a constant impedance in the range of negligible attenuation and a decreasing impedance with increasing frequency in the attenuation range. An example of one type of peaking circuitry is illustrated in RCA Review, volume XVII, Number 2, March 1956, page 485. It should be particularly noted that by operating in the attenuation range there is a highly undesirable reduction in the signal to noise ratio. Although high peaking circuits provide means by which a constant voltage output is obtained throughout the entire frequency range they do not improve the low signal to noise ratio at frequencies in the attenuation region.
The present invention provides a system wherein a con- 3,136,848 Patented June 9, 1964 stant voltage output is obtained through the entire operating frequency range and the maximum available signal to noise ratio is realized. This is accomplished by applying the output signal of a low current source directly to a current sensitive low impedance transistor. Since the input impedance of the transistor is very low (0-200 ohms) and shunts the unavoidable circuit capacitance, there results a greatly extended high frequency response since attenuation occurs at a much higher frequency. Because operation is not in the attenuation region, a constant voltage output is obtained throughout the entire frequency range, without the necessity of employing high peaking circuits, and the maximum available signal to noise ratio is realized.
An object of the present invention is to provide amplification of high frequency low current signals without attenuation resulting from the use of a load resistor in series with the signal source.
Another object is to provide a low input impedance transistor amplifier for amplification of low current high frequency signals wherein maximum available signal to noise ratio is obtained throughout the operating frequency range.
Another object is to connect the output of a low current high frequency source directly to a current sensitive amplifier wherein the input impedance of the amplifier is very low and shunts all input circuit capacitances, resulting in greatly extended high frequency response.
Other objects and many of the attendant advantages of this invention will become readily appreciated as the same becomes better understood by reference to the following detailed description when considered in connection with the accompanying drawings wherein:
FIG. 1 is a schematic diagram of a conventional amplifier of low current high frequency signals;
FIG. 2 is a log-log plot of the output signal of the circuits shown in FIGS. 1, 3 and 4;
FIG. 3 is a schematic diagram of the conventional amplifier of FIG. 1 and a conventional high peaking circuit to compensate for attenuation of the amplifier; and
FIG. 4 is a schematic diagram of the low impedance amplifier of the present invention for amplification of low current high frequency signals.
In order to clearly understand the unique advantages of the instant invention it is well to consider it in conjunction with conventional systems and the problems associated therewith. In the following analysis it should be particularly noted that the amplifier is used to amplify a high frequency low current signal. It is considered that the hereinafter described invention is uniquely applicable to a signal of this type and it has overcome major disadvantages of prior systems.
When the signal to be amplified has a current of a relatively large order of magnitude, the prior systems have proved adequate since the frequency at which attenuation occurs may be increased by decreasing the value of the resistance in series with the signal. Likewise, the prior systems have proved adequate when the current is small and the frequency is low since at low frequencies there will be negligible attenuation. However, when the current is small and the frequency is high, the prior systems have proved inadequate since there is considerable attenuation and reduction of the signal to noise ratio.
Referring now to FIG. 1 there is shown a typical transistor circuit that has been used for amplification of high frequency low current signals. As shown, transistor 11 is of the emitter-follower type wherein the output voltage e appears across load resistor 13. Load resistor 13 is comparatively large and, including the base-emitter resistance, corresponds to the grid-cathode impedance of a vacuum tube; shunted by resistor 17, the resultant voltage division with device 15 makes the base signal of suflEicient magnitude. The high frequency low current output signal of device 15 is applied to the base of transistor 11 and in parallel with resistor 17. It should be particularly noted that resistor 17 is unavoidably shunted by the output capacitance of device 15, the input capacitance of transistor 11 and the internal capacitance of the lead wires. These capacitances are shown in dotted lines as equivalent capacitor 19. The signal applied to the base of transistor 11 must have a magnitude of approximately .05 volt in order to provide an acceptable signal to noise ratio and since the current output of device 15 is approximately 1 10 amperes it is necessary that resistor 17 have a value of approximately 50,000 ohms.
In FIG. 2 is shown a log-log plot of the output voltage e versus the output frequency of signal device 15. Curve A is a plot of the FIG. 1 device which can be determined from actual measurements or from the relation and C is the value of effective capacitor 19.
The relationship (2) 27rR C defines that frequency where the impedance of R and C are equal and where the current through R is .707 times the current therethrough if capacitor 19 (C) were not included in the circuit. The attenuation at this frequency is indicated at point (1" in the attenuation region of curve A of FIG. 2.
By virtue of the above and especially by examination of Equation 2 it can be seen that as the value of resistor 17 (R decreases, attenuation occurs at higher frequencies as indicated by curve B of FIG. 2. It is undesirable to operate in the attenuation region and especially at frequencies much above that at point a since the reduced signal to noise ratio becomes critical. However, the frequency at which attenuation occurs cannot be appreciably increased since the value of resistor 17 of the circuit of FIG. 1 cannot be appreciably reduced because it is necessary to provide sufiicient voltage at the base of transistor 11 as previously explained.
A high peaking circuit of the type generally indicated at 21 in FIG. 3 is frequently employed to correct for the attenuation of the FIG. 1 device. High peaking circuit 21 includes resistors 23 and 25 and capacitor 27 and provides a voltage-frequency plot as indicated by curve C of FIG. 2. The logarithmic addition of the voltages of curves A and C results in curve D which is linear and provides a constant voltage e at frequencies below the maximum operating frequency which in this case is approximately 5 megacycles.
The values of resistors 23 and capacitor 27 are selected so that where 6 is the value of capacitor 27, R is the value of resistor 23, C is the value of effective capacitor 19 and Z is the input impedance at the base of transistor 11. At low frequencies capacitor 27 is not effective since the capacitive reactance of capacitor 2'7 is large compared to the resistance of resistor 23 and the output voltage s is e i R -i- R2 where R is the value of resistor 25 and e is the voltage applied to the input of the high peaking circuit. From this it can be seen that transistor 11 must amplify the incoming signal to a. value e in the non-attenuated as well as attenuated regions, which is considerably greater than the output signal a since there is a voltage drop across the peaking circuit. This means either the value of resistor 17 must be increased above the FIG. 1 device to maintain the same value of c which decreases the frequency at which attenuation occurs, or the output voltage :2 will be decreased.
Another type of high peaking circuit is shown on page 485 of the RCA Review, volume XVII, March 1956 published by RCA Laboratories wherein the fourth, fifth and sixth stage transistors include emitter peaking circuits to provide improved rising characteristics to the peaking circuits between the fourth and fifth stages. In order to compensate for the impedance variations of the stage, which may result from variable temperatures, the interstage peaking circuits are provided with variable capacitors which are adjusted to compensate for impedance changes.
The present invention provides a unique system wherein there is no attenuation of low current high frequency signals and as a result there is the best possible signal to noise ratio, the output voltage is constant over the entire requency range and high peaking circuits have been eliminated.
Referring to FIG. 4, the system there shown includes the low current high frequency device 15. The output signal of this device has a frequency which may vary from kilocycles to 100 megacycles and a current of approximately l 10* amperes. As shown, device 15 is connected in series with a positive D.C. potential through D.C. bias resistor 28 and is provided an A0. path through D.C. blocking capacitor 29. The AC. signal is applied from one side of resistor 28 through coupling and D.C. blocking capacitor 30 to the base of transistor 11. The collector of transistor 11 is connected through load resistor 31 to a negative potential source and the emitter is connected directly to ground. A positive potential is connected through resistor 35 to the base of transistor 11 to provide a positive D.C. bias voltage. ransistor 37 has the base connected directly to the collector of transistor 11, the collector connected to A.C. ground (via the battery connected to the V terminal), and the emitter connected through resistor 39 to the base of transistor 11. There is a phase shift across transistor 11, and no phase shift across transistor 37 since it is an emitterfollower, which results in degenerative feedback wherein currents I and I are 180 out of phase. Output voltage terminal 41 is connected through D.C. blocking capacitor 43 to the emitter of transistor 37.
The impedance which device 15 looks into is defined by the relation wherein Z, is the impedance looking into the base of transistor 11 with degeneration feedback, e is the voltage drop across the base-emitter junction of transistor 11, I is the output current of device 15, I, is the current flow into the base of transistor 11 and Z is the commonemitter input impedance of transistor 11.
From the above relation it can be seen that the ratio I /I decreases with degenerative feedback and the degenerative feedback therefore decreases the effective input impedance. Normally the impedance (Z across the base-emitter junction is from approximately 500 to 1000 ohms. With generative feedback, the effective impedance (Z is reduced to a range of from approximately 0 to 200 ohms.
The input impedance of transistor 11 is very low and it is therefore unnecessary to provide a separate resistor in series with the output signal of device 15 since the voltage at the base of the transistor is sufiicient without employing a voltage divider as was necessary in the prior devices shown in FIGS. 1 and 3.
It can therefore be seen that the present invention obviates high frequency losses of low current sources. By
examination of Equation 2 and by taking a typical value of 20 micromicrofarads as the unavoidable shunting capacitance, an attenuation frequency (f,,) of approximately 150 kilocycles is obtained when the shunting impedance is 50,000 ohms, as in the devices of FIGS. 1 and 3, and a frequency of 53 megacycles when the shunting impedance is 200 ohms as in the FIG. 4 device of the present invention. Therefore the high-frequency roll-off or attenuation is far higher than when a load resistor is employed. The following table shows by way of example, the values of resistance, bias voltage and capacitance in one embodiment of the network of the present invention:
Reference Unit Value Reference Unit Value Numeral Numeral Ohms 500,000 30 Mierofarads. 2. do rr rr 8,200 43 do 2. 0 do 15,000 +V Volts 0 to do ,200 +V do +12 Microfarads. 8. 0 V (lo 12 Obviously many modifications and variations of the present invention are possible in the light of the above teachings. It is to be understood that a degenerative feedback reduces the efiective input impedance of the amplifier; however, it should not be limited to this feature in that the low impedance of the first stage transistor alone, greatly extends the unattenuated frequency range of the amplifier. It is therefore to be understood that within the scope of the appended claims the invention may be practiced otherwise than as specifically described.
What is claimed is:
1. A vidicon and amplifier circuit combination for providing extended flat frequency response without deterioration of signal-to-noise ratio, said combination comprising:
(a) a vidicon circuit;
(b) a transistor amplifier characterized by low input impedance of value no greater than the order of two hundred ohms; and
(c) capacitor means establishing a signal current path between said vidicon circuit and said transistor amplifier, said capacitor having suitably large microfarad value to present negligible impedance, relative to said transistor amplifier input impedance, at signal frequencies as great as the order of fifty megacycles per second.
2. A combination as defined in claim 1, wherein said transistor amplifier comprises a common-emitter commoncollector pair provided with sufiicient degenerative feedback to decrease the effective input impedance of said amplifier to fall within a range extending to an upper value no greater than two hundred ohms.
References Cited in the file of this patent UNITED STATES PATENTS 2,863,957 Hamilton Dec. 9, 1958 2,889,519 Montgomery June 2, 1959 2,901,556 Chapman Aug. 25, 1959 3,025,472 Greatbatch Mar. 13, 1962 3,040,264 Weidner June 19, 1962 FOREIGN PATENTS 1,127,774 France, Dec. 24, 1956

Claims (1)

1. A VIDICON AND AMPLIFIER CIRCUIT COMBINATION FOR PROVIDING EXTENDED FLAT FREQUENCY RESPONSE WITHOUT DETERIORATION OF SIGNAL-TO-NOISE RATIO, SAID COMBINATION COMPRISING: (A) A VIDICON CIRCUIT; (B) A TRANSISTOR AMPLIFIER CHARACTERIZED BY LOW INPUT IMPEDANCE OF VALUE NO GREATER THAN THE ORDER OF TWO HUNDRED OHMS; AND (C) CAPACITOR MEANS ESTABLISHING A SIGNAL CURRENT PATH BETWEEN SAID VIDICON CIRCUIT AND SAID TRANSISTOR AMPLIFIER, SAID CAPACITOR HAVING SUITABLY LARGE MICROFARAD VALUE TO PRESENT NEGLIGIBLE IMPEDANCE, RELATIVE TO SAID TRANSISTOR AMPLIFIER INPUT IMPEDANCE, AT SIGNAL FREQUENCIES AS GREAT AS THE ORDER OF FIFTY MEGACYCLES PER SECOND.
US42729A 1960-07-13 1960-07-13 Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio Expired - Lifetime US3136848A (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
US42729A US3136848A (en) 1960-07-13 1960-07-13 Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
US42729A US3136848A (en) 1960-07-13 1960-07-13 Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio

Publications (1)

Publication Number Publication Date
US3136848A true US3136848A (en) 1964-06-09

Family

ID=21923447

Family Applications (1)

Application Number Title Priority Date Filing Date
US42729A Expired - Lifetime US3136848A (en) 1960-07-13 1960-07-13 Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio

Country Status (1)

Country Link
US (1) US3136848A (en)

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3244995A (en) * 1961-07-07 1966-04-05 Westinghouse Electric Corp Amplifier including a common emitter and common collector transistor providing regenerative feedback
US3329904A (en) * 1965-11-22 1967-07-04 Blonder Tongue Elect Wide-band transistor amplifier system employing impedance mismatch and high frequency peaking
US3714498A (en) * 1970-02-16 1973-01-30 Us Navy Television camera
US4363035A (en) * 1980-04-24 1982-12-07 Robert Bosch Gmbh Method and apparatus for signal pick-up from semiconductor image or line sensors
US5126846A (en) * 1988-08-08 1992-06-30 Kabushiki Kaisha Toshiba Non-linear amplifier and non-linear emphasis/deemphasis circuit using the same

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR1127774A (en) * 1954-07-22 1956-12-24 Thomson Houston Comp Francaise Improvements to junction type transistron amplifiers
US2863957A (en) * 1958-03-10 1958-12-09 Ryan Aeronautical Co Triad transistor amplifier
US2889519A (en) * 1958-04-23 1959-06-02 Montgomery George Franklin Clamp-type current transducer
US2901556A (en) * 1954-02-10 1959-08-25 Int Standard Electric Corp Semi-conductor amplifiers
US3025472A (en) * 1956-12-11 1962-03-13 Taber Instr Corp Transistor amplifier with temperature compensation
US3040264A (en) * 1959-05-29 1962-06-19 Ibm Transistorized amplifier

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US2901556A (en) * 1954-02-10 1959-08-25 Int Standard Electric Corp Semi-conductor amplifiers
FR1127774A (en) * 1954-07-22 1956-12-24 Thomson Houston Comp Francaise Improvements to junction type transistron amplifiers
US3025472A (en) * 1956-12-11 1962-03-13 Taber Instr Corp Transistor amplifier with temperature compensation
US2863957A (en) * 1958-03-10 1958-12-09 Ryan Aeronautical Co Triad transistor amplifier
US2889519A (en) * 1958-04-23 1959-06-02 Montgomery George Franklin Clamp-type current transducer
US3040264A (en) * 1959-05-29 1962-06-19 Ibm Transistorized amplifier

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3244995A (en) * 1961-07-07 1966-04-05 Westinghouse Electric Corp Amplifier including a common emitter and common collector transistor providing regenerative feedback
US3329904A (en) * 1965-11-22 1967-07-04 Blonder Tongue Elect Wide-band transistor amplifier system employing impedance mismatch and high frequency peaking
US3714498A (en) * 1970-02-16 1973-01-30 Us Navy Television camera
US4363035A (en) * 1980-04-24 1982-12-07 Robert Bosch Gmbh Method and apparatus for signal pick-up from semiconductor image or line sensors
US5126846A (en) * 1988-08-08 1992-06-30 Kabushiki Kaisha Toshiba Non-linear amplifier and non-linear emphasis/deemphasis circuit using the same

Similar Documents

Publication Publication Date Title
US3286189A (en) High gain field-effect transistor-loaded amplifier
US2550518A (en) Control of impedance of semiconductor amplifier circuits
US2694142A (en) Signal-to-noise energy detection unit
US3136848A (en) Vidicon with low impedance amplifier for extended high frequency response and improved signal to noise ratio
US2903522A (en) Transistor amplifier
US3223938A (en) Emitter follower transistor amplifier
US3135934A (en) Variable reactance attenuation network controlled by control voltage
US2208665A (en) Amplifier circuits with controlled gain
US2822434A (en) Amplifying apparatus
US3027518A (en) Automatic gain control system
US3619797A (en) Operational amplifier
US3174111A (en) Twin-t filter with negative feedback
US2936424A (en) Transistor amplifier
US3119077A (en) Automatic gain control circuit using variable attenuators in the signal path
US3176236A (en) Drift stabilized amplifier
US2396531A (en) Electrical coupling circuits
US3466559A (en) Bandpass voltage amplifier
US3121201A (en) Direct coupled negative feedback hybrid amplifier
US3343099A (en) Audio compressor circuit
US3289088A (en) Automatic non-linear gain control circuit
US2527737A (en) Variable frequency band width amplifier
US2974288A (en) D.-c. amplifier and filter therefor
US2864002A (en) Transistor detector
US3375326A (en) Video d.c. insertion circuit
US3334308A (en) Simplified compressor amplifier circuit utilizing a field effect transistor feedbackloop and a auxiliary solid state components