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Feedback wave translating system

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US2255804A US15357437A US2255804A US 2255804 A US2255804 A US 2255804A US 15357437 A US15357437 A US 15357437A US 2255804 A US2255804 A US 2255804A
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Nils J Oman
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RCA Corp
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RCA Corp
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    • H03F1/00Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/34Negative-feedback-circuit arrangements with or without positive feedback
    • H03F1/36Negative-feedback-circuit arrangements with or without positive feedback in discharge-tube amplifiers
    • H03F3/00Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/26Push-pull amplifiers; Phase-splitters therefor
    • H03F3/28Push-pull amplifiers; Phase-splitters therefor with tubes only
    • Y10S604/00Surgery
    • Y10S604/906Artificial insemination


Sept. 16, 1941. N. J. OMAN V FEEDBACK WAVE TRANSLATING SYSTEM 4 Sheets-Sheet 1 Filed July 14, 1957 iigia W k I Qnnentor v (Ittomeg s t. 16,1941. N J, WAN 2,255,804

FEEDBACK WAVE TRANSLATING SYSTEM Filed July 14, 1937 4 Sheets-Sheet? Sept 16, 1941. N J OMAN FEEDBACK WAVE TRANSLATING .SYSTEM Filed July 14, 1937 4 Sheets-Sheet s 27' J1 w p fif .162

home-" Gttorneg Sept. 16,1941. N. J. OMAN FEEDBACK WAVE TRANSLATING SYSTEM Filed July 14, 1937 4 Sheets-Sheet 4 2 4 i I SEC-L 0,90

C/Kll/T REHEC TED IKCI" lboomcv Smaentor Gttorneg Patented Sept. 16,1941

FEEDBACK WAVE'TRANSLATING SYSTE Nils 3. Oman, Haddon Heights, N. 1., assignor to Radio Corporation of America, a corporation oi Delaware Application July 14, 1937, Serial No. 153,574

My invention relates to feedback wave translating systems and more particularly to negative feedback circuits in electric wave transmission amplifiers.

It has been well recognized that with the present day thermionic discharge amplifiers designed for handling large power output, the tendency to distortion, as well as hum or other noise interference from the power supply energizing circuits, has greatly increased. Several negative or degenerative feedback arrangements have been developed for reducing this difilculty but there has still remained certain disadvantages to be overcome, e. g., objectionable phaseshift at certain frequencies, undue complication of circuit apparatus and general instability or unreliability over an extended frequency range.

'Heretoiore a certain type of amplifier has been I tion amplifier adapted to high level modulation of a radio frequency amplifier in a transmitting system.

It is a further object of my invention to provide a system making it possible to use a class B audio amplifier biased to plate current cut-oil for high operating efllciency without the usual I attendant bad distortion.

It is another object of my invention to provide a new and simplified negative feedback circuit for space discharge amplifiers that shall cause the latter to be substantially free of distortion V and power supply circuit hum or noise overthe operating range.

It is another object of my invention to provide an improved high gain negative feedback coupling circuit that is characterized by substantially constant attenuation and freedom from phase shiftfrom zero to veryhigh frequencies, e. g., several megacycles.

It is still another object ofmy inventionto improve the operation of push-pull resistance coupled amplifiers in regard to maintaining equality of voltage on the two halves and with degree phase relation, as well as to prevent longitudinal transmission of signals in the amplifier, in other words 'to preserve a balanced amplifier relation.

In accordance with one aspect of my invention, a portion of the voltage from the output stage of an' audio amplifier, coupled by phase correcting networks, is fed back into a preceding stage in such manner as to obtain substantially constant attenuation and freedom from phase shift from zero to several megacycles, providing a full equalized feedback system. For this purpose I have provided a frequency or phase compensated potential divider network across the output of a cascade amplifier, and direct current feedback connections, free of impedance elements, from points on-the output potential divider network to the input circuits of the firstgeneration depends on whether the loop has phase reversing elements other than the tubes.

One or more of the stages are resistance coupled and, particularly with a class B amplifier, final stage and the above cross connection, it is desirable to obtaincoupling between halves of the push-pull resistance coupled tubes in order to maintain equal voltages with 180 degree phase relation. This coupling is obtained by means of a common cathode-to-ground resistor without th usual by-pass condenser. The use of this particular arrangement has the further advantage of reducing any tendency toward transmission longitudinally of a signal through the amplifier. By this means it is possible to use degenerative feedback over an even number of stages with the last stage using class B operation. Longitudinal transmission refers to the case where both sides of'a, push pull, or balanced circuit, vary in signal potential together or with in-phase operation, with respect to ground, as

compared to opposed phase, or balanced operation, to ground.

In my negative feedback amplifier-the distortion and noise or hum is reduced in proportion to the reduction in gain effected by the feedback, This means that the system'is designed to have much greater amplification than normal and then it is reduced by the inverse feedback means, the amount of this excess gain depending upon the amount of reduction in noise and/ or distortion that is required. Heretofore the amount of gain and usable feedback has been I limited by too much total phase shift in the loop.

I have effected a substantial improvement by.

eliminating phase shift in the feedback path.

, I amaware that prior developments have been made in negative feedback systems, some with phase compensating circuits, reference being a made to Nyquist 1,915,440, Barnes 1,994,457, Black 2,003,282 and 2,011,566 patents fora more com plete understanding of the background of feedback compensation. Very early work on regenerative feedback in cascade D. C. coupled ampli- 1 fiers was done by Hartley, Patent No. 1,218,650,

March 13, 1917.

An important aspect of my invention is the production of a system using pure class B amplification with its attendant high efiiciency of operation. Heretofore, a compromise has been used because of inherent distortion in pure class B .operation. In class B work, each tube handles one half of the wave and functions only when the grid swings in a positive direction with respect to its bias position. See Colpitts 1,128,292 by way of historical interest in the early development of class B amplification. For true class B amplification, the tubesshould draw no plate current in It has been between the characteristics of the two tubes.

has partaken of class A amplifier functions.

Figures 5, 7 and 8 .are modifications of othe features of my invention,

Figure 6 is a simplified explanatory diagram of circuit of Fig. 2 showing operation at high frequencies, v

Figure 9 is a characteristic curve of substantially pure class B amplifier operation,

Figures 10, 10a, 12, 12a, 14, 14a, 16, 16a, 18, 18a and 19 are characteristic curves showing the characteristics of various portions of the system,

Figures 11, 11a, 13, 13a, 15, a, 17 and 17a are simplified diagrams illustrating circuits used in deriving the curves,.the.diagrams being shown immediately adjacent the curves pertaining thereto.

The modulator amplifier, built in accordance.

with my invention, difiers from that usually employed for high level modulation of a high power transmitter in several respects:

(1) It makes use of inverse feedback from the plates of class B modulator tubes to thegrids of the first audio tubes.

(2) The feedback connection is unique in that it can be made to have negligible phase and frequency characteristic from zero to extremely high frequencies.

(3) The feedback is coupled over an even number of stages from the plate of each modulator tube to the diagonally opposite grid of each input tube taking advantage of the 180 degree voltage relation between the two sides of the push-pull circuit.

(4) This practice, places thefeedback voltage in a channel terminating in a modulator grid that is biased beyond cut-off by virtue of class B operation of the final or modulator stage. It is therefore necessary to provide coupling means between the halves of the push-pull circuit to ensure oper- Y ation of the feedback principle. This coupling is obtainedthrough the use of unbypassed common cathode resistors in the first three stages of the This method of coupling is substantially free of frequency characteristic or phase 1 Reference is made to Barton Patent 2,084,180,

June 15, 1937, for a more complete understanding 3 of class B operation in its various forms. Refer: ence is also made to Loughren Patent 1,699,110,

January 15, 1929, on a species of class B, the grid circuit of which has more in common with the present class B amplifier. While this residual or no load plate current is allowablein radio receiving sets, small amplifiers and the like, it becomes a serious power los in high power radio i be apparent from the following description and claims.

In the accompanying drawings,

Figure 1 is aschematic diagram of the circuit of a modulating amplifier embodying my invention, i

Figure 2 shows a circuit diagram of a modification of a portion. of the system in Fig. l-, also made in accordance with my invention, and is preferable to the said portion of Fi 1,

Figures 3 and 4 are simplified diagrams of a portion of the system of Fig. 1 used to analyze an I 75 aspect of my invention,

shift over the frequency range necessary for satis- J factory operation of the feedback circuit of the modulator.

,(5) Simplified and effective means for con-' trolling phase and gain or voltage. amplification high level modulating amplifier system for a high power broadcast transmitter designed and built in accordance with my invention, a pair of class A push-pull screen grid indirectly heated cathode tubes 3 and 5 (type RCA-1603) are connected to a source of audio frequency signals by means of a push-pull transformer l. The transformer is provided with the usual iron core and has a grounded shield 9 between windings. The secondaries ll of the transformer are connected to grids l3 and each winding is shunted by a resistor l5 and a condenser ll. Thev grids are .biased by means of a tapped or adjustable resistor l9, of a value large enough to supply the requiredoperating bias, say two volts negative,

and additional voltage to compensate for positive bias, of the orderof twenty volts, fed back from resistors I9 and 35.

the voltage divider -16, hereinafter described.

The anodes of the tubes 3 and I are supplied with direct current operating voltage, from 9. rectified A. C. source, through resistors 2|, which serve u part of the coupling network to the next stage. A resistor 23 in series with condenser 25 connected from each anode to ground forms a high frequency phase correcting network, more where given. The correct amount of operating bias voltage for the grids is taken from a portion of the bias and coupling resistor 35, through filter resistor 33 and leak resistors 3 I Resistor 30 provides a greater amount of resistance than is rewinding I6, the primary of which is not shown.

It also energizes the cathode heaters of stage I603.

The output of stage 801 is coupled to the input or signal grids of push-pull parallel connected amplifiers SI and 53 (type RCA-845i triodes with directly heated cathodes, through stopping capacitors. 43 and low frequency phase correcting networks including resistors 65 and 41 similar to'those in the input of stage 000.

The cathodes are energized from the second- I ary winding 22 with raw alternating current. Grid bias is obtained through self bias resistor 55 in the cathode circuit between winding 22 and ground, the resistor being unbypassed for same reasons as given elsewhere in connection with for reasons given below. The 891-R tubes are air cooled by means of blowers 10. The grids are negatively biased to about 1100 volts from the bias rectified A. C. source through a grid filter comprising resistor 09 and bypass condenser II. The cathodes are fed by raw alternating current from separate windings I2 and I4 and the anodes are supplied with very high voltage of the order of 8500 volts from the rectified three phase source shown. The output audio frequency power from the anodes is used for high level modulation of the plate circuit of an RF power output amplifier, not shown. For this purpose, the anodes of tubes 65 and 08 are connected through resistors 8| to'a primary 31 of an A. F. transformer, the secondary 00 of which is connected through a stopping condenser M to a modulating reactor 93 in series in the power supply lead to the plate of said R. F. power tube. As to one function of resistors 8|, see Barton 2,023,506.

The feedback potential divider, described elsewhere, isconnected from plate to plate of class B tubes "IR, the midpoint being grounded in balanced relation. It is desirable that resistors 8| be connected in the anode leads between the divider and the primary 81, across each half of which is connected frequency correcting networks, including condensers 85 and resistors 03 in series. Resistors III are parasitic surge prevention devices and preferably are non-inductive for use in combination with my feedback system.

The operating bias on' the grids of the 891R tubes is preferably adjusted to an amount, of theorder of 1100 volts, that the plate current is practically, but not entirely, out off. While the tubes could operate biased beyond the region of zero plate current, it is preferable to operate them practically at cut-oil! in the present case. This point is determined by readings on ammeters A in the cathode return leads, and in order to be certain of this operating condition, the adjustments should be such that a small plate current flows, say 20 to 30 milliamperes, an, amount that just moves the indicator slightly from-zero. I

The curves in Fig. 9 represent the operation of tubes 65 and 06, although the curves were not taken by direct readings from the present system. The operating bias? line corresponds to the 1100 volts bias used. The distance from the abscissa to the points of intersection of the-bias line by either of the plate current Ip curves represents the above residual plate current of around 25 mills. The extent to which the Ip curves of the tubes 05 and t6 depart from the ideal load line represents serious distortion tendency. Now,

The anodes are supplied if the conditions of operation were made such that the curves substantially follow this load line,

as in Fig. 2 of Barton Patent 2,084,180, to mini-.

mize distortion, the intersections by the I curves with the operating bias line would be quite a bit farther from the abscissa. Instead of 25 mills, the anodes. would each draw about 350 mills of current for the condition of low distortion. This would mean a substantial waste of power in apparatus of the present kind. For both tubes,

by way of example, twice 350 mills multipled by- 8500 volts equals 5,950 watts or 5.95 kw., this compared with only 500 watts as at present.

In a transmitting system of 5 kw. rated output, Y

this would mean that the total power consumption of the equipment would be 22 kw. instead of '16 kw., the power consumption of, the present usual type involving the drawing of grid current over a part or all of the grid swing, or complete half cycle, may be used. The grids each swing from the 1100 volts bias as far as zero bias dividers.

1 applied to the grids of the first tubes .of the amplifier.

feedback circuit will compensate against the resulting tendency to distortion in such a case.

Regardless of the kind of grid circuit characteristics used, the more important feature is that my feedback circuit makes possible the zero signal, zero plate current class B operation. Furthermore, the feedback system removes the A. C.

hum which would be prohibitive with theuse of raw A. C. on the cathodes.

In order to assist those skilled in' the art-to i practice my invention, I have given below desirable values for the variouscircuit elements.

The values are merely illustrative of a particular modulating amplifier that has been designed and built, in accordance with my invention, for a 5 kw. air cooled, broadcast transmitter, 5D, and are not to be considered as limiting the invention. In the table, it stands for resistance in ohms and C for capacity in mlcrofarads:

Element I R Element 0 Referring to the feedback coupling circuit, the high A. C. voltages at the plate of each modulator tube 65 and 66 are reduced to a suitable value for 40 1 introduction into the grid circuits of the first tubes 3 and 5 by means of resistance 15 and It capacity 11 and I9 voltage dividers with distributed constants. There is a separate divider for each modulator plate. Each divider has ground potential at its bottom end. At medium and high frequencies these units are capacity The values of capacities are-so chosen that the will be large compared to any stray capacities that might ailect the performance of the divider. The capacities must not; be so large as to seriously affect the impedance of the load 1 into which the modulator tube works. It is necessary to have control over the D. C. voltage It is therefore necessary to parallel the lower sections of the dividers with resistors '56. These resistors would affect the performance f of the divider at low frequencies; however,- by

placing resistances across all the capacities of the divider it is possible to make the voltage ratio oi the resistance divider the same as that of the capacity divider. The system will then be A portion of the D. C. modulator plate potential then appears on the grids of the first tubes. This voltage mustindependent of frequency.

be considered when choosing the value of the self bias resistor It for tubes 3 and 5, as above explained. The use of many series condensers for the upper arms of the voltage dividers reduces the heat to be dissipated by any one resistor and makes practical the use of small carbon units in parallel with each of these capacitors. While onl two resistor capacity sectiom it and II are 15 shown in the upper arm of each divider, in actual practice I have used ten sections.

The input transformer I used to couple the source of signal to the modulator is designed with two secondary windings ll symmetrically placed so as to insure identical characteristics for each winding. The capacity from the grid end of each winding to ground is made as low as possible.

An external capacity I1 is connected across each secondary winding. These capacities are made as large as possible without their having a serious effect on the frequency characteristic of the transformer. They must be large compared to the capacity from the grid end of each secondary to-ground.

- There are two reasons for keeping the capacity 1 from the secondary grid connections to ground low in the input to stage I603. First: at high frequencies this capacity will allow A. C.'currents from the divider to flow through the secondary winding H to ground. The impedance of the winding II is not constant with frequency, and the variation of impedance would upset phase and amplitude of the feedback voltage. Second:

the'capacity from the grid connection to ground forms with the capacity across the secondary another voltage divider across the bottom section of the main dividers "and I8. If the capacity across the winding is large compared to the capacity from the grid connection to ground this undesired potential divider actionwill have little effect on the feedback voltage. 1

The capacity from the low side of each secondary winding II to shield 9 and ground is of no consequence as it may be included in the capacity 19 at the bottom of the voltage divider. Otherwise, this would cause bad phase shift, if

the voltage divider were resistive.

The capacity of the leads I8 conducting the feedback voltage from the voltage divider tap to the low side of each secondary winding H may also be included in the capacity 19 of the bottom section of 'the voltage divider. Since the poten- 45 tial divider network is capacitive, and the lowest sections of large capacity, the capacity of the feedback conductors is but a small part of this capacity. Itis possible to enclose the leads 18 in agrounded sheath as of lead or other metallic cable shield. This prevents the pickup from extraneous fields, R. F. or high A. F., and capacity coupling to leads in other parts of the amplifier, which disturbances would cause singing or other trouble. I

It is desirable that the leads between the two arms of each voltage divider be kept short. Long leads may have enough inductance to resonate with the large capacity is of the bottom section of the divider at some high frequency .within the frequency range of interest for operation of the'feedback circuit.

Referring to the phase and gain control features in the feedback loop, in order to have stability in a feedback amplifier it is necessary that the gain around the feedback loop should be less than unity when the feedback vector is inphase with the input voltage to the amplifier; when this iii-phase condition exists, the amplie fier is regenerative and the amplification of the system may rise. It is desirable to have the frequencies at which this condition occurs so. far removed from the working frequency band that,

it will not adversely aifect the operation of the amplifier over the transmission band of frequency required to modulate the transmitter.

It is also desirable to keep the feedback loop gain high over the transmission range of the amplifier and for a considerably higher range of frequencies so as to ensure reduction of harmonies generated in the amplifier that may fall above the required transmission range. In the present modulator amplifier these high frequency harmonics are further attenuated by the filter Fig-11a, at high frequencies. It is apparent that considerable phase shift takes place before anpreciable attenuation occurs. In this amplifier there are four such circuits. If they all could be made equally good, each would contribute an equal amount. of attenuation at any frequency. If the amplifier were to-utilize 30 db. of feedback it might be desirable to have 40 db. loss in loop gain when the accumulative phase shift becomes 180 degrees to ensure a margin of safety Y for stability. Each stage would then be required to contribute 10 db. loss at this frequency. This, stage shown on Fig. 10a has a loss of 10 db. at 900 kc. The phase shift is 72 degrees. The total phase shift for a 40 db. loss with four of these' circuits would be 4X72=288. This amplifier would not be stable with 30 db. feedback. The maximum phase shift per stage may not exceed .l80' o 4 45 This occurs at 300 kc. with a loss per circuit of only 3 db. ,Such an amplifier would be stable if the feedback did not exceed 12 db. This would not allow any margin of safety to take care of possible gain variation in the amplifier caused by changes in tubes and operating voltages, etc.

If the resistor23 and condenser 25 shown connected between points a and b, Fig. 11a, is added to the circuit, the phase shift and loss will change to conform to the solid curves. From these curves it is obvious that it will be possible to obtain a loss in loop gain of 10 db. per stage with a phase shift of only 16 degrees at 250 kc. The

loss would then be the required 40 db. with a total cumulative phase shift of only 4 16=64" At th-e frequency of 45 degrees "phase shift per circuit, the loss per circuit is more than 18 db.,

- or a total loss of 72 db. Such an amplifier would have more than ample margin of safety for stagrids of the modulator tubes. This transformer has a primary winding 55 and a secondary winding 51, both balanced to ground in all respects.

, from primary to secondary taps.

The secondary winding 5'! has more turns than the primary, in the ratio of 1.4 to 1.0, but has taps so it is possible to obtain a '1 to 1 turn ratio densers are connected from each end to the primary winding to the secondary tap of corresponding polarity and voltage, causing the transformer to be, in effect, an auto-transformer at high frequencies. The step-up section of the secondary winding is shunted by a resistance 6| and capacity 63 in series. circuit is somewhat similar to that of the high frequency circuit of Fig. 11a. Without this cirhave a final limit of 270 degrees instead of 90 bility with 30 db. of feedback. Such an amplifier should be operative with 11 such high frequency circuits and have a limit of feedback of nearly 200 db., as governed, by high frequency characteristics.

Results similar to those obtained withthe circuit described on Fig. 11a may be obtained by having a tap on the 10,000 ohm resistoril 'and connecting a capacity 28 across a section of the resistor as shown in Fig. 5, element C rep-- resenting input and output capacities of the tubes plus the distributed capacity to ground of the circuit elements.

The 80! stage, Figs. 13 and 13a does not have any high frequency control circuits.

The 845 stage, see Figs. 15 and 15a. uses a transformer to couple its plate circuit to the cuit the phase shift of the transformer would degrees and it would be impossible to meet requirements for stability with 30 db. of feedback.

My computations have shown that the phase shift of a circuit as used to connect the driver transformer may be made less if the connections are asshown in'Fig. 2, analyzed in Fig. 6. The

circuitin Fig. 2 is an improvement over the v corresponding portion in Fig. 1 in that there is provided a low impedance path around the trans-' former, i. e.-, from the plates of the 845 stage to the grids of the 89 [R stage, by means of the path directly through condensers 59' and 63'. The leakage reactance between the 1:1 sections is effectively shorted by condenser 63'. The modification in Fig. 2, while preferable to the corre-- sponding portion in Fig. 1, is not shown'there becauseof the desirability of making Fig. 1 correspond to the system actually used in the 5 kw. transmitter, type 5D.

The final stage of the modulator may be represented as shown in Figs. 17 and 17a. The feedback voltage is taken off at the plate of the modulator tubes 65 and '86. The modulator load is. fed through a 100 ohm resistance 8|. The distributed. capacity across the primary 8! would cause the phaseshift of this stage to reach 90 degrees at less than 500 kilocycles. The 100 ohm resistor 8i limits thisphase shift to degrees and causes the phase shift at very high frequen: cies to return to nearly.0.

The performance of the low frequency phase control circuits used in the 801 and i603 stages is illustrated by Figs. 12 and 10, respectively. The results of using these circuits are similar to those obtained with the high frequency circuit, Fig. 11a. A shunt inductance 30 ona portion of the plate resistor 2|, 'as shown in Fig. 7,

show the limiting values of phase shift and attenuation, depending on primary inductance of the transformer which is. a function of signal voltage. a

Fig. 16 shows a similar set of curves for the modulator stage.

and losses of each of the 4. circuits to obtain the total for the modulator feedback loop at low frequencies.

Fig. 18:: shows the same for high frequencies.

Large con- The action of this The use of the ohm resistors 8| limits the .phase shift of this stage to This is a condition that is impossible. I ever, if Re is made large, the value of the frac- Fig. 19 gives the results of FIE. 18 and Fig. 180 in polar coordinates.

Referring to Fig. 3, I have shown a common cathode resistor method of obtaining couplin between halves of a push-pull resistance coupled circuit. It is desirable to have a substantially larger coupling resistor in the cathode circuit than is req'uired'for bias in order to properly couple both halves'of the push-pull stage.

31 and 39 are two like tubes each having constants and Rp (internal plate resistance).

7 Let the grid 31 be connected to its cathode.

Let a voltage e be introduced in series with Re.

A current will flow in Re causing a voltage to appear in series with e. 7

Let the sum of these voltages e0, that is, the

voltage from cathode to ground.

eo=e+ERe whereERs induced voltage in Re.

Referring to Fig. 4:

I Now introduce a voltage e1 from grid of 31 to cathode. A voltage 11-91 will appear in the plate circuit of tube 31.

only at high frequencies. Inasmuch as the total feed back is 30 db. this is not serious.

From the foregoing it is apparent that I have provided an improved negative feedback amplifying system, characterized by high efficiency and stability of operation, high fidelity and freedom from hum and noise. While the system has pare ticular utility as a modulating amplifier in a high power radio transmission system, it may be used as an amplifier for many other purposes where high fidelity and emciency are required, e. g.,

public address and sound film amplifier systems. When used as a high level modulator in a 5 kw. broadcast transmitter, by way of example, it makes possible the following: a substantial yearly saving in cost of power; lower cost of tubes because of longer life resulting from pure class B operation; high fidelity over the entire audio frequency band (less than 3% R. M. -S.) as distinguished from feedbacksystems heretofore used and tend to increase high frequencydistortion; freedom from further adjustments of the negative feedback circuit after installation, even in case of a change in power from 1 to 5 kw. or the replacement of tubes; stable operation of amplifier although having db. or more of degenerative feedback. Although the feedback acts from the anodes of the class B modulator tubes back to the audio input A part of this voltage corresponding to e of the.

previous discussion will appear in Re- ReI l (v) R;+'R .+RZ-

. This voltage will be altered by tube 3! to give To have perfect push-pull voltage in the plate circuit load resistors R1. and Ro when a voltage is impressed from only one grid to ground e0, must equal 01 and from (VII) tion can be made to approach 1.

In the first audio stage of the amplifier used in the modulator the coupling is approximately 80%. Another fact obvious from the above discussion is that the amplification of the stage is onlyone half for unbalance voltage because the input unbalance is divided nearly equally between grid to cathode and cathode to ground. This will result in a 6 db. loss in feedback at a frequency where no voltage is fed back from one modulator plate.

The modulation transformer furnishes exceltransformer, in a high level modulation system with which the audio amplifier is used, the feedback device will compensate for distortion and hum ripple occurring in the class C radio frequency amplifier portion of the transmitter, not

shown, and also tends to compensate the modu-' latlng voltage of the class 0 stage.

While I have shown and described certain embodiments of my invention, it will, of course, be understood that I do not wishto be limited thereto, since modifications may be made in the circuits and apparatus employed without departing from the spirit and scope of my invention lent coupling between the halves of the circuit at low frequencies. At high frequencies it fails because of'leakage reactance between halves of its primary. This loss of feedback will occur correcting network connected between another as set forth in the appended claims.

I claim as my invention:

1. In a negative feedback audio frequency amplifier, a plurality of amplifier devices, coupling means provided with phase correcting networks, means forming. a feedback loop circuit with said amplifier devices and said coupling means for compensating against distortion, one of said coupling means comprising a transformer having other than unity ratio of turns between a pair of primary and secondary windings, a capacitor con-v nected between a high potential terminal of the lower impedance winding to a tap on the higher impedance winding so that the turn ratio with respect to said tap is substantially unity, a resistor connected at one end to a high potential terminal of said secondary winding in series relation therewith, and a second capacitor connected between said tap and the other end-of said resistor, whereby a low impedance path for high pair of said amplifier devices, and a. degenerative feedback circuit characterized by absence of phase shifts connected between said output and input circuits, said circuit forming with said amplifier devices and coupling means afeedback loop that is equalized over a range from-verylow to very high frequencies substantially exceeding in extent said transmission band.

3. In an electric wave transmission system, a balanced input transformer having a primary and a pair of secondaries, a balanced amplifier having cathode, output and input electrodes connected to said secondaries, a second balanced amplifier, resistance coupling means between said amplifiers provided with phase correcting network, a third balanced amplifier coupled to the output of said second amplifier, transformer coupling means connected to the input electrodes of said third amplifier and provided with phase correctin'g'networks, a frequency equalized potential divider connected betweetn the output electrodes of said third amplifier and ground, said divider comprising a plurality of resistors connected in series and shunted by a' plurality of capacitors, respectively, means forming a negative feedback loop comprising a, pair of shielded conductors directly connected between potential points of said divider on either side of the ground potential point and the low potential ends of said input secondaries, respectively.

4. The invention as set forth in claim 3 wherein negative bias means is connected in circuit with said first balanced amplifier to supply oper-.-' ating bias for the input electrodes and to compensate against a substantial amount of positive bias introduced by said feedback conductor connections.

5. An electrical wave amplifying system comprising a cascade amplifier, input and output coupling means therefor, a frequency compensated potential divider network including capacity connected in parallel with said output coupling means and to ground, said output coupling means being connected to ground exclusively of said network, an inverse feedback connection substantially free of impedance elements and phase shiftincluding a, conductor connected from a low potential point on said divider network and a low potential side of said input coupling means, and a grounded shield for said conductor, the capacity between said conductor and shield being included in a portion of the capacity of said divider network.

electrical wave amplifying system com prising first'and last amplifiers in cascade, said clusively of said divider network, said input coupling means being connected to an input electrode of said first amplifier, and a ground return path for said input coupling means including a low impedance connection from a low potential side of said input coupling means to a low potential point on said divider network.

'7. An electrical wave amplifying system comprising a plurality of push-pull stages in cascade relation, balanced input and output coupling means for said system, a frequency compensated network connected in parallel with said output coupling means and to ground, said output coupling'means being connected to ground exclusively of said network, and a pair of ground return paths for said input coupling means incl-uding low impedance connections from low potential sides of said input coupling means to points, I respectively, on said divider network and through low potential portions of said network to ground.

8. An audio frequency amplifying system fora wide transmission band comprising input and output balanced circuits, an even number of balanced amplifier stages, one of said stages comprising a pair of electric discharge devices having its'in'put resistance coupled in push-pull relation resistance means common to said devices in a mid-branch for ensuring symmetry of pushpull operation, the last of said stages comprising a pair of electric discharge devices connected for in cascade, at least one of said stages being pushpull resistance coupled, the amplifiers of said stages having input, output and cathode electrodes, balanced input and output coupling means for said system, degenerative feedback paths connected from both sides of said output to opposite sides of said input coupling means, respectively, and resistance means connected in a circuit common to both sides of said push-pull stage, said means having sufiicient magnitude of resistance -to currents of audio signal frequency in said band to maintain substantially equal signal voltages and degree phase relation between both sides of said resistance coupled stage.

10. In an audio frequency amplifying system for a wide band of frequencies having input and output balanced circuits, a plurality of pairs of electric discharge devices connected'in balanced cascade relation and having input, output and cathode electrodes, one of said pairs being resistance coupled in push-pull relation, resistance means connected in a path common to the oathode circuits of said resistance coupled pair for ensuring symmetry of push-pull action, a driving transformer having its secondary windings connected to input electrodes of a following pair of discharge devices, respectively, means affording high frequency paths connected between points on the primary and secondary windings of substantially equal points of signal voltage, respectively, for minimizing phase shift at high frequencies, and means forming afeedback loop connected between one side of said balanced output circuit and one side of said balanced input circuit including insaid loop said transformer, said loop being equalized over a range from low to high frequencies substantially exceeding said band. i

11. An audio frequency amplifying system for wide band signal transmission comprising balanced input and output circuits, a plurality of amplifier stages coupled in cascade relation including a stage arranged for class B operation, said stages including electric discharge devices having input, output and cathode electrodes, inductive coupling means'connected to the input electrodes of said class B stage, a feedback circuit connected from at least one side of said bal- 8 anced output circuit to the opposite side of said balanced input circuit and forming a loop including said stages, push-pull resistance coupling means connected to the input electrodes "of a balanced amplifier stage preceding said class B 12-. The invention as set forth in claim 2,

wherein said transformer coupling means has a ratio of transformation ,other'than unity, and a stage, resistance means in a common branch cir.-

path for high frequency currents connected between points of substantially equal potential on primary and secondary windings, respectively.

.13. The invention as set forth in claim 2, wherein said transformer coupling means has a ratio of transformation and turns other than unity, a path for high frequency currents con- I ,nected between points of substantially equal ratio of transformation, respectively, and a resistor capacitor circuit connected from one of said points to a high voltage point on one of said windings having the greaternumber of turns.


US2255804A 1937-07-14 1937-07-14 Feedback wave translating system Expired - Lifetime US2255804A (en)

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BE429159A BE429159A (en) 1937-07-14
US2255804A US2255804A (en) 1937-07-14 1937-07-14 Feedback wave translating system
GB2089238A GB516935A (en) 1937-07-14 1938-07-14 Improvements in electrical wave amplifying systems

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US2835749A (en) * 1954-06-17 1958-05-20 Garrett Corp Feedback amplifiers

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US2835749A (en) * 1954-06-17 1958-05-20 Garrett Corp Feedback amplifiers

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BE429159A (en) grant
GB516935A (en) 1940-01-16 application

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