US20100066450A1 - High-Speed Low-Power Differential Receiver - Google Patents

High-Speed Low-Power Differential Receiver Download PDF

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US20100066450A1
US20100066450A1 US12/524,525 US52452508A US2010066450A1 US 20100066450 A1 US20100066450 A1 US 20100066450A1 US 52452508 A US52452508 A US 52452508A US 2010066450 A1 US2010066450 A1 US 2010066450A1
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voltage
current
transistor
common
coupled
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Robert E. Palmer
John W. Poulton
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Rambus Inc
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Rambus Inc
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Priority to PCT/US2008/001826 priority patent/WO2008100494A2/en
Assigned to RAMBUS INC. reassignment RAMBUS INC. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: PALMER, ROBERT E., MR., POULTON, JOHN W., MR.
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    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00Baseband systems
    • H04L25/02Details ; Arrangements for supplying electrical power along data transmission lines
    • H04L25/06Dc level restoring means; Bias distortion correction decision circuits providing symbol by symbol detection
    • H04L25/061Dc level restoring means; Bias distortion correction decision circuits providing symbol by symbol detection providing hard decisions only; arrangements for tracking or suppressing unwanted low frequency components, e.g. removal of dc offset
    • H04L25/063Setting decision thresholds using feedback techniques only
    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F3/00Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/45Differential amplifiers
    • H03F3/45071Differential amplifiers with semiconductor devices only
    • H03F3/45076Differential amplifiers with semiconductor devices only characterised by the way of implementation of the active amplifying circuit in the differential amplifier
    • H03F3/45179Differential amplifiers with semiconductor devices only characterised by the way of implementation of the active amplifying circuit in the differential amplifier using MOSFET transistors as the active amplifying circuit
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00Baseband systems
    • H04L25/02Details ; Arrangements for supplying electrical power along data transmission lines
    • H04L25/0264Arrangements for coupling to transmission lines
    • H04L25/0278Arrangements for impedance matching
    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F2203/00Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/45Indexing scheme relating to differential amplifiers
    • H03F2203/45008Indexing scheme relating to differential amplifiers the addition of two signals being made by a resistor addition circuit for producing the common mode signal
    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F2203/00Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/45Indexing scheme relating to differential amplifiers
    • H03F2203/45306Indexing scheme relating to differential amplifiers the common gate stage implemented as dif amp eventually for cascode dif amp
    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F2203/00Indexing scheme relating to amplifiers with only discharge tubes or only semiconductor devices as amplifying elements covered by H03F3/00
    • H03F2203/45Indexing scheme relating to differential amplifiers
    • H03F2203/45588Indexing scheme relating to differential amplifiers the IC comprising offset compensating means
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L7/00Arrangements for synchronising receiver with transmitter
    • H04L7/04Speed or phase control by synchronisation signals
    • H04L7/041Speed or phase control by synchronisation signals using special codes as synchronising signal
    • H04L7/046Speed or phase control by synchronisation signals using special codes as synchronising signal using a dotting sequence

Abstract

A low-voltage differential communication system includes a low- and programmable-swing voltage-mode transmitter that delivers a low-voltage differential signal to a receiver via differential channel. The receiver employs two input transistors, each in a common-gate configuration, to recover the low-voltage differential signal. A current source in the receiver biases the input transistors such that their source voltages are nominally biased at the common-mode voltage of the differential signal, and their gate-source voltages remain essentially constant with common-mode-voltage fluctuations.

Description

    FIELD
  • The subject matter disclosed herein relates generally to the field of communications, and more particularly to high speed electronic signaling within and between integrated circuit devices.
  • BACKGROUND
  • Present-day computing systems require very high off-chip communications bandwidth, and high-speed serial links for chip-to-chip interconnect are now ubiquitous. Many of these links have channels that present modest attenuation, crosstalk, and reflections. Chip-to-chip serial links, however, have generally evolved from backplane transceivers that must deal with comparatively difficult channels, and so often dissipate far more power than necessary for short-haul chip-to-chip links. Such channels are therefore relatively power inefficient.
  • Many systems would benefit from links that require less power to move information at high data rates. For example, laptop computers quickly drain expensive batteries, and power dissipated as heat can be uncomfortable and often necessitates noisy fans and/or complex power-management schemes. Perhaps more important, power requirements significantly impact the usage time and performance of handheld devices. There is therefore a demand for systems and methods for communicating data at high speeds using minimal power. The term “specific power” is a helpful performance metric that considers the power required to move data at speed, and is typically expressed in units of Watts per Gigabit per second (W/Gb/s). The demand for efficient high-speed signaling can thus be phrased as a need for systems with very low specific-power requirements.
  • BRIEF DESCRIPTION OF THE DRAWINGS
  • The subject matter disclosed is illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings and in which like reference numerals refer to similar elements and in which:
  • FIG. 1 depicts a differential, low-voltage system 100 in accordance with one embodiment.
  • FIG. 2 details a current source 200 that can be used as an embodiment of current source 190 of FIG. 1.
  • FIG. 3 details a transmitter 300 for use in one embodiment.
  • FIG. 4 details an embodiment of a receiver 400 for use in one embodiment.
  • DETAILED DESCRIPTION
  • FIG. 1 depicts a differential, low-voltage system 100 in accordance with one embodiment. System 100 includes a transmitter 105 and a receiver 110 interconnected by a differential communication channel 115. Transmitter 105 efficiently delivers a low-voltage differential signal to receiver 110. Receiver 110 amplifies the differential signal while providing excellent common-mode noise rejection. Both the transmitter and receiver have very low specific-power requirements, and can thus be used to advantage in applications that require both power efficiency and speed performance.
  • Transmitter 105 includes a pair of voltage regulators 120 and 122, a complementary pre-driver 125, a differential stacked nMOS transmit stage 130, and a pair of complementary output pads 135/140. Output driver 130 is powered by regulator 122 and ground potential. Regulator 122 can adjust swing voltage Vo such that the output signal TX[P,N] is anywhere between 50 and 300 mVppd (millivolts peak-to-peak, differential).
  • Pre-driver 125 splits incoming data stream Din into complementary bit streams DP/DN to the gates of the nMOS transistors within output driver 130. When signal DP is high and DN low, the upper left and lower right nMOS transistors are turned on. In that case, current flows from power node Vo to ground via channel 115 and receiver 110. Conversely, when signal DP is low and DN high, the upper right and lower left nMOS transistors are turned on. Once again, current flows from power node Vo to ground via channel 115 and receiver 110, albeit in the opposite direction through receiver 110. The power efficiency of this signaling scheme is evident in the fact that substantially all the current between supply nodes Vo and ground is used to convey signals to receiver 110.
  • Pre-driver 125 is powered by regulator 120 and ground potential, so the signals DP/DN applied to the nMOS gates alternate between regulated voltage Vr and ground. Regulator 120 adjusts voltage Vr to set the sum of the pull-up and pull-down impedances of stage 130 approximately equal to the differential impedance of channel 115. The relative sizes of the pull-up and pull-down transistors may be fixed at design time to give equal impedance at an assumed operating point.
  • The differential impedance looking into the transmitter output terminals 135/140 is fairly independent of the common-mode voltage Vcm of signal TX[P,N] because the pull-up transistor is operating as a source follower while the pull-down transistor is common-source. As common-mode voltage Vcm increases, the pull-down's small-signal impedance increases, while the pull-up's small-signal impedance decreases.
  • To provide high-quality termination, the capacitance shunting the termination should be small. In both the transmitter and receiver, shunt capacitance is generally dominated by wiring and ESD clamps 145/150. Using a low voltage Vo and referencing the output signal TX[P,N] to ground allows ESD clamps 145/150 to include forward-biased diodes. The overall ESD protection strategy is detailed later in connection with FIG. 3. Regulator 122 keeps voltage Vo low enough to prevent the forward-biased ESD diode from turning on (i.e., Vo<Vtd), and can be adjusted between 50 and 300 mVppd in one embodiment. That is, the common-mode voltage Vcm can be adjusted between 25 and 150 mV. Regulator 120 then regulates voltage Vr such that the nMOS transistors of output driver 130, when biased on, exhibit a desired resistance and remain in the triode region. In one embodiment, the desired on-resistance for the nMOS transistors is fifty ohms.
  • In a typical example, assume it is desired that transmitter 105 convey a differential signal TX[P,N] having an output swing voltage of about 200 mVppd and exhibiting a differential output impedance of 100 ohms. Regulator 122 may be set such that output voltage Vo is about 200 mV. The output swing of each half of the differential output driver 130 is equal to half the regulated supply voltage Vo, centered at a common-mode voltage equal to half the regulated supply voltage Vo. Regulator 120 may then set voltage Vr to a level that provides, on each of the complementary transmitter outputs, an output impedance of the desired 50 ohms. There are a number of methods and circuits for controlling Vr to obtain a desired impedance, some of which are discussed below. A single-ended transmitter that is in many respects similar to transmitter 105 is detailed in “Self Terminating Low Voltage Swing CMOS Output Driver,” by Tom Knight and Alex Krymm (CICC 1987).
  • The transistors within output driver 130 double as termination impedances. As such, the “on” impedance of each transistor should be matched to the channel and should behave linearly (like a resistor) over the range of output voltages. To accomplish this, each transistor is biased to remain in the “linear” region (aka the “triode” region) when conducting. When a transistor is in the linear region the current through the transistor changes linearly with changes in drain-source voltage Vds: in other words, the transistor behaves like a resistor.
  • As is well known, a transistor is in the linear region when its gate/source voltage Vgs is greater than the threshold voltage Vt of the transistor (Vgs>Vt) and its drain-source voltage Vds is less than or equal to Vgs−Vt. Stated mathematically, a transistor is in the linear region when:

  • Vgs>Vt&Vds≦Vgs−Vt  (1)
  • In FIG. 1, output driver 130 is referenced to ground, or zero volts. The voltage Vgs for a given transistor is therefore either zero volts (transistor off) or Vr (transistor on), and Vds is at most 0.75×Vo. We can ignore the case where a transistor is off because the impedance of a transistor biased off is very large relative to the impedance of the transistor biased on. Equation (1) therefore simplifies to:

  • 0.75*Vo≦Vr−Vt.  (2)
  • which can be rearranged to derive the voltage Vr that keeps the transistors of output driver 130 in the triode region when active:

  • Vr≧0.75*Vo+Vt  (3)
  • The transistors in driver 130 are selected such that regulator 120 can provide a voltage Vr that produces a desired output impedance over the expected range of output-voltage swings. The voltage Vt in equations 1 through 3 relates to the transistors within driver 130, and is assumed to be the same or similar for each nMOS transistor in this example. Voltage Vt may be the same or different from diode threshold Vtd.
  • A waveform diagram 155 inset over channel 115 in FIG. 1 shows output signal TX[P,N] as a differential signal between output voltage Vo and ground, and shows that output voltage Vo is well below Vtd and regulated voltage Vr is above Vtd. In a typical example, voltage Vo is 200 mV, Vcm is 100 mV, signal TX[P,N] is 100 mV peak-to-peak (200 mVppd) centered about voltage Vcm, threshold voltages Vt and Vtd are respectively about 400 mV and 600 mv, and regulated voltage Vr is somewhere above 700 mV and is selected such that the output impedance of an nMOS transistor in driver 130 is 50 ohms when biased on.
  • Receiver 110 receives the low-voltage differential signal TX[P,N] as a differential, low-common-mode, signal RX[P,N] on complementary input pads 165/170. ESD clamps 175/180 similar to clamps 145 and 150 protect these differential input nodes and the input devices of receiver 110 from damage due to electrostatic discharge. Pads 165 and 170 drive a common-gate amplifier 185 that provides both gain and level conversion for the incoming differential signal RX[P,N]. In this example, the inputs of amplifier 185 are terminated for the differential mode only, and about 25% of the termination impedance is represented by the amplifier's input impedance.
  • A voltage divider consisting of two equivalent resistors R1 and R2 (e.g., 75 ohms each) extends between input nodes RXP and RXN. The node common to resistors R1 and R2 is a tap from which amplifier 185 extracts the common-mode voltage Vcm of the incoming signal. A first nMOS transistor T1 has its source coupled to input node 165 and, via a resistor R3, to ground; its drain tied to supply voltage Vdd via a resistor R4; and its gate tied to a bias voltage Vcg (for “voltage common-gate”). A second nMOS transistor T2 has its source coupled to input node 170 and, via a resistor R5, to ground; its drain tied to supply voltage Vdd via a resistor R6; and its gate tied to bias voltage Vcg. A third nMOS transistor T3 has its source coupled to the common-mode-voltage tap Vcm; its drain coupled to a current source 190 that delivers a bias current Ibias; and its gate tied to both its drain and to bias voltage Vcg.
  • Common-gate amplifier 185 works well for relatively low input voltages, and the bias scheme provided by current source 190 and transistor T3 affords considerable common-mode rejection without compromising the amplifier bandwidth.
  • Common-mode voltage fluctuations appear on the sources of transistors T1 and T2, and at the common-mode-voltage tap Vcm between resistors R1 and R2. The gate-source voltage Vgs3 of transistor T3 depends upon the value of bias current Ibias. Because bias current Ibias is constant, so too is voltage Vgs3. Common-gate voltage Vcg is the sum of Vcm and Vcg3, so common-gate voltage Vcg rises and falls with voltage Vcm. Common-mode-voltage fluctuations thus appear on both the sources and the gates of transistors T1 and T2. As a consequence, the gate-source voltages Vgs1 and Vgs2 of input transistors T1 and T2 remain constant in the face of common-mode noise. It follows that the currents through resistors R4 and R6 and respective output voltages DinP and DinN also remain constant. Common-gate amplifier 185 thus provides effective common-mode noise rejection. Further, the common-mode rejection circuitry, which includes current source 190 and transistor T3 in this embodiment, is outside of the differential signal paths between nodes 165/170 and the sources of respective transistors T1 and T2. Transistor T3 is diode-connected in this embodiment, and functions to convert bias current Ibias into a stable gate-source voltage Vgs. Transistor T3 might be replaced with e.g. a diode or resistor in other embodiments.
  • In some embodiments the source voltages Vs1 and Vs2 of transistors T1 and T2, the inputs of common-gate amplifier 185, are biased to the nominal common-mode voltage. In this example, voltage Vcm should be at or very near output voltage Vo divided by two (Vcm=Vo/2). Bias current Ibias is thus carefully controlled such that the gate-source voltage Vgs3 of transistor T3 sets voltage Vcg to a level that biases the source voltages Vs1 and Vs2 of transistors T1 and T2 to the nominal common-mode voltage Vo/2.
  • FIG. 2 details a current source 200 that can be used as an embodiment of current source 190 of FIG. 1. Current source 200 includes a voltage reference 202, a replica common-gate amplifier 205, a level converter 210, an operational amplifier 215, and some bias circuitry 220 to create the appropriate bias levels for converter 210 and op-amp 215. The purpose of current source 200, as noted above in connection with current source 190 of FIG. 1, is to set current Ibias to a level that ensures transistors T1 and T2 (FIG. 1) are biased such that their source voltages Vs1 and Vs2 are set to the nominal common-mode voltage Vo/2.
  • Voltage reference 202 is a simple voltage divider, in this example, that divides output voltage Vo by two. Voltage Vo or Vo/2 can be sourced on or off chip, and can be fixed or adjustable. In embodiments in which receiver 110 has one or more neighboring transmitter like transmitter 105 of FIG. 1, voltage Vo can be taken from one of those. Alternatively, voltage Vo/2 can be derived from the common-mode voltage Vcm at the receiver. Many other methods of providing an appropriate reference common-mode voltage can be used, as will be readily apparent to those of skill in the art.
  • Replica amplifier 205 includes transistors T1′ and T3′ and resistors R3′ and R4′, each of which is fabricated to behave in the same manner as the like-named elements of FIG. 1. Transistors T1′ and T3 and resistors R3′ and R4′ can thus be expected to behave in the same manner as transistor T1 and resistors R3 and R4, respectively.
  • Level converter 210 raises voltages Vo/2 and Vcm′ to Vo1 and Vcm1 respectively. Op-amp 215 then amplifies the difference between signals Vo1 and Vcm1 and presents the resulting output voltage to the gate of an nMOS transistor T4, and consequently controls a bias current Ibias′. A current minor comprised of PMOS transistors T5, T6, and T7 replicates current Ibias′, by distributing a bias voltage Pbias, to produce current Ibias of FIG. 1 and Ibias″ in replica amplifier 205. (The transistor T8 below transistor T6 is off when the receiver is operational.)
  • The feedback from the output of op-amp 215 to transistor T7 forces voltage Vcg′ to change until the current through transistor T1′ and resistor R3′ causes voltage Vcm′, the source voltage of transistor T1′, to match the nominal common-mode voltage Vo/2. The current Ibias″ dictates voltage Vcg′, so the identical bias current Ibias sets voltage Vcg of FIG. 1 equal to Vcg′. Because transistor T1′ and resistors R3′ and R4′ replicate transistor T1 and resistors R3 and R4, applying a gate voltage Vcg that matches the replica control voltage Vcg′ sets the source of transistor T1 to the nominal common-mode voltage Vo/2. The same holds true for transistor T2. Current source 200 thus ensures that the source voltages of transistors T1 and T2 are nominally biased at voltage Vo/2.
  • Returning to FIG. 2, a pair of inverters 250 and 255 generates signals ponL and ponH from a power-on signal Pon. Asserting signal Pon places current source 200 in an operational mode, whereas deasserting signal Pon places current source 200 in a low-power state. Voltage Vo may be pulled to ground upon power down, or reference voltage 202 can otherwise be disabled in a manner that reduces or eliminates current flow when current source 200 is disabled.
  • FIG. 3 details a transmitter 300 for use in one embodiment. Transmitter 300 is in many ways similar to transmitter 105 of FIG. 1, with like-labeled elements being the same or similar. Output terminals TX[P,N] are driven by an N-over-N output driver 130 operating between ground GND and a regulated supply voltage Vo, which is generated by an on-chip regulator 305 in this embodiment. A pre-driver 125 operating between ground GND and a second regulated voltage Vr drives the gates of the nMOS transistors in driver 130. The second regulated voltage Vr determines the output impedance of output driver 130 and is generated by an on-chip regulator 310 in this embodiment. The output voltage Vo, and thus the output swing, are adjusted digitally via a DAC 315 that sets the currents both regulators 305 and 310. In systems with a number of transmitters, one or both of regulators 305 and 310 may be shared among the transmitters. Providing each transmitter with a dedicated regulator or regulators, however, allows the output impedance and/or output swing to be adjusted on a per-transmitter basis.
  • Voltage regulator 310 is a replica-bias circuit that forces a replica transmitter 320 to have the same impedance as a scaled resistor. This method sets the sum of the pull-up and pull-down impedances of driver 130 approximately equal to the line impedance of an associated channel (e.g., channel 115 of FIG. 1). The relative sizes of the pull-up and pull-down transistors are fixed at design time to give equal impedance at an assumed operating point.
  • The differential impedance looking into the transmitter output terminals (e.g., nodes TX[P,N]) is fairly independent of common-mode voltage Vcm because the active pull-up transistor is configured as a source follower while the active pull-down transistor is in a common-source configuration. As the common-mode voltage Vcm increases, the pull-down transistor's small-signal impedance increases, while the pull-up transistor's impedance decreases. During a data transition, the output impedance depends on the details of the trajectories of the drive voltages at the gates of the output transistors. Simulation indicates that the small-signal differential output impedance varies by less than 15% during a transition.
  • The capacitance Cbyp rejects noise and contributes to the termination impedance. Line currents into terminals TX[P,N] flow through the pull-up/pull-down impedances in series with the regulator 305 output impedance, which is dominated by the reactance XCbyp of capacitance Cbyp at high frequencies. Capacitance Cbyp, about 36 pF in one example, may be implemented in the 2.5-V “native” nMOS device and occupy about 8400 um2. Capacitance Cbyp can be implemented as a thick-oxide device for improved reliability, but this capacitor is charged to less than 300 mV, so the oxide is not heavily stressed in any case. Use of the 1-V native device would have provided 2.5 the capacitance in the same area.
  • As noted previously, pre-driver 125 is powered from the regulated termination-control Voltage Vr. The paired inverters ensure that transition times for both data-edge polarities will be equal. The pre-driver provides fanout, which allows a 2:1 multiplexer 325 at the input of pre-driver 125 to be drawn quite small, thereby minimizing the load on a half-bit-rate distributed clock Clk[P,N]. Since pre-driver 125 is powered from a regulated supply, it is fairly immune to power supply noise and introduces very little timing jitter. The variation of regulated voltage Vr across cases tends to make the edge rate of the gate control signals driving the output stage nearly constant across PVT variation, and the edge rate of the transmitter output is also nearly constant.
  • With reference to both voltage regulators, since the voltages that must be compared within the regulator circuitry are near GND, common-gate (CG) nMOS amplifiers are a good choice for the gain blocks. A Vt-referenced current source 330 produces a bias voltage Vb for pMOS DAC 315. The current in the reference is proportional to Vtn/R, so the voltage across the resistor load in the DAC is Vref=KVtn, independent of R, where K can be adjusted digitally via Vs_set. The diode-connected nMOS in DAC 315 is drawn the same size as the nMOS common-gate devices in regulators 305 and 310, and, assuming high loop gain, the voltage appears at all four of the input terminals of the CG amplifiers.
  • Regulator 305 is a simple series regulator that forces output voltage Vo equal to reference voltage Vref (Vo=Vref), thereby allowing Vo to be set directly by Vo_set. Regulator 305 has two poles in its loop transfer function: one set by the output impedance of the CG amplifier and the input capacitance of the pMOS series regulator, and the second by the load impedance and bypass capacitor Cbyp in output driver 130. An additional compensating capacitor Ccomp makes the amplifier pole the dominant one. The addition of capacitor Ccomp saves power in the CG error amp of regulator 305 and avoids an extremely large bypass capacitor, but compromises the power supply rejection for regulator 305, allowing half of the supply noise to get through the regulator between 10-100 MHz. A production version of this link may have the output pole be dominant, which may require a larger bypass capacitance and higher power consumption in the CG amplifier of regulator 305. For example, with capacitor Cbyp implemented in the thin-oxide native nMOS in the same 8200 um2 area, it appears possible to build a greatly improved regulator for about 1 mW of additional power dissipation.
  • Regulator 310 is a two-stage design. The first stage generates a “master” copy Vm of the Vr control voltage. Since the load current for the pMOS stage is near zero, it is easy to make the output pole for this two-pole regulator the dominant one, and power supply rejection is quite good. The second stage is a simple series regulator with a gain of one, and it serves to isolate the “master” voltage Vm from the time-varying load of the transmitter's pre-driver inverters. The transmitter replica 320 used to set the impedance is drawn very small ( 1/16th scale); mismatch between the replica devices and the main transmitter contributes about a 5% variation in output impedance.
  • All of the P+/poly de-salicided resistors are digitally trimmable by ±20% to account for process variation. Trim is performed using a bench measurement in some embodiments; a production link may use e.g. a resistor trim cell and an external reference resistor.
  • A half-bit-rate distributed clock Clk[P,N] is received and buffered in two CMOS inverter stages to drive the final 2:1 multiplexer (mux) 325 and a clock divider stage 335 that generates one-quarter, one-eighth, and one-sixteenth-rate clocks for a 16:2 mux stage 340. Since the clock loading is relatively small, the fanout of the two-stage clock buffer is set to two to avoid introducing jitter from power supply noise, while still dissipating relatively little power in the clock buffer.
  • An embodiment of ESD clamps 145/150 is detailed in the lower right corner of FIG. 3. Similar ESD claims can be used in receiver 110. In this example, a cross-coupled diode pair 445 extends between the output pad (e.g., 140) and ground. The left-most diode is forward biased, which is to say its anode is connected to the output terminal and its cathode is connected to ground. The ESD claim will therefore shunt current to ground if the voltage on node 140 rises above a diode threshold voltage Vtd. The series-connected nMOS transistors and the resistor represent the output from one leg of driver 130. Additional diodes 450 establish shunt paths between ground and supply line Vdd. Other suitable low-voltage ESD clamps are detailed by John Poulton in published U.S. Application No. 20060132996.
  • FIG. 4 details a receiver 400 for use in one embodiment. Input nodes RX[P,N] drive a common-gate amplifier that provides both gain and level conversion for a second-stage amplifier/equalizer 405. The input is terminated for the differential mode only, and about 25% of the termination impedance is represented by the common-gate amplifier's input. Correctly terminating the common mode in the common-gate amplifier would have required burning five times the power to maintain the proper input common-mode bias. This embodiment instead relies on the transmitter's common-mode termination along with the receiver's common-mode rejection.
  • Large nFET switches 410 enabled by a control signal dgate allow the receiver to be disconnected from the line for offset trim. The bias voltage Vcg may be developed in a replica bias arrangement as detailed previously in connection with FIG. 2. Voltage Vcg sets the input common-mode voltage of the receiver to the nominal output common-mode voltage of the transmitter; alternatively, voltage Vcg can be programmed externally.
  • The second-stage amplifier 405 is a fairly conventional source-degenerated differential amplifier. The amount of degeneration is controlled digitally via the eq_set input to a decoder 412. In an embodiment designed for 6.25 Gb/s operation in 90 nm CMOS, the maximum EQ setting in this amplifier provides about 8.7 dB of peaking at the Nyquist frequency relative to dc gain, with a slope of three dB/octave. The offset of the two input amplifier stages can be trimmed, separately from the samplers, by adjusting a trim signal eqtrim to a DAC 415. The EQ amplifier drives a bank of four samplers, two data and two edge, derived from StrongARM flip-flops, e.g. of the type described in J. Montanaro et al., “A 160-MHz, 32-b, 0.5-W CMOS RISC microprocessor,” IEEE J. Solid-State Circuits, vol. 31, no. 11, pp. 1703-1714, November 1996.
  • Each sampler has an associated 8-bit DAC 420 to cancel input offset to within about 1 mV. Each sampler pair (data and edge) drives identical 2:16 demultiplexers 425 composed of a binary tree of 1:2 stages. The samplers are equipped with an enable input, which, when signal enab_edge is de-asserted, prevent the samplers from toggling. Edge samplers are enabled infrequently by a software CDR and adaptive EQ algorithm, neither of which is detailed here. When disabled, the edge samplers and their associated 2:16 demultiplexer dissipate essentially no power. A 7-bit digitally programmable phase rotator 430, implemented as a PLL in one embodiment, accepts the distributed half-bit-rate clock clk[P,N] as input and generates four quadrature half-bit-rate clocks eclkN, eclkP, dclkN, and dclkP to drive the four samplers.
  • The input offset of receiver 400 may be trimmed iteratively. First, both input amplifiers are powered down, allowing each sampler to be trimmed with its inputs at Vdd. Next, the input amplifiers are powered on, with inputs RX[P,N] disconnected from the inputs of the common-gate amplifier by de-asserting dgate. The resulting offset, averaged over the samplers, is then trimmed by adjusting signal eqtrim to DAC 415. Finally, any residual offset is removed at each sampler. The maximum required offset trim across 32 receivers (128 samplers) was about 45 mV for the samplers and 25 mV for the input amplifiers in one example.
  • The above-described transmitters and receivers are designed to operate at a single speed, but could be modified to operate over a range of speeds, though providing serial links that work over a wide range of data rates typically requires more power consumption. Whether fixed or adjustable, the frequency of operation can be chosen to best utilize the process technology used to fabricate the devices.
  • In the foregoing description and in the accompanying drawings, specific terminology and drawing symbols are set forth to provide a thorough understanding of the present invention. In some instances, the terminology and symbols may imply specific details that are not required to practice the invention. For example, the interconnection between circuit elements or circuit blocks may be shown or described as multi-conductor or single conductor signal lines. Each of the multi-conductor signal lines may alternatively be single-conductor signal lines, and each of the single-conductor signal lines may alternatively be multi-conductor signal lines. Signals and signaling paths shown or described as being single-ended may also be differential, and vice-versa. Similarly, signals described or depicted as having active-high or active-low logic levels may have opposite logic levels in alternative embodiments.
  • With reference to FIG. 1, output driver 130 and amplifier 185 used nMOS transistors, which are well suited for low-voltage operation when configured as shown. However, other types of n- and p-type transistors might also be used. For example, transistors T1, T2, and T3 might be implemented using n-type bipolar junction transistors (BJTs). A BJT configured like transistor T1 would be in a common-base configuration, the current-handling terminals would be called the collector and emitter instead of the drain and source, and the BJT control terminal would be called the base instead of the gate.
  • An output of a process for designing an integrated circuit, or a portion of an integrated circuit, comprising one or more of the circuits described herein may be a computer-readable medium such as, for example, a magnetic tape or an optical or magnetic disk. The computer-readable medium may be encoded with data structures or other information describing circuitry that may be physically instantiated as an integrated circuit or portion of an integrated circuit. Although various formats may be used for such encoding, these data structures are commonly written in Caltech Intermediate Format (CIF), Calma GDS II Stream Format (GDSII), or Electronic Design Interchange Format (EDIF). Those of skill in the art of integrated circuit design can develop such data structures from schematic diagrams of the type detailed above and the corresponding descriptions and encode the data structures on computer readable medium. Those of skill in the art of integrated circuit fabrication can use such encoded data to fabricate integrated circuits comprising one or more of the circuits described herein.
  • While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. For example, the above-described links can be bidirectional, as would be required e.g. for memory systems, and might operate with a non-terminated receiver in channels with little crosstalk and reflections. Moreover, some components are shown directly connected to one another while others are shown connected via intermediate components. In each instance the method of interconnection, or “coupling,” establishes some desired electrical communication between two or more circuit nodes, or terminals. Such coupling may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description. Only those claims specifically reciting “means for” or “step for” should be construed in the manner required under the sixth paragraph of 35 U.S.C. §112.

Claims (23)

1. An amplifier comprising:
a voltage divider having a common-mode-voltage tap between differential first and second inputs;
a first transistor having a first current-handling terminal, coupled to the first input, and a first control terminal;
a second transistor having a second current-handling terminal, coupled to the second input, and a second control terminal coupled to the first control terminal;
a third transistor having a third current-handling terminal, coupled to the tap, a third control terminal coupled to the first control terminal, and a fourth current-handling terminal; and
a current source having a bias-current output node coupled to the fourth current-handling terminal to provide a bias current through the third transistor.
2. The amplifier of claim 1, wherein the first, second, and third transistors are nMOS transistors.
3. The amplifier of claim 1, wherein the bias current through the fourth current-handling terminal of the third transistor fixes a voltage between the tap and the first and second control terminals.
4. The amplifier of claim 1, wherein the current source includes a replica transistor having a replica control terminal and a replica current-handling terminal.
5. The amplifier of claim 4, wherein the current source is to hold the replica current-handling terminal at a reference voltage.
6. The amplifier of claim 5, wherein the reference voltage approximates a common-mode voltage on the tap.
7. The amplifier of claim 1, further comprising a diode having an anode connected to the first input and a cathode connected to a ground terminal at a ground potential, wherein the diode exhibits a threshold voltage.
8. The amplifier of claim 7, wherein a common-mode voltage on the tap is referenced to the ground potential, and wherein the common-mode voltage is less than half the threshold voltage.
9. A communication system comprising:
a differential transmitter having first and second transmitter nodes to convey a differential signal referenced to a ground node at ground potential;
first and second diodes each having an anode connected to one of the first and second transmitter nodes and a cathode connected to the ground node, wherein each of the first and second diodes conducts current from the anode to the cathode when the anode-to-cathode voltage exceeds a diode threshold voltage, and wherein the differential signal exhibits a common-mode voltage, referenced to the ground potential, that is less than half of the diode threshold voltage; and
a differential receiver having:
a voltage divider having a common-mode-voltage tap between the first and second transmitter nodes;
a first transistor having a first current-handling terminal, coupled to the first transmitter node, and a first control terminal;
a second transistor having a second current-handling terminal, coupled to the second transmitter node, and a second control terminal coupled to the first control terminal;
a third transistor having a third current-handling terminal, coupled to the tap, a third control terminal coupled to the first control terminal, and a fourth current-handling terminal; and
a current source having a bias-current output node coupled to the fourth current-handling terminal to provide a bias current through the third transistor.
10. The system of claim 9, wherein the first, second, and third transistors are nMOS transistors.
11. The system of claim 9, wherein the bias current through the fourth current-handling terminal of the third transistor fixes a second voltage between the tap and the first and second control terminals.
12. The system of claim 9, wherein the current source includes a replica transistor similar to the first transistor and having a replica control terminal and a replica current-handling terminal.
13. The amplifier of claim 12, wherein the current source is to hold the replica current-handling terminal at a reference voltage.
14. The amplifier of claim 13, wherein the reference voltage approximates the common-mode voltage.
15. A method for receiving first and second complementary signals on respective complementary input pads and exhibiting a common-mode voltage, wherein each of the complementary input pads is coupled to a supply node via a respective diode that exhibits a diode threshold voltage, the method comprising:
biasing a source or emitter of each of a first and second n-type transistors to a reference voltage less than half of the diode threshold voltage and a gate or base of the first and second transistors to a gate-bias voltage above the reference voltage;
applying the first and second complementary signals to the sources or emitters of the respective first and second n-type transistors;
extracting the common-mode voltage from the first and second complementary signals; and
changing the gate-bias voltage with changes in the extracted common-mode voltage, wherein the gate-bias voltage rises and falls with the extracted common-mode voltage.
16. The method of claim 15, further comprising generating the gate-bias voltage by passing a bias current through a third transistor having a gate or base connected to the gate or base of the first and second n-type transistors.
17. The method of claim 16, further comprising passing a current through a replica of the first transistors to establish the bias current.
18. An amplifier comprising:
complementary first and second inputs to receive a differential signal that exhibits a common-mode voltage;
a first transistor having a first current-handling terminal coupled to the first input, a second current-handling terminal, and a first control terminal;
a second transistor having a third current-handling terminal coupled to the second input, a fourth current-handling terminal, and a second control terminal coupled to the first control terminal;
means for biasing the first and third current-handling terminals at the common-mode voltage, and the first and second control terminals at a gate-bias voltage above the common-mode voltage; and
means for changing the gate-bias voltage with changes in the common-mode voltage.
19. A computer-readable medium having stored thereon a data structure defining an amplifier, the data structure comprising:
first data representing a voltage divider having a common-mode-voltage tap between differential first and second inputs;
second data representing a first transistor having a first current-handling terminal, coupled to the first input, and a first control terminal;
third data representing a second transistor having a second current-handling terminal, coupled to the second input, and a second control terminal coupled to the first control terminal;
fourth data representing a third transistor having a third current-handling terminal, coupled to the tap, a third control terminal coupled to the first control terminal, and a fourth current-handling terminal; and
fifth data representing a current source having a bias-current output node coupled to the fourth current-handling terminal to provide a bias current through the third transistor.
20. A receiver comprising:
complementary first and second differential inputs to receive a differential signal that exhibits a common-mode voltage;
complementary first and second signal paths that extend from the respective first and second differential inputs;
a differential amplifier having first and second amplifier inputs coupled to the respective first and second differential inputs via the first and second signal paths; and
common-mode rejection circuitry outside of the first and second differential signal paths.
21. The receiver of claim 20, further comprising a voltage divider extending between the first and second inputs and including a common-mode-voltage tap coupled to the common-mode rejection circuitry.
22. The receiver of claim 20, wherein the differential amplifier includes a first transistor and a second transistor, wherein the first amplifier input is a first source of the first transistor, and wherein the second amplifier input is a second source of the second transistor.
23. The receiver of claim 22, wherein the common-mode rejection circuitry includes a diode-connected third transistor, and wherein the first, second, and third transistors have interconnected gate terminals.
US12/524,525 2007-02-12 2008-02-11 High-Speed Low-Power Differential Receiver Abandoned US20100066450A1 (en)

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US8199866B2 (en) 2012-06-12
WO2008100843A2 (en) 2008-08-21
US20100220828A1 (en) 2010-09-02
WO2008100523A2 (en) 2008-08-21
US20100085100A1 (en) 2010-04-08
WO2008100494A4 (en) 2009-02-12
EP2111709A2 (en) 2009-10-28
WO2008100494A3 (en) 2008-12-24
WO2008100843A3 (en) 2009-02-12
JP2010518749A (en) 2010-05-27
CN101606365A (en) 2009-12-16
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WO2008100494A2 (en) 2008-08-21
US8310294B2 (en) 2012-11-13

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