US20030091100A1  Method and device for radio signal reception  Google Patents
Method and device for radio signal reception Download PDFInfo
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 US20030091100A1 US20030091100A1 US10312058 US31205802A US2003091100A1 US 20030091100 A1 US20030091100 A1 US 20030091100A1 US 10312058 US10312058 US 10312058 US 31205802 A US31205802 A US 31205802A US 2003091100 A1 US2003091100 A1 US 2003091100A1
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 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04B—TRANSMISSION
 H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00  H04B13/00; Details of transmission systems not characterised by the medium used for transmission
 H04B1/69—Spread spectrum techniques
 H04B1/707—Spread spectrum techniques using direct sequence modulation
 H04B1/7097—Interferencerelated aspects
 H04B1/7103—Interferencerelated aspects the interference being multiple access interference
 H04B1/7107—Subtractive interference cancellation
 H04B1/71075—Parallel interference cancellation

 H—ELECTRICITY
 H04—ELECTRIC COMMUNICATION TECHNIQUE
 H04B—TRANSMISSION
 H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00  H04B13/00; Details of transmission systems not characterised by the medium used for transmission
 H04B1/69—Spread spectrum techniques
 H04B1/707—Spread spectrum techniques using direct sequence modulation
 H04B1/709—Correlator structure
 H04B1/7093—Matched filter type
Abstract
Description
 [0001]The present invention relates to digital radio communication techniques using Code Division Multiple Access (CDMA).
 [0002]It is more especially aimed at multiuser detection procedures which are sometimes employed in these techniques to improve the reception performance.
 [0003]
 [0004]where n(t) is additive noise and U is the number of channels multiplexed on the CDMA carrier, the general expression for whose contributions y^{u}(t) is:
$\begin{array}{cc}{y}^{u}\ue8a0\left(t\right)=\sum _{i}\ue89e\text{\hspace{1em}}\ue89e{b}_{i}^{u}\xb7{s}_{i}^{u}\ue8a0\left(t\mathrm{iT}\right)& \left(2\right)\end{array}$  [0005]where:
 [0006]b_{i} ^{u }is the information symbol of rank i transmitted on the uth channel;
 [0007]s_{i} ^{u}(t) is a generalized code given by the convolution of the impulse response of the uth channel with the portion corresponding to the symbol b_{i} ^{u }of the spreading code c^{u }assigned to the channel.
 [0008]The number U corresponds to the number of users if each user involved utilizes a single channel. There may however be several channels per user (for example traffic and control).
 [0009]The spreading codes c^{u }are sequences of discrete samples called “chips”, with real values (±1) or complex values (±1 ±j), having a given chip rate. The symbols b_{i} ^{u }also have real values (±1) or complex values (±1 ±j). The duration of a symbol on a channel is a multiple of the chip duration, the ratio between the two being the spreading factor Q of the channel.
 [0010]In certain systems, the spreading factor may vary from one channel to another. In such a case, a common spreading factor Q is considered, equal to the greatest common divisor (GCD) of the U spreading factors Q^{u}. A symbol on the channel u is then regarded as the concatenation of Q^{u}/Q consecutive symbols b_{i} ^{u }whose values are identical.
 [0011]The duration of the generalized response s_{i} ^{u}(t) corresponds to Q+W−1 chips if W denotes the length of the impulse response expressed as a number of chips.
 [0012]By sampling at the chip rate the CDMA signal y(t) received for a block of n symbols on each of the channels, the receiver obtains complex samples that can be modeled by a vector Y of n×Q+W−1 components:
 Y=A.b+N (3)
 [0013]where:
 [0014]b denotes a column vector of size n×U, which can be decomposed into b^{T}=(b_{1} ^{T}, b_{2} ^{T}, . . . , b_{n} ^{T}), (.)^{T }representing the transposition operation, the vectors b_{i }being of size U for 1≦i≦n, with b_{i} ^{T}=(b_{i} ^{1}, b_{i} ^{2}, . . . , b_{i} ^{U});
 [0015]N is a random noise vector of size n×Q+W−1;
 [0016]A=(A_{1}, A_{2}, . . . , A_{n}) is a matrix of generalized codes of size (n×Q+W−1)×(n×U) which can be subdivided into n submatrices A_{i }of size (n×Q+W−1)×U. In the matrix A_{i }(1≦i≦n), the uth column (1≦u≦U) is a convolution of the impulse response of the uth channel and of the Q samples of the spreading code of the uth channel corresponding to the ith symbol of the block.
 [0017]Stated otherwise, the matrices A_{i }may be written:
 A _{i}=(Ω_{i} ^{1}, Ω_{i} ^{2}, . . . , Ω_{i} ^{U}) (4)
 [0018]with
 Ω_{i} ^{u} =M _{i} ^{u} .H _{i} ^{u} (5)
 [0019]where M_{i} ^{u }is a Toeplitz matrix of size (n×Q+W−1)×(n×Q+W−Q) obtained from the values c_{i} ^{u}(q) of the chips of the spreading code cu of the uth channel over the duration of the ith bit of the block:
$\begin{array}{cc}{M}_{i}^{u}=\left(\begin{array}{cccc}{c}_{i}^{u}\ue8a0\left(1\right)& 0& \cdots & 0\\ {c}_{i}^{u}\ue8a0\left(2\right)& {c}_{i}^{u}\ue8a0\left(1\right)& \u22f0& \vdots \\ \text{\hspace{1em}}& {c}_{i}^{u}\ue8a0\left(2\right)& \u22f0& 0\\ {c}_{i}^{u}\ue8a0\left(Q\right)& \text{\hspace{1em}}& \u22f0& {c}_{i}^{u}\ue8a0\left(1\right)\\ 0& {c}_{i}^{u}\ue8a0\left(Q\right)& \text{\hspace{1em}}& {c}_{i}^{u}\ue8a0\left(2\right)\\ \vdots & \u22f0& \u22f0& \vdots \\ 0& \cdots & 0& {c}_{i}^{u}\ue8a0\left(Q\right)\end{array}\right)& \left(6\right)\end{array}$  [0020]and H_{i} ^{u }is a column vector of size (n−1)×Q+W which, when the U channels are received synchronously, contains (i−1)×Q zeroes, followed by the W samples of the impulse response of the uth channel relating to the ith symbol b_{i} ^{u}, and followed by (n−i)×Q other zeroes. The time offsets in reception along the various channels, in numbers of chips, result in corresponding offsets of the W samples of the impulse response of the channels along the vector H_{i} ^{u}.
 [0021]The most commonly used receiver uses one or more matched filters to estimate the value of the symbols transmitted on each channel. This receiver estimates the impulse response of the channel along a propagation path or several propagation paths (“rake receiver”).
 [0022]The operation performed by such receivers amounts to performing the matrix product:
 Z=Â*.Y (7)
 [0023]where Â* is the conjugate transpose of an estimate Â=(Â_{1}, Â_{2}, . . . , Â_{n}) of the matrix A=(A_{1}, A_{2}, . . . , A_{n}), the matrices Â_{i }stemming from the impulse responses estimated by applying relations (4) and (5).
 [0024]The n×U components Z_{i} ^{u }of the vector Z are respective soft estimates of the n×U symbols b_{i} ^{u }of the vector b. If the decoding operations performed downstream admit soft estimates as input, the components of the vector Z can be used directly. Otherwise, the sign of these components is taken to form the hard estimates of the symbols.
 [0025]The matched filter receiver is optimal when the generalized codes (vectors Ω_{i} ^{u}) are pairwise orthogonal, i.e. when the matrix A*.A is diagonal. In general, the systems adopt pairwise orthogonal spreading codes having good autocorrelation properties, this making it possible to satisfy this condition to a first approximation.
 [0026]However, when the impulse response of the channel is taken into account, the orthogonality condition is no longer fulfilled. The above approximation becomes poor especially in the presence of multiple propagation paths.
 [0027]An object of the present invention is to compensate for the nonoptimal nature of the receiver with matched filter in these situations.
 [0028]The invention thus proposes a method of receiving a radio signal comprising contributions from a plurality of channels multiplexed by respective spreading codes, wherein a matched filter receiver is allocated to each multiplexed channel to estimate an impulse response of the channel and to provide first soft estimates of symbols transmitted on the channel. According to the invention, the first soft estimates provided by the matched filter receivers are processed to obtain a corrected soft estimate of at least one symbol transmitted on a channel. by subtracting from the first soft estimate of said symbol at least one term equal to the product of a decision value assigned to another symbol transmitted on another channel, determined from a soft estimate obtained previously for said other transmitted symbol, and a correlation between two generalized codes associated respectively with said symbol and with said other symbol, the generalized code associated with a symbol transmitted on a channel being a convolution of the estimated impulse response of said channel and Q samples of the spreading code of the channel which correspond to said symbol.
 [0029]The method corrects the soft estimates provided by the matched filter receiver by making allowance for the particular form of the interference caused in a channel by the presence of the other multiple access channels. This interference is here called MAI (“multiple access interference”). This allowance improves the performance of the receiver in terms of signaltonoise ratio.
 [0030]The symbols are typically transmitted on U multiplexed channels in the form of respective blocks of n symbols, n and U being numbers greater than 1. At the output of the matched filter receivers, the first soft estimate of the ith symbol of a block transmitted on the uth channel (1≦i≦n, 1≦u≦U) is given by the ((i−1)×U+u)th component of a vector Z=Â*.Y, where Y is a vector of size n×Q+W−1 composed of complex samples of a baseband signal obtained from the received radio signal, Q is the number of samples per symbol in the spreading codes, W is the number of samples in the estimates of the impulse responses, and Â* is the conjugate transpose of a matrix Â=(Â_{1}, Â_{2}, . . . , Â_{n}) subdivided into n submatrices Â_{i }of size (n×Q+W−1)×U with 1≦i≦n, the uth column of the matrix Â_{i }for 1≦u≦U being a convolution of the estimated impulse response of the uth channel and of the Q samples of the spreading code of the uth channel corresponding to the ith symbol of the block. The corrected soft estimate of at least one ith symbol of a block transmitted on a uth channel (1≦i≦n, 1≦u≦U) may then be obtained by subtracting from the first soft estimate of said symbol at least one term given by {tilde over (R)}_{i,0} ^{u}.{circumflex over (b)}_{i}, where {tilde over (R)}_{i,0} ^{u }is the uth row of a matrix of size U×U whose diagonal components are zero and whose other components are respectively from a correlation matrix {circumflex over (R)}_{i,0}=Â_{i}*.Â_{i}, and {circumflex over (b)}_{i }is a column vector of size U composed of decision values assigned to the ith symbols of the U blocks, determined respectively from soft estimates obtained previously for said ith symbols.
 [0031]The term {tilde over (R)}_{i,0} ^{u}.{circumflex over (b)}_{i }represents, to a first approximation, an estimate of the interference caused in the uth channel by the presence of the U−1 other channels. This approximation may suffice in cases where the intersymbol interference (ISI) is weak.
 [0032]The row vector {tilde over (R)}_{i,0} ^{u }is neglected in traditional receivers (which assume that A*.A is diagonal), so that the MAI is regarded as included in the additive noise (vector N of relation (3)). This degrades the performance of the receiver in terms of signaltonoise ratio.
 [0033]The above method exploits knowledge which may be obtained regarding the structure of the contributions from the MAI in the signal received, so as to avoid or at least attenuate this degradation.
 [0034]In an advantageous implementation of the method, said corrected soft estimate of the ith symbol of the block transmitted on the uth channel is obtained by further subtracting from the first soft estimate of said symbol at least one term of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j}, where j is a nonzero integer, and {circumflex over (R)}_{i,j} ^{u }is the uth row of a matrix of size U×U whose components are respectively from a correlation matrix {circumflex over (R)}_{i,j}=Â_{i}*.Â_{i+j}. Preferably, each of the terms of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j }for −m≦j≦−1 and 1≦j≦m is subtracted from the first soft estimate of said symbol to obtain its corrected soft estimate, m being the integer equal to or immediately greater than the number (Q+W−1)/Q.
 [0035]The estimate of the contribution from the MAI is thus sharpened and the intersymbol interference (ISI), whose integer m represents the degree, is taken into account in the same way.
 [0036]Typically, the corrected soft estimates will be determined sequentially for a plurality of symbols of the U blocks. To form said soft estimates obtained previously for certain of the symbols, serving to determine the decision values, one then advantageously takes the corrected soft estimates of said symbols if the former have previously been determined, and the first soft estimates of said symbols otherwise. The corrections already performed on the soft estimates are thus taken into account recursively, this making it possible to further improve accuracy.
 [0037]Another aspect of the present invention pertains to a device for receiving a radio signal comprising contributions from a plurality of channels multiplexed by respective spreading codes, comprising matched filter receivers each allocated to a respective multiplexed channel to estimate an impulse response of the channel and to provide first soft estimates of symbols transmitted on the channel, and means for processing these first soft estimates provided by the matched filter receivers to obtain a corrected soft estimate of at least one symbol transmitted on a channel in the manner indicated above.
 [0038]Such a device can in particular be incorporated into a base station of a CDMA radio communication system.
 [0039]Other features and advantages of the present invention will become apparent in the description hereinbelow of nonlimiting exemplary implementations, with reference to the appended drawings, in which:
 [0040][0040]FIG. 1 is a schematic diagram of a receiving device according to the invention; and
 [0041][0041]FIG. 2 is a diagram of a matched filter receiver of the device.
 [0042]The device represented in FIG. 1 forms part of the receiving stage of a radio communication station able to communicate with several remote stations 1.
 [0043]The uplink channels used by these distant stations 1 are multiplexed by the CDMA technique, so that the radio signal picked up by the antenna 2, and converted to baseband, can be represented in the form (1)(2) for U multiplexed channels originating from V stations (1≦V≦U).
 [0044]The station incorporating the device is for example a base station of a thirdgeneration cellular radio communication system of the UMTS (“Universal Mobile Telecommunication System”) type.
 [0045]In FIG. 1, the unit 3 diagrammatically represents the modules performing in a conventional manner the signal reception preprocessing (amplification, filterings, conversion to baseband, chip rate sampling). This unit 3 delivers blocks Y of n×Q+W−1 samples, corresponding to blocks of n symbols transmitted simultaneously on the U channels. If the blocks of n symbols follow one another without interruption on the channels, there is an overlap of W samples (chips) between the successive blocks Y, corresponding to the duration of the impulse response.
 [0046]The receivedsignal blocks Y are provided in parallel to U matched filter receivers 4 ^{u }operating with respective channel codes c^{u }produced by pseudorandom code generators 5 ^{u }(1≦u≦U).
 [0047][0047]FIG. 2 illustrates the wellknown structure of a matched filter receiver 4 ^{u }of “rake” type.
 [0048]This receiver 4 ^{u }comprises a channel probing unit 6 which evaluates the impulse response of the uth channel by searching for K propagation paths (K≧1), for the K “fingers” of the receiver. Each path K is characterized by a delay t_{k} ^{u }expressed as a number of chips and a complex response r_{k} ^{u }(1≦k≦K). By way of example, the signal transmitted on each channel by a remote station may comprise sequences of known training symbols. By searching, over a window of length W chips, for the K correlations of largest amplitude between the received signal Y and these known sequences modulated by the channel spreading code c^{u}, the unit 6 obtains the delays t_{k} ^{u }(time offsets of the maxima) and the responses r_{k} ^{u }(values of the maxima).
 [0049]In each finger k of the receiver 4 ^{u}, the spreading code cu produced by the generator 5 ^{u }(or its conjugate if the codes are complex) is delayed by a unit 7 which applies the delay of t_{k} ^{u }chips thereto. Each code thus delayed is multiplied by the received signal Y (multiplier 8) and by the conjugate of the complex response r_{k} ^{u }(multiplier 9). The K results of these multiplications are summed by an adder 10 to form the block Z^{u }of n soft estimates for the uth channel. At the output of the matched filter receiver 4 ^{u}, the ith component of the block Z^{u }is the soft estimate of the symbol b_{i} ^{u}. If the symbols b_{i} ^{u }are signed bits (±1), the soft estimates of the block Z^{u }are the real parts of the summed contributions of the K fingers. If the symbols b_{i} ^{u }are pairs of signed bits, they are complex numbers equal to these summed contributions.
 [0050]As shown in FIG. 1, the soft estimates Z^{u }can be transformed into hard estimates {circumflex over (b)}^{u }by decision modules 12 ^{u }at the output of the receivers 4 ^{u}. When the symbols b_{i} ^{u }are signed bits, the modules 12 ^{u }simply apply the sign function to the real components of the vectors Z^{u}. When these are pairs of signed bits, the modules 12 ^{u }apply the sign function to the real parts and to the imaginary parts of the components of the vectors Z^{u}.
 [0051]If the set of components of the U vectors Z^{u }is ordered by grouping together the estimates of the symbols of like rank i, then the vector Z defined above of size n×U is obtained, in which the soft estimate of the symbol b_{i} ^{u }is the ((i−1)×U+u)th component. The column vector Z can also be decomposed into n vectors Z_{i }of size U, according to Z^{T}=(Z_{1} ^{T}, Z_{2} ^{T}, . . . , Z_{n} ^{T}). A similar ordering of the components of the U vectors {circumflex over (b)}^{u }gives the column vector of hard estimates {circumflex over (b)} of size n×U, which can be decomposed into {circumflex over (b)}^{T}=({circumflex over (b)}_{1} ^{T}, {circumflex over (b)}_{2} ^{T}, . . . , {circumflex over (b)}_{n} ^{T}), the vectors {circumflex over (b)}_{i }being of size U for 1≦i≦n and representing respective estimates of the vectors b_{i}.
 [0052]The matrix product (7) performed by the matched filter receivers can also be written (cf. (3)):
 Z=Â*.A.b+A*.N (8)
 [0053]If we assume that the impulse responses have been estimated correctly, then the matrices A and Â may be regarded as equal, and we see that the relation between the vector b of the symbols and the vector Z of the soft estimates of these symbols is characterized by the matrix A*.A, which can be expanded as:
$\begin{array}{cc}{A}^{*}\xb7A=\left(\begin{array}{ccccccc}{R}_{1,0}& {R}_{1,+1}& \cdots & {R}_{1,+m}& 0& \cdots & 0\\ {R}_{2,1}& {R}_{2,0}& \text{\hspace{1em}}& \text{\hspace{1em}}& \u22f0& \u22f0& \vdots \\ \vdots & \text{\hspace{1em}}& \u22f0& \text{\hspace{1em}}& \text{\hspace{1em}}& \u22f0& 0\\ {R}_{m+1,m}& \text{\hspace{1em}}& \text{\hspace{1em}}& \u22f0& \u22f0& \text{\hspace{1em}}& {R}_{nm,+m}\\ 0& \u22f0& \text{\hspace{1em}}& \u22f0& \u22f0& \text{\hspace{1em}}& \vdots \\ \vdots & \u22f0& \u22f0& \text{\hspace{1em}}& \text{\hspace{1em}}& {R}_{n1,0}& {R}_{n1,+1}\\ 0& \cdots & 0& {R}_{n,m}& \cdots & {R}_{n,1}& {R}_{n,0}\end{array}\right)& \left(9\right)\end{array}$  [0054]with, for 1≦i≦n and −m≦j≦+m:
 R _{i,j} =R _{i+j,−i} *=A _{i} *.A _{i+j} (10)
 [0055]Each matrix R_{i,j }of size U×U contains the correlations of the generalized codes between the ith symbols and the (i+j)th symbols of the blocks relating to the U channels.
 [0056]
 [0057]where Z_{i }is a vector of size U containing the soft estimates of the ith symbols of the U blocks and N_{i }a corresponding noise vector.
 [0058]In the particular case where all the channels are synchronized and where the ISI is negligible (m=0), a simplification arises:
 Z _{i} =R _{i,0} .b _{i} +N _{i} (12)
 [0059]The method according to the invention comprises a postprocessing of the soft estimates of the vectors Z_{i}, which is carried out in the module 13 represented in FIG. 1. The algorithm used is called MFPIC (“Matched Filter Parallel Interference Cancellation”).
 [0060]The first step of this algorithm consists in obtaining, for 1≦i≦n and −m≦j≦+m, the components of the matrices {circumflex over (R)}_{i,j}=Â_{i}*.Â_{i+j }which are the estimates of the correlation matrices R_{i,j }based on the impulse responses estimated by the probing units 6. Relations (10) show that the required quantity of calculations can be reduced by virtue of the symmetry properties of the matrices R_{i,j}.
 [0061]Each component {circumflex over (R)}_{i,j} ^{u,v}={circumflex over (Ω)}_{i} ^{u}*.{circumflex over (Ω)}_{i,j} ^{v }of a matrix {circumflex over (R)}_{i,j }is the scalar product of two vectors {circumflex over (Ω)}_{i} ^{u }and {circumflex over (Ω)}_{i+j} ^{v }corresponding to the columns u and v of the matrices Â_{i }and Â_{i+j}. Each vector {circumflex over (Ω)}_{i} ^{u }contains the convolution of the estimated impulse response of channel u and of the Q samples of the spreading code of this channel corresponding to the ith symbol of the block, and is defined as in relation (5), the matrix M_{i} ^{u }being determined according to (6) in accordance with the code cU provided by the generator 5 ^{u}, and the vector H_{i} ^{u }being replaced by an estimated response vector Ĥ_{i} ^{u }containing the complex responses r_{k} ^{u }estimated by the probing units 6, positioned according to the corresponding delays t^{u} _{k}.
 [0062]According to one implementation of the invention, the module 13 executes the following operation (13) sequentially for i going from 1 to n:
$\begin{array}{cc}{X}_{i}={Z}_{i}{\stackrel{~}{R}}_{i,0}\xb7{\hat{b}}_{i}\sum _{j=1}^{m}\ue89e\text{\hspace{1em}}\ue89e\left({\hat{R}}_{i,j}\xb7{\hat{b}}_{ij}+{\hat{R}}_{i,j}\xb7{\hat{b}}_{i+j}\right)& \left(13\right)\end{array}$  [0063]
 [0064]where the matrix {tilde over (R)}_{i,0 }is equal to the correlation matrix {circumflex over (R)}_{i,0 }in which the diagonal components are set to zero. In general the block of n symbols is preceded and followed by other symbols, whose hard estimates are placed in the vectors {circumflex over (b)}_{i,j }for i−j≦1 and {circumflex over (b)}_{i+j }for i+j>n. Otherwise, these vectors may be set to zero.
 [0065]The vectors X_{i }thus obtained are corrected soft estimates taking account of relations (11). This correction exploits the decisions taken in the hard estimates {circumflex over (b)}_{i}, and hence a certain structure of the MAI and of the ISI, which is not the same as that of the Gaussian noise N.
 [0066]Additional hard estimates {circumflex over ({circumflex over (b)})}_{i }may in their turn be deduced from the soft estimates X_{i}, typically by applying the sign function to the real components (where the symbols are bits) or to each of the real and imaginary parts of the complex components (where the symbols are pairs of bits): {circumflex over ({circumflex over (b)})}=sign[X_{i}].
 [0067]At the output of the postprocessing module 13, the soft estimates of the vectors X_{i }and/or the hard estimates of the vectors {circumflex over ({circumflex over (b)})}_{i }(1≦i≦n) are redistributed among the U channels, as indicated by the vectors X^{u }and {circumflex over ({circumflex over (b)})}^{u }in FIG. 1 (1≦u≦U), so as to provide the estimates useful to the decoders in the downstream channel processing pathways.
 [0068]In a preferred variant of the invention, the module 13 executes the following operations (13) and (14) sequentially for i going from 1 to n:
$\begin{array}{cc}{X}_{i}={Z}_{i}{\stackrel{~}{R}}_{i,0}\xb7{\hat{b}}_{i}\sum _{j=1}^{m}\ue89e\text{\hspace{1em}}\ue89e\left({\hat{R}}_{i,j}\xb7{\hat{b}}_{ij}+{\hat{R}}_{i,j}\xb7{\hat{b}}_{i+j}\right)& \left(13\right)\end{array}$  {circumflex over (b)} _{i}=sign[X_{i}] (14)
 [0069]In this case, the corrected estimates X_{i }which have already been calculated are taken into account recursively in the decision taken in the operation (14), thereby further improving the estimates.
 [0070]It should be noted that the operations (13) and (14) above could be executed in an order other than that of increasing indices i. For example, they could be executed in an order determined by an energy criterion. One possibility is first to correct the estimates of the symbols of least energy in the signal received, that is to say to proceed in the order of the indices i for which the diagonal terms of the correlation matrix {circumflex over (R)}_{i,0 }are decreasing.
 [0071]In other alternative embodiments:
 [0072]the function applied (in the decision modules 12 ^{u}) to deduce from the soft estimates Z_{i }the estimates {circumflex over (b)}_{i}used in formula (13) is, rather than the sign function, a function generally increasing between −1 and +1. For example, a function with three values (−1 for Z_{i} ^{u}<−T, 0 for −T≦Z_{i} ^{u}≦+T and +1 for Z_{i} ^{u}>+T) makes it possible to do away with making corrections based on estimates with low likelihood relative to a threshold T. The function can also increase continuously from −1 to +1. The advantage of the sign function is mainly in terms of complexity since it avoids the multiplications in formula (13);
 [0073]likewise, a function increasing between −1 and +1 can generally be used in the operation (14) to obtain the estimate {circumflex over (b)}_{i }used in the following iterations of operation (13);
 [0074]the MFPIC algorithm is applied to only some of the channels received, this amounting to taking, in the expression for the algorithm, a value of U (>1) which is smaller than the number of rake receivers. Here again, energy criteria can govern which channels are subjected to the correction of MAI;
 [0075]the MFPIC algorithm is applied in a station receiving in diversity mode with the aid of d distinct antennas (or antenna sectors). It is then sufficient to replace A*.A by
$\underset{p=1\ue89e\text{\hspace{1em}}}{\overset{d\ue89e\text{\hspace{1em}}}{\sum \text{\hspace{1em}}}}\ue89e\text{\hspace{1em}}\ue89e{A}^{*}\ue8a0\left(p\right)\xb7A\ue8a0\left(p\right)\ue89e\text{\hspace{1em}}\ue89e\mathrm{and}\ue89e\text{\hspace{1em}}\ue89e{A}^{*}\xb7Y\ue89e\text{\hspace{1em}}\ue89e\mathrm{by}\ue89e\text{\hspace{1em}}\ue89e\sum _{p=1}^{d}\ue89e\text{\hspace{1em}}\ue89e{A}^{*}\ue8a0\left(p\right)\xb7Y\ue8a0\left(p\right)$  [0076]in the expression for the algorithm, the index p referring to the various antennas.
 [0077]The MFPIC algorithm offers multiuser detection having good performance, especially in respect of the relatively low spreading factors Q. As soon as the binary error rate is less than 15%, it affords a significant gain in terms of signaltonoise ratio, as compared with the simple rake receiver. Its limitations seem to come only from the uncertainties in the estimates of the impulse responses of the channels.
Claims (16)
 1. A method of receiving a radio signal comprising contributions from a plurality of channels multiplexed by respective spreading codes, wherein a matched filter receiver (4 ^{1}, 4 ^{2}, . . . , 4 ^{U}) is allocated to each multiplexed channel to estimate an impulse response of the channel and to provide first soft estimates of symbols transmitted on the channel, and wherein the first soft estimates provided by the matched filter receivers are processed to obtain a corrected soft estimate of at least one symbol transmitted on a channel by subtracting from the first soft estimate of said symbol at least one term equal to the product of a decision value assigned to another symbol transmitted on another channel, determined from a soft estimate obtained previously for said other transmitted symbol, and a correlation between two generalized codes associated respectively with said symbol and with said other symbol, the generalized code associated with a symbol transmitted on a channel being a convolution of the estimated impulse response of said channel and Q samples of the spreading code of the channel which correspond to said symbol.
 2. The method as claimed in
claim 1 , wherein the symbols are transmitted on U multiplexed channels in the form of respective blocks of n symbols, n and U being numbers greater than 1, wherein, at the output of the matched filter receivers, the first soft estimate of the ith symbol of a block transmitted on the uth channel (1≦i≦n, 1≦u≦U) is given by the ((i−1)×U+u)th component of a vector Z=Â*.Y, where Y is a vector of size n×Q+W−1 composed of complex samples of a baseband signal obtained from the received radio signal, Q is the number of samples per symbol in the spreading codes, W is the number of samples in the estimates of the impulse responses, and Â* is the conjugate transpose of a matrix Â=(Â_{1}, Â_{2}, . . . , Â_{n}) subdivided into n submatrices Â_{i }of size (n×Q+W−1)×U with 1≦i≦n, the uth column of the matrix Â_{i }for 1≦u≦U being a convolution of the estimated impulse response of the uth channel and of the Q samples of the spreading code of the uth channel corresponding to the ith symbol of the block, and wherein the corrected soft estimate of at least one ith symbol of a block transmitted on a uth channel (1≦i≦n, 1≦u≦U) is obtained by subtracting from the first soft estimate of said symbol at least one term given by {tilde over (R)}_{i,0} ^{u}.{circumflex over (b)}_{i}, where {tilde over (R)}_{i,0} ^{u }is the uth row of a matrix of size U×U whose diagonal components are zero and whose other components are respectively from a correlation matrix {circumflex over (R)}_{i,0}=Â_{i}*.Â_{i}, and {circumflex over (b)}_{i }is a column vector of size U composed of decision values assigned to the ith symbols of the U blocks, determined respectively from soft estimates obtained previously for said ith symbols.  3. The method as claimed in
claim 2 , wherein said corrected soft estimate of the ith symbol of the block transmitted on the uth channel is obtained by further subtracting from the first soft estimate of said symbol at least one term of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j}, where j is a nonzero integer, and {circumflex over (R)}_{i,j} ^{u }is the uth row of a matrix of size U×U whose components are respectively from a correlation matrix {circumflex over (R)}_{i,j}=Â_{i}*.Â_{i+j}.  4. The method as claimed in
claim 3 , wherein said corrected soft estimate of the ith symbol of the block transmitted on the uth channel is obtained by subtracting from the first soft estimate of said symbol each of the terms of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j }for −m≦j≦−1 and 1≦j≦m, where m is the integer equal to or immediately greater than the number (Q+W−1)/Q.  5. The method as claimed in any one of the preceding claims, wherein corrected soft estimates are obtained sequentially for a plurality symbols of the U blocks, and wherein each decision value assigned to a symbol transmitted on a channel is determined either from the first soft estimate of said symbol, if a corrected soft estimate of said symbol has not been obtained, or from the corrected soft estimate of said symbol.
 6. The method as claimed in any one of the preceding claims, wherein the symbols are bits, the soft estimates are real numbers, and the decision values are determined from soft estimates obtained previously by applying a function increasing between −1 and +1 to said soft estimates.
 7. The method as claimed in any one of
claims 1 to5 , wherein the symbols are bit pairs, the soft estimates are complex numbers, and the decision values are determined from the soft estimates obtained previously by applying a function increasing between −1 and +1 to each of the real and imaginary parts of said soft estimates.  8. The method as claimed in
claim 6 or 7, wherein said function increasing between −1 and +1 is the sign function.  9. A device for receiving a radio signal comprising contributions from a plurality of channels multiplexed by respective spreading codes, comprising matched filter receivers (4 ^{1}, 4 ^{2}, . . . , 4 ^{U}) each allocated to a respective multiplexed channel to estimate an impulse response of the channel and to provide first soft estimates of symbols transmitted on the channel, and means (13) for processing the first soft estimates provided by the matched filter receivers to obtain a corrected soft estimate of at least one symbol transmitted on a channel by subtracting from the first soft estimate of said symbol at least one term equal to the product of a decision value assigned to another symbol transmitted on another channel, determined from a soft estimate obtained previously for said other transmitted symbol, and a correlation between two generalized codes associated respectively with said symbol and with said other symbol, the generalized code associated with a symbol transmitted on a channel being a convolution of the estimated impulse response of said channel and Q samples of the spreading code of the channel which correspond to said symbol.
 10. The receiving device as claimed in
claim 9 , comprising at least U matched filter receivers (4 ^{1}, 4 ^{2}, . . . , 4 ^{U}) each assigned to a respective multiplexed channel to estimate an impulse response of the channel and to obtain first soft estimates of a block of n symbols transmitted on the channel, n and U being numbers greater than 1, wherein, at the output of the matched filter receivers, the first soft estimate of the ith symbol of a block transmitted on the uth channel (1≦i≦n, 1≦u≦U) is given by the ((i−1)×U+u)th component of a vector Z=Â*.Y, where Y is a vector of size n×Q+W−1 composed of complex samples of a baseband signal obtained from the received radio signal, Q is the number of samples per symbol in the spreading codes, W is the number of samples in the estimates of the impulse responses, and Â* is the conjugate transpose of a matrix Â=(Â_{1}, Â_{2}, . . . , Â_{n}) subdivided into n submatrices Â_{i }of size (n×Q+W−1)×U with 1≦i≦n, the uth column of the matrix Â_{i }for 1≦u≦U being a convolution of the estimated impulse response of the uth channel and of the Q samples of the spreading code of the uth channel corresponding to the ith symbol of the block, and wherein the corrected soft estimate of at least one ith symbol of a block transmitted on a uth channel (1≦i≦n, 1≦u≦U) is obtained by subtracting from the first soft estimate of said symbol at least one term given by {tilde over (R)}_{i,0} ^{u}.{circumflex over (b)}_{i}, where {tilde over (R)}_{i,0} ^{u }is the uth row of a matrix of size U×U whose diagonal components are zero and whose other components are respectively from a correlation matrix {circumflex over (R)}_{i,0}=Â_{i}*.Â_{i}, and {circumflex over (b)}_{i }is a column vector of size U composed of decision values assigned to the ith symbols of the U blocks, determined respectively from soft estimates obtained previously for said ith symbols.  11. The receiving device as claimed in
claim 10 , wherein the processing means (13) are arranged to obtain said corrected soft estimate of the ith symbol of the block transmitted on the uth channel by further subtracting from the first soft estimate of said symbol at least one term of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j}, where j is a nonzero integer, and {circumflex over (R)}_{i,j} ^{u }is the uth row of a matrix of size U×U whose components are respectively from a correlation matrix {circumflex over (R)}_{i,j}=Â_{i}*.Â_{i+j}.  12. The receiving device as claimed in
claim 11 , wherein the processing means (13) are arranged to obtain said corrected soft estimate of the ith symbol of the block transmitted on the uth channel by subtracting from the first soft estimate of said symbol each of the terms of the form {circumflex over (R)}_{i,j} ^{u}.{circumflex over (b)}_{i+j }for −m≦j≦−1 and 1≦j≦m, where m is the integer equal to or immediately greater than the number (Q+W−1)/Q.  13. The receiving device as claimed in any one of
claims 8 to10 , wherein the processing means (13) are arranged to obtain corrected soft estimates sequentially for a plurality of symbols of the U blocks, and wherein each decision value assigned to a symbol transmitted on a channel is determined either from the first soft estimate of said symbol, if a corrected soft estimate of said symbol has not been obtained, or from the corrected soft estimate of said symbol.  14. The receiving device as claimed in any one of
claims 9 to13 , wherein the symbols are bits, the soft estimates are real numbers, and the decision values are determined from soft estimates obtained previously by applying a function increasing between −1 and +1 to said soft estimates.  15. The receiving device as claimed in any one of
claims 9 to13 , wherein the symbols are bit pairs, the soft estimates are complex numbers, and the decision values are determined from the soft estimates obtained previously by applying a function increasing between −1 and +1 to each of the real and imaginary parts of said soft estimates.  16. The receiving device as claimed in
claim 14 or 15, wherein said function increasing between −1 and +1 is the sign function.
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Cited By (18)
Publication number  Priority date  Publication date  Assignee  Title 

US20050053121A1 (en) *  20011206  20050310  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20050053165A1 (en) *  20011206  20050310  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20050058180A1 (en) *  20011206  20050317  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20050069020A1 (en) *  20011206  20050331  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20050068932A1 (en) *  20000925  20050331  Ismail Lakkis  Ultrawideband communication system and methods 
US20050152483A1 (en) *  20011206  20050714  Ismail Lakkis  Systems and methods for implementing path diversity in a wireless communication network 
US20050233710A1 (en) *  20011206  20051020  Ismail Lakkis  High data rate transmitter and receiver 
US20060274817A1 (en) *  20000925  20061207  Lakkis Ismail A  Method and apparatus for wireless communications 
US20070286317A1 (en) *  20040908  20071213  Stentiford Frederick W  High Data Rate Demodulation System 
US20080008234A1 (en) *  20011206  20080110  Ismail Lakkis  Systems and methods for equalization of received signals in a wireless communication network 
US20080043654A1 (en) *  20011206  20080221  Lakkis Ismail A  Systems and methods for wireless communication over a wide bandwidth channel using a plurality of subchannels 
US20080056186A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication systems and methods 
US20080056333A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20080056332A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication systems and methods 
US20080109696A1 (en) *  20011206  20080508  Ismail Lakkis  Systems and methods for forward error correction in a wireless communication network 
US20080107199A1 (en) *  20011206  20080508  Ismail Lakkis  Systems and methods for recovering bandwidth in a wireless communication network 
US20090003492A1 (en) *  20040908  20090101  Michael Robert Fitch  High Data Rate Demodulation System 
US20120328060A1 (en) *  20100728  20121227  Yoonoh Yang  Method and apparatus for channel estimation in multipath channel 
Families Citing this family (1)
Publication number  Priority date  Publication date  Assignee  Title 

FR2850501B1 (en) *  20030129  20050408  Nortel Networks Ltd  Method and device for receiving a radio signal 
Citations (6)
Publication number  Priority date  Publication date  Assignee  Title 

US20020067761A1 (en) *  20001201  20020606  Ning Kong  Method and system for canceling multiple access interference in CDMA wireless communication system 
US6498804B1 (en) *  19980130  20021224  Matsushita Electric Industrial Co., Ltd.  Method of directional reception using array antenna, and adaptive array antenna unit 
US20030043893A1 (en) *  20010606  20030306  Alexandre Jard  Signal processing method and apparatus for a spread spectrum radio communication receiver 
US6570864B1 (en) *  19981116  20030527  Electronics And Telecommunications Research Institute  Integrated receiving apparatus of subtractive interference cancellation receiver and adaptive MMSE receiver 
US20040223537A1 (en) *  20030129  20041111  Nortel Networks Limited.  Method and device for receiving a radio signal 
US6931050B1 (en) *  19981203  20050816  Ericsson Inc.  Digital receivers and receiving methods that scale for relative strengths of traffic and pilot channels during soft handoff 
Family Cites Families (2)
Publication number  Priority date  Publication date  Assignee  Title 

JP3335900B2 (en) *  19980227  20021021  松下電器産業株式会社  Interference cancellation apparatus and an interference removal method 
EP0964530A1 (en) *  19980605  19991215  Siemens Aktiengesellschaft  Radio communications receiver and interference cancellation method 
Patent Citations (7)
Publication number  Priority date  Publication date  Assignee  Title 

US6498804B1 (en) *  19980130  20021224  Matsushita Electric Industrial Co., Ltd.  Method of directional reception using array antenna, and adaptive array antenna unit 
US6570864B1 (en) *  19981116  20030527  Electronics And Telecommunications Research Institute  Integrated receiving apparatus of subtractive interference cancellation receiver and adaptive MMSE receiver 
US6931050B1 (en) *  19981203  20050816  Ericsson Inc.  Digital receivers and receiving methods that scale for relative strengths of traffic and pilot channels during soft handoff 
US6882678B2 (en) *  20001201  20050419  Ning Kong  Method and system for canceling multiple access interference in CDMA wireless communication system 
US20020067761A1 (en) *  20001201  20020606  Ning Kong  Method and system for canceling multiple access interference in CDMA wireless communication system 
US20030043893A1 (en) *  20010606  20030306  Alexandre Jard  Signal processing method and apparatus for a spread spectrum radio communication receiver 
US20040223537A1 (en) *  20030129  20041111  Nortel Networks Limited.  Method and device for receiving a radio signal 
Cited By (30)
Publication number  Priority date  Publication date  Assignee  Title 

US20050068932A1 (en) *  20000925  20050331  Ismail Lakkis  Ultrawideband communication system and methods 
US20080225963A1 (en) *  20000925  20080918  Ismail Lakkis  Ultrawideband communication systems and methods 
US7339955B2 (en) *  20000925  20080304  PulseLink, Inc.  TDMA communication method and apparatus using cyclic spreading codes 
US20060274817A1 (en) *  20000925  20061207  Lakkis Ismail A  Method and apparatus for wireless communications 
US20050053121A1 (en) *  20011206  20050310  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20050152483A1 (en) *  20011206  20050714  Ismail Lakkis  Systems and methods for implementing path diversity in a wireless communication network 
US20050233710A1 (en) *  20011206  20051020  Ismail Lakkis  High data rate transmitter and receiver 
US20050069020A1 (en) *  20011206  20050331  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US8744389B2 (en)  20011206  20140603  Intellectual Ventures Holding 73 Llc  High data rate transmitter and receiver 
US20080008234A1 (en) *  20011206  20080110  Ismail Lakkis  Systems and methods for equalization of received signals in a wireless communication network 
US20080043654A1 (en) *  20011206  20080221  Lakkis Ismail A  Systems and methods for wireless communication over a wide bandwidth channel using a plurality of subchannels 
US20080043653A1 (en) *  20011206  20080221  Lakkis Ismail A  Systems and methods for wireless communication over a wide bandwidth channel using a plurality of subchannels 
US20080049652A1 (en) *  20011206  20080228  Lakkis Ismail A  Systems and methods for wireless communication over a wide bandwidth channel using a plurality of subchannels 
US20080049827A1 (en) *  20011206  20080228  Ismail Lakkis  Systems and methods for implementing path diversity in a wireless communication network 
US20050058180A1 (en) *  20011206  20050317  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US7929596B2 (en)  20011206  20110419  PulseLink, Inc.  Ultrawideband communication apparatus and methods 
US20080056333A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20080056332A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication systems and methods 
US20080069256A1 (en) *  20011206  20080320  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US20080109696A1 (en) *  20011206  20080508  Ismail Lakkis  Systems and methods for forward error correction in a wireless communication network 
US20080107199A1 (en) *  20011206  20080508  Ismail Lakkis  Systems and methods for recovering bandwidth in a wireless communication network 
US20050053165A1 (en) *  20011206  20050310  Ismail Lakkis  Ultrawideband communication apparatus and methods 
US8532586B2 (en)  20011206  20130910  Intellectual Ventures Holding 73 Llc  High data rate transmitter and receiver 
US20080056186A1 (en) *  20011206  20080306  Ismail Lakkis  Ultrawideband communication systems and methods 
US8045935B2 (en)  20011206  20111025  PulseLink, Inc.  High data rate transmitter and receiver 
US8085881B2 (en) *  20040908  20111227  British Telecommunications Public Limited Company  High data rate demodulation system 
US20090003492A1 (en) *  20040908  20090101  Michael Robert Fitch  High Data Rate Demodulation System 
US20070286317A1 (en) *  20040908  20071213  Stentiford Frederick W  High Data Rate Demodulation System 
US20120328060A1 (en) *  20100728  20121227  Yoonoh Yang  Method and apparatus for channel estimation in multipath channel 
US8934580B2 (en) *  20100728  20150113  Lg Electronics Inc.  Method and apparatus for channel estimation in multipath channel 
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