TW201414353A - Driving circuits, methods and controllers for driving light source - Google Patents

Abstract

A driving circuit includes a converter, a transformer, a first sensor and a second sensor. The converter is coupled to a switch that operates in a first state or in a second state. The converter receives an input voltage and provides a regulated voltage. The transformer transforms the regulated voltage to an output voltage to power a light source. Both a first current through the converter and a second current through the transformer further flow through the switch when the switch operates in the first state. The first sensor coupled between the switch and a first reference node provides a first sense signal indicating a combined current of the first current and the second current. The second sensor coupled between the first reference node and a second reference node provides a second sense signal indicating only the second current.

Description

Light source driving circuit, method and controller

The invention relates to a power supply circuit, in particular to a power supply circuit, a power converter and a power supply method for a light-emitting diode light source.

FIG. 1 is a schematic diagram of a conventional light source driving circuit 100. The light source driving circuit 100 is for driving a light source (for example, the light emitting diode string 108). The light source driving circuit 100 supplies an input voltage VIN from a power source 102 to power the driving circuit 100. The light source driving circuit 100 includes a buck converter that provides an adjusted voltage VOUT to the LED string 108 under the control of a controller 104. The buck converter includes a diode 114, an inductor 112, a capacitor 116, and a switch 106. A resistor 110 is coupled in series with the switch 106. When the switch 106 is turned on, the resistor 110 is coupled to the inductor 112 and the LED string 108, and generates a feedback signal to indicate the current flowing through the inductor 112. When the switch 106 is turned off, the resistor 110 is disconnected from the inductor 112 and the LED string 108, so no current flows through the resistor 110.

Switch 106 is controlled by controller 104. When the switch 106 is turned on, a current flows through the LED string 108, the inductor 112, the switch 106, and the resistor 110 to ground. This current gradually increases under the action of the inductor 112. When the current increases to a predetermined peak current level, the controller 104 turns off the switch 106. When the switch 106 is turned off, a current flows through the LED string 108, the inductor 112, and the diode 114. The controller 104 can turn the switch 106 on again after a period of time. Therefore, the controller 104 controls the buck converter based on the preset peak current level. However, the average current level flowing through the inductor 112 and the LED string 108 varies with the inductance of the inductor 112, the input voltage VIN, and the voltage VOUT across the LED string 108, thus flowing through the inductor 112. The average current level (i.e., the average current flowing through the LED string 108) cannot be accurately controlled.

In order to solve the above technical problem, the present invention provides a driving circuit comprising: a converter that receives an input voltage and provides a regulated voltage; a switch coupled to the converter, the switch operating in a first state alternately And a second state; a transformer coupled to the converter and the switch, converting the regulated voltage to an output voltage to supply power to a load, wherein when the switch operates in the first state, flowing through the a first current of the converter and a second current flowing through the transformer flow through the switch; a first inductor coupled between the switch and a first reference node, providing the first current and the a first sensing signal of a combined current of the second current; and a second inductor coupled between the first reference node and a second reference node to provide a second sensing indicating only the second current signal.

The present invention also provides a control load power controller, comprising: an output pin, generating a drive signal to cause a switch to alternately operate in a first state and a second state; a converter coupled to the switch, An input voltage is converted into a regulated voltage; a transformer coupled to the switch, the regulated voltage is converted into an output voltage to supply power to a load, and when the switch operates in the first state, a flow through the converter a first current and a second current flowing through the transformer flow through the switch; a protection pin coupled to a protection circuit, the protection circuit transmitting a total of a first sensor and a second sensor A first inductor is coupled between the switch and a first reference node, and the second inductor is coupled to the first reference node and the first inductor And a sensing pin coupled to the first reference node to sense the second current by monitoring a voltage on the second inductor, wherein the controller receives the second current according to the sensing pin A signal and a reception signal of the protection pin of the driving control signal.

The invention also provides a load power control method, comprising: converting a input voltage into a regulated voltage by using a converter; converting the regulated voltage into an output voltage by using a transformer to supply power to a load; The switch is alternately operated in a first state and a second state, and when the switch operates in the first state, a first current flowing through the converter and a second current flowing through the transformer flow through The first sensor is coupled to receive a first sensing signal indicating a combined current of the first current and the second current by monitoring a total voltage on a first inductor and a second inductor, the first inductor being coupled On the switch and a first reference section Between the points, the second sensor is coupled between the first reference node and a second reference node; and receiving a second sensing signal indicating the second current by monitoring a voltage on the second sensor And controlling the driving signal according to the first sensing signal and the second sensing signal to adjust a current flowing through the load.

100‧‧‧Light source drive circuit

102‧‧‧Power supply

104‧‧‧ Controller

106‧‧‧Switch

108‧‧‧Lighting diode strings

110‧‧‧resistance

112‧‧‧Inductance

114‧‧‧dipole

116‧‧‧ Capacitance

200‧‧‧ drive circuit

202‧‧‧Power supply

204‧‧‧Rectifier

206‧‧‧Power Converter

208‧‧‧Lighting diode strings

210‧‧‧ Controller

212‧‧‧ filter

214‧‧‧ Energy storage components

218‧‧‧resistance

278‧‧‧ Current sensor

288‧‧‧load

300‧‧‧Light source drive circuit

302, 304‧‧‧Inductance

308‧‧‧ Capacitance

314‧‧‧ diode

316‧‧‧ switch

318‧‧‧ Capacitance

320‧‧‧resistance

322‧‧‧ Capacitance

324‧‧‧ Capacitance

333‧‧‧Common node

402‧‧‧Error amplifier

404‧‧‧ Comparator

408‧‧‧Pulse width modulation signal generator

602‧‧‧Error amplifier

604‧‧‧ Comparator

606‧‧‧Sawtooth signal generator

608‧‧‧Reset signal generator

610‧‧‧Pulse width modulation signal generator

800‧‧‧Light source drive circuit

802‧‧ ‧ Zener diode

804‧‧‧ switch

900‧‧‧Light source drive circuit

902‧‧‧Sawtooth signal generator

906‧‧‧Power Converter

910‧‧‧ Controller

912‧‧‧Power cord

920‧‧‧ filter

960‧‧‧Sawtooth signal

962‧‧‧Drive signal

1000‧‧‧Light source drive circuit

1008‧‧‧ input capacitor

1012‧‧‧resistance

1014‧‧‧ Capacitance

1016‧‧‧resistance

1018‧‧‧dipole

1024‧‧‧ output filter

1300‧‧‧flow chart

1302~1312‧‧‧Steps

1400‧‧‧Light source drive circuit

1402‧‧‧ Current filter

1406‧‧‧Power Converter

1408‧‧‧Light source

1410‧‧‧ Controller

1420‧‧‧voltage filter

1422‧‧‧Transformer

1424‧‧‧Switch

1462‧‧‧Drive signal

1464‧‧‧Induction signal

1466‧‧‧Monitoring signal

1500‧‧‧Light source drive circuit

1502‧‧‧ magnetic core

1504‧‧‧Primary winding

1506‧‧‧Secondary winding

1508‧‧‧Auxiliary winding

1512‧‧‧Inductance

1602‧‧‧Sawtooth signal generator

1660‧‧‧Sawtooth signal

1700‧‧‧Flowchart

1702, 1704, 1706, 1708, 1710, 1712‧‧ steps

1800‧‧‧Light source drive circuit

1808‧‧‧Light source

1810‧‧‧ Controller

1820‧‧‧ converter

1822‧‧‧Transformer

1824‧‧‧Primary winding

1826‧‧‧Secondary winding

1828‧‧‧Auxiliary winding

1830‧‧‧ magnetic core

1832‧‧ ‧ Voltage divider

1834‧‧‧Switch

1836‧‧‧Protection circuit

1838‧‧‧ sensor

1840‧‧‧Clamp circuit

1842‧‧‧ sensor

1850‧‧‧ drive signal

1852‧‧‧Induction signal

1854‧‧‧Monitoring signal

1856‧‧‧Induction signal

1860‧‧‧ Waveform

1900‧‧‧ waveform

2002‧‧‧ Acquisition circuit

2004‧‧‧State detector

2006‧‧‧Switch

2012‧‧‧Operational Amplifier

2014‧‧‧Sawtooth generator

2016‧‧‧ Comparator

2018‧‧‧buffer

2020‧‧‧Operational Transducer

2022‧‧‧ Capacitance

2050‧‧‧Signal Generator

2052‧‧‧ drive

2060‧‧‧Switch control signal

2062‧‧‧ square wave signal

2100‧‧‧ Waveform

2200‧‧‧Electronic system

2202‧‧‧Light source drive circuit

2204‧‧‧AC thyristor fluid (TRIAC) dimmer

2206‧‧‧TRIAC components

2208‧‧‧Variable resistor

2210‧‧‧ Capacitance

2212‧‧‧Diode AC Switch

2214‧‧‧Drainage path

2216‧‧‧Lower circuit

2218‧‧‧ Controller

2220‧‧‧Low signal

2222‧‧‧Detection signal

2250‧‧‧ drive signal

2402‧‧‧ Capacitance

2404‧‧‧Switch

2502‧‧‧TRIAC monitor

2602‧‧‧ Comparator

2604‧‧‧Filter

2606‧‧‧ Comparator

2608‧‧‧Division signal

2610‧‧ ‧ Voltage divider

2612‧‧‧ square wave signal

2700‧‧‧Flowchart

2702, 2704, 2706, 2708, 2710, 2712‧ ‧ steps

The technical method of the present invention will be described in detail below in conjunction with the accompanying drawings and specific embodiments to make the features and advantages of the present invention more obvious. Wherein: Figure 1 shows a schematic diagram of a conventional light source driving circuit.

2 is a schematic diagram of a driving circuit in accordance with an embodiment of the present invention.

3 is a circuit diagram of a light source driving circuit according to an embodiment of the invention.

4 is a schematic diagram of the controller shown in FIG. 3 in accordance with an embodiment of the present invention.

Figure 5 is a waveform diagram of the controller shown in Figure 4 in accordance with an embodiment of the present invention.

FIG. 6 is a block diagram showing another architecture of the controller shown in FIG. 3 according to an embodiment of the invention.

Figure 7 is a waveform diagram of the controller shown in Figure 6 in accordance with an embodiment of the present invention.

FIG. 8 is a schematic diagram of a light source driving circuit light source driving circuit according to another embodiment of the present invention.

FIG. 9A is a schematic diagram of a light source driving circuit according to another embodiment of the present invention.

Fig. 9B is a diagram showing signal waveforms in the light source driving circuit of Fig. 9A according to an embodiment of the present invention.

Figure 10 is a schematic illustration of a light source driving circuit in accordance with yet another embodiment of the present invention.

Figure 11 is a block diagram showing the structure of the controller of Figure 9A in accordance with an embodiment of the present invention.

Figure 12 is a diagram showing signal waveforms generated or received by a light source driving circuit in accordance with an embodiment of the present invention.

Figure 13 is a flow chart of a method for driving a drive circuit of a load in accordance with an embodiment of the present invention.

FIG. 14A is a block diagram showing a light source driving circuit according to another embodiment of the present invention. Figure.

Figure 14B is a diagram showing signal waveforms generated or received by the light source driving circuit shown in Figure 14A in accordance with the present invention.

Figure 15 is a circuit diagram showing a light source driving circuit according to another embodiment of the present invention.

Figure 16 is a block diagram showing the structure of the controller shown in Figure 14A in accordance with an embodiment of the present invention.

17 is a flow chart of a method of driving a light source in accordance with an embodiment of the present invention.

FIG. 18A is a circuit diagram showing a light source driving circuit according to another embodiment of the present invention.

Figure 18B is a diagram showing signal waveforms generated or received by a light source driving circuit in accordance with one embodiment of the present invention.

Figure 19 is a diagram showing signal waveforms generated or received by a light source driving circuit in accordance with one embodiment of the present invention.

Figure 20 is a block diagram showing the structure of the controller of Figure 18A in accordance with one embodiment of the present invention.

Figure 21 is a diagram showing signal waveforms generated or received by the controller of Figure 18A in accordance with an embodiment of the present invention.

Figure 22 is a schematic illustration of an electronic system in accordance with another embodiment of the present invention.

Figure 23 is a diagram showing signal waveforms generated or received by the TRIAC dimmer of Figure 22, in accordance with one embodiment of the present invention.

Figure 24 is a circuit diagram of the light source driving circuit of Figure 22, in accordance with one embodiment of the present invention.

Figure 25 is a block diagram showing the structure of the controller of Figure 22 in accordance with one embodiment of the present invention.

Figure 26 is a block diagram showing the structure of the TRIAC monitor of Figure 25 in accordance with one embodiment of the present invention.

27 is a flow chart of a method of driving a load in accordance with an embodiment of the present invention.

A detailed description of the embodiments of the present invention will be given below. Although the invention The invention will be described in connection with the embodiments, but it should be understood that this is not intended to limit the invention. Rather, the invention is to cover various modifications, equivalents, and equivalents of the invention as defined by the scope of the appended claims.

In addition, in the following detailed description of the embodiments of the invention However, it will be understood by those of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well-known methods, procedures, components, and circuits have not been described in detail in order to facilitate the invention.

2 is a schematic diagram of a drive circuit 200 in accordance with an embodiment of the present invention. Light source drive circuit 200 includes a rectifier 204 that receives an input voltage from a power source 202 and provides an adjusted voltage to power converter 206. Power converter 206 receives the adjusted voltage and provides an output power to load 288. In an embodiment, power converter 206 can be a buck converter or a boost converter. In one embodiment, power converter 206 includes an energy storage component 214 and a current sensor 278 (eg, a resistor) for sensing the power condition of energy storage component 214. Current sensor 278 provides a first signal ISEN to controller 210 to indicate the instantaneous current flowing through energy storage element 214. The drive circuit 200 also includes a filter 212 that generates a second signal IAVG for indicating an average current flowing through the energy storage element 214 based on the first signal ISEN. In one embodiment, the controller 210 receives the first signal ISEN and the second signal IAVG and controls the average current flowing through the energy storage element 214 to be a target current value level.

FIG. 3 is a circuit diagram of a light source driving circuit 300 according to an embodiment of the invention. Elements in Figure 3 having the same reference numerals as in Figure 2 have similar functions. In the example of FIG. 3, the light source driving circuit 300 includes a rectifier 204, a power converter 206, a filter 212, and a controller 210. Rectifier 204 can be a bridge rectifier including diodes D1-D4. Rectifier 204 adjusts the voltage from power source 202. Power converter 206 receives the voltage adjusted by rectifier 204 and provides an output power to power the load (e.g., LED string 208).

In the example of FIG. 3, power converter 206 is a buck converter that includes capacitor 308, switch 316, diode 314, current inductor (eg, resistor 218), The inductor 302 and the inductor 304 and the capacitor 324 are coupled to each other. The diode 314 is coupled between the switch 316 and the ground of the light source driving circuit 300. The capacitor 324 is coupled in parallel with the LED string 208. In an embodiment, the inductor 302 and the inductor 304 are electromagnetically coupled to each other. More specifically, the inductor 302 and the inductor 304 are coupled to a common node 333. In the example of FIG. 3, the common node 333 is interposed between the resistor 218 and the inductor 302. However, the invention is not limited to this architecture, and the common node 333 can also be located between the switch 316 and the resistor 218. The common node 333 provides a reference ground for the controller 210. In an embodiment, the reference ground of the controller 210 is different from the ground of the light source driving circuit 300. By turning on and off switch 316, the current flowing through inductor 302 can be adjusted to adjust the power supplied to light emitting diode string 208. Inductor 304 senses the power condition of inductor 302, for example, monitoring whether the current flowing through inductor 302 drops to a predetermined current level.

One end of the resistor 218 is coupled to a node between the switch 316 and the cathode of the diode 314, and the other end of the resistor 218 is coupled to the inductor 302. When switch 316 is turned "on" and "off", resistor 218 provides a first signal ISEN indicative of the instantaneous current flowing through inductor 302. In other words, the resistor 218 senses the instantaneous current flowing through the inductor 302 regardless of whether the switch 316 is turned "on" or "off". Filter 212 is coupled to resistor 218 and produces a second signal IAVG indicative of the average current flowing through inductor 302. In an embodiment, filter 212 includes a resistor 320 and a capacitor 322.

The controller 210 receives the first signal ISEN and the second signal IAVG and controls the average current flowing through the inductor 302 to be a target current level by turning on or off the switch 316. The capacitor 324 filters out the chopping current flowing through the LED string 208, thereby smoothing the current flowing through the LED string 208 and substantially equal to the average current flowing through the inductor 302. Thus, the current flowing through the LED string 208 can be substantially equal to the target current. Here, "substantially equal to the target current" means that the current flowing through the LED string 208 may be slightly different from the target current, but is still within an allowable range, so that the circuit component is not considered to be undesirable. The power transmitted from the inductor 304 to the controller 210 can be ignored and can be ignored.

In the example of FIG. 3, the endpoints of controller 210 include ZCD, GND, DRV, VDD, CS, COMP, and FB. The endpoint ZCD is coupled to the inductor 304 for receiving A detection signal AUX indicating the power condition of the inductor 302 (eg, whether the current flowing through the inductor 302 drops to a predetermined current level, for example, "0"). The detection signal AUX can also indicate whether the LED string 208 is in an open state. The terminal DRV is coupled to the switch 316 and generates a drive signal (eg, pulse width modulation signal PWM1) to turn the switch 316 on or off. Endpoint VDD is coupled to inductor 304 and receives power from inductor 304. The terminal CS is coupled to the resistor 218 and receives a first signal ISEN indicative of an instantaneous current flowing through the inductor 302. The terminal COMP is coupled to the reference ground of the controller 210 through the capacitor 318. The terminal FB is coupled through the filter 212 coupled to the resistor 218 to receive a second signal IAVG indicative of the average current flowing through the inductor 302. In the example of FIG. 3, the terminal GND (ie, the reference ground of the controller 210) is coupled to a common node 333 between the resistor 218, the inductor 302, and the inductor 304.

Switch 316 can be an N-channel metal oxide semiconductor field effect transistor (NMOSFET). The conduction state of switch 316 is determined based on a voltage difference between the gate voltage of switch 316 and the voltage at terminal GND (i.e., the voltage at common node 333). Therefore, the pulse width modulation signal PWM1 output from the terminal DRV determines the on or off state of the switch 316. When the switch 316 is turned on, the voltage level of the reference ground of the controller 210 is higher than the voltage level of the ground of the light source driving circuit 300, and thus the circuit of the present invention can be applied to a power source having a relatively high voltage.

In operation, when switch 316 is turned on, a current flows through switch 316, resistor 218, inductor 302, and LED string 208 to ground of light source drive circuit 300. When switch 316 is open, a current flows through resistor 218, inductor 302, LED string 208, and diode 314. The inductor 304 is magnetically coupled to the inductor 302 to detect the power condition of the inductor 302, for example, to detect if the current flowing through the inductor 302 has dropped to a preset current level. Therefore, the controller 210 monitors the current flowing through the inductor 302 through the detection signal AUX, the first signal ISEN, and the second signal IAVG, and controls the switch 316 through the pulse width modulation signal PWM1 to control the average current flowing through the inductor 302. A target current level. Therefore, the current flowing through the LED string 208 after being filtered by the capacitor 324 can also be substantially equal to the target current level.

In an embodiment, the controller 210 determines to send based on the detection signal AUX. Whether the photodiode string 208 is in an open state. If the LED string 208 is open, the voltage across the capacitor 324 increases. When the switch 316 is in the off state, the voltage across the inductor 302 increases, and the voltage of the detection signal AUX also increases accordingly. As a result, the current flowing into the controller 210 through the terminal ZCD increases. Accordingly, the controller 210 monitors the detection signal AUX, and if the switch 316 is turned off and the current flowing into the controller 210 increases beyond a current threshold, the controller 210 determines that the LED string 208 is in an open state.

The controller 210 can also determine whether the light emitting diode string 208 is in a short circuit state based on the voltage at the terminal VDD. If the LED string 208 is short-circuited, when the switch 316 is in the off state, since both ends of the inductor 302 are coupled to the ground of the light source driving circuit 300, the voltage across the inductor 302 will decrease. The voltage across inductor 304 and the voltage at terminal VDD are also reduced accordingly. Therefore, when the switch 316 is in the off state, if the voltage at the terminal VDD is lower than a voltage threshold, the controller 210 determines that the LED string 208 is in a short circuit state.

4 is a schematic diagram of the controller 210 shown in FIG. 3 in accordance with an embodiment of the present invention. FIG. 5 is a waveform diagram of the controller 210 shown in FIG. 4 in accordance with an embodiment of the present invention. Figure 4 will be described in conjunction with Figures 3 and 5.

In the example of FIG. 4, controller 210 includes an error amplifier 402, a comparator 404, and a pulse width modulation signal generator 408. The error amplifier 402 generates an error signal VEA based on a voltage difference between a reference signal SET and a second signal IAVG. The reference signal SET can indicate the target current level. The second signal IAVG is received through the terminal FB to indicate the average current flowing through the inductor 302. The error signal VEA can be used to adjust the average current flowing through the inductor 302 to the target current level. The comparator 404 is coupled to the error amplifier 402 and compares the error signal VEA with the first signal ISEN. The first signal ISEN is received through the terminal CS indicating the instantaneous current flowing through the inductor 302. The sense signal AUX is received through the endpoint ZCD indicating whether the current flowing through the inductor 302 has dropped to a predetermined current level (eg, reduced to zero). The pulse width modulation signal generator 408 is coupled to the comparator 404 and the terminal ZCD, and generates a pulse width modulation signal PWM1 based on the output of the comparator 404 and the detection signal AUX. The pulse width modulation signal PWM1 controls the conduction state of the switch 316 through the terminal DRV.

The pulse width modulation signal generator 408 generates a pulse width modulation signal PWM1 having a first level (eg, logic 1) to turn on the switch 316. When the switch 316 is turned on, a current flows through the switch 316, the resistor 218, the inductor 302, and the LED string 208 to the ground of the light source driving circuit 300. The current flowing through the inductor 302 gradually increases, so that the voltage of the first signal ISEN gradually increases. In an embodiment, when the switch 316 is turned on, the voltage of the detection signal AUX is a negative value. In one embodiment, within controller 210, comparator 404 compares error signal VEA with first signal ISEN. When the voltage of the first signal ISEN exceeds the voltage of the error signal VEA, the comparator 404 outputs a logic 0, otherwise the comparator 404 outputs a logic 1. In other words, the output of comparator 404 is a series of pulses. The pulse width modulation signal generator 408 generates a pulse width modulation signal PWM1 having a second level (e.g., logic 0) in response to the negative going output of the comparator 404, thereby turning off the switch 316. When the switch 316 is turned off, the voltage of the detection signal AUX becomes a positive value. When switch 316 is open, a current flows through resistor 218, inductor 302, LED string 208, and diode 314. The current flowing through the inductor 302 gradually decreases, so the voltage of the first signal ISEN gradually decreases. When the current flowing through the inductor 302 is reduced to a predetermined current level (eg, reduced to zero), the voltage of the detection signal AUX will generate a negative edge, and the pulse width modulation signal generator 408 will have the first state ( For example, the pulse width modulation signal PWM1 of logic 1) turns on the switch 316.

In one embodiment, the duty cycle ratio of the pulse width modulation signal PWM1 is determined by the error signal VEA. If the voltage of the second signal IAVG is less than the voltage of the reference signal SET, the error amplifier 402 increases the voltage of the error signal VEA to increase the duty cycle ratio of the pulse width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 increases until the voltage of the second signal IAVG increases to the voltage level of the reference signal SET. If the voltage of the second signal IAVG is greater than the voltage of the reference signal SET, the error amplifier 402 reduces the voltage of the error signal VEA to reduce the duty cycle ratio of the pulse width modulation signal PWM1, thereby reducing the average current flowing through the inductor 302 until The voltage of the second signal IAVG is lowered to the voltage level of the reference signal SET. Therefore, the average current flowing through the inductor 302 can be maintained to be equal to the target current level.

Figure 6 shows the controller shown in Figure 3 in accordance with an embodiment of the present invention. Another architectural schematic of 210. FIG. 7 is a waveform diagram of the controller 210 shown in FIG. 6 in accordance with an embodiment of the present invention. Figure 6 will be described in conjunction with Figures 3 and 7.

In the example of FIG. 6, the controller 210 includes an error amplifier 602, a comparator 604, a sawtooth signal generator 606, a reset signal generator 608, and a pulse width modulation signal generator 610. The error amplifier 602 generates an error signal VEA based on a voltage difference between a reference signal SET and a second signal IAVG. The reference signal SET indicates a target current level. The second signal IAVG receives an average current indicative of the flow through the inductor 302 through the terminal FB. The error signal VEA can be used to adjust the average current flowing through the inductor 302 to be equal to the target current level. The sawtooth signal generator 606 generates a sawtooth signal SAW. The comparator 604 is coupled to the error amplifier 602 and the sawtooth signal generator 606, and compares the error signal VEA with the sawtooth signal SAW. The reset signal generator 608 generates a reset signal RESET and provides a reset signal RESET to the sawtooth signal generator 606 and the pulse width modulation signal generator 610. In response to the reset signal RESET, the switch 316 is turned "on". The pulse width modulation signal generator 610 is coupled to the comparator 604 and the reset signal generator 608, and generates a pulse width modulation signal PWM1 based on the output of the comparator 604 and the reset signal RESET. The pulse width modulation signal PWM1 controls the conduction state of the switch 316 through the terminal DRV.

In one embodiment, the reset signal RESET is a pulse signal having a fixed frequency. In another embodiment, the reset signal RESET is a pulse signal that causes the switch 316 to be in an off state for a constant period of time. The reset signal RESET is such that the time when the pulse width modulation signal PWM1 in FIG. 5 is logic 0 is a constant.

In operation, pulse width modulation signal generator 610 generates a pulse width modulation signal PWM1 having a first state (eg, logic 1) to turn on switch 316 and to respond to reset signal RESET. When the switch 316 is turned on, a current flows through the switch 316, the resistor 218, the inductor 302, and the LED string 208 to the ground of the light source driving circuit 300. The voltage of the sawtooth wave signal SAW generated by the sawtooth signal generator 606 is increased from an initial level INI in response to the pulse of the reset signal RESET. When the voltage of the sawtooth signal SAW increases to the voltage of the error signal VEA, the pulse width modulation signal generator 610 generates a pulse width modulation signal PWM1 having a second state (for example, logic 0) to be turned off. Off 316, and the voltage of the sawtooth signal SAW is reset to the initial level INI until the sawtooth signal generator 606 receives the next pulse of the reset signal RESET. Upon receiving the next pulse of the reset signal RESET, the voltage of the sawtooth signal SAW will gradually increase from the initial level INI again in response to the pulse.

In one embodiment, the duty cycle ratio of the pulse width modulation signal PWM1 is determined by the error signal VEA. If the voltage of the second signal IAVG is less than the voltage of the reference signal SET, the error amplifier 602 increases the voltage of the error signal VEA to increase the duty cycle ratio of the pulse width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 increases until the voltage of the second signal IAVG increases to the voltage level of the reference signal SET. If the voltage of the second signal IAVG is greater than the voltage level of the reference signal SET, the error amplifier 602 reduces the voltage of the error signal VEA to reduce the duty cycle ratio of the pulse width modulation signal PWMI. Accordingly, the average current flowing through the inductor 302 decreases until the voltage of the second signal IAVG drops to the voltage level of the reference signal SET. Therefore, the average current flowing through the inductor 302 can be maintained to be equal to the target current level.

FIG. 8 is a schematic diagram of a light source driving circuit light source driving circuit 800 according to another embodiment of the present invention. Elements in Figure 8 having the same reference numerals as in Figures 2 and 3 have similar functions.

The terminal VDD of the controller 210 is coupled to the rectifier 204 through the switch 804 and receives the output voltage adjusted by the rectifier 204. A Zener diode 802 coupled between the switch 804 and the reference ground of the controller 210 is used to maintain the voltage at the terminal VDD substantially constant. In the example of FIG. 8 , the terminal ZCD of the controller 210 is electrically coupled to the inductor 302 and receives a detection signal AUX indicating the power condition of the inductor 302 . The detection signal AUX may indicate whether the current flowing through the inductor 302 has decreased to a preset current level (eg, whether it is reduced to zero). The common node 333 can provide a reference ground for the controller 210.

In summary, the present invention provides a circuit that controls a power converter to power a load. In one embodiment, the power converter provides a substantially constant current to a load, such as a string of light emitting diodes. In another embodiment, the power converter provides a current to charge the battery. The current provided by the circuit of the present invention to the load or battery can be more accurately controlled than the conventional circuit of FIG. Moreover, the circuit of the present invention is applicable to A voltage source with a relatively high voltage.

FIG. 9A is a block diagram showing a light source driving circuit 900 in accordance with another embodiment of the present invention. Elements in Figure 9A numbered the same as Figures 2 and 3 have similar functions. In an embodiment, light source drive circuit 900 includes a filter 920 coupled to power source 202, a rectifier 204, a power converter 906, a load 288, a sawtooth signal generator 902, and a controller 910. The power supply 202 produces an AC input voltage V _AC (eg, the AC input voltage V _AC has a sinusoidal signal) and an AC input current I _AC . The AC input current I _AC flows into the filter 920. Current I _AC ' flows out of filter 920 and flows into rectifier 204. The rectifier 204 receives the AC input voltage V _AC through the filter 920 and provides a rectified voltage V _IN and a rectified current I _IN on the power line 912. The power line 912 is coupled between the rectifier 204 and the power converter 906. Power converter 906 converts rectified voltage V _IN to an output voltage V _OUT to provide power to load 288. The controller 910 is coupled to the power converter 906 for controlling the power converter 906 to regulate the current I _OUT flowing through the load 288 and correct the power factor of the light source driving circuit 900.

Controller 910 generates a drive signal 962. In one embodiment, power converter 906 includes a switch 316. Drive signal 962 controls switch 316 to regulate current I _OUT flowing through load 288. Power converter 906 also generates an induced signal IAVG indicative of current I _OUT flowing through load 288.

In one embodiment, the sawtooth signal generator 902 coupled to the controller 910 generates a sawtooth signal 960 based on the drive signal 962. For example, drive signal 962 can be a pulse width modulated signal. In one embodiment, the sawtooth signal 960 is increased when the drive signal 962 is at a logic high level; when the drive signal 962 is at a logic low level, the sawtooth signal 960 is lowered to a predetermined voltage value (eg, reduced to 0V).

Advantageously, controller 910 generates drive signal 962 based on sawtooth signal 960 and sense signal IAVG. The drive signal 962 controls the switch 316 to maintain the current I _OUT flowing through the load 288 at the target current value to improve the accuracy of the current control. In addition, the drive signal 962 controls the switch 316, and the average current I _{IN_AVG of the} regulated rectified current I _IN is substantially in phase with the rectified voltage V _IN to correct the power factor of the light source drive circuit 900. In the present invention, substantially in phase means that the two waveforms are theoretically in phase, however, in practical applications, due to the presence of capacitance in the circuit, there is a slight phase difference between the two waveforms. The operation of light source drive circuit 900 will be further described in Figure 9B.

Figure 9B is a waveform diagram of signals in the light source driving circuit 900 of Figure 9A in accordance with one embodiment of the present invention, and Figure 9B will be described in conjunction with Figure 9A. FIG. 9B depicts waveforms of the input AC voltage V _AC , the rectified voltage V _IN , the rectified current I _IN , the average current I _{IN — AVG of the} rectified current, the current I _AC ', and the input AC current I _AC .

For convenience of description, the input AC voltage V _AC is a sinusoidal waveform, but is not limited thereto. The rectifier 204 rectifies the input AC voltage V _AC . In the embodiment of Figure 9B, the rectified voltage V _IN has a rectified sinusoidal waveform, i.e., the forward waveform of the input AC voltage V _AC remains, and its negative waveform is converted to a corresponding forward waveform.

In one embodiment, the drive signal 962 generated by the controller 910 controls the rectified current I _IN . The rectified current I _{IN increases} from a predetermined value (for example, 0 amps). After the rectified current I _IN reaches a value proportional to the rectified voltage V _IN , the rectified current I _IN drops to a preset value. As shown in FIG. 9B, the waveform of the average current I _{IN_AVG} of the rectified current I _IN is substantially in phase with the waveform of the rectified voltage V _IN .

The rectified current I _IN flows out of the rectifier 204 and flows into the power converter 906. The rectified current I _IN is the current rectified by the current I _AC ' flowing into the rectifier 204. As shown in FIG. 9B, when the input AC voltage V _AC is positive, the forward waveform of the current I _AC ' is similar to the forward waveform of the rectified current I _IN ; when the input current voltage V _AC is negative, the current I _AC The negative waveform of ' corresponds to the waveform of the rectified current I _IN .

In one embodiment, the input AC current I _AC is equal or proportional to the average of the current I _AC ' through a filter 920 coupled between the power source 202 and the rectifier 204. Therefore, as shown in FIG. 9B, the waveform of the input alternating current I _AC is substantially in phase with the waveform of the input alternating current voltage V _AC . In theory, the input AC current I _{AC is in} phase with the input AC voltage V _AC . However, in practical applications, there may be a slight phase difference between the input alternating current I _AC and the input alternating voltage V _AC due to the presence of capacitance in the filter 920 and the power converter 906. In addition, the input AC current I _AC is also substantially similar to the input AC voltage V _AC waveform. Therefore, the power factor of the light source driving circuit 900 is corrected, thereby improving the power supply quality of the light source driving circuit 900.

FIG. 10 is a schematic diagram of a light source driving circuit 1000 in accordance with still another embodiment of the present invention. Elements in Figure 10 that are numbered the same as Figures 2, 3, and 9A have similar functions. Figure 10 will be described in conjunction with Figures 4, 5 and 9A.

In the example of FIG. 10, the light source driving circuit 1000 includes a filter 920 coupled to the power source 202, a rectifier 204, a power converter 906, a load 288, a sawtooth signal generator 902, and a controller 910. In one embodiment, load 288 includes a light emitting diode string 208 (eg, a light emitting diode chain). The invention is not limited in this regard, and the load 288 may include other types of light sources or other types of loads (eg, battery packs). The filter 920 can be an inductor-capacitor filter including a pair of inductors and a pair of capacitors, but is not limited thereto. In one embodiment, controller 910 includes a plurality of ports, for example, ZCD埠, GND埠, DRV埠, VDD埠, FB埠, COMP埠, and CS埠.

In one embodiment, power converter 906 includes an input capacitor 1008 coupled to power line 912. The input capacitor 1008 reduces the chopping of the rectified voltage V _IN to smooth the waveform of the rectified voltage V _IN . In one embodiment, input capacitor 1008 has a relatively small capacitance value (eg, less than 0.5 microfarads) to help eliminate or reduce distortion of the rectified voltage V _IN waveform. Additionally, in one embodiment, since the capacitance of the input capacitor 1008 is small, the current flowing through the input capacitor 1008 can be ignored. Thus, when switch 316 is turned "on", current I ₂₁₄ flowing through switch 316 is substantially equal to the rectified current I _IN flowing from rectifier 204.

Power converter 906 is similar to the operation of power converter 206 in FIG. In one embodiment, the energy storage component 214 includes an inductor 302 and an inductor 304 that is electromagnetically coupled to the inductor 304. Inductor 302 is coupled to switch 316 and light emitting diode string 208. Therefore, current I ₂₁₄ flows through inductor 302 depending on the on state of switch 316. More specifically, in one embodiment, controller 910 generates a drive signal 962 (e.g., a pulse width modulation signal) on DRV to control switch 316 to be turned "on" or "off". When switch 316 is closed, current I ₂₁₄ flows from power line 912, through switch 316 and inductor 302, and increases. Current I ₂₁₄ can be derived from equation (1): ΔI ₂₁₄ = (V _IN - V _OUT ) * T _ON / L ₃₀₂ (1)

Where T _ON represents the time when the switch 316 is turned on, ΔI ₂₁₄ represents the amount of change of the current I ₂₁₄ , and L ₃₀₂ represents the inductance value of the inductor 302. In one embodiment, controller 910 controls drive signal 962 such that T _ON is a constant value. Therefore, when the output voltage V _OUT is substantially constant, in a time interval T _ON, the current I ₂₁₄ is the amount of change △ I of the rectified voltage ₂₁₄ is proportional to V _IN. In one embodiment, when current I _{214 is} lowered to a preset value (eg, 0 amps), switch 316 is closed. Therefore, the peak value of the current I ₂₁₄ is proportional to the rectified voltage V _IN .

When switch 316 is open, current I ₂₁₄ flows from ground and flows through diode 314 and inductor 302 into current LED string 208. Accordingly, current I _{214 is} decreased according to equation (2): ΔI ₂₁₄ = (-V _OUT ) * T _OFF / L ₃₀₂ (2) where T _OFF represents the off time of switch 316.

In one embodiment, current I _{IN is} equal to current I ₂₁₄ when switch 316 is on, and current I _IN is equal to 0 amps when switch 316 is open.

Inductor 304 senses the condition of inductor 302, for example, whether the current flowing through inductor 302 drops to a preset current value, such as 0 amps. In conjunction with FIG. 5, in one embodiment, the monitor signal AUX is low when the switch 316 is closed and the monitor signal AUX is high when the switch 316 is open. When the current I ₂₁₄ flowing through the inductor 302 is lowered to a preset current value, the voltage of the monitor signal AUX produces a negative edge. The ZCD of the controller 910 is coupled to the inductor 304 for receiving the monitoring signal AUX.

In one embodiment, power converter 906 includes an output filter 1024. Output filter 1024 can be a capacitor having a relatively large capacitance value (eg, greater than 400 microfarads). Therefore, the current I _OUT flowing through the LED string 208 represents the average value of the current I ₂₁₄ .

Resistor 218 produces a current sense signal ISEN indicative of the current flowing through inductor 302. In one embodiment, filter 212 is a resistor-capacitor filter comprising a resistor 320 and a capacitor 322. The filter 212 removes the chopping in the current sense signal ISEN to generate an average current sense signal IAVG of the current sense signal ISEN. Therefore, in the embodiment of FIG. 10, the average current sense signal IAVG represents the current I _OUT flowing through the LED string 208. The 埠FB of the controller 910 is used to receive the average current sense signal IAVG.

The sawtooth signal generator 902 is coupled to the DRV埠 and CS埠. The sawtooth signal generator 902 generates a sawtooth signal 960 on CS埠 based on the DRV drive signal 962. For example, the sawtooth signal generator 902 includes a resistor 1016 and a diode 1018 coupled between the DRV埠 and the CS埠 and connected in parallel with each other, and further includes a resistor 1012 and a capacitor coupled between the CS埠 and the ground and connected in parallel with each other. 1014. In operation, the sawtooth signal 960 varies according to the drive signal 962. More specifically, in one embodiment, drive signal 962 is a pulse width modulated signal. When the drive signal 962 is at a logic high level, the current I1 flows out of the DRV, passes through the resistor 1016, and flows into the capacitor 1014. Therefore, the capacitor 1014 is charged, and the voltage V _{960 of the} sawtooth signal 960 is increased. When the drive signal 962 is at a logic low level, the current I2 flows out of the capacitor 1014, passes through the diode 1018, and flows into the DRV. Therefore, the capacitor 1014 is discharged and the voltage V _{960 is} lowered to 0 volts. The sawtooth signal generator 902 may also include other components and is not limited to the embodiment shown in FIG.

In one embodiment, controller 910 is integrated on an integrated circuit die. Resistors 1016 and 1012, diode 1018, and capacitor 1014 are peripheral circuit components of the integrated circuit die. In another embodiment, the sawtooth signal generator 902 and controller 910 can also be integrated on an integrated circuit die. In this embodiment, the CS埠 can be omitted, thereby reducing the size and cost of the light source driving circuit 1000. The power converter 906 can also have other configurations and is not limited to the embodiment shown in FIG.

FIG. 11 is a block diagram showing the structure of the controller 910 of FIG. 9A according to an embodiment of the present invention. Elements in Figure 11 that are numbered the same as Figures 4 and 9A have similar functions. Figure 11 will be described in conjunction with Figures 4, 5, 9A and 10.

In one embodiment, controller 910 has a similar structure to controller 210 of FIG. 4, except that CS埠 receives sawtooth signal 960 instead of current sense signal ISEN. The controller 910 generates a drive signal 962 based on the sawtooth signal 960, the average current sense signal IAVG, and the monitor signal AUX. Controller 910 includes error amplifier 402, comparator 404, and pulse width modulation signal generator 408. The error amplifier 402 generates an error signal VEA based on the difference between the average current sense signal IAVG and the reference signal SET indicating the target current value. Comparator 404 compares sawtooth signal 960 and error signal VEA to produce comparison signal S. The pulse width modulation signal generator 408 generates a drive signal 962 based on the comparison signal S and the monitor signal AUX.

In one embodiment, when the monitor signal AUX indicates that the current I ₂₁₄ flowing through the inductor 302 drops to a preset value (eg, 0 amps), the drive signal 962 switches to a first potential (eg, a logic high) to close Switch 316. When the sawtooth signal 960 reaches the error signal VEA, the drive signal 962 switches to a second potential (eg, a logic low) to turn off the switch 316. Advantageously, since CS埠 receives the sawtooth signal 960 instead of the current sense signal ISEN, the peak value of the current I ₂₁₄ flowing through the inductor 302 is not limited by the error signal VEA. Therefore, as described in equation (1), the current I ₂₁₄ flowing through the inductor 302 changes according to the rectified voltage V _IN . For example, the peak value of current I ₂₁₄ is proportional to the rectified voltage V _IN rather than to the error signal VEA.

The controller 910 controls the drive signal 962 to maintain the current I _OUT at the target current value represented by the reference signal SET. For example, if the current I _{OUT is} greater than the target current value (eg, due to a change in the rectified voltage V _IN ), the error amplifier 402 reduces the error signal VEA to shorten the time T _ON at which the switch 316 is closed. Therefore, the average current of current I ₂₁₄ is reduced to reduce current I _OUT . Similarly, if the current I _{OUT is} less than the target current value, the controller 910 extends the time T _{ON at} which the switch 316 is closed to increase the current I _OUT .

Figure 12 is a diagram showing signal waveforms generated or received by a light source driving circuit (e.g., light source driving circuit 900 or 1000) in accordance with an embodiment of the present invention. Figure 12 will be described in conjunction with Figures 4, 9A, 9B and 10. Figure 12 depicts the rectified voltage V _IN, the rectified current I _IN, the rectified current I _IN average current _I IN_AVG, flowing through the light-emitting diode current I _OUT string 208, the current flowing through the inductor 302 represents the inductive signal ISEN ₂₁₄ of I The error signal VEA, the sawtooth signal 960, and the drive signal 962.

As shown in FIG. 12, the rectified voltage V _IN is a rectified sine wave signal. At time t1, drive signal 962 becomes a logic high. Thus, switch 316 is closed, current flows through the inductor 302 represents the increase I of the sensing signal ISEN _214. At the same time, the sawtooth signal 960 is increased in accordance with the drive signal 962.

At time t2, the sawtooth signal 960 is added to the error signal VEA. Accordingly, controller 910 adjusts drive signal 962 to a logic low level and sawtooth signal 960 to a voltage of zero volts. The drive signal 962 turns off the switch 316, so the sense signal ISEN drops. In other words, the sawtooth signal 960 and the error signal VEA determine the time T _{ON at which} the drive signal 962 is logic high.

At time t3, current I _{214 is} reduced to a preset current value (eg, 0 amps), whereby controller 910 adjusts drive signal 962 to a logic high to close switch 316.

In one embodiment, the current I _OUT flowing through the LED string 208 is equal or proportional to the average of the current I ₂₁₄ during one cycle of the rectified voltage V _IN . In conjunction with the description of FIG. 11, controller 910 adjusts current I _OUT to a target current value represented by reference signal SET. In addition, as shown in FIG. 12, the induced signal ISEN indicating the current I ₂₁₄ has the same waveform during the period from t1 to t4 and during the period from t5 to t6. Therefore, the average value of the current I ₂₁₄ during t1 to t4 is equal to the average value during t5 to t6. Therefore, the current I _{OUT is} maintained at the target current value. In one embodiment, T _{ON is} determined by the sawtooth signal 960 and the error signal VEA. Since the time during which the sawtooth signal 960 rises from 0 volts to the error signal VEA is equal during each period of the drive signal 962, T _ON is constant. According to equation (1), within a time T _ON, the current I ₂₁₄ is the amount of change △ I _{214 IN} proportional to the rectified voltage V. Therefore, as shown in FIG. 12, the peak value of the induced signal ISEN is proportional to the input voltage V _IN .

In one embodiment, when switch 316 is closed, the waveform of current I _IN is similar to the waveform of current I ₂₁₄ , and when switch 316 is open, current I _IN is equal to 0 amps. In the period t1 to t6, and the average current _I IN_AVG substantial rectifying the rectified voltage V _IN is in phase with current I _IN. As described in connection with FIG. 9B, the input current I _{AC is} substantially in phase with the input voltage V _AC , thereby correcting the power factor of the light source driving circuit, thereby improving the power supply quality.

13 is a flowchart 1300 of a method for driving a load driving circuit (eg, light source driving circuit 900 or 1000 for driving light emitting diode string 208) in accordance with an embodiment of the present invention. FIG. 13 will be described in conjunction with FIGS. 9A through 12. The specific steps covered in Figure 13 are only examples. That is, the present invention is also applicable to the steps of performing other reasonable steps or improving FIG.

In step 1302, an input voltage (eg, rectified voltage V _IN ) and an input current (eg, rectified current I _IN ) are received. In step 1304, the input voltage is converted to an output voltage to provide electrical energy to a load (eg, a light emitting diode source). In step 1306, current flowing through the energy storage element (eg, energy storage element 214) is controlled in accordance with a drive signal (eg, drive signal 962) to regulate the current flowing through the load.

In step 1308, a first induced signal (eg, average current sense signal IAVG) representative of the current flowing through the load is received. In one embodiment, the first sensed signal is obtained by filtering a second sensed signal indicative of current flowing through the energy storage element. In step 1310, a sawtooth signal is generated based on the drive signal.

In step 1312, the driving signal is controlled by the sawtooth wave signal and the first sensing signal to adjust the current flowing through the load to the target current value, and the average current through the control input current is substantially in phase with the input voltage to correct the light source driving circuit. Power factor. In one embodiment, an error signal is generated based on a difference between the first sensed signal and the reference signal, the reference signal representing a target current value flowing through the light emitting diode source. The sawtooth signal and the error signal are compared and a monitoring signal indicative of the condition of the energy storage element is received. If the monitoring signal indicates that the current flowing through the energy storage element is reduced to a preset value, the driving signal is switched to the first state, and the driving signal is switched to the second state according to the comparison value of the sawtooth wave signal and the error signal. When the drive signal is in the first state, the current flowing through the energy storage element is increased; and when the drive signal is in the second state, the current flowing through the energy storage element is reduced. In one embodiment, if the current flowing through the light emitting diode source is maintained at the target current value, the time during which the sawtooth signal is increased from the preset value to the error signal is constant.

FIG. 14A is a block diagram showing a light source driving circuit 1400 according to another embodiment of the present invention. Elements in Figure 14A numbered the same as Figures 2, 3 and 9A have similar functions. Figure 14B is a diagram showing signal waveforms generated or received by the light source driving circuit 1400 shown in Figure 14A in accordance with the present invention. 14A and 14B will be described in conjunction with Figs. 9A and 9B.

In the example of FIG. 14A, light source drive circuit 1400 includes a current filter 1402 coupled to a power source 202, a rectifier 204, a power converter 1406, a light source 1408, and a controller 1410. Power source 202 produces an AC input voltage V _AC (eg, V _AC has a sinusoidal signal) and an AC input current I _AC . The AC input current I _AC flows into the current filter 1402. Current I _AC ' flows out of current filter 1402 and flows into rectifier 204. Rectifier 204 receives AC input voltage V _AC through current filter 1402 and provides rectified voltage V _IN and rectified current I _IN on power line 912. The power line 912 is coupled between the rectifier 204 and the power converter 1406.

In one embodiment, power converter 1406 includes voltage filter 1420, transformer 1422, and switch 1424. Voltage filter 1420 receives voltage V _IN and filters voltage V _IN to produce a regulated voltage V _REG . For example, harmonic components having a relatively high frequency in voltage V _IN are excluded or eliminated. Therefore, as shown in FIG. 14B, the waveform of the stable voltage V _REG is more stable than the waveform of the voltage V _IN . Transformer 1422 converts regulated voltage V _REG to output voltage V _OUT to provide power to light source 1408. Therefore, the waveform of the output voltage V _OUT is not affected by variations in the input voltage V _IN (eg, sine wave). Correspondingly, the chopping of the current I _OUT flowing through the light source 1408 caused by the change of the input voltage V _IN is reduced or eliminated, thereby further reducing the horizontal frequency interference of the light emitted by the light source 1408.

The controller 1410 generates a drive signal 1462 to control the switch 1424 to operate in a first state or a second state, thereby further controlling the input current I _IN flowing into the voltage filter 1420 and the output current I _OUT flowing through the light source 1408. In one embodiment, transformer 1422 provides an inductive signal 1464 indicative of output current _IOUT . Based on the sense signal 1464, the controller 1410 controls the ratio of the on time T _ON and the off time T _OFF of the switch 1424 to adjust the output current I _OUT to a target value.

In one embodiment, during operation of switch 1424 in the first state, input current I _IN increases, and during operation of switch 1424 in the second state, input current I _IN decreases. During the second state, the controller 1410 controls the duration of the second state to allow the input current I _{IN to} decrease to a preset value (eg, zero). The controller 1410 further controls the duration of the first state to allow the input current to increase from a preset value to a value proportional to the input voltage V _IN . Input current I _{IN I} IN_AVG average current and input voltage V _IN substantial phase. Similar to the discussion of FIG. 9B, the input current I _AC voltage V _AC with substantial phase. Ideally, the AC input voltage V _AC and the AC input current I _AC are in phase. However, in practical applications, due to the presence of a capacitor in the current filter 1402 and the power converter 1406, a slight phase difference may result. Further, the waveform of the AC input current I _AC is similar to the waveform shape of the AC input voltage V _AC . Therefore, the power factor of the light source driving circuit 1400 can be corrected.

Advantageously, by switching the switch 1424 alternately between the first state and the second state, the power factor of the light source drive circuit 1400 can be corrected and the output current I _{OUT can be} adjusted to a target value. Therefore, the accuracy of the power quality control of the light source driving circuit 1400 is improved. Since only the switch 1424 is controlled, the size and cost of the light source driving circuit 1400 are reduced.

FIG. 15 is a circuit diagram of a light source driving circuit 1500 according to another embodiment of the present invention. Elements in Figure 15 that are numbered the same as Figures 2, 3, 9A and 14A have similar functions. Figure 15 will be described in conjunction with Figures 14A and 14B. In one embodiment, controller 1410 includes multiple endpoints, such as VIN endpoints, COMP endpoints, GND endpoints, DRV endpoints, CS endpoints, VDD endpoints, ZCD endpoints, and FB endpoints.

In one embodiment, voltage filter 1420 includes an inductor 1512, diodes D15 and D16, and a capacitor C15. Transformer 1422 can be a flyback converter including primary winding 1504, secondary winding 1506, auxiliary winding 1508, and magnetic core 1502. The switch 1424 is coupled to the diode D16 and the primary winding 1504 and operates in a first state (eg, an on state) and a second state (eg, an off state) to control an input current I _IN flowing through the inductor 1512. And an output current I _OUT flowing through the light source 1408.

In one embodiment, controller 1410 generates drive signal 1462 (eg, a pulse width modulation signal) to control switch 1424. More specifically, in one embodiment, when the drive signal 1462 has a logic high potential (eg, during the on state, T _ON ), the switch 1424 is turned on, the diode D15 is reverse biased, and the diode D16 is positive. Offset. The regulated voltage V _REG supplies power to the transformer 1422. Current I _PRI flows through primary winding 1504, switch 1424 to ground. The current I _{PRI is} increased to store energy in the magnetic core 1502. In addition, the input current I _IN flows through the inductor 1512, the diode D16, and the switch 1424, and the input current I _IN increases to charge the inductor 1512. The input current I _IN can be derived from equation (3): ΔI _IN =V _IN * T _CH /L ₁₅₁₂ (3)

Where T _CH represents the charging time during which the inductor 1512 is charged during the on state of the switch 1424. ΔI _IN represents the amount of change in the input current I _IN , and L ₁₅₁₂ represents the inductance value of the inductor 1512. In one embodiment, when switch 1424 is turned on, duration _TCH is equal to duration _TON .

When the drive signal 1462 has a logic low (eg, during the off state _TOFF ), the switch 1424 is turned off, the diode D15 is forward biased, and the diode D16 is reverse biased. The discharge of the transformer 1422 provides electrical energy to the LED string 208. Therefore, the current I _SE flowing through the secondary winding 1506 is reduced. In addition, the input current I _IN flows through the inductor 1512, the diode D15, and the capacitor C15, and according to equation (4), the input current I _IN decreases and discharges to the inductor 1512: ΔI _IN = (V _IN - V _REG )* T _DISCH /L ₁₅₁₂ (4)

Where T _DISCH represents the duration of discharge of the inductor 1512 during the off state of the switch 1424. Since the discharge of the inductor 1512 is terminated once the input current I _{IN is} reduced to 0 amps, the time T _DISCH and the time T _OFF may be different for the off state.

In one embodiment, inductor 1512 and capacitor C15 form an inductive-capacitor filter. The inductor-capacitor filter filters the high frequency harmonic components of the input voltage V _IN . Thus, the _{chopping of the} waveform of the stable voltage V _REG caused by the change in the input voltage V _IN is thus reduced. The transformer 1422 converts the stable voltage V _REG into an output voltage V _OUT .

In one embodiment, the auxiliary winding 1508 is coupled to the controller 1410 through the ZCD terminal. The auxiliary winding 1508 provides a monitoring signal 1466 indicating whether the current I _SE drops to a preset value (eg, 0 amps). The FB endpoint of controller 1410 receives an inductive signal 1464 indicating the output current I _OUT flowing through LED string 208. In one embodiment, the controller 1410 controls the duty cycle of the drive signal 1462 based on the monitor signal 1466 and the sense signal 1464 to adjust the output current I _OUT flowing through the LED string 208 to the target current value. The operation of controller 1410 will be further described in FIG.

In one embodiment, controller 1410 also controls the duration of drive signals 1462 T _ON and T _OFF to correct the power factor of light source drive circuit 1500. More specifically, in one embodiment, the controller 1410 will shut down duration T _OFF state is set to a time greater than the threshold value T _TH. By rewriting equation (4), the discharge time of inductor 1512 can be derived from equation (5): T _DISCH = _{ΔI IN} * L ₁₅₁₂ / (V _IN _V _REG ) (5)

As shown in FIG. 14B, ΔI _IN may be different in different cycle periods of the drive signal 1462. In one embodiment, the time threshold T _TH may be set to be equal to or greater than the amount of the maximum discharge time T _{DISCH — MAX} of the inductor 1512. Thus, the duration of the switch 1424 in the off state is sufficient to allow the input current I _{IN to} decrease to 0 amps. In addition, controller 1410 maintains the duration of T _ON at the same value. Thus, according to equation (3), the input current I _IN is increased from a preset value to a peak proportional to the input voltage V _IN . Therefore, as described in FIGS. 14A and 14B, the power factor of the light source driving circuit 1500 is corrected to improve the power supply quality of the light source driving circuit 1500.

Figure 16 is a block diagram showing the structure of the controller 1410 shown in Figure 14A, in accordance with an embodiment of the present invention. Elements in Figure 16 that are numbered the same as Figures 4 and 9A have similar functions. Figure 16 will be described in conjunction with Figures 4, 5, 10 and 11.

In one embodiment, controller 1410 has a similar structure to controller 910 of FIG. 11 except that it includes sawtooth signal generator 1602 that produces sawtooth signal 1660. In one embodiment, the operation of the sawtooth signal generator 1602 is similar to the sawtooth signal generator 902 shown in FIG. When the drive signal 1462 turns on the switch 1424, the sawtooth signal 1660 ramps up, and when the drive signal 1462 turns off the switch 1424, the sawtooth signal 1660 drops to 0 amps.

Controller 1410 generates drive signal 1462 based on sawtooth signal 1660, sense signal 1464, and monitor signal 1466. Controller 1410 also includes error amplifier 402, comparator 404, and pulse width modulation (PWM) signal generator 408. Error amplifier 402 amplifies the difference between sensed signal 1464 and reference signal SET indicative of the target current value to produce error signal VEA. Comparator 404 compares sawtooth signal 1660 with error signal VEA to produce a comparison signal S. The pulse width modulation signal generator 408 generates a drive signal 1462 based on the comparison signal S and a monitor signal 1466. T _ON corresponds to the time when the sawtooth signal 1660 is increased from the preset value to the error signal VEA.

In one embodiment, when the monitor signal 1466 indicates that the current I _SE flowing through the secondary winding 1506 has dropped to a preset value (eg, 0 amps), the drive signal 1462 has a logic high potential to turn on the switch 1424. When the sawtooth signal 1660 reaches the error signal VEA, the drive signal 1462 has a logic low to turn off the switch 1424.

The controller 1410 controls the drive signal 1462 to maintain the output current _IOUT at the target current value represented by the reference signal SET. For example, if the output current I _{OUT is} greater than the target value (eg, due to undesired noise), the error amplifier 402 will reduce the error signal VEA to shorten the on-state duration T _{ON of the} switch 1424. Therefore, the duty cycle of the drive signal 1462 is reduced to reduce the output current I _OUT . Likewise, if the output current I _{OUT is} less than the target value, the controller 1410 will increase the duty cycle of the drive signal 1462 to increase the output current I _OUT . In one embodiment, if the output current I _{OUT is} maintained at the target value, the duration T _{ON is} maintained at a constant value.

17 is a flow chart 1700 of a method of driving a light source in accordance with an embodiment of the present invention. Figure 17 will be described in conjunction with Figures 14A-16. The specific steps covered in Figure 17 are by way of example only. That is, the present invention is also applicable to the steps of performing other reasonable steps or improving FIG.

In step 1702, an input current (for example, input current I _IN ) and an input voltage (eg, input voltage V _IN ) are received. In step 1704, the input voltage is filtered to provide a regulated voltage (eg, a regulated voltage V _REG ). In step 1706, the regulated stable voltage is an output voltage (eg, output voltage _VOUT ) to provide electrical energy to the light source. In step 1708, a drive signal (eg, drive signal 1462) is generated to control the switch (eg, switch 1424) to alternate between the first state and the second state. The input current increases during the first state and decreases during the second state. In step 1708, step 1710 can be further included.

In step 1710, controlling the duration of the operation in the first state and the duration of operation in the second state such that the input current is reduced to a preset value (eg, 0 amps) during the second state operation, and at During a state operation, it increases from a preset value to a peak that is proportional to the input voltage.

In step 1712, the time ratio of the first state to the second state is controlled to adjust the output current through the light source to a target value.

FIG. 18A is a circuit diagram of a light source driving circuit 1800 according to another embodiment of the present invention. Elements in Figure 18A numbered the same as Figures 2 and 9A have similar functions. Figure 18A will be described in conjunction with Figure 14A.

In the example of FIG. 18A, the light source driving circuit 1800 includes a power source 202, a filter 920, a rectifier 204, a converter 1820, a transformer 1822, a sensor 1838, an inductor 1842, a switch 1834, a protection circuit 1836, and a light source 1808 (eg, an LED). And controller 1810. Power source 202 produces an AC input voltage V _AC (eg, V _AC has a sinusoidal signal) and an AC input current I _AC . The AC input current I _AC flows into the filter 920. Current I _AC ' flows out of filter 920 and flows into rectifier 204. Rectifier 204 receives AC input voltage V _AC through filter 920 and provides rectified voltage V _IN and rectified current I _C to converter 1820. Converter 1820 provides regulated voltage V _REG to transformer 1822. Transformer 1822 converts regulated voltage V _REG to output voltage V _{OUT to} power light source 1808. The controller 1810 controls the output current I _OUT to maintain the brightness of the light source 1808 at a target value and controls the rectified current I _C to correct the power factor of the light source driving circuit 1800. In one embodiment, the controller 1810 includes a plurality of pins, such as a pin DRV, a pin COMP, a pin CS, a pin FB, a pin GND, and a pin VDD.

In one embodiment, the converter 1820 coupled to the switch 1834 includes an inductor 1512, a diode D15, a diode D16, and a capacitor C18. In one embodiment, the transformer 1822 coupled to the switch 1834 can be a flyback transformer including a primary winding 1824, a secondary winding 1826, an auxiliary winding 1828, and a magnetic core 1830. The rectifier 204 has a reference ground GND1. Secondary winding 1826 has a reference ground GND2. The controller 1810 has a reference ground GND3. Converter 1820, primary winding 1824, auxiliary winding 1828, protection circuit 1836, and clamp circuit 1840 share reference ground GND3 with controller 1810. In one embodiment, the reference grounds GND1, GND2, and GND3 have different reference voltage values.

In one embodiment, controller 1810 generates drive signal 1850 at pin DRV to cause switch 1834 to alternately operate in a first state (eg, an on state) and a second state (eg, an off state). Thus, switch 1834 controls the rectified current I _C flowing through converter 1820 and the current I _PR flowing through primary winding 1824, thereby controlling the output current I _OUT flowing through source 1808.

Figure 18B shows a signal waveform diagram 1860 generated or received by light source drive circuit 1800 in accordance with one embodiment of the present invention. Fig. 18B will be described in conjunction with Fig. 18A. FIG. 18B shows waveforms of drive signal 1850, rectified current I _C flowing through converter 1820, current I _PR flowing through primary winding 1824, sense signal 1852, monitor signal 1854, and sense signal 1856.

In the example of FIG. 18B, the drive signal 1850 is a PWM signal. Within the on time T _ON (eg, from time t1 to t2, from t3 to t4, or from t5 to t6), the drive signal 1850 has a first state (eg, a high potential); during the off time _TOFF (eg, The drive signal 1850 has a second state (eg, a low potential) from time t2 to t3, or from t4 to t5).

When the drive signal 1850 is at a high potential, such as during the on time T _ON , the switch 1834 is turned on, the diode D15 is reverse biased, and the diode D16 is forward biased. Transformer 1822 is powered by regulated voltage V _REG . Current I _PR flows through capacitor C18, primary winding 1824, and switch 1834. As shown in Fig. 18B, the current I _{PR is} increased to transfer power from the converter 1820 to the magnetic core 1830 as shown in equation (6): ΔI _PR = V _REG * T _ON / L ₁₈₂₄ (6)

Here, ΔI _PR represents the amount of change in the current I _PR , and L ₁₈₂₄ represents the inductance value of the primary winding 1824. The current I _PR reaches a peak value I _PK at the moment the switch 1834 is turned off. In addition, the current I _C flowing through the inductor 1512, the diode D16, and the switch 1834 is increased to charge the inductor 1512 as shown in the equation (7): ΔI _C = V _IN * T _ON / L ₁₅₁₂ (7)

Here, ΔI _C represents the amount of change in the current I _C , and L ₁₅₁₂ represents the inductance value of the inductor 1512 . Therefore, when switch 1834 is turned on, both current I _C and current I _PR flow through switch 1834.

When the drive signal 1850 is low, as in the off time _TOFF , the switch 1834 is turned off, the diode D15 is forward biased, and the diode D16 is reverse biased. The current I _SE flowing through the secondary winding 1826 drops to transfer electrical energy from the magnetic core 1830 to the light source 1808 as shown in equation (8): ΔI _SE = (-V _OUT )* T _DIS /L ₁₈₂₆ (8)

Where T _DIS represents the time at which the current I _SE falls, and L ₁₈₂₆ represents the inductance value of the secondary winding 1826. Further, the current I _C flows from the rectifier 204 through the inductor 1512, the diode D15, and the capacitor C18, and flows to the reference ground GND3. As shown in Fig. 18B, the current I _C drops, causing the inductor 1512 to discharge, as shown in equation (9): _{ΔI C} = (V _IN - V _REG ) * T _DISCH / L ₁₅₁₂ (9)

Where T _DISCH represents the discharge time of the inductor 1512. Since the inductor 1512 stops discharging when the current I _{C is} reduced to zero amps, the discharge time T _DISCH can be different from the off time T _OFF . As shown in FIG. 18B, the discharge time T _{DISCH is} smaller than the off time T _OFF .

In one embodiment, the auxiliary winding 1828 provides a monitoring signal 1854. The monitor signal 1854 indicates whether the transformer 1822 is operating in a preset state. In one embodiment, the FB pin of the controller 1810 is coupled to the auxiliary winding 1828 via a voltage divider 1832 (eg, the voltage divider 1832 is a series coupled resistor R1 and R2) that receives the monitor signal 1854. More specifically, in one embodiment, when switch 1834 is in the off state and current I _{SE is} falling (eg, within time T _DIS ), the voltage across auxiliary winding 1828 is a positive voltage value. Thus, monitoring the voltage value having a positive signal 1854 in FIG. 18B V _3. When the monitoring signal of a voltage value of 1854 V _3, indicating that the transformer operates in 1822 by default. When the current I _SE drops to a predetermined value (e.g., zero amps), 1828 across the auxiliary winding voltage is zero volts, then the monitor signal 1854 has a voltage value V ₄ (e.g., zero volts). When the switch 1834 is in the ON state, the current I _PR rises 1828 across auxiliary winding voltage is a negative voltage value, then the monitor signal 1854 having a negative voltage value V _5. When the monitoring signal 1854 has a voltage value of V ₄ or V ₅ , it indicates that the transformer 1822 is not operating in the preset state.

In one embodiment, due to the electrical connection between the switch 1834, the reference ground GND1, the reference ground GND3, and the inductors 1838 and 1842, even the current I _C and the current I _PR flow through the switch during the on time T _ON . In 1834, an inductive signal 1852 indicating only the current I _PR may also be provided. Controller 1810 utilizes sense signal 1852 to obtain information regarding the output current I _OUT flowing through light source 1808. Therefore, the inductor on the secondary side of the light source driving circuit 1800 and the isolator between the primary side and the secondary side of the light source driving circuit 1800 can be omitted.

More specifically, in one embodiment, the inductors 1838 and 1842 are resistors, and the inductor 1838 is coupled between the switch 1834 and the reference ground GND1. The inductor 1842 is coupled between the reference ground GND1 and the reference ground GND3. In one embodiment, since the inductor 1838 is coupled in series to the switch 1834, both the current I _C and the current I _PR flow through the inductor 1838 during the on time T _ON . Therefore, the inductor 1838 senses the combined current I _{COMBINE of the} current I _C and the current I _PR . In one embodiment, the reference ground GND1 is also coupled to the current path of the current I _C . For example, current I _C flows through rectifier 204 and inductor 1512 without flowing through inductor 1842. However, since the reference ground GND3 is coupled to the capacitor C18, the current I _PR flows through the inductor 1842. Therefore, when the switch 1834 is turned on, the current I _C flows from the rectifier 204 through the inductor 1512, the diode D16, the switch 1834, the inductor 1838, the reference ground GND1, and flows back to the rectifier 204. Current I _PR flows from capacitor C18 through primary winding 1824, switch 1834, inductor 1838, ground reference GND1, inductor 1842, and back to capacitor C18. Thus, the inductor 1838 senses the combined current I _COMBINE (eg, the current value of the combined current I _COMBINE is equal to the sum of the current I _C and the current I _PR ). In addition, the inductor 1842 senses only the current I _PR .

In one embodiment, the pin CS of the controller 1810 is coupled to the reference ground GND1. Since the controller 1810 has the reference ground GND3, the controller 1810 can receive the sensing signal 1852 indicating the current _IPR at the pin CS. In one embodiment, the sense signal 1852 can be represented by a voltage on the inductor 1842. In one embodiment, the protection circuit 1836 is coupled to a common node between the switch 1834 and the inductor 1838 and receives an inductive signal 1856 indicative of the combined current I _COMBINE . In one embodiment, the sense signal 1856 can be represented by the total voltage V _TO on the inductor 1838 and the inductor 1842, as shown in equation (10): V _TO = I _C * R ₁₈₃₈ + I _PR * (R ₁₈₃₈ + R ₁₈₄₂ ) (10)

Wherein, R ₁₈₃₈ represents the resistance of the inductor 1838, and R ₁₈₄₂ represents the resistance of the inductor 1842. In another embodiment, the protection circuit 1836 includes a pair of pins coupled to both ends of the inductor 1838. Thus, the pin of protection circuit 1836 receives an inductive signal represented only by the voltage on inductor 1838, for example, I _COMBINE * R ₁₈₃₈ .

As shown in FIG. 18B, both the current I _C and the current I _PR rise during the on-time T _ON . Accordingly, the sense signal 1852 indicating the current I _PR rises, and the sense signal 1856 indicating the combined current I _COMBINE of the current I _C and the current I _PR rises. During the off time T _OFF , the current I _C flows from the capacitor C18 through the inductor 1842 to the reference ground GND1. Since the voltage value of the sense signal 1852 is equal to the voltage on the inductor 1842, the sense signal 1852 is a negative value and is inversely proportional to the current I _C .

In one embodiment, light source drive circuit 1800 also includes a clamp circuit 1840. Clamp circuit 1840 clamps voltage V _{1852 of} sense signal 1852 to a predetermined voltage value to prevent voltage V _{1852 from} falling below a preset threshold V _TH1 . In one embodiment, the clamp circuit 1840 includes a diode D17 and a resistor R3. Preset threshold V _TH1 D17 may be related threshold (e.g., -0.7 volts) and the diode. If voltage V _{1852 is} greater than V _TH1 , diode D17 is reverse biased. At this time, the voltage V _{1852 is} determined by the voltage on the inductor 1842. If the voltage V _{1852 is} less than V _TH1 , the diode D17 is forward biased and conducts current through the diode D17 and the resistor R3. Since a voltage drop across the diode D17 occurs, the voltage V ₁₈₅₂ is clamped at a predetermined voltage value, such as minus 0.7 volts. Therefore, in one embodiment, as shown in FIG. 18B, when the voltage on the inductor 1842 is less than the voltage V _TH1 , the sensing signal 1852 is clamped at the preset voltage value V _TH1 ; when the voltage on the inductor 1842 is greater than the voltage V _{At TH1} , the sense signal 1852 increases in inverse proportion to the current I _C . In addition, no current flows through the inductor 1838. Thus, when switch 1834 is turned off, the voltage value of sense signal 1856 is equal to the voltage value on inductor 1842.

The controller 1810 receives the monitor signal 1854 through the FB pin and receives the sense signal 1852 indicating the current I _PR flowing through the primary winding 1824 through the pin CS. In one embodiment, based on the sense signal 1852 and the monitor signal 1854, the controller 1810 monitors the output current I _OUT flowing through the light source 1808. Based on the sense signal 1852 and the monitor signal 1854, the controller 1810 generates a square wave signal. The average voltage of the square wave signal is proportional to the output current I _OUT (described further in Figures 20 and 21). Accordingly, controller 1810 generates drive signal 1850 to control switch 1834 to regulate output current I _OUT to target current value I _TARGET .

Advantageously, the controller 1810 is capable of sensing the output current I _OUT based on the sensed signal 1852 and the monitor signal 1854 generated by the primary side circuit of the light source drive circuit 1800. Therefore, the sensing circuit on the secondary side of the light source driving circuit 1800 and the isolation circuit coupled between the primary side and the secondary side of the light source driving circuit 1800 can be omitted, saving the size and cost of the light source driving circuit 1800.

In one embodiment, the protection circuit 1836 is coupled to the pin COMP of the controller 1810. The protection circuit 1836 compares the sense signal 1856 with the threshold value V _TH2 and pulls the voltage at the pin COMP to a preset voltage value, such as the voltage of the reference ground GND3, according to the comparison result. More specifically, in one embodiment, the protection circuit 1836 can be a transistor (not shown), but the invention is not limited thereto. The gate of the transistor receives the sensing signal 1856, the drain is coupled to the pin COMP, and the source is coupled to the reference ground GND3. If the voltage of the sense signal 1856 is greater than the threshold V _TH2 (eg, a threshold associated with the transistor), the transistor conducts between the pin COMP and the reference ground GND3. Accordingly, the controller 1810 controls the drive signal 1850 to prevent the light source drive circuit 1800 from entering an overcurrent condition. In one embodiment, if the voltage at pin COMP is pulled to the voltage of reference ground GND3, controller 1810 keeps switch 1834 in an open state, as further described below in FIG. At this time, both the current I _C and the current I _PR are cut off. Advantageously, the sense signal 1856 indicates the combined current I _COMBINE rather than the separate current I _C or I _PR . Therefore, the overcurrent state of either of the current I _C or the current I _PR will trigger the protection circuit 1836 to pull down the voltage at the pin COMP to prevent the light source driving circuit 1800 from being damaged.

Figure 19 is a diagram 1900 of a signal waveform generated or received by a light source driving circuit 1800 in accordance with one embodiment of the present invention. Figure 19 will be described in conjunction with Figures 14B, 18A and 18B. FIG 19 shows the rectified voltage V _IN, the average adjustment waveform voltage V _REG, the output voltage V _OUT, the current I _C, current I _{C I} C_AVG and the drive signal 1850.

As depicted in Figures 18A and 18B, when switch 1834 is turned on, current I _C rises. When switch 1834 is turned off, current I _C drops. The waveform of the current I _C is similar to the waveform of the rectified current I _IN in Fig. 14B. Thus, similar to the discussion of Fig. 14B, the AC input current I _AC and the AC input voltage V _AC in phase with the substance. Further, the waveform shape of the AC input current I _AC is similar to the waveform shape of the AC input voltage V _AC . Therefore, the power factor of the light source driving circuit 1800 is corrected, and the power supply quality of the light source driving circuit 1800 is improved. Further, the waveform of the adjustment voltage V _REG is more stable than the waveform of the rectified voltage V _IN . Therefore, the waveform of the output voltage V _OUT is not affected by the change in the rectified voltage V _IN (for example, a sine wave). Correspondingly, since the chopping of the output current I _OUT flowing through the light source 1408 caused by the change of the rectified voltage V _IN is reduced or eliminated, the line frequency interference of the light source 1808 is further reduced.

Figure 20 is a block diagram showing the structure of the controller 1810 of Figure 18A in accordance with one embodiment of the present invention. Elements in Figure 20 that are numbered the same as Figure 18A have similar functions. Figure 20 will be described in conjunction with Figure 18A.

Controller 1810 includes a signal generator 2050 and a driver 2052. The signal generator 2050 is coupled to the pin CS and the pin FB to receive the sense signal 1852 and the monitor signal 1854. Based on the sense signal 1852 and the monitor signal 1854, the signal generator 2050 generates a square wave signal 2062. The driver 2052 generates a driving signal 1850 on the pin DRV according to the square wave signal 2062 to control the turning on and off of the switch 1834, thereby controlling the output current I _OUT .

In one embodiment, signal generator 2050 includes acquisition circuitry 2002, state detector 2004, and a multiplexer (eg, switch 2006). The acquisition circuit 2002 is coupled to the pin CS to receive the sense signal 1852. Acquisition circuit 2002 collects peak I _PK of current I _PR flowing through primary winding 1824 based on current sense signal 1852. In one embodiment, the acquisition circuit 2002 has a sample hold function to generate a peak signal V _PK . That is, the acquisition circuit 2002 may be sampled current value of the current I _PR and holding the peak current I _PR I _PK. Therefore, the acquisition circuit 2002 outputs a peak signal V _{PK that is} proportional to the peak value I _{PK of the} current I _PR . In one embodiment, after the peak I _PK1 occurs at the current I _PR , the peak signal V _{PK is} constant to a voltage value V _PK1 that is proportional to I _PK1 until another peak occurs in the current I _PR .

In one embodiment, the switch 2006 includes a first pin, a second pin, and a third pin. The first pin of the switch 2006 is coupled to the output of the acquisition circuit 2002 for receiving the peak signal V _PK . The second pin of the switch 2006 is connected to the reference ground GND3 for receiving a preset voltage signal V _PRE (for example, V _PRE is zero volts). A third pin of switch 2006 is coupled to the input of driver 2052 for providing a square wave signal 2062. In another embodiment, the second pin of the switch can also be connected to other signal generators to receive a preset constant reference voltage.

State detector 2004 is coupled to pin FB to receive monitoring signal 1854. State detector 2004 determines whether transformer 1822 is operating in a preset state based on monitor signal 1854 and generates switch control signal 2060 to control the switch. More specifically, in one embodiment, when the monitoring signal 1854 has a voltage value V ₃ (indicated transformer 1822 operates in the default state), the switch control signal 2060 has a first state (e.g., high). At this time, the first pin and the third pin of the switch are turned on. Thus, the square wave signal 2062 is equal to the peak signal V _PK . When the monitor signal 1854 has a voltage value of V ₄ or V ₅ (indicating that the transformer 1822 is not operating in a preset state), the switch control signal 2060 has a second state (eg, a low potential). At this time, the second pin and the third pin of the switch are turned on. Thus, the square wave signal 2062 is equal to the preset voltage signal V _PRE .

21 is a signal waveform diagram 2100 generated or received by controller 1810 of FIG. 18A, in accordance with an embodiment of the present invention. 21 will be described in conjunction with FIGS. 18A, 18B, and 20. 21 shows waveforms of a square wave signal 2062, a current I _SE , a current I _PR , a monitor signal 1854, and a drive signal 1850.

In the embodiment of Figure 21, the drive signal period T _S 1850 is a PWM signal. During the time interval T _ON (eg, t1 to t2, t3 to t4, and t5 to t6), the drive signal 1850 has a first state (eg, a high potential). Therefore, the switch 1834 is in an on state. During the time interval T _OFF (eg, t2 to t3, t4 to t5, and t6 to t7), the drive signal 1850 has a second state (eg, a low potential). Therefore, the switch 1834 is in an off state.

When the monitoring signal 1854 has a voltage value V ₃ (indicated transformer 1822 is in the preset state), the square wave signal 2062 having a voltage value and the current peak value I _PR I V _{PK PK} proportional as shown in Equation (I1) as shown: V _PK = A * I _PK (11)

Where A represents the scaling factor between the voltage value V _PK and the peak value I _PK . When the monitor signal 1854 has a voltage value of V ₄ or V ₅ (indicating that the transformer 1822 is not operating in a preset state), the square wave signal 2062 is switched to a preset voltage value V _PRE (eg, zero volts).

According to the principle of conservation of energy, the average value I _{SE_AVG} of the current I _SE flowing through the secondary winding 1826 in the time interval T _DIS is proportional to the average value I _{PR_AVG} of the current IPR flowing through the primary winding 1824 in the time interval T _ON , as in the equation ( 12) indicates: I _{SE_AVG} = I _{PR_AVG} *(N _PR /N _SE )=1/2 * I _PK *(N _PR /N _SE ) (12)

Where N _PR /N _SE represents the turns ratio between the primary winding 1824 and the secondary winding 1826. Further, the average voltage V _{SQ_AVG of the} square wave signal 2062 can be expressed by equation (13): V _{SQ_AVG} = V _PK *(T _DIS /T _S ) (13)

In addition, the average current I _{OUT_AVG of the} output current I _OUT is equal to the average value I _{SE_AVG} of the current I _SE in the period T _S as expressed by equation (14): I _{OUT_AVG} = I _{SE_AVG} *(T _DIS /T _S ) (14)

Combined with equations (11), (12), (13), and (14), the average voltage V _{SQ_AVG of the} square wave signal 2062 can be expressed as: V _{SQ_AVG} = (2 * A / (N _PR / N _SE )) * I _{OUT_AVG} (15)

Therefore, according to equation (15), the average voltage V _{SQ — AVG} of the square wave signal 2052 generated by the light source driving circuit of the present invention is proportional to the average current I _{OTU — AVG} of the output current I _OUT flowing through the light source 1808.

Returning to FIG. 20, in one embodiment, the driver 2052 includes an operational amplifier 2012, a sawtooth generator 2014, a comparator 2016, and a buffer 2018. In one embodiment, operational amplifier 2012 includes an operational transconductance amplifier (OTA) 2020 and a capacitor 2022. The non-inverting input of the operational transduction amplifier 2020 receives the square wave signal 2062, and the inverting input receives the reference signal REF. Wherein, the reference signal REF represents the target current value I _{TARGET of the} output current I _OUT . The operational transconductance amplifier 2020 generates a current I ₂₀₂₀ at the output based on the difference between the square wave signal 2062 and the reference signal REF to charge or discharge the capacitance 2022, thereby generating an error signal 2064. Since the capacitor 2022 filters the chopping on the error signal 2064, the error signal 2064 is determined by the difference between the average voltage V _{SQ — AVG of the} square wave signal 2062 and the reference signal REF. In another embodiment, capacitor 2022 is external to controller 1810 and is coupled to operational transconductance amplifier 2020 via a pin of the controller.

The sawtooth generator 2014 generates a sawtooth wave signal SAW. The comparator 2016 compares the error signal 2064 with the sawtooth signal SAW and produces a comparison signal. Buffer 2018 receives the comparison signal and produces a drive signal 1850 (eg, a pulse width modulation signal). In the embodiment of FIG. 20, if the average voltage V _{SQ — AVG of the} square wave signal 2062 increases, the error signal 2064 increases, and the sawtooth signal SAW requires more time to increase to the error signal 2064. Thereby, the duty cycle of the drive signal 1850 is reduced. Similarly, if the average voltage V _{SQ — AVG of the} square wave signal 2062 decreases, the duty cycle of the drive signal 1850 increases.

In conjunction with Figures 18A and 20, controller 1810 and transformer 1822 form a negative feedback loop. More specifically, the drive signal duty cycle determines the average current 1850 _I OUT_AVG output current I _OUT. Also, the average voltage V _{SQ — AVG of the} square wave signal 2062 is _proportional to the average current I _{OUT — AVG} . In addition, the average voltage V _{SQ — AVG of the} square wave signal 2062 determines the duty cycle of the drive signal 1850. Thus, the negative feedback loop including controller 1810 and transformer 1822 can maintain the average voltage V _{SQ — AVG of} square wave signal 2062 equal to reference signal REF, thereby adjusting average current I _{OUT — AVG} to target current value I _TARGET .

For example, if the average voltage of the square wave signal _V SQ_AVG 2062 is greater than the reference signal REF (represented by the output current I _OUT is greater than the average current _I OUT_AVG target current value I _TARGET), increasing the operational amplifier 2012 to reduce the error signal, the drive signal 2064 The duty cycle of 1850, in turn, reduces the average current I _{OUT_AVG of the} output current I _OUT until the average voltage V _{SQ_AVG of the} square wave signal 2062 decreases to the reference signal REF. Similarly, if the average voltage of the square wave signal 2062 _V SQ_AVG less than the reference signal REF (represented by the average current output current I _OUT is less than the target current value _I OUT_AVG I _TARGET), operational amplifier 2012 to reduce the error signal to increase the driving signal 2064 The duty cycle of 1850, which in turn increases the average current I _{OUT_AVG of the} output current I _OUT , until the average voltage V _{SQ_AVG of the} square wave signal 2062 increases to the reference signal REF. Thus, the average current of the output current I _OUT can be adjusted to be equal to the target current value I _TARGET .

As described in FIG. 18A, when the light source driving circuit 1800 is in an overcurrent state, the protection circuit 1836 pulls the voltage at the pin COMP to a preset voltage value, such as the reference ground GND3. As shown in FIG. 20, if the voltage at the pin COMP is pulled to GND3, the comparator 2016 and the buffer 2018 keep the switch 1834 open, thereby cutting off the current I _C and the current I _PR . Controller 1810 can have other configurations and is not limited to the embodiment shown in FIG.

FIG. 22 is a circuit diagram of an electronic system 2200 in accordance with another embodiment of the present invention. Elements in Figure 22 that are numbered the same as Figures 2, 9A and 18A have similar functions. Figure 22 will be described in conjunction with Figure 18A.

In the example of FIG. 22, electronic system 2200 includes a power source 202, an AC thyristor fluid (TRIAC) dimmer 2204, and a light source drive circuit 2202. In one embodiment, light source drive circuit 2202 includes filter 920, rectifier 204, converter 1820, transformer 1822, switch 1834, light source 1808, drain path 2214, pull-down circuit 2216, and controller 2218. The power source 202 generates an AC input voltage V _AC between the live line and the neutral. The TRIAC dimmer 2204 converts the AC input voltage V _AC to an AC voltage V _TRIAC . Rectifier 204 receives AC voltage V _TRIAC through filter 920 and provides rectified voltage V _IN to converter 1820. Converter 1820 provides regulated voltage V _REG to transformer 1822. Transformer 1822 converts regulated voltage V _REG to output voltage V _{OUT to} power light source 1808. The controller 2218 controls the output current I _OUT to maintain the brightness of the light source 1808 as a target value.

The TRIAC dimmer 2204 can be a push button switch or a knob switch mounted on a wall or on a lamp holder. In one embodiment, the TRIAC dimmer 2204 includes a TRIAC element 2206 coupled between the power source 202 and the filter 920. The TRIAC component 2206 has a pin A1, a pin A2, and a gate G. The TRIAC dimmer 2204 also includes a variable resistor 2208 and a capacitor 2210 coupled in series, and a diode AC switch (DIAC) 2212. One end of the diode AC switch 2212 is coupled to the capacitor 2210 and the other end is coupled to the gate G of the TRIAC component 2206. The TRIAC component 2206 is a bidirectional switch that conducts current in either direction once triggered. The TRIAC element 2206 can be triggered by a positive or negative current applied to the gate G. Once triggered, the TRIAC component 2206 will remain conductive until the current flowing through pin A1 and pin A2 drops to a threshold (eg, hold current _IH ).

Figure 23 is a diagram showing signal waveforms generated or received by the TRIAC dimmer 2204 of Figure 22, in accordance with one embodiment of the present invention. Figure 23 will be described in conjunction with Figure 22. Figure 23 shows the AC input voltage V _AC , the voltage V _A2-A1 between the TRIAC component 2206 pin A1 and the pin A2, the current I _DIAC flowing through the diode AC switch 2212, the AC voltage V _TRIAC and the rectified voltage V The waveform of _IN .

In the example of FIG. 23, the AC input voltage V _AC has a sinusoidal waveform. Between time T ₀ and time T ₁ , the TRIAC element 2206 is turned off, and the voltage V _A2-A1 between the pin A1 and the pin A2 increases as the AC input voltage V _AC increases. At this time, the charging current I _CH flows through the resistor 2208 and the capacitor 2210 to charge the capacitor 2210. The voltage on capacitor 2210 rises accordingly. When the voltage across the capacitor at time T ₁ 2210 reaches exchange switch 2212 and diode voltage associated threshold, diode AC switch 2212 is turned on, thereby generating a current pulse is applied to the gate electrode of the TRIAC element 2206 G's. The TRIAC element 2206 is triggered to conduct by the current pulse. Accordingly, the current I ₁ flowing through the line of fire from the TRIAC element 2206, a filter 920, discharge path 2214 to neutral (Neutral). In addition, current I ₂ flows from the live line through TRIAC element 2206, filter 920, to rectifier 204. Thus, the current I flowing through a TRIAC element 2206 is equal to the current I and I _3, and ₁₂ of the. Between time T ₁ and time T ₂ , the bleed path 2214 conducts current I ₁ to maintain the current I ₃ flowing through the TRIAC element 2206 greater than the holding current I _H . Thus, at time T ₁ to time T ₂ Room, 2206 a TRIAC element is kept turned on. Therefore, the waveform of the AC voltage V _TRIAC coincides with the waveform of the AC input voltage V _AC between the time T ₁ and the time T ₂ .

At time T ₂ near the end of the first half cycle of the AC input voltage V _AC , since the current I ₃ flowing through the TRIAC element 2206 drops below the holding current I _{H of the} TRIAC element 2206, the TRIHC element 2206 is turned off.

During the second half cycle of the AC input voltage V _AC , when the voltage on the capacitor 2210 turns on the diode AC switch 2212 at time T ₃ , the TRIAC element 2206 is again turned on. Similarly, between time T ₃ and time T ₄ , the waveform of the alternating voltage V _TRIAC coincides with the waveform of the alternating current input voltage V _AC .

In one embodiment, the user can adjust the resistance R _{2208 of the} variable resistor 2208, for example, by rotating the knob of the TRIAC dimmer 2204 to adjust the resistance R _{2208 of the} variable resistor ₂₂₀₈ . The resistance R _{2208 of the} variable resistor 2208 determines the turn-on timing of the TRIAC element 2206 during each half cycle of the AC input voltage V _AC . More specifically, in one embodiment, if the resistance R _{2208 of the} variable resistor increases, the average value of the charging current I _CH that charges the capacitor 2210 after time T ₀ decreases. Therefore, the voltage on capacitor 2210 requires more time to reach the voltage threshold associated with diode AC switch 2212. Therefore, the conduction time of a TRIAC element 2206 is delayed, for example, later than time T _1. Similarly, if the resistance R _{2208 of the} variable resistor is decreased, the turn-on timing of the TRIAC element 2206 is advanced, for example, earlier than the time T ₁ . Therefore, by adjusting the resistance R _{2208 of the} variable resistor 2208, the conduction timing of the TRIAC element 2206 is adjusted correspondingly in each half cycle, for example, the conduction time is delayed or advanced. The TRIAC dimmer 2204 can have other configurations and is not limited to the embodiments of Figures 22 and 23. In another embodiment, if the resistance R _{2208 of the} variable resistor 2208 changes, for example, the resistance R ₂₂₀₈ is adjusted by the user, the turn-off timing of the TRIAC component 2206 is adjusted during each half cycle. By way of example, in the following description, the TRIAC dimmer 2204 adjusts the turn-on timing of the TRIAC element 2206. However, the present invention is not limited thereto, and the TRIAC dimmer 2204 of the present invention is also suitable for adjusting the turn-off timing of the TRIAC element 2206.

In one embodiment, rectifier 204 can be a bridge rectifier. Reserved rectifier 204 converts the negative portion of the AC voltage V _TRIAC and the positive portion of the AC voltage V _TRIAC into a corresponding value, thereby generating the rectified voltage V _IN. In some cases, since the capacitive elements in the converter 1820 and the transformer 1822 are capable of storing electrical energy, which in turn causes distortion of the waveform of the rectified voltage V _IN , at the end of the half cycle of the AC input voltage V _AC , the rectified voltage V _IN may Can't drop to zero volts. In one embodiment, the controller 2218 compares the detection signal 2222 indicative of the rectified voltage V _IN with the threshold voltage V _TH3 and produces a pull down signal 2220 based on the comparison. The pull-down circuit 2216 pulls the rectified voltage V _IN to a preset value in response to the pull-down signal 2220, for example, reference ground GND1. In one embodiment, at the end of each half cycle (eg, when the rectified voltage V _{IN is} below the threshold voltage V _TH3 ), the rectified voltage V _IN is pulled low. Therefore, the waveform distortion of the rectified voltage V _IN caused by the capacitive elements in the converter 1820 and the transformer 1822 is eliminated or avoided.

Returning to Figure 22, the operation of converter 1820, transformer 1822, and switch 1834 is similar to the operation of the corresponding components of Figure 18A. Advantageously, the controller 2218 receives the detection signal 2222 indicative of the rectified voltage V _IN and thereby detects the conduction state of the TRIAC element 2206. Controller 2218 generates drive signal 2250 based on the conduction state of TRIAC element 2206. The drive signal 2250 causes the switch 1834 to alternately operate in a first state (eg, an on state) and a second state (eg, an off state), thereby adjusting an average current flowing through the source 1808. More specifically, in one embodiment, controller 2218 detects the turn-on instant of TRIAC element 2206 in each cycle based on detection signal 2222. If the resistance R _{2208 of the} variable resistor 2208 increases, the turn-on timing of the TRIAC element 2206 is delayed in each cycle. Thus, controller 2218 controls switch 1834 to reduce the average current flowing through light source 1808. Similarly, if the resistance R _{2208 of the} variable resistor 2208 is reduced, the controller 2218 controls the switch 1834 to increase the average current flowing through the source 1808. Thus, controller 2218 implements dimming control of light source 1808 in accordance with the operation of TRIAC dimmer 2204. The operation of controller 2218 will be further described in conjunction with FIG.

Figure 24 is a circuit diagram of the light source driving circuit 2202 of Figure 22, in accordance with one embodiment of the present invention. Elements in Figure 22 that are numbered the same as Figures 2, 9A, 18A and 22 have similar functions. Figure 24 will be described in conjunction with Figures 18A and 22.

The light source driving circuit 2202 and the light source driving circuit 1800 in FIG. 18A have a similar structure except for the drain path 2214, the pull-down circuit 2216, and the controller 2218. In one embodiment, the bleed path 2214 includes a resistor R4 and a capacitor 2402 coupled in series. The pull-down circuit 2216 includes a switch 2404 and a resistor R5 coupled in series.

The controller 2218 includes a plurality of pins, for example, a pin CLP, a pin HV, a pin DRV, a pin COMP, a pin CS, a pin FB, a pin GND, and a pin VDD. In one embodiment, the controller 2218 receives the sensing signal 1852 indicating the current I _PR through the pin CS, and receives the sensing signal 1856 indicating the combined current I _COMBINE of the current I _C and the current I _PR through the pin COMP, through the pin FB. A monitoring signal 1854 indicating whether the transformer 1822 is operating in a preset state is received, a detection signal 2222 indicating the rectified voltage V _IN is received through the pin HV, a driving signal 2250 is generated through the pin DRV, and a pull-down signal 2220 is generated through the pin CLP.

Figure 25 is a block diagram showing the structure of the controller 2218 of Figure 22, in accordance with one embodiment of the present invention. Elements in Figure 25 that are numbered the same as Figures 20 and 24 have similar functions. FIG. 25 will be described in conjunction with FIGS. 20 and 24.

In the example of FIG. 25, controller 2218 includes a signal generator 2050, a TRIAC monitor 2502, and a driver 2052. The driver 2052 is coupled to the signal generator 2050 and the TRIAC monitor 2502. Signal generator 2050 generates a monitoring signal (eg, square wave signal 2062). The average voltage of the monitored signal is proportional to the average current flowing through the source 1808 (eg, the average current I _{OUT — AVG} ). The TRIAC monitor 2502 generates a reference signal REF based on the detection signal 2222. The reference signal REF is indicative of a target current value (eg, target current value I _TARGET ) of the average current flowing through the source 1808. Accordingly, driver 2052 generates drive signal 2250 based on square wave signal 2062 and reference signal REF. Similar to the discussion of FIG. 20, signal generator 2050, driver 2052, and transformer 1822 form a negative feedback loop. The negative feedback loop maintains the average voltage of the square wave signal 2062 equal to the reference signal REF, thereby maintaining the average current flowing through the source 1808 equal to the target current value I _TARGET .

Advantageously, the TRIAC monitor 2052 can adjust the reference signal REF according to the TRIAC dimmer 2204. More specifically, in one embodiment, the TRIAC monitor 2502 increments the reference signal REF if the detection signal 2222 indicates that the turn-on instant of the TRIAC component 2206 is advanced in each cycle. Thereby, the average current flowing through the light source 1808 increases. Similarly, if the detection signal 2222 indicates that the TAPAC element 2206 is delayed at the turn-on instant in each cycle, the TRIAC monitor 2502 reduces the reference signal REF. Thereby, the average current flowing through the light source 1808 is reduced. Controller 2218 can have other configurations and is not limited to the embodiment of FIG.

Figure 26 is a block diagram showing the structure of the TRIAC monitor 2502 of Figure 25 in accordance with one embodiment of the present invention. Figure 26 will be described in conjunction with Figure 25. In the example of FIG. 26, TRIAC monitor 2502 includes a comparator 2602, a comparator 2606, a voltage divider 2610, and a filter 2604. In one embodiment, voltage divider 2610 includes a resistor R6 and a resistor R7 coupled in series. Voltage divider 2610 receives detection signal 2222 and provides a divided signal 2608 indicative of rectified voltage V _IN . The comparator 2606 compares the divided voltage signal 2608 with the threshold voltage _VTH4 and generates a square wave signal 2612 based on the comparison result. Filter 2604 filters square wave signal 2612 to produce reference signal REF.

More specifically, in one embodiment, during the on time T _{TRI_ON} from time T ₁ to time T ₂ , the divided signal 2608 is greater than the threshold voltage V _TH4 (eg, zero volts) and the square wave signal 2612 is switched. To high potential. In the off time T _{TRI_OFF} from time T ₂ to time T ₃ , the divided voltage signal 2608 is less than the threshold voltage V _TH4 , and the square wave signal 2612 is switched to the low potential. When the turn-on timing of the TRIAC element 2206 changes, the average voltage of the square wave signal 2612 changes accordingly. Filter 2604 filters square wave signal 2612, which in turn provides a reference signal REF that is proportional to the average voltage of square wave signal 2612. Therefore, the average current through the light source 1808 can be rectified by adjusting the reference signal REF, thereby achieving dimming control of the light source 1808 according to the TRIAC dimmer 2204.

In addition, the comparator 2602 compares the detection signal 2222 indicating the rectified voltage V _IN with the threshold voltage V _TH3 to control the pull-down signal 2220 of the pin CLP. More specifically, in one embodiment, when the detection signal 2222 is below the threshold voltage _VTH3 , the pull-down signal 2220 has a high potential to turn on the switch 2404. When the detection signal 2222 is above the threshold voltage _VTH3 , the pull-down signal 2220 has a low potential to turn off the switch 2404. The TRIAC monitor 2502 can have other configurations and is not limited to the embodiment of FIG.

It is worth noting that although the above embodiment is to drive the LED light source The present invention is described by way of example, but the invention is not limited thereto, and the drive circuit of the present invention can also drive other loads, such as other types of light sources or battery packs.

27 is a flow chart 2700 of a method for driving a load (eg, light source 1808) in accordance with an embodiment of the present invention. Figure 27 will be described in conjunction with Figures 18A through 26. The specific steps covered in Figure 27 are by way of example only. That is, the present invention is also applicable to the steps of performing other reasonable steps or improving FIG.

In step 2702, a converter (eg, converter 1820) converts an input voltage (eg, rectified voltage V _IN ) to a regulated voltage (eg, regulated voltage V _REG ).

In step 2704, a transformer (eg, transformer 1822) converts the regulated voltage to an output voltage (eg, voltage _VOUT ) to power the load (eg, light source 1808).

In step 2706, a switch (eg, switch 1834) is alternately operated in a first state (eg, an on state) and a second state (eg, an off state) based on a drive signal (eg, drive signal 1850 or drive signal 2250) ). When the switch is operating in the first state, a first current flowing through the converter (eg, current I _C ) and a second current flowing through the transformer (eg, current I _PR ) flow through the switch. In one embodiment, the transformer includes a primary winding (eg, primary winding 1824) and a secondary winding (eg, secondary winding 1826). When the switch operates in the first state, the second current flowing through the primary winding rises. When the switch is operating in the second state, the third current (eg, current I _SE ) flowing through the secondary winding drops until it drops to a preset value (eg, zero amps). In one embodiment, when the switch is operating in the second state and the third current is decreasing (eg, in time interval T _DIS ), the transformer operates in a preset state.

In step 2708, a combined current indicative of the first current and the second current (eg, current) is received by monitoring a total voltage across the first inductor (eg, inductor 1838) and the second inductor (eg, inductor 1842) The first sensing signal of I _COMBINE ) (eg, sensing signal 1856). The first inductor is coupled between the switch and the first reference node. The second inductor is coupled between the first reference node and the second reference node. In one embodiment, the first reference node is the reference ground of the rectifier and the second reference node is the reference ground of the controller, wherein the rectifier is for generating an input voltage and the controller is for controlling the drive signal.

In step 2710, a second sensed signal (eg, sensed signal 1852) that only indicates the second current is received by monitoring the voltage on the second inductor. In one embodiment, the first square wave signal (eg, square wave signal 2062) is provided based on the second induced signal, and the average voltage of the first square wave signal (eg, the average voltage value of the square wave signal V _{SQ — AVG} ) flows through The average current of the load (the average current I _{OUT_AVG of the} output current I _OUT ) is proportional. In one embodiment, the first square wave signal is adjusted to a first voltage value (eg, voltage value V _PK ) that is proportional to the peak of the second current when the transformer is operating in the preset state. When the transformer is not operating in the preset state, the first square wave signal is adjusted to a second voltage value (eg, a preset voltage value V _PRE ).

In step 2712, the drive signal is controlled based on the first sensed signal and the second sensed signal to adjust the current flowing through the load. In one embodiment, the drive signal is controlled based on the first square wave signal to adjust the average current flowing through the load to a target current value (eg, target current value I _TARGET ). In one embodiment, the first sensed signal and the first threshold (eg, threshold V _TH2 ) are compared, and the voltage at the controller's pin (eg, pin COMP) is pulled to the pre-predict based on the comparison. Set the voltage value (for example, ground reference GND3). When the voltage at the pin is pulled to a preset voltage value, the drive signal is controlled to keep the switch operating in the second state. In one embodiment, an AC input voltage (eg, AC input voltage V _AC ) is converted to an AC voltage (eg, AC voltage V _TRIAC ) by a TRIAC component (eg, TRIAC component 2206) by a rectifier (eg, rectifier 920) The AC voltage is converted to an input voltage (eg, rectified voltage V _IN ). The conduction state of the TRIAC element is detected based on a detection signal (eg, detection signal 2222) indicating the input voltage. A reference signal (eg, reference signal REF) indicating a target current value of the average current flowing through the load is generated according to the detection signal. The drive signal is controlled in accordance with the reference signal to cause the switch to alternately operate in the first state and the second state, thereby controlling the average current flowing through the load. In one embodiment, it is detected based on the detection signal whether the conduction time of the TRIAC element changes in each cycle of the AC input voltage. The reference signal is adjusted according to the change in the conduction time.

Embodiments of the present invention provide a drive circuit that drives a load (eg, a light source). The drive circuit includes a converter, a transformer, a first inductor, and a second inductor. The converter receives the input voltage and provides a regulated voltage. The transformer converts the regulated voltage to an output voltage to power the load. When the switch is operating in the first state, both the first current flowing through the converter and the second current flowing through the transformer flow through the switch. A first inductor coupled between the switch and the first reference node provides a first sensed signal indicative of a combined current of the first current and the second current. A second inductor coupled between the first reference node and the second reference node provides a second sensing signal indicative of only the second current. This hair The load driving circuit, method and controller of the invention can not only save the inductor on the secondary side of the circuit and the isolator between the primary side and the secondary side of the circuit, reduce the size and cost of the circuit, and correct the power of the circuit. The factor improves the quality of the power supply.

The above detailed description and the accompanying drawings are only typical embodiments of the invention. It is apparent that various additions, modifications and substitutions are possible without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood by those of ordinary skill in the art that the present invention may be applied in the form of the form, structure, arrangement, ratio, material, element, element, and other aspects in the actual application without departing from the invention. Changed. Therefore, the embodiments disclosed herein are intended to be illustrative and not limiting, and the scope of the invention is defined by the scope of the appended claims and their legal equivalents.

202‧‧‧Power supply

204‧‧‧Rectifier

920‧‧‧ filter

1512‧‧‧Inductance

1800‧‧‧Light source drive circuit

1808‧‧‧Light source

1810‧‧‧ Controller

1820‧‧‧ converter

1822‧‧‧Transformer

1824‧‧‧Primary winding

1826‧‧‧Secondary winding

1828‧‧‧Auxiliary winding

1830‧‧‧ magnetic core

1832‧‧ ‧ Voltage divider

1834‧‧‧Switch

1836‧‧‧Protection circuit

1838‧‧‧ sensor

1840‧‧‧Clamp circuit

1842‧‧‧ sensor

1850‧‧‧ drive signal

1852‧‧‧Induction signal

1854‧‧‧Monitoring signal

1856‧‧‧Induction signal

Claims (34)

A driving circuit comprising: a converter receiving an input voltage and providing a regulating voltage; a switch coupled to the converter, the switch alternately operating in a first state and a second state; a transformer, coupled Connecting the converter and the switch, converting the regulated voltage to an output voltage to supply power to a load, wherein when the switch operates in the first state, a first current flowing through the converter and flowing through a second current flowing through the switch; a first inductor coupled between the switch and a first reference node to provide a first current indicating a combined current of the first current and the second current And a second sensor coupled between the first reference node and a second reference node to provide a second sensing signal indicating only the second current. The driving circuit of claim 1, further comprising: a controller coupled to the switch to generate a driving signal to alternately operate the switch in the first state and the second state; and a protection circuit coupled Receiving a pin of the controller, receiving the first sensing signal, the protection circuit comparing the first sensing signal with a first threshold, and pulling a voltage of the pin to a first according to a comparison result A preset voltage value. The driving circuit of claim 2, wherein when the voltage of the pin is maintained at the first predetermined voltage value, the controller controls the driving signal to keep the switch operating in the second state. The driving circuit of claim 1, wherein when the switch operates in the first state, the first current and the second current flow through the first inductor, and the second current only flows through the The second sensor. The driving circuit of claim 1, wherein the first reference node And coupling a current path of the first current, the first current does not flow through the second inductor. The driving circuit of claim 1, wherein a voltage value of the first sensing signal is equal to a sum of a voltage on the first inductor and a voltage on the second inductor, the second sensing signal A voltage value is equal to the voltage on the second inductor. The driving circuit of claim 1, further comprising: a rectifier for providing the input voltage; and a controller for generating a driving signal for alternately operating the switch in the first state and the second state, the rectifier And the controller has a different reference ground, the first reference node is a reference ground of the rectifier, and the second reference node is a reference ground of the controller. The driving circuit of claim 1, further comprising: a controller coupled to the switch to generate a driving signal to cause the switch to alternately operate in the first state and the second state, the controller comprising receiving the a pin of the second sensing signal; and a clamping circuit coupled between the first reference node and the pin, when a voltage on the second inductor drops below a second threshold The clamp circuit clamps a voltage at the pin to a second predetermined voltage value. The driving circuit of claim 1, further comprising: a controller coupled to the switch, and providing a first square wave signal based on the second sensing signal, an average voltage of the first square wave signal flowing through The average current of the load is proportional, and the controller provides a drive signal based on the first square wave signal to control the switch to control the average current. The driving circuit of claim 9, wherein the first square wave signal has a peak value of the second current when the transformer operates in a predetermined state. a first voltage value of the ratio; the first square wave signal has a second voltage value when the transformer is not operating in the preset state. The driving circuit of claim 10, wherein the transformer comprises a primary winding and a primary winding, wherein in the first state of the switch, the second current flowing through the primary winding rises; In the second state of the switch, a third current flowing through the secondary winding is decreased, wherein the transformer operates in the predetermined state when the third current flowing through the secondary winding drops. The driving circuit of claim 9, wherein the controller comprises: a driver that generates the driving signal to control the switch; and an acquisition circuit that acquires the peak value of the second current according to the second sensing signal, and Generating a peak signal having a first voltage value proportional to the peak; and a multiplexer transmitting the peak signal to the driver if the transformer is operating in the predetermined state; The transformer is not operating in the preset state, and the multiplexer transmits a preset signal to the driver, the preset signal having a second voltage value. The driving circuit of claim 9, wherein the controller and the transformer form a negative feedback loop, and the negative feedback loop maintains the average voltage of the first square wave signal equal to a reference signal to maintain the current The average current through the load is equal to a target current value. The driving circuit of claim 1, further comprising: an AC thyristor fluid, receiving an AC voltage, the AC 矽 control fluid alternately turning on and off during each cycle of the AC voltage, generating the The driving circuit further includes: a controller receiving a detection signal indicating the input voltage, the controller detecting an on state of the AC 矽 control fluid according to the detection signal, and generating a driving according to the conduction state a signal that causes the switch to alternately operate in the first state State and the second state. The driving circuit of claim 14, wherein the controller further comprises: a signal generator for generating a monitoring signal, an average voltage of the monitoring signal being proportional to an average current flowing through the load; The thyristor fluid monitor generates a reference signal according to the detection signal, the reference signal indicating a target current value of the average current flowing through the load; and a driver coupled to the signal generator and the AC sluice gate The fluid monitor generates the driving signal based on the monitoring signal and the reference signal to control the switch to adjust the average current to the target current value. The driving circuit of claim 15 , wherein the AC 矽 control fluid monitor detects, according to the detection signal, whether a conduction time of the AC 矽 control fluid in each cycle of the AC input voltage changes, and The reference signal is adjusted according to the change in the conduction time. The driving circuit of claim 15 , wherein the AC 矽 control fluid monitor generates a second square wave signal according to the detection signal, and filters the second square wave signal to generate the second square wave signal An average voltage is proportional to the reference signal. A control load power controller includes: an output pin that generates a drive signal to cause a switch to alternately operate in a first state and a second state; a converter coupled to the switch to convert an input voltage into a voltage adjustment; a transformer coupled to the switch, converting the regulated voltage to an output voltage to supply a load, and a first current and current flowing through the converter when the switch operates in the first state A second current flowing through the transformer flows through the switch; a protection pin is coupled to a protection circuit, and the protection circuit monitors a first induction And a total voltage on the second inductor is coupled to the combined current of the first current and the second current, the first inductor being coupled between the switch and a first reference node, the second sensing The sensor is coupled between the first reference node and a second reference node; and an inductive pin coupled to the first reference node to sense the second current by monitoring a voltage on the second inductor, The controller controls the driving signal according to a signal received by the sensing pin and a signal received by the protection pin. The control load power controller of claim 18, further comprising: a feedback pin coupled to an auxiliary winding of the transformer, the signal received by the feedback pin indicating whether the transformer operates at a preset a state, the controller generates a first square wave signal based on the signal received by the sensing pin and the signal received by the feedback pin, an average voltage of the first square wave signal and an average flowing through the load The current is proportional. The control load power controller of claim 19, wherein the first square wave signal has a first voltage value proportional to a peak value of the second current when the transformer operates in the preset state; The first square wave signal has a second voltage value when the transformer is not operating in the preset state. The controller of claim 19, wherein the transformer comprises a primary winding and a primary winding, and in the first state of the switch, the second current flowing through the primary winding rises; In the second state, a third current flowing through the secondary winding is decreased, wherein the transformer operates in the predetermined state when the third current flowing through the secondary winding drops. The control load power controller of claim 18, wherein when the total voltage on the first inductor and the second inductor is greater than a first threshold, the The protection circuit pulls a voltage at the protection pin to a first preset voltage value; if the voltage at the protection pin is pulled to the first preset voltage value, the controller controls the driving signal to maintain The switch operates in the second state. The control load power controller of claim 18, wherein the first reference node is a reference ground of a rectifier, the rectifier generates the input voltage, and the second reference node is a reference ground of the controller. The control load power controller of claim 18, further comprising: an AC 矽 control fluid, receiving an AC voltage, the AC 矽 control fluid alternately turning on and off in each cycle of the AC input voltage Breaking, generating the input voltage; the controller further comprising: a detecting pin, receiving a detecting signal indicating the input voltage, the controller detecting a conductive state of the alternating current control fluid according to the detecting signal, and according to the The on state controls the drive signal. The control load power controller of claim 24, further comprising: a signal generator for generating a monitoring signal, an average voltage of the monitoring signal being proportional to an average current flowing through the load; The thyristor monitor generates a reference signal according to the detection signal, the reference signal indicating a target current value of an average current flowing through the load; and a driver generating the driving signal based on the monitoring signal and the reference signal to The switch is controlled to adjust the average current to the target current value. The control load power controller of claim 25, wherein the controller detects, according to the detection signal, whether an alternating current control fluid changes at a conduction time in each cycle of the alternating voltage, and according to the conduction The change in time adjusts the reference signal. A load power control method includes: Converting an input voltage into a regulated voltage by using a converter; converting the regulated voltage into an output voltage by using a transformer to supply power to a load; and operating the switch alternately in a first state and a first according to a driving signal a second state, when the switch operates in the first state, a first current flowing through the converter and a second current flowing through the transformer flow through the switch; and a first sensor and a first a total voltage on the two inductors receives a first sensing signal indicating a combined current of the first current and the second current, the first inductor being coupled between the switch and a first reference node, The second inductor is coupled between the first reference node and a second reference node; receiving a second sensing signal indicating only the second current by monitoring a voltage on the second inductor; and according to the first An inductive signal and the second inductive signal control the drive signal to regulate a current flowing through the load. The method of claim 27, further comprising: providing a first square wave signal based on the second sensing signal, an average voltage of the first square wave signal being proportional to an average current flowing through the load; The driving signal is controlled based on the first square wave signal to adjust the average current to a target current value. The method of claim 28, further comprising: adjusting the first square wave signal to a first voltage value when the transformer is operating in a predetermined state, the first voltage value and one of the second current The peak is proportional; and when the transformer is not operating in the preset state, the first square wave signal is adjusted to a second voltage value. The method of claim 28, wherein the transformer comprises a primary winding and a primary winding, and in the first state of the switch, flowing through the primary winding The second current rises; in the second state of the switch, a third current flowing through the secondary winding drops until a predetermined value is reached, the method further comprising: if in the second state of the switch When the third current drops, it is determined that the transformer operates in the preset state. The method of claim 27, wherein the first reference node is a reference ground of a rectifier, the rectifier generates the input voltage, the second reference node is a reference ground of a controller, and the controller controls the driving signal. The method of claim 27, further comprising: comparing the first sensing signal with a first threshold; and pulling a voltage of a pin of a controller to a first predetermined voltage according to a comparison result And if the voltage of the pin is pulled to the first predetermined voltage value, the driving signal is controlled to keep the switch operating in the second state. The method of claim 27, further comprising: converting an alternating current voltage into the input voltage by using an alternating current control fluid; and detecting a conducting state of the alternating current control fluid according to a detection signal indicating the input voltage. Generating a reference signal according to the detection signal, the reference signal indicating a target current value of an average current flowing through the load; and controlling the driving signal according to the reference signal to control the switch to alternately operate in the first state and The second state. The method of claim 33, further comprising: detecting, according to the detection signal, whether an on-time of the alternating current control fluid in each period of the alternating voltage changes; and adjusting the reference according to the change of the conduction time signal.