RU2258940C1 - Satellite system for detection of watercrafts and aircrafts in state of emergency - Google Patents

Abstract

FIELD: detection of emergency radio buoys.

SUBSTANCE: system has two emergency radio buoys, satellite and information receipt station. Satellite has four antennae, three receipt devices, two memory devices and transmitter with antenna. information receipt station has antenna, receiver, two information processing devices, device for connecting to communication networks, control device and communication device of special rescue organizations. Third receipt device of satellite has five receipt antennae, seven mixers, six amplifiers of first intermediate frequency, two heterodynes, second intermediate frequency amplifier, block for detecting phase-manipulated signal, phase doubling block, two means for measuring spectrum width, comparison block, threshold block, delay line, two keys, phase-manipulated signal demodulator, ten multipliers, five narrow-band filters, lower frequency filter, four phase detector, five adders, four band filters, three phase inverters, two phase rotators for 90° and amplitude detector.

EFFECT: higher precision, higher interference resistance.

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Description

The proposed system is a modified COSPAS-SARSAT system and is designed to determine the location of emergency beacons (ARB) transmitting distress signals at a frequency of 121.5 MHz and in the frequency range 406-406.1 MHz.

The following satellite systems are known:

- the INMARSAT satellite system, which provides various types of services for use in the Global Maritime Communications System, including distress alerts and telephony, letterpress, and facsimile communications;

- INTELSAT-VI satellite communication system, consisting of ten independent repeaters - one for each beam of the communication antenna;

- GLOMAR system - a promising satellite communications system with moving objects in the frequency range 1.5-1.6 MHz;

- LOKSTAR system designed for positioning of moving objects and relaying radio messages;

- a system of global automatic control of vehicles under normal and extreme conditions (RF patent No. 2.158.003, G 01 S 7/00, 2000);

- satellite systems for determining the location of ships and aircraft in an accident (RF patents No. 2.027.195, G 01 S 5/12, 1992; No. 2.201.601, G 01 S 5/04, 2001; Global Maritime Distress Communication System and to ensure security. M: Transport, 1989, p.30, Fig. 12) and others.

Of the known satellite systems in the base quantity, the “Satellite system for determining the location of ships and aircraft that have crashed” was selected (RF patent No. 2.201.601, G 01 S 5/04, 2001), which is a modified COSPAS-SARSAT system. The latter is a joint international satellite search and rescue system for victims of disasters, developed and currently managed by organizations from Canada, France, the USA, Russia, Australia and Japan. The COSPAS-SARSAT system during the period of testing and practical application confidently demonstrated the possibility of using low-orbit satellites launched in near-polar orbits for global detection and location of emergency beacons (ARB).

However, in the known system, the same value of the first intermediate frequency w _up can be obtained by receiving signals at two frequencies w _s and w _s , i.e.

w _up1 = w _with -w _G1 and w _up1 = w _G1 -w _s .

Therefore, if the frequency w _{s is} taken as the main reception channel, then along with it there will be a mirror receiving channel, the frequency w _{s of} which differs from the frequency w _s by 2w _up1 and is located symmetrically (mirror) with respect to the frequency w _{Г1 of the} first local oscillator.

Conversion on the mirror channel of the reception occurs with the same conversion coefficient K _ol as on the main channel. Therefore, it most significantly affects the selectivity and noise immunity of the third receiving device (figure 2).

In addition to the mirror, there are other additional (combinational and intermodulation) reception channels.

In general terms, any Raman receive channel occurs when the following condition is met

,

,

where w _ki is the frequency of the i-th Raman reception channel;

m, n, i are positive integers.

The most harmful combinational reception channels are those generated by the interaction of the carrier frequency of the received signal with the harmonics of the local oscillator frequency of the second order (second, third, etc.), since the sensitivity of the third receiver on these channels is close to the sensitivity of the main channel. So, two combinational channels with m = 1 and n = 2 correspond to frequencies

w _k1 = 2w _G1 -w _up1 and w _k2 = 2w _r1 + w _up1.

If the interference frequency is close to or equal to the first intermediate frequency (w _P = w _up1 ), then a direct channel is formed. For this interference, the elements of the frequency converter perform the function of simple transmission links.

Intermodulation channels appear when two signals interacting with each other simultaneously appear in a free channel. The nature of intermodulation interference is as follows.

If two large amplitude signals with frequencies w ₁ and w ₂ (Fig. 3) simultaneously fall at the input of the third receiver, then they form a series of intermodulation frequencies on any nonlinear elements defined by the expression:

mw ₁ ± nw ₂ = w _mn .

The sum (difference) of the coefficients m and n is called the order, i.e. the intermodulation frequency w _mn is called a frequency of the order of m ± n. With an increase in the order of magnitude, the noise amplitudes quickly decrease.

If the interference frequency falls into the passband Δw _{P of the} receiving device, it is received as a useful signal.

The presence of false signals (interference) received through the mirror, Raman, intermodulation channels and the direct channel leads to a decrease in noise immunity and direction finding accuracy of ships and aircraft that have crashed.

An object of the invention is to increase the noise immunity and direction finding accuracy of beacons installed on ships and aircraft that have crashed by suppressing false signals (interference) received via additional channels.

The problem is achieved in that a satellite system for determining the location of ships and aircraft that have crashed, containing emergency beacons on ships and aircraft, at the information receiving points, a receiver with an antenna, a first information processing device, a device for interfacing with communication networks, a second input which through the second information processing device is connected to the output of the receiving device, the monitoring and control device and the communication device of the search and rescue body For example, on artificial satellites of the Earth, a second receiver with a second antenna, a first storage device and a transmitter with an antenna, the second input of which is connected to the output of the first receiver with the first antenna, and the third input with the output of the second receiver, the third receiver a device with third, fourth and fifth antennas and a second storage device, the output of which is connected to the fourth input of the transmitter, the fifth input of which is connected to the output t of the receiving device, while the first and second antennas are also connected to the third receiving device, complex signals with phase shift keying are used to transmit alarm information from emergency beacons, the third receiving device contains measuring and four direction finding channels, the measuring channel consists of the first antenna and in series included the first mixer, the second input of which is connected through the first local oscillator to the output of the search unit, and the first amplifier of the first intermediate frequency, followed by the phase doubler, the second spectral width meter, a comparison unit, the second input of which is connected to the output of the first spectral width meter, a pore block, the second input of which is connected to its output through the delay line, the first key, the second mixer, the second input of which is connected to the output the second local oscillator, the amplifier of the second intermediate frequency, the first multiplier, the second input of which is connected to the output of the low-pass filter, a narrow-band filter, the second multiplier, the second input of which is connected to the amplifier of the second intermediate frequency, the low-pass filter and the first adder, the output of which is the output of the third receiving device, while the control input of the unit, each direction finding channel consists of a series antenna, a mixer, the second input of which is connected to the output of the first local oscillator, the amplifier of the first intermediate frequency, multiplier, the second input of which is connected to the output of the amplifier of the second intermediate frequency, a narrow-band filter and a phase detector, the output of which is connected to the corresponding input of the first adder, while the second inputs of the phase detectors of the first and third direction-finding channels are connected to the outputs of the narrow-band filters of the first and third direction-finding channels, respectively, the receiving antennas of the measuring and direction-finding channels are placed in the form of a symmetrical geometric cross, at the intersection of which there is a receiving antenna of the measuring channel, the measuring channel is equipped with a second, third, fourth and fifth adders, four bandpass filters, three phase inverters with two 90 ° phase shifters, a second switch, a third multiplier, a second amplifier of the first intermediate frequency, a third mixer and an amplitude detector, and the first band-pass filter, the first phase inverter, the second adder, the second input of which is connected to the output, are connected in series to the output of the first receiving antenna the first receiving antenna, the second bandpass filter, the second phase inverter, the third adder, the second input of which is connected to the output of the second adder, the third bandpass filter, the third phase inverter and the fourth the adder, the second input of which is connected to the output of the third adder, and the output is connected to the first input of the first mixer, to the second output of the first local oscillator, the first 90 ° phase shifter, the third mixer, the second input of which is connected to the output of the fourth adder, the second amplifier of the first intermediate frequency, the second phase shifter 90 °, the fifth adder, the second input of which is connected to the output of the first amplifier of the first intermediate frequency, the third multiplier, the second input of which is connected to the output of the four rtogo adder, a fourth bandpass filter and amplitude detector of the second key, the second input of which is connected to the output of the fifth adder, and an output connected to the inputs doubler phase first measuring the spectral width and a first key.

The structural diagram of a satellite system for determining the location of ships and aircraft that have crashed, is presented in figure 1. Frequency diagrams explaining the principle of formation of additional receiving channels are shown in FIGS. 2 and 3. A block diagram of a third, second receiving device is shown in FIG. The relative position of the airborne receiving antennas is shown in Fig.5. Timing diagrams explaining the principle of the system are shown in Fig.6.

The system contains the first 1 and second 2 emergency beacons, artificial Earth satellite (AES) 3 and item 16 for receiving information.

The satellite equipment 3 includes in series the first receiving antenna 4, the first receiving device 9 and the transmitter 14 with the antenna 15, the second receiving antenna 5 in series, the second receiving device 10 and the first storage device 12, the output of which is connected to the second input of the transmitter 14, the third the input of which is connected to the output of the second receiving device 10, connected in series to the receiving antennas 4-8, the third receiving device 11 and the second storage device 13, the output of which is connected to the four th input of the transmitter 14, a fifth input connected to the output of the third receiving device 11.

The information receiving point 16 comprises in series a receiving antenna 17, a receiving device 18, a first information processing device 19, a communication network interface device 21, the second input of which is connected to the output of the receiving device 18 through a second information processing device 20, a monitoring and control device 22, and device 23 communications search and rescue organizations.

The third receiving device 11 consists of a measuring and four direction finding channels.

The measuring channel contains in series a first receiving antenna 4, a first bandpass filter 65, a first phase inverter 66, a second adder 67, the second input of which is connected to the output of the antenna 4, a second bandpass filter 68, a second phase inverter 69, a third adder 70, the second input of which is connected to the output of the second adder 67, the third bandpass filter 71, the third phase inverter 72 and the fourth adder 73, the second input of which is connected to the output of the adder 70, the first mixer 24, the second input of which is connected to the first output of the first local oscillator 30 and the first amplifier 31 of the first intermediate frequency, connected in series to the second output of the local oscillator 30, the first phase shifter 74 by 90 °, the third mixer 75, the second input of which is connected to the output of the adder 73, the second amplifier 76 of the first intermediate frequency, the second phase shifter 77 by 90 °, the fifth adder 78, the second input of which is connected to the output of the amplifier 31 of the first intermediate frequency, the third multiplier 79, the second input of which is connected to the output of the adder 73, the fourth band-pass filter 80, the amplitude detector 81, the second switch 82, W a swarm input which is connected to the output of the adder 78, a phase doubler 37, a second spectral width meter 39, a comparator 40, a second input of which is connected to an output of a key 82 through a first spectral width meter 38, a threshold unit 41, whose second input is connected via a delay line 42 with its output, the first key 43, the second input of which is connected to the output of the key 82, the second mixer 45, the second input of which is connected to the output of the second local oscillator 44, the amplifier 46 of the second intermediate frequency, the first multiplier 48, the second input of which is connected to the output m of a low-pass filter 51, a narrow-band filter 50, a second multiplier 49, the second input of which is connected to the output of the second intermediate-frequency amplifier 46, a low-pass filter 51 and a first adder 64, while the control input of the first local oscillator 30 is connected to the pore output through the search unit 29 block 41, phase doubler 37, spectral width meters 38 and 39, comparison block 40, threshold block 41, delay line 42, and first key 43 form a phase-shift signal detection block 36. Multipliers 48 and 49, a narrow-band filter 50 and a low-pass filter 51 form a demodulator 47 of the phase-shifted signal.

Each direction-finding channel consists of a series-connected and a receiving antenna 5 (6, 7, 8), a mixer 25 (26, 27, 28), the second input of which is connected to the first output of the first local oscillator 30, amplifier 32 (33, 34, 35) of the first intermediate frequency, multiplier 52 (53, 54, 55), the second input of which is connected to the output of the amplifier 46 of the second intermediate frequency, narrow-band filter 56 (57, 58, 59) and phase detector 60 (61, 62, 63), the output of which is connected with the corresponding input of the first adder 64, and the second inputs of the phase detectors 60 and 62 of the first and third bearing ion channels are connected to the output of the second local oscillator 44, the second inputs of the phase detectors 61 and 63 of the third and fourth direction-finding channels are connected to the outputs of the narrow-band filters 56 and 58 of the first and third direction-finding channels, respectively.

The suppression of false signals (interference) received on the Mirror channel at a frequency w _s is provided by an "outer ring" consisting of mixers 24 and 75, a local oscillator 30, phase shifters 74 and 79 by 90 °, amplifiers 31 and 76 of the first intermediate frequency and adder 78 and implements phase compensation method.

The suppression of false signals (interference) received via Raman channels at frequencies w _k1 and w _k2 is provided by the "inner ring" consisting of a multiplier 79, a bandpass filter 80, an amplitude detector 81, a key 82 and implementing the method of narrow-band filtering.

The suppression of false signals (interference) received through the direct channel at the frequency w _up is provided by a filter plug consisting of a bandpass filter 65, a phase inverter 66, an adder 67 and implementing a phase compensation method.

The suppression of false signals (interference) received via intermodulation channels in the frequency band Δw _П1 and Δw _П2 is ensured by filter plugs consisting of bandpass filters 68 and 71, phase inverters 69 and 72, adders 70 and 73, and implements a phase compensation method.

The system operates as follows.

The modified system (normal configuration) includes four satellites, two of which are presented and supported by the COMPASS side and two by the SARSAT side. Currently, there are three types of ARBs: aviation, marine and portable (for use on land) that emit distress signals detected and received by satellites of the modified CASPAS-SARSAT system for the purpose of subsequent relay to ground stations - information receiving points (PPI) for processing and location of emergency beacons. The service area of the system in real time is determined by the number and geographical location of the PPI. Each PPI serves an area with a radius of approximately 2,500 km. The system includes 15 PPI deployed in seven countries. In Russia, PPI are located in Moscow, Arkhangelsk, Novosibirsk and Vladivostok.

Distress messages and the coordinates of the emergency facility are transmitted through the system control center (NCC) either to the national rescue coordination center, or to another NSC or the corresponding search and rescue service in order to deploy a search and rescue operation.

The system uses ARB 1 operating at a frequency of 121.5 MHz - the international aviation emergency frequency - and in the frequency range 406-406.1 MHz, where ARB 2, which are technically more complex than ARB 1, are used.

An important feature of the new ARB provision is the inclusion in its radiation of a digital message that carries information on the ARB (country) affiliation, the identification number of the vessel or aircraft and the type of distress. The ARB message installed on ships may also include information on the location of the ship, entered manually or automatically from ship's radio navigation devices.

Two modes of operation are used to detect ARB signals and determine their location: the mode of receiving and transmitting information in real time and the mode of reception with storing information on board the satellite and its subsequent transmission to the point of reception of information when the satellite is in the radio-visibility zone of the PPI. ARB 1 can only be used in direct transmission mode, while ARB 2 can be used in both operating modes.

The tuning frequency of the receiving device 9 is 121.5 MHz, the tuning frequency of the receiving devices 10 and 11 is in the frequency range 406-406.1 MHz.

The receiving devices 9 and 10 perform the following functions:

- demodulation of digital messages received from ARB 1 and ARB 2;

- measurement of the frequency of the received signal;

- binding of time stamps to the given measurements.

The receiving device 11 performs the following functions:

- suppression of false signals (interference) received via additional channels;

- detection and selection of phase-shifted (PSK) signals in a given frequency range;

- synchronous detection of PSK signals;

- accurate and unambiguous direction finding ARB 2 phase method;

- linking the results of measurements to time stamps.

To determine the coordinates of ARB 2, the proposed system uses the phase direction finding method, which is characterized by a contradiction between the requirements of accuracy and uniqueness of determining the angular coordinates of ARB 2.

In order to eliminate this contradiction, two reference scales are used in each plane: large - accurate, but ambiguous and small - rough, but unequivocal:

.

.

In this case, a smaller measuring base d forms a rough but unambiguous direction finding scale, and a large measuring base 2d forms a thin but ambiguous direction finding scale.

Viewing this frequency range Df and searching for the PSK signals of the ARB 2 is performed using the search unit 29, which periodically with a period T _P changes the frequency of the first local oscillator 30 according to a sawtooth law. A sawtooth voltage generator can be used as the search unit 29.

Received QPSK signals:

u ₁ (t) = υ _c · cos [(w _с ± Δw) t + φ _к (t) + φ ₁ ],

u ₂ (t) = υ _c · cos [(w _с ± Δw) t + φ _к (t) + φ ₂ ],

u ₃ (t) = υ _c · cos [(w _с ± Δw) t + φ _к (t) + φ ₃ ],

u ₄ (t) = υ _c · cos [(w _с ± Δw) t + φ _к (t) + φ ₄ ],

u ₅ (t) = υ _c · cos [(w _with ± Δw) t + φ _к (t) + φ ₅ ], 0≤t≤T _c ,

where υ _c , w _s , T _s , φ ₁ -φ ₅ is the amplitude, carrier frequency, duration and initial phases of the signals;

± Δw is the instability of the carrier frequency due to the Doppler effect and other destabilizing factors;

φ _к (t) = {0, π} is the manipulated component of the phase that displays the law of phase manipulation in accordance with the modulating code M (t), and φ _к (t) = const for kτ _e <t <(k + 1) τ _e and can change abruptly at t = kτ _e , i.e. at the boundaries between elementary premises (k = 1, 2, ..., N-1);

τ _e , N is the duration and number of chips that make up a signal of duration T _s (T _s = N · τ _e ), from the outputs of the receiving antennas 4-8 directly and through adders 67, 70 and 73, for which only one shoulder, are fed to the first inputs of the mixers 25-28, 24 and 75, the second inputs of which are supplied with the voltage of the first local oscillator 30 of a ramp frequency:

_G1 u (t) = υ _G1 · cos (w ² + πγt _G1 + φ _r1)

_G1 u '(t) = υ _G1 · cos (w ² + πγt _G1 + φ _r1 + 90 °), 0≤t≤T _P,

where υ _Г1 , w _Г1 , φ _Г1 - amplitude, initial frequency and initial phase of the local oscillator voltage 30;

local oscillator frequency change rate (tuning rate):

;

;

T _P - the period of perestroika.

At the outputs of the mixers 24, 75, 25-28, voltages of combination frequencies are generated. Amplifiers 31, 76, 32-35 distinguish the voltage of the first intermediate (difference) frequency:

u _up1 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up1 ],

u _up2 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up1 -90 °],

u _up3 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up2 ],

u _up4 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up3 ],

u _up5 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up4 ],

u _up6 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up5 ], 0≤t≤T _c ,

К₁·υ_с·υ_Г1;where υ _pr =

K ₁ · υ _s · υ _G1 ;

w _up1 = w _c -w _G1 is the first intermediate frequency;

φ _up1 = φ ₁ -φ _G1 ; φ _up2 = φ ₂ -φ _G1 ;

φ _up3 = φ ₃ -φ _G1 ; φ _up4 = φ ₄ -φ _G1 ;

φ _up5 = φ ₅ -φ _G1 ;

which are complex signals with combined phase shift keying and linear frequency modulation (FMN-LUM) at an intermediate frequency.

Voltages u _up3 (t) -u _up6 (t) are supplied to the first inputs of the multipliers 52-55.

The voltage u _up2 (t) with the output of the amplifier 76 of the first intermediate frequency is supplied to the input of the phase shifter 77 by 90 °, at the output of which a voltage is generated

u _up7 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up1 + 90 ° -90 °] =

υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up1 ].

Voltages u _up2 (t) and u _up7 (t) are supplied to two inputs of adder 78, at the output of which the total voltage is formed

u _Σ1 (t) = υ _Σ1 · cos [(w _up1 ± Δw) t + φ _к (t) -πγt ² + φ _up1 ], 0≤t≤T _c ,

where υ _Σ1 = 2υ _pr ,

which is fed to the first input of the multiplier 79, to the second input of which the received PSK signal u ₁ (t) is supplied from the output of the adder 73. A harmonic voltage is generated at the output of the multiplier 79

₆ u (t) = υ ₆ · cos (w t + πγt _r1 ² + φ _r1], 0≤t≤T _c,

;Where

;

K ₂ - transfer coefficient of the multiplier.

The tuning frequency w _{H1 of the} bandpass filter 65 is chosen equal to

.

.

The tuning frequency w _{H2 of the} bandpass filter 68 is chosen equal to

.

.

The tuning frequency w _{H3 of the} band-pass filter 71 is chosen equal to

.

.

The tuning frequency w _{H4 of the} bandpass filter 80 is chosen equal to

w _H4 = w _G1 .

The harmonic voltage u ₆ (t) is extracted by a band-pass filter 80, detected by an amplitude detector 81 and supplied to the control input of the key 82, opening it. In the initial state, keys 43 and 82 are always closed. The voltage u _Σ1 (t) from the output of the adder 78 through the public key 82 is supplied to the input of the detector 36, consisting of a phase doubler 37, the first 38 and second 39 spectral width meters, a comparison unit 40, a threshold block 41, a delay line 42, and a key 43.

At the output of the phase doubler 37, which can be used as a multiplier, the two inputs of which receive the same signal, a voltage is generated

u ₇ (t) = υ ₇ · cos [2 (w _up1 ± Δw) t-2πγt ² + φ _up1 ], 0≤t≤T _c ,

Where

Since 2φ _K (t) = {0.2π}, phase manipulation is already absent in the indicated voltage.

The width of the spectrum Δƒ _{2 of the} second harmonic of the signal is determined by the signal duration T _c (Δƒ ₂ = 1 / T _c ), while the width of the spectrum of the QPSK signal is determined by the duration τ _{e of} its elementary premises (Δƒ _c = 1 / τ _e ), i.e. the spectrum width of the second harmonic of the signal is N times smaller than the spectrum width of the input signal:

Therefore, when the phase of the QPSK signal is doubled, its spectrum collapses N times. This circumstance makes it possible to detect the QPSK signal even when its power at the input of the receiving device 11 is less than the power of noise and interference.

The width of the spectrum Δƒ _{from the} input signal is measured using a meter 38, and the width of the spectrum Δƒ _{2 of} its second harmonic is measured using a meter 39. Voltages υ ₁ and υ ₂ proportional to Δƒ _s and Δƒ ₂ , from the outputs of the meters 38 and 39 the width of the spectrum of the signals on two inputs of block 40 comparison. Since υ ₁ ≫υ ₂ , a positive voltage is generated at the output of the comparison unit 40, which exceeds the threshold level υ _then in the threshold unit 41. The latter is chosen so that it is exceeded by random noise. The threshold level υ _{then is} exceeded only when the detection of the PSK signal. When the threshold voltage υ _pores is exceeded, a constant voltage is generated in the threshold block 41, which is fed to the input of the delay line 42, to the control input of the key 43, opening it, and to the control input of the search unit 29, putting it into stop mode.

From this point in time, viewing the specified frequency range Дƒ and searching for PSK signals stops for the processing time of the detected PSK signal, which is determined by the delay time τ _{s of} the delay line 42.

Upon termination of the tuning of the local oscillator 30 amplifiers 34, 76, 32-35 are allocated the following voltage:

u _up8 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up1 ],

u _up9 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up1 -90 °],

u _up10 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up2 ],

u _up11 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up3 ],

u _up12 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up4 ],

u _up13 (t) = υ _pr · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up5 ], 0≤t≤T _c .

The output of the adder 78 in this case, the total voltage

u _Σ2 (t) = υ _Σ1 · cos [(w _up1 ± Δw) t + φ _к (t) + φ _up1 ], 0≤t≤T _c ,

which through the public key 43 enters the input of the mixer 45, the second input of which receives the voltage of the second local oscillator 44

u _Г2 (t) = u _Г2 · cos (w _Г2 t + φ _Г2 )

with a stable frequency w _G2 .

At the input of the mixer 45, voltages of combination frequencies are generated. The amplifier 46 is allocated the voltage of the second intermediate frequency (Fig.6, b)

u _up14 (t) = υ _pr1 · cos [(w _up2 ± Δw) t + φ _к (t) + φ _up6 ], 0≤t≤T _c ,

Where

w _up2 = w _up1 -w _G2 - second intermediate frequency;

φ _up6 = φ _up1 -φ _G2 .

This voltage is supplied to the input of the demodulator 47 QPSK signal, consisting of the first 48 and second 49 multipliers, a narrow-band filter 50 and a low-pass filter 51.

The second input of the multiplier 49 is fed from the output of the narrow-band filter 50 reference voltage (Fig.6, c)

u ₀ (t) = υ ₀ · cos [(w _up2 ± Δw) t + φ _up6 ].

As a result of multiplication, the resulting voltage is formed

u _Σ (t) = υ _Σ · cosφ _to (t) + υ _Σ · cos [2 (w _up2 ± Δw) t + φ _to (t) + 2φ _up6 ],

Where

Analog modulating code (Fig.6, g)

u _H (t) = υ _Σ · cosφ _K (t)

is allocated by the low-pass filter 51 and is fed to the first input of the adder 64 and to the second input of the multiplier 48. A harmonic voltage is generated at the output of the latter

u ₀ (t) = υ ₀ · cos [(w _up2 ± Δw) t + φ _up6 ], 0≤t≤T _c ,

Where

The operation of the phase demodulator 47 described above corresponds to a stationary (stable) mode of operation. It is preceded by a transient mode of operation at the moment the device is turned on, when a harmonic voltage u ₆ (t) appears in the passband of the narrow-band filter 50. This is due to transients, accompanied by the appearance of a large number of spectral components, among which there will also be harmonic oscillation with a frequency (w _up2 ± Δw).

The phase demodulator 47 is free from the phenomenon of "reverse work".

The voltage u _up14 (t) from the output of the amplifier 46 of the second intermediate frequency is simultaneously supplied to the second inputs of the multipliers 52-55 direction finding channels. The outputs of the multipliers 52-55 form the following harmonic voltages:

u ₈ (t) = υ ₈ · cos (w _Г2 t + φ _Г2 + Δφ ₁ ),

u ₉ (t) = υ ₈ · cos (w _Г2 t + φ _Г2 -Δφ ₂ ),

u ₁₀ (t) = υ ₈ cos (w _Г2 t + φ _Г2 + Δφ ₃ ),

u ₁₁ (t) = υ ₈ · cos (w _Г2 t + φ _Г2 -Δφ ₄ ), 0≤t≤T _c ,

Where

α, β are the angular coordinates of ARB 2 (azimuth and elevation),

which are allocated by narrow-band filters 56-59 and fed to the first inputs of phase detectors 60-63, respectively. The second inputs of the phase detectors 60 and 62 are supplied with voltage u _Г2 (t) of the second local oscillator 44, and the second inputs of the phase detectors 61 and 63 are supplied with harmonic voltages u ₈ (t) and u ₁₀ (t) from the outputs of the narrow-band filters 60 and 62, respectively.

The signs "+" and "-" before the phase shifts correspond to diametrically opposite positions of the receiving antennas 5 and 6, 7 and 8 relative to the receiving antenna 4.

At the outputs of the phase detectors 60-63, constant voltages are formed:

u _H1 (α) = υ _H1 cosΔφ ₁ ,

u _H2 (α) = υ _H2 cosΔφ ₅ ,

u _H3 (β) = υ _H1 cosΔφ ₃ ,

u _H4 (β) = υ _H2 cosΔφ ₆ ,

Where

K ₃ - transfer coefficient of phase detectors;

which are supplied to the respective inputs of the adder 64. The output of the adder 64 is the output of the third receiving device 11.

Knowing the satellite altitude h and measuring the azimuth α and elevation angle β with high accuracy and unambiguity, it is possible to determine the location of the emergency beacon and, consequently, the ship or plane that crashed. The analogue of the modulating code u _H (t) (Fig.6d), extracted from the received FMN signal, contains all the necessary information about the injured object (object type, object class, country, airline, company, owner, etc. )

Receiving antennas 4-8 are arranged so that the measuring bases form a symmetrical geometric cross, at the intersection of which is placed the receiving antenna 4 of the measuring channel, common to antennas 5-8 direction finding channels located in the azimuth and elevation planes (Fig. 5). Moreover, in each plane, smaller bases of d form coarse, but unambiguous direction finding scales, and large bases of 2d form accurate, but ambiguous direction finding scales, between which the following inequality holds:

.

.

So, it is supposed to use the phase method of direction finding of emergency beacons using five receiving antennas arranged in the form of a symmetrical geometric cross. In this case, the receiving antennas 4-8 are located on special panels similar to solar panels, which, after the spacecraft is put into orbit, open and are located towards the Earth's surface (Fig. 5).

The delay time τ _{s of} the delay line 42 is selected so that it is possible to process the detected QPSK signal. After this time, the voltage from the output of the delay line 42 is supplied to the reset input of the threshold block 41 and resets it to the initial (zero) state. In this case, the search unit 29 is transferred to the tuning mode, and the key 43 is closed, i.e. is reset.

If the next QPSK signal is detected at a different carrier frequency emitted by the emergency beacon of another object that has crashed, the system works in a similar way.

All information received on board the satellite 3 from emergency beacons is included in the format of the digital message transmitted to PPI 16. The generated digital message is transmitted at a speed of 2400 bps in real time after preprocessing and is simultaneously recorded in storage devices 12 and 13. Transmission information from storage devices 12 and 13 is produced in the same format and at the same speed as in real time, as a result of which PPI 16 receives stored on-board storage devices Islands 12 and 13 posts ARB 2, accumulated during a full revolution around the Earth's satellite.

If at the time of the transmission of information from the memorable devices 12 and 13 to the input of the receiving devices 10 or 11 of the satellite receives a signal from the ARB 2, the transmission is interrupted to process the signal, information about which after processing is included in the message format for transmission to PPI 16.

The corresponding binary number is included in the message, indicating the type of transmission: the real time scale or from the storage devices, in addition, the transmission time of the last message from the storage devices is identified.

Information from the receiving devices 9, 10, 11 and memory devices 12, 13 is fed to the input of the on-board transmitter 14. The radiation power of the transmitter 14 can be adjusted from the ground-based system control complex. The transmitter 14 also uses phase shift keying of the oscillation of the carrier frequency by a composite code in the stages of its formation (before multiplication). Then the oscillation is transferred to a frequency of 1544.5 MHz, amplified to the required level and radiated by the antenna 15 in the direction of PPI 16.

At PPI 16, the received signal from the output of the receiving device 18 is fed to information processing devices 19 and 20. Moreover, the device 19 provides the processing of information from the ARB 1, and the device 20 provides the processing of information from the ARB 2. The processed information is coupled to communication networks, by which the necessary information is communicated to search and rescue organizations.

The system uses complex signals with phase shift keying as alarm signals.

From the point of view of detection, complex signals with phase shift keying have energetic and structural secrecy.

The energy secrecy of these signals is due to their high compressibility in time and spectrum with optimal processing, which reduces the instantaneous radiated power. As a result, a complex QPSK signal at the receiving point may be masked by noise and interference. Moreover, the energy of a complex QPSK signal is by no means small; it is simply distributed over the time-frequency domain so that at each point of this region the signal power is less than the power of noise and interference.

The structural secrecy of complex QPSK signals is caused by a wide variety of their forms and significant ranges of parameter changes, which makes it difficult to optimize or at least quasi-optimal processing of QPSK signals of an a priori unknown structure in order to increase the sensitivity of receiving devices.

In addition, complex QPSK signals allow the use of a new type of selection - structural selection. This means that there is a new opportunity to separate signals operating in the same frequency band and at the same time intervals. Fundamentally, you can abandon the traditional method of dividing the operating frequencies of the used range between operating emergency beacons and selecting them on board an artificial satellite using frequency filters. It can be replaced by a new method based on the simultaneous operation of each ARB in the entire frequency range by phase-shift signals with the allocation by the receiving devices 10 and 11 of the signal of the required ARB through its structural selection.

The system is invariant to the type of modulation and instability of the carrier frequency of the received signals, since the direction finding of emergency beacons is carried out at a stable frequency w _{Г2 of the} second local oscillator 44. Therefore, these factors influence phase measurements.

In addition, through the use of panels on which receiving antennas 4-8 are located, the technical implementation of the phase method of direction finding on board an aircraft is greatly simplified.

The above-described operation of the third on-board receiving device 11 of the system corresponds to the case of receiving useful QPSK signals on the main channel at a frequency w _s (FIG. 2).

If a complex signal (interference) is received on the mirror channel at a frequency w _s

u ₃ (t) = υ _z · cos (w _z t + φ _z ), 0≤t≤T _z ,

the amplifiers 31 and 76 of the first intermediate frequency are allocated the following voltages:

u _up15 (t) = υ _pr15 · cos (w _up1 t + φ _up15 ),

u _up16 (t) = υ _pr15 · cos (w _up1 t + φ _up15 + 90 °), 0≤t≤T _s ,

Where

w _up1 = w _G1 -w _s - the first intermediate frequency;

φ _up15 = φ _s -φ _G1.

The voltage u _up16 (t) from the output of the amplifier 76 is supplied to the input of the phase shifter 77 by 90 °, at the output of which a voltage is generated

u _up17 (t) = υ _pr15 · cos (w _up1 t + φ _up15 + 90 ° + 90 °) = - υ _pr15 · cos (w _up1 t + φ _up15 ), 0≤t≤T _s ,

The _voltages u _up15 (t) and u _up17 (t) supplied to the two inputs of the adder 78 are compensated at its output.

Therefore, a false signal (interference) received on the mirror channel at a frequency w _s is suppressed.

For a similar reason, the phase-compensation method also suppresses a complex signal (interference) received via the first combination channel at a frequency w _K1 .

If a false signal (interference) is received on the second combinational channel at a frequency w _K2

u _К2 (t) = υ _к2 · cos (w _К2 t + φ _К2 ), 0≤t≤Т _К2 ,

the amplifiers 31 and 76 of the first intermediate frequency are allocated the following voltages:

u _up18 (t) = υ _pr18 cos (w _up1 t + φ _up18 ),

u _up19 (t) = υ _pr18 · cos (w _up1 t + φ _up18 -90 °), 0≤t≤T _K2 ,

Where

w _up1 = w _K2 -2w _G1 - the first intermediate frequency;

φ _up18 = φ _K2 -φ _G1 .

The voltage u _up19 (t) from the output of the amplifier 76 of the first intermediate frequency is supplied to the input of the phase former 77 by 90 °, at the output of which a voltage is generated

u _up20 (t) = υ _pr18 · cos (w _up1 t + φ _up18 ), 0≤t≤T _K2 ,

Voltages u _up18 (t) and u _up20 (t) are supplied to two inputs of adder 78, at the output of which the total voltage is formed

u _Σз (t) = υ _Σз · cos (w _up1 t + φ _up18 ), 0≤t≤T _K2 ,

where υ _Σз = 2υ _пр18 .

The voltage u _Σз (t) from the output of the adder 78 is supplied to the first input of the multiplier 79, the second input of which from the output of the adder 73 receives the received false signal (interference) u _K2 (t). The output of the latter produces a harmonic voltage

₁₂ u (t) = υ ₁₂ · cos (2w _T1 t + φ _r1) 0≤t≤T _K2

Where

which does not fall into the passband of the bandpass filter 80.

Key 82 does not open and a false signal (interference) received on the second combination channel at a frequency w _K2 is suppressed.

If a false signal (interference) is received on the direct channel at a frequency w _up1

u _up (t) = υ _pr · cos (w _up1 t + φ _up ), 0≤t≤T _up ,

then it is allocated by a band-pass filter 65 and fed to the input of the bass reflex 66, where the next voltage is formed

u _up '(t) = - υ _pr · cos (w _up1 t + φ _up ), 0≤t≤T _up .

The voltages u _up (t) and u _up '(t) supplied to the two inputs of the adder 67 are compensated at its output.

Therefore, a false signal (interference) received on the direct channel at a frequency w _up is suppressed.

The suppression of false intermodulation signals (interference) received in the frequency band Δw _P1 is provided by a filter plug consisting of a bandpass filter 68, a phase inverter 69 and an adder 70 and implementing a phase compensation method. The suppression of false intermodulation signals (interference) received in the frequency band Δw _P2 is provided by a filter plug consisting of a band-pass filter 71, a phase inverter 72 and an adder 73 and implementing a phase compensation method.

The above cases are described when the input of the third receiving device receives false signals (interference) in the form of harmonic oscillations. This is done to simplify recording.

If the input of the third receiving device receives false signals (interference), for example with phase shift keying, then the device works in a similar way.

Thus, the proposed system in comparison with the prototype and other technical solutions for a similar purpose provides increased noise immunity and direction finding accuracy of beacons installed on ships and aircraft that have crashed. This is achieved by suppressing false signals (interference) received through the mirror, Raman and intermodulation channels, as well as through the direct channel. Moreover, to suppress false signals (interference) received through the mirror, the first Raman and intermodulation channels, as well as through the direct channel, the phase-compensation method is used. And to suppress false signals (interference) received on the second combinational channel, the method of narrow-band filtering is used.

Claims (1)

A satellite system for determining the location of ships and aircraft that have crashed, containing emergency beacons on ships and aircraft, for the transmission of alarm information from which complex signals with phase shift keying are used, at the information receiving points, a receiving device with an antenna, a first information processing device, a device are connected in series interfacing with communication networks, the second input of which through the second information processing device is connected to the output of the receiving device, a control device and formations and communication device of search and rescue organizations, on artificial Earth satellites, a second receiver with a second antenna, a first storage device and a transmitter with an antenna, the second input of which is connected to the output of the first receiver with the first antenna and the third input to the output of the second a receiving device, a third receiving device with third, fourth and fifth antennas and a second storage device, the output of which is connected to the fourth input, are connected in series a transmitter, the fifth input of which is connected to the output of the third receiving device, while the first and second antennas are also connected to the third receiving device, the third receiving device contains a measuring and four direction finding channels, the measuring channel consists of a first mixer connected in series, the first input of which is connected to the first antenna, and the second input of which is connected through the first local oscillator to the output of the search unit, and the first amplifier of the first intermediate frequency, doubled in series a phase meter, a second spectral width meter, a comparison unit, the second input of which is connected to the output of the first spectral width meter, a threshold block, the second input of which is connected to its output via a delay line, the first key, the second mixer, the second input of which is connected to the output of the second local oscillator amplifier of the second intermediate frequency, the first multiplier, the second input of which is connected to the output of the low-pass filter, a narrow-band filter, the second multiplier, the second input of which is connected to the output of the amplifier w a swarm of an intermediate frequency, a low-pass filter and a first adder, the input of which is the output of the third receiving device, while the control input of the search unit is connected to the output of the threshold unit, each direction-finding channel consists of a series-connected mixer, the first input of which is connected to the corresponding antenna, and the second the input of which is connected to the output of the first local oscillator, the amplifier of the first intermediate frequency, the multiplier, the second input of which is connected to the output of the amplifier of the second intermediate often s, a narrow-band filter and a phase detector, the output of which is connected to the corresponding input of the first adder, while the second inputs of the phase detectors of the first and third direction-finding channels are connected to the output of the second local oscillator, the second inputs of the phase detectors of the second and fourth direction-finding channels are connected to the outputs of the narrow-band filters of the first and third direction finding channels, respectively, receiving antennas of direction finding channels are placed in the form of a symmetrical geometric cross, at the intersection of which the receiving antenna of the measuring channel is replaced, characterized in that the measuring channel is equipped with adders, four bandpass filters, three phase inverters, a second key, a third multiplier, a second intermediate frequency amplifier, a third mixer, two 90 ° phase shifters and an amplitude detector, and to the output of the first receiving the antennas are connected in series with the first bandpass filter, the first phase inverter, the second adder, the second input of which is connected to the output of the first receiving antenna, the second bandpass filter , a second phase inverter, a third adder, the second input of which is connected to the output of the second adder, a third bandpass filter, a third phase inverter and a fourth adder, the second input of which is connected to the output of the third adder, and the output is connected to the first input of the first mixer, to the second output of the first local oscillator in series connected the first phase shifter 90 °, the third mixer, the second input of which is connected to the output of the fourth adder, the second amplifier of the first intermediate frequency, the second phase shifter 90 °, the fifth adder, the second input of which is connected to the output of the first amplifier of the first intermediate frequency, the third multiplier, the second input of which is connected to the output of the fourth adder, the fourth bandpass filter, the amplitude detector and the second switch, the second input of which is connected to the output of the fifth adder, and the output is connected to the inputs of the phase doubler , the first spectral width meter and the first key.

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