JPWO2018012083A1 - Switching circuit, automatic gain control circuit and phase synchronization circuit - Google Patents

Abstract

In an AGC circuit including a transistor that shifts to an on state or an off state according to the level of a signal, leakage current is reduced while reducing low-pass noise.
The switching circuit comprises a first bipolar transistor, a second bipolar transistor, and a comparing unit. The second bipolar transistor is cascode connected to the first bipolar transistor at a connection point, and a constant bias voltage not lower than the threshold voltage is applied to the base. The comparison unit compares the input signal with a predetermined reference signal, and outputs a pair of differential output signals indicating the comparison result to the base of the first bipolar transistor and the connection point.

Description

  The present technology relates to a switching circuit, an automatic gain control circuit, and a phase synchronization circuit. More particularly, the present invention relates to a switching circuit, an automatic gain control circuit, and a phase synchronization circuit including a transistor that transitions to an on state or an off state according to the level of a signal.

  2. Description of the Related Art Conventionally, in order to maintain the amplitude of a signal constant in an audio device, a communication device, etc., an AGC (Automatic Gain Control) circuit that dynamically controls the gain for the signal is used. The AGC circuit generally includes a detection circuit that outputs a control voltage according to the amplitude of the signal, and a variable gain amplifier that amplifies the signal with a gain according to the control voltage. The detection circuit is provided with, for example, a comparator that compares an input signal and a reference signal, a transistor that turns on and off according to the comparison result, a recovery current source, and an attack current source. The recovery current source and a capacitor that charges and discharges the control current from the attack current source controlled by the comparison result to generate a control voltage are provided (for example, see Patent Document 1). One end of this capacitor is connected to the connection point of the recovery current source and the transistor and the variable gain amplifier.

Unexamined-Japanese-Patent No. 2007-255909

  The time constant Tc of the AGC circuit described above is proportional to the capacitance C of the capacitor and inversely proportional to the control current Icon from the transistor. If the time constant Tc is increased, the high frequency cutoff frequency can be reduced to reduce the AC component in the signal, and the signal quality can be enhanced. In order to increase the time constant Tc while the capacitance C is constant, the control current Icon may be reduced. However, when the control current Icon is reduced, the leak current flowing when the transistor is in the off state becomes relatively large. Then, due to the increase of the leak current, the control voltage for controlling the gain becomes a value different from the design value, and there is a problem that the gain deviates from the expected value.

  Here, if a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) having a leakage current lower than that of a bipolar transistor is used as a transistor, the influence of the leakage current can be alleviated. However, in the MOSFET, low-pass noise (1 / f noise) is larger than that of the bipolar transistor. As described above, in the above-described conventional technology, it is difficult to suppress the leakage current while reducing low-pass noise.

  The present technology has been created in view of such a situation, and in an AGC circuit including a transistor that shifts to an on state or an off state according to the level of a signal, leakage current is reduced while low-pass noise is reduced. The purpose is to

  The present technology has been made to solve the above-mentioned problems, and the first aspect is cascode-connected to a first bipolar transistor and the first bipolar transistor at the connection point to set a threshold voltage. A second bipolar transistor, the base of which is applied a constant bias voltage not falling below, an input signal and a predetermined reference signal, and a pair of differential output signals indicating the comparison result It is a switching circuit provided with the comparison part output to a base and the said connection point. This brings about the effect that the cascode-connected first and second bipolar transistors are controlled by the pair of differential output signals.

  In addition, in the first aspect, the input stage of the comparison unit may include a pair of differential transistors. This brings about the effect that a pair of differential output signals are outputted by a pair of differential transistors.

  In the first aspect, the pair of differential transistors may be bipolar transistors. This brings about the effect | action that a pair of differential output signal is output by a pair of bipolar transistor.

  In the first aspect, the pair of differential transistors may be MOS (Metal Oxide Semiconductor) transistors. This brings about the effect that a pair of differential output signals are outputted by a pair of MOS transistors.

  Further, according to a second aspect of the present technology, there is provided a second bipolar transistor, wherein the first bipolar transistor and the first bipolar transistor are cascode-connected at their connection points, and a constant bias voltage not lower than the threshold voltage is applied to the base. A comparator configured to compare a bipolar transistor, a pair of differential output signals indicating the comparison result by comparing an input signal and a predetermined reference signal, to the base of the first bipolar transistor and the connection point; An automatic gain comprising: a recovery current source for supplying current; a capacitor connected to the recovery current source and the second bipolar transistor; and a variable gain amplifier for changing a gain according to a charge / discharge voltage of the capacitor It is a control circuit. This brings about the effect that the gain changes according to the charge / discharge voltage of the capacitor connected to the cascode-connected bipolar transistor and the recovery current source.

  In the second aspect, the input of the comparison unit may be an input of the variable gain amplifier.

  In the second aspect, the input of the comparison unit may be an output of the variable gain amplifier.

  Further, according to a third aspect of the present technology, there is provided a detector that detects a phase difference and a frequency difference between an input signal and a feedback signal and outputs first and second output signals indicating the detection result, A bipolar transistor, a second bipolar transistor cascode-connected to the first bipolar transistor at a first connection point and having a base applied with a constant bias voltage not lower than a threshold voltage, and the first output signal A first comparison in which a pair of differential output signals indicating the comparison result is compared with a signal obtained by inverting the first output signal to the base of the first bipolar transistor and the first connection point. , A third bipolar transistor, and the third bipolar transistor in cascode connection at a second connection point, and a base of a constant bias voltage not lower than a threshold voltage A pair of differential output signals indicating the comparison result by comparing the applied fourth bipolar transistor, the second output signal, and a signal obtained by inverting the second output signal are the third bipolar transistor. A second comparison unit for outputting to the second base and the second connection point, a capacitor connected to the connection point between the second and fourth bipolar transistors, and a frequency corresponding to the charge / discharge voltage of the capacitor A phase locked loop circuit comprising: a voltage controlled oscillator for outputting a periodic signal; and a frequency divider for dividing the periodic signal and feeding it back to the detector as the feedback signal. This brings about the effect that a periodic signal of a frequency according to the charge / discharge voltage of the capacitor connected to the cascode-connected bipolar transistor is generated.

  According to the present technology, in the AGC circuit including the transistor, it is possible to achieve an excellent effect that the leakage current can be reduced while reducing the low frequency noise. In addition, the effect described here is not necessarily limited, and may be any effect described in the present disclosure.

It is a circuit diagram showing an example of 1 composition of a current switching circuit in a 1st embodiment of this art. It is a block diagram showing an example of 1 composition of an electronic device in a 2nd embodiment of this art. It is a circuit diagram showing an example of 1 composition of an AGC circuit in a 2nd embodiment of this art. It is a circuit diagram showing an example of 1 composition of a peak detection circuit in a 2nd embodiment of this art. It is a figure which shows an example of the state of a switching circuit when the input voltage in the 2nd Embodiment of this technique is higher than a reference voltage. It is a figure which shows an example of the state of a switching circuit in case the input voltage in 2nd Embodiment of this technique is the reference voltage or less. It is a circuit diagram showing an example of 1 composition of an AGC circuit in a comparative example of this art. It is a timing chart which shows an example of change of an input voltage and an output voltage in a 2nd embodiment of this art. It is a timing chart which shows an example of operation of the AGC circuit in a 2nd embodiment of this art. It is a flow chart which shows an example of operation of the AGC circuit in a 2nd embodiment of this art. It is a block diagram showing an example of 1 composition of an AGC circuit in a modification of a 2nd embodiment of this art. It is a block diagram showing an example of 1 composition of an electronic device in a 3rd embodiment of this art. It is a circuit diagram showing an example of 1 composition of a charge pump in a 3rd embodiment of this art. It is a timing chart which shows an example of operation of a charge pump in a 3rd embodiment of this art.

Hereinafter, modes for implementing the present technology (hereinafter, referred to as embodiments) will be described. The description will be made in the following order.
1. First embodiment (example of cascode connection of transistors in a current switching circuit)
2. Second embodiment (example of cascode connection of transistors in AGC circuit)
3. Third embodiment (example of cascode connection of transistors in charge pump)
<1. First embodiment>
[Configuration Example of Current Switching Circuit]

  FIG. 1 is a circuit diagram showing a configuration example of the current switching circuit 240 in the first embodiment. The current switching circuit 240 includes a comparing unit 250 and bipolar transistors 241 and 242. These bipolar transistors 241 and 242 are, for example, npn type.

  A constant bias voltage Vb is applied to the base of the bipolar transistor 241. The bias voltage Vb is set to, for example, a value that does not fall below the threshold voltage of the bipolar transistor 241.

  The collector of the bipolar transistor 241 is connected to the output terminal of the current switching circuit 240. The bipolar transistor 241 is cascode connected to the bipolar transistor 242. The bipolar transistor 242 is an example of the first transistor described in the claims. The bipolar transistor 241 is an example of a second transistor described in the claims.

  The comparison unit 250 compares the input voltage Vin with the reference voltage Vref. The comparison unit outputs a differential output signal indicating the comparison result to the base of the bipolar transistor 242 and the connection point of the bipolar transistors 241 and 242. Assuming that the input terminal of the input voltage Vin is a positive input terminal and the input terminal of the reference voltage Vref is a negative input terminal, the output voltage Vop of the positive differential output signal is output to the base of the bipolar transistor 242. Further, the output voltage Von of the negative differential output signal is output to the connection point of the bipolar transistors 241 and 242.

  In addition, comparison unit 250 includes constant current source 260, switches 270 and 280, and bipolar transistors 251 and 252. These bipolar transistors 251 and 252 are, for example, npn type.

  The constant current source 260 supplies a constant current. The switch 270 transitions to either the conductive state (ie, on state) or the non-conductive state (ie, off state) depending on the value of the input voltage Vin with respect to the reference voltage Vref. For example, when the input voltage Vin is higher than the reference voltage Vref, the switch 270 shifts to the off state. The switch 280 operates exclusively with the switch 270. The switches 270 and 280 are an example of a pair of differential transistors described in the claims.

  The emitter of the bipolar transistor 251 is grounded, and the collector is connected to the switch 270. Also, the collector and the base of the bipolar transistor 251 are shorted (so-called diode connection). The base of the bipolar transistor 251 is also connected to the base of the bipolar transistor 242.

  The emitter of the bipolar transistor 252 is grounded and the collector is connected to the switch 280. The bipolar transistor 252 is also diode-connected, and its base is also connected to the connection point of the bipolar transistors 241 and 242.

  According to the above-described configuration, bipolar transistors 241 and 242 are turned off when input voltage Vin is higher than reference voltage Vref. On the other hand, when the input voltage Vin is lower than or equal to the reference voltage Vref, the bipolar transistors 241 and 242 are turned on. Since the bipolar transistors 241 and 242 are cascode-connected, the output resistance seen from the output terminal of the current switching circuit 240 is higher than that of the comparative example when switching with only one bipolar transistor. Thereby, the leak current can be reduced. Further, since switching is performed by the bipolar transistors 241 and 242, 1 / f noise can be reduced as compared with the case of using a MOS transistor.

  As described above, according to the first embodiment of the present technology, since the two bipolar transistors turned on and off by the comparison unit 250 are cascode-connected, the output resistance becomes higher than the case where only one transistor is provided. . Thereby, the leak current of those transistors can be reduced.

<2. Second embodiment>
[Configuration Example of Electronic Device]
FIG. 2 is a block diagram showing one configuration example of the electronic device 100 in the second embodiment. The electronic device 100 is a device that processes an analog signal, and includes an analog signal output unit 110, an AGC circuit 200, and a signal processing unit 120. As the electronic device 100, for example, a wireless communication device, an audio device, a video reproduction device, etc. are assumed.

  The analog signal output unit 110 outputs an analog signal of the input voltage Vin to the AGC circuit 200 via the signal line 119. As the analog signal output unit 110, for example, an antenna or a sensor (such as an image sensor or a microphone) in a wireless communication apparatus is assumed.

  The AGC circuit 200 increases or decreases the input voltage Vin of the signal according to the gain corresponding to the amplitude of the analog signal. The AGC circuit 200 supplies the amplitude-adjusted voltage as an output voltage Vout to the signal processing unit 120 via the signal line 209. The AGC circuit 200 is an example of the automatic gain control circuit described in the claims.

  The signal processing unit 120 performs predetermined processing on an analog signal of the output voltage Vout. The signal processing unit 120 performs, for example, AD conversion processing, decoding processing for decoding a wireless signal, and the like.

[Configuration example of AGC circuit]
FIG. 3 is a circuit diagram showing a configuration example of the AGC circuit 200 in the second embodiment. The AGC circuit 200 includes a variable gain amplifier 210 and a peak detection circuit 220. The peak detection circuit 220 further includes a reference voltage supply unit 221, a capacitor 222, a bias voltage supply unit 230, and a current switching circuit 240. The configuration of the current switching circuit 240 is the same as that of the first embodiment except that a recovery current source 290 for supplying a constant current is added. The recovery current source 290 is connected to the bipolar transistor 241 and the output terminal.

  The reference voltage supply source 221 supplies the lower limit threshold level of the input voltage signal Vin to the comparison unit 250 in order to obtain a desired amplitude of the output signal Vout. The bias voltage supply unit 230 supplies a constant bias voltage Vb to the current switching circuit 240.

  One end of the capacitor 222 is connected to the output terminal of the current switching circuit 240 and the variable gain amplifier 210. The capacitor 222 charges and discharges the control current Icon from the output terminal of the current switching circuit 240 to generate a control voltage Vc.

  The variable gain amplifier 210 changes the gain according to the control voltage Vc. For example, as the control voltage Vc is higher, the gain is larger. The variable gain amplifier 210 increases or decreases the input voltage Vin with a gain corresponding to the control voltage Vc, and outputs the increased or decreased voltage to the signal processing unit 120 as an output voltage Vout.

[Configuration example of peak detection circuit]
FIG. 4 is a circuit diagram showing a configuration example of the peak detection circuit 220 in the second embodiment. In peak detection circuit 220, constant current source 260 includes, for example, bipolar transistors 261 and 262. Further, as the switch 270, a bipolar transistor 271 is used, and as the switch 280, a bipolar transistor 281 is used. Further, the bias voltage supply unit 230 includes bipolar transistors 231 and 233 and a resistor 232. The recovery current source 290 includes a MOS (Metal Oxide Semiconductor) transistor 291.

  The bipolar transistors 231, 261, 262, 271 and 281 are, for example, pnp type, and the bipolar transistor 233 is, for example, npn type. The MOS transistor 291 is, for example, P-type.

  The emitter of bipolar transistor 261 is connected to the power supply, and the base is connected to the bases of bipolar transistors 262 and 231. The bipolar transistor 261 is diode-connected, and its collector is connected to a constant current source (not shown) that supplies a reference current Iref.

  The emitter of bipolar transistor 262 is connected to the power supply, and the collector is connected to the emitters of bipolar transistors 271 and 281. The reference current Iref is replicated by such a current mirror circuit.

  The input voltage Vin is input to the base of the bipolar transistor 271, and the collector is connected to the bipolar transistor 251. The reference voltage Vref is input to the base of the bipolar transistor 281, and the collector is connected to the bipolar transistor 252.

  The emitter of the bipolar transistor 231 is connected to the power supply, and the collector is connected to the resistor 232 and the base of the bipolar transistor 241. The bipolar transistor 233 is diode-connected and inserted between the resistor 232 and the ground terminal. The bipolar transistor 231 generates a current that is a duplicate of the reference current Iref, and the current and the resistor 232 and the bipolar transistor 233 generate a bias voltage Vb.

  The MOS transistor 291 is diode-connected, and its drain is connected to the bipolar transistor 241 and the capacitor 222.

  As described above, the comparator 250 generates the output voltages Vop and Von using the current obtained by duplicating the reference current Iref. Similarly, the bias voltage supply unit 230 generates a bias voltage Vb using a current obtained by duplicating the reference current Iref. Therefore, even if the reference current Iref varies due to conditions (PVT conditions) such as process and voltage or temperature, the bias voltage Vb and the output voltages Vop and Von increase and decrease to the same degree according to the current.

  FIG. 5 is a diagram showing an example of the state of the current switching circuit 240 when the input voltage Vin is higher than the reference voltage Vref in the second embodiment. The current flowing to switch 270 and bipolar transistor 251 decreases, and the current flowing to switch 280 and bipolar transistor 252 increases. As a result, the base potential (Vop) of the bipolar transistor 242 is lowered, and the emitter potential (Von) of the bipolar transistor 241 is raised.

  The lowering of the base potential shifts the bipolar transistor 242 to the off state. Further, since the bias voltage Vb is applied to the base of the bipolar transistor 241, the base-emitter voltage of the bipolar transistor 241 is less than the threshold voltage. Therefore, the bipolar transistor 241 also shifts to the off state.

  Since the bipolar transistors 241 and 242 are off, the current Ir of the recovery current source 290 is output almost as it is as the control current Icon, and the capacitor 222 is charged and discharged by the current. As a result, the control voltage Vc rises with the passage of time.

  FIG. 6 is a diagram showing an example of the state of the current switching circuit 240 when the input voltage Vin is lower than or equal to the reference voltage Vref in the second embodiment. The current flowing to switch 270 and bipolar transistor 251 increases, and the current flowing to switch 280 and bipolar transistor 252 decreases. As a result, the base potential (Vop) of the bipolar transistor 242 is increased, and the emitter potential (Von) of the bipolar transistor 241 is decreased.

  The rise of the base potential shifts the bipolar transistor 242 to the on state. Further, since the bias voltage Vb is applied to the base of the bipolar transistor 241, the base-emitter voltage of the bipolar transistor 241 becomes equal to or higher than the threshold voltage. Therefore, the bipolar transistor 241 also shifts to the on state.

  The accumulated charge of the capacitor 222 is extracted by the attack current Ia flowing to the bipolar transistors 241 and 242 in the on state, and the control voltage Vc decreases with the passage of time.

FIG. 7 is a circuit diagram showing one configuration example of the AGC circuit in the comparative example. In this comparative example, only one transistor that is turned on and off by the output of the comparison unit is provided. Here, the time constant Tb of the AGC circuit is generally expressed by the following equation, where the control current from the transistor is Icon and the capacitance of the capacitor is C.
Tc = C / Icon

  In the comparative example, in order to increase the time constant Tc, the control current Icon may be decreased or the capacitance C may be increased according to the above equation. When the capacitance C is increased, the size of the capacitor is increased, and the layout area of the circuit is increased. In order to increase the time constant Tc without increasing the capacitance C, the control current Icon may be reduced. However, when the control current Icon is reduced, the leak current flowing when the transistor is in the off state may be relatively increased. Then, due to the increase of the leak current, the attack current Ia for drawing the current from the capacitor becomes smaller than the desired value, and the operating point of the AGC circuit becomes a value different from the design value.

  On the other hand, in the current switching circuit 240, the cascode-connected bipolar transistors 241 and 242 are provided at the output stage of the comparing unit 250. The output resistance seen from the output terminal of the current switching circuit 240 is higher than that of the comparative example when switching with only one bipolar transistor. Thereby, the leak current can be made smaller than that of the comparative example. Further, since switching is performed by the bipolar transistors 241 and 242, 1 / f noise can be reduced as compared with the case of using a MOS transistor.

  The transistors in the AGC circuit 200 are preferably bipolar transistors from the viewpoint of reducing 1 / f noise, but some of them may be MOS transistors.

[Operation example of AGC circuit]
FIG. 8 is a timing chart showing an example of fluctuations of the input voltage Vin and the output voltage Vout in the second embodiment. As illustrated in the figure, it is assumed that the amplitude of the input voltage Vin is reduced from time T1 to time T2. In this case, the AGC circuit 200 increases or decreases the input voltage Vin with a relatively large gain and outputs it as the output voltage Vout. Then, it is assumed that the amplitude of the input voltage Vin is increased after time T2. In this case, the AGC circuit 200 increases or decreases the input voltage Vin with a relatively small gain and outputs it as the output voltage Vout. Such control can make the amplitude of the output voltage Vout constant.

  FIG. 9 is a timing chart showing an example of the operation of the AGC circuit 200 in the second embodiment. In the period from time t0 to time t1, the input voltage Vin is higher than the reference voltage Vref. In this period, the AGC circuit 200 charges and discharges the capacitor 222 by the current Ir from the recovery current source, and the control voltage Vc rises with the passage of time.

  In the period from time t1 to time t2, the input voltage Vin is lower than or equal to the reference voltage Vref. In this period, the AGC circuit 200 discharges the capacitor 222 by the attack current Ia, and the control voltage Vc decreases with the lapse of time.

  Further, in the period from time t2 to time t3, the input voltage Vin becomes higher than the reference voltage Vref, and the control voltage Vc rises with the passage of time.

Assuming that the period from time t0 to t3 is a cycle P of the input voltage Vin and the period from time t1 to t2 is P1, the following equation is established by the repetition of the above-described operation.
Ir × P = Ia × P1

  A control voltage Vc according to the integral value of the current represented by the right side or the left side of the above equation is generated by the capacitor 222. If the leak current can be reduced, the control current Icon can be reduced, and a large time constant Tc can be realized even with a small capacitance C.

  FIG. 10 is a flow chart showing an example of the operation of the AGC circuit 200 in the second embodiment. This operation starts, for example, when an analog signal is input to the AGC circuit 200.

  The AGC circuit 200 determines whether the input voltage Vin is higher than the reference voltage Vref (step S901). If the input voltage Vin is less than or equal to the reference voltage Vref (step S901: No), the AGC circuit 200 gradually reduces the control voltage Vc (step S902). The gain decreases as the control voltage Vc decreases.

  On the other hand, when the input voltage Vin is higher than the reference voltage Vref (step S901: Yes), the AGC circuit 200 gradually raises the control voltage Vc (step S903). The gain increases as the control voltage Vc rises. After step S902 or S903, the AGC circuit 200 increases or decreases the input voltage Vin by the gain (step S904).

  As described above, according to the second embodiment of the present technology, since the two bipolar transistors turned on and off by the comparison unit 250 are cascode-connected, the output resistance becomes higher than the case where only one transistor is provided. . Thereby, the leak current of those transistors can be reduced.

[Modification]
In the second embodiment described above, the peak detection circuit 220 performs feedforward control that generates the control voltage Vc from the input voltage Vin before the increase or decrease, but can be realized by feedback control.

  FIG. 11 is a block diagram showing a configuration example of an AGC circuit 200 in a modification of the second embodiment. In the AGC circuit 200, the variable gain amplifier 210 inputs the output voltage Vout to the peak detection circuit 220 (in other words, feedback). Then, the peak detection circuit 220 compares the output voltage Vout with the reference voltage Vref to generate the control voltage Vc.

  As described above, according to the modification of the second embodiment of the present technology, since the variable gain amplifier 210 feeds back the output voltage Vout to the peak detection circuit 220, feedback control can be performed.

<3. Third embodiment>
In the first embodiment described above, cascode-connected transistors are provided in the AGC circuit 200, but in the case of a circuit that performs switching operation, cascode-connected transistors are provided in circuits other than the AGC circuit 200, such as charge pumps. It can also be done. The electronic device according to the second embodiment is different from the first embodiment in that cascode-connected transistors are provided in the charge pump.

  FIG. 12 is a block diagram showing one configuration example of the electronic device 101 in the third embodiment. The electronic device 101 includes a phase synchronization circuit 130 and a signal processing unit 140.

  The phase synchronization circuit 130 is used to stabilize the oscillation frequency of the voltage control oscillator using the external clock CLKin with low phase noise as a reference signal. The clock signal CLKin is generated by a crystal oscillator or the like. The phase synchronization circuit 130 includes a phase / frequency detector 131, a charge pump 300, a voltage control oscillator 132, and a frequency divider 133.

  The phase / frequency detector 131 detects the phase difference and frequency difference between the clock signal CLKin and the clock signal CLKfb fed back from the frequency divider 133. The phase / frequency detector 131 supplies the charge pump 300 with output signals Voa and Vob indicating the detection result. The phase / frequency detector 131 is an example of a detector described in the claims.

  The charge pump 300 generates the control voltage Vc according to the phase difference and the frequency difference. The charge pump 300 supplies the control voltage Vc to the voltage control oscillator 132.

  The voltage control oscillator 132 generates a signal of a frequency corresponding to the control voltage Vc as the periodic signal VCOout. The voltage control oscillator 132 supplies the periodic signal VCOout to the signal processing unit 140 and the frequency divider 133.

  The divider 133 divides the periodic signal VCOout at a predetermined dividing ratio. The divider 133 feeds the divided signal back to the phase / frequency detector 131 as a clock signal CLKfb.

  The signal processing unit 140 executes predetermined signal processing in synchronization with the periodic signal VCOout.

  FIG. 13 is a circuit diagram showing a configuration example of charge pump 300 in the third embodiment. Charge pump 300 includes inverters 310 and 311, constant current sources 321 and 335, bipolar transistors 322 to 325, and bipolar transistors 331 to 334. Charge pump 300 also includes bipolar transistors 341 to 344 and a capacitor 350.

  The bipolar transistors 322, 324, 331, 333, 341, 342 are, for example, pnp type, and the bipolar transistors 323, 325, 332, 334, 343, and 344 are, for example, npn type.

  The emitters of the bipolar transistors 322 and 324 are connected to a constant current source 321. The base of the bipolar transistor 322 is connected to the output terminal of the inverter 310, and the output signal Vob is input to the base of the bipolar transistor 324. The collector of bipolar transistor 322 is connected to the collector of bipolar transistor 323, and the collector of bipolar transistor 324 is connected to the collector of bipolar transistor 325. The inverter 310 inverts the output signal Vob.

  Also, the collector of the bipolar transistor 323 is connected to the base of the bipolar transistor 323 itself and the base of the bipolar transistor 344. The collector of bipolar transistor 325 is connected to the base of bipolar transistor 325 itself and the connection point of bipolar transistors 343 and 344. The emitters of bipolar transistors 323 and 325 are grounded.

  The emitters of bipolar transistors 332 and 334 are connected to a constant current source 335. The base of the bipolar transistor 332 is connected to the output terminal of the inverter 311, and the output signal Voa is input to the base of the bipolar transistor 334. The collector of the bipolar transistor 332 is connected to the collector of the bipolar transistor 331, and the collector of the bipolar transistor 334 is connected to the collector of the bipolar transistor 333. The inverter 311 inverts the output signal Voa.

  The collector of the bipolar transistor 333 is connected to the base of the bipolar transistor 333 itself and the base of the bipolar transistor 341. The collector of bipolar transistor 331 is connected to the base of bipolar transistor 331 itself and the connection point of bipolar transistors 341 and 342. The emitters of bipolar transistors 331 and 333 are connected to the power supply.

  Bipolar transistors 341 and 342 are cascode connected, and bipolar transistors 343 and 344 are also cascoded. The emitter of the bipolar transistor 341 is connected to the power supply, and the emitter of the bipolar transistor 344 is grounded. The collector of bipolar transistor 342 is connected to the collector of bipolar transistor 343. The bias voltage Vb1 is applied to the base of the bipolar transistor 342, and the bias voltage Vb2 is applied to the base of the bipolar transistor 343. A circuit (not shown) for supplying these bias voltages is disposed in charge pump 300. One end of capacitor 350 is connected to the connection point of bipolar transistors 342 and 343, and the output terminal of charge pump 300, and the other end is grounded.

  In the configuration described above, the circuit formed of constant current source 321, bipolar transistors 322 to 325, and bipolar transistors 343 and 344 has the same function as that of current switching circuit 240 in FIG. Also, the circuit including the constant current source 335, the bipolar transistors 331 to 334, and the bipolar transistors 341 and 342 also has the same function as the current switching circuit 240 of FIG. Each of these current switching circuits is turned on and off according to the values of output signals Voa and Vob. The bipolar transistors 341 to 344 are examples of the first to fourth transistors described in the claims.

Assuming that the current flowing in bipolar transistors 341 and 342 is Id1 and the current flowing in bipolar transistors 343 and 344 is Id2, a control current Icon represented by the following equation is output to capacitor 222.
Icon = Id1-Id2

  The capacitor 222 is charged and discharged by the control current Icon, and the charge and discharge voltage is output as the control voltage Vc.

  FIG. 14 is a timing chart showing an example of the operation of the charge pump 300 in the third embodiment. At timing T10, the output signal Voa shifts from low level to high level, and at timing T11 thereafter, the output signal Vob shifts from low level to high level. Further, at the subsequent timing T12, the output signals Voa and Vob are assumed to be shifted to the low level.

  In this case, bipolar switches 341 and 342 are turned on between timing T10 and timing T12, and current Id1 increases. On the other hand, bipolar switches 343 and 344 are turned on between timing T11 and timing T12, and current Id2 increases.

  Here, since the bipolar transistors 341 and 342 are cascode-connected, the leakage current is reduced while they are in the off state, and the current Id1 becomes substantially zero. Similarly, since bipolar transistors 343 and 344 are also cascode-connected, the leakage current is reduced while they are off, and current Id2 becomes substantially zero. On the other hand, in the comparative example which is not cascode-connected, a leak current is generated. The dotted lines in FIG. 14 are the current and voltage of the comparative example.

  Further, the control current Icon of the difference between the currents Id1 and Id2 increases from the timing T10 to the timing T11. In the comparative example, the amount of increase in the control current Icon decreases due to the occurrence of the leak current.

  The capacitor 350 is charged by the control current Icon that flows from the timing T10 to the timing T11, and the control voltage Vc rises. In the comparative example, the amount of increase in the control voltage Vc becomes small due to the generation of the leak current.

  As described above, according to the third embodiment of the present technology, since the cascode-connected transistor is provided in the charge pump 300, the leak current of the charge pump 300 can be reduced.

  Note that the above-described embodiment shows an example for embodying the present technology, and the matters in the embodiment and the invention-specifying matters in the claims have correspondence relationships. Similarly, the invention specific matter in the claims and the matter in the embodiment of the present technology with the same name as this have a correspondence relation, respectively. However, the present technology is not limited to the embodiments, and can be embodied by variously modifying the embodiments without departing from the scope of the present technology.

  In addition, the effect described here is not necessarily limited, and may be any effect described in the present disclosure.

The present technology can also be configured as follows.
(1) a first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a junction and having a constant bias voltage applied to the base that does not fall below a threshold voltage;
A switching circuit comprising: a comparison unit that compares an input signal and a predetermined reference signal and outputs a pair of differential output signals indicating the comparison result to the base of the first bipolar transistor and the connection point.
(2) The switching circuit according to (1), wherein the input stage of the comparison unit includes a pair of differential transistors.
(3) The switching circuit according to (2), wherein the pair of differential transistors are bipolar transistors.
(4) The switching circuit according to (2), wherein the pair of differential transistors are MOS (Metal Oxide Semiconductor) transistors.
(5) a first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a junction and having a constant bias voltage applied to the base that does not fall below a threshold voltage;
A comparator comparing the input signal with a predetermined reference signal and outputting a pair of differential output signals indicating the comparison result to the base of the first bipolar transistor and the connection point;
A recovery current source for supplying a constant current;
A capacitor connected to the recovery current source and the second bipolar transistor;
An automatic gain control circuit comprising: a variable gain amplifier that changes a gain according to a charge / discharge voltage of the capacitor.
(6) The automatic gain control circuit according to (5), wherein the input of the comparison unit is an input of the variable gain amplifier.
(7) The automatic gain control circuit according to (5), wherein the input of the comparison unit is an output of the variable gain amplifier.
(8) A detector for detecting a phase difference and a frequency difference between an input signal and a feedback signal and outputting first and second output signals indicating the detection result,
A first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a first connection point and having a constant bias voltage applied to the base that is not lower than a threshold voltage;
A pair of differential output signals indicating the comparison result by comparing the first output signal with a signal obtained by inverting the first output signal, the base of the first bipolar transistor, and the first connection point A first comparison unit that outputs to
A third bipolar transistor,
A fourth bipolar transistor cascode connected to the third bipolar transistor at a second connection point and having a constant bias voltage applied to the base that is not lower than a threshold voltage;
A pair of differential output signals indicating the comparison result by comparing the second output signal with a signal obtained by inverting the second output signal, the base of the third bipolar transistor, and the second connection point A second comparison unit that outputs
A capacitor connected to a connection point of the second and fourth bipolar transistors;
A voltage control oscillator that outputs a periodic signal of a frequency corresponding to the charge and discharge voltage of the capacitor;
A frequency divider that divides the periodic signal and feeds it back to the detector as the feedback signal.

DESCRIPTION OF SYMBOLS 100, 101 Electronic device 110 Analog signal output part 120 Signal processing part 130 Phase-locking circuit 131 Phase / frequency detector 132 Voltage control oscillator 133 Divider 140 Signal processing part 200 AGC circuit 210 Variable gain amplifier 220 Peak detection circuit 221 Reference voltage Supply source 222, 350 Capacitor 230 Bias voltage supply section 231, 233, 241, 242, 251, 252, 261, 262, 271, 281, 322 to 325, 331 to 334, 341 to 344 Bipolar transistor 232 Resistance 240 Current switching circuit 250 comparison unit 260, 321, 335 constant current source 270, 280 switch 290 recovery current source 291 MOS transistor 300 charge pump 310, 311 inverter

Claims (8)

A first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a junction and having a constant bias voltage applied to the base that does not fall below a threshold voltage;
A switching circuit comprising: a comparison unit that compares an input signal and a predetermined reference signal and outputs a pair of differential output signals indicating the comparison result to the base of the first bipolar transistor and the connection point. The switching circuit according to claim 1, wherein the input stage of the comparison unit comprises a pair of differential transistors. The switching circuit according to claim 2, wherein the pair of differential transistors are bipolar transistors. The switching circuit according to claim 2, wherein the pair of differential transistors are MOS (Metal Oxide Semiconductor) transistors. A first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a junction and having a constant bias voltage applied to the base that does not fall below a threshold voltage;
A comparator comparing the input signal with a predetermined reference signal and outputting a pair of differential output signals indicating the comparison result to the base of the first bipolar transistor and the connection point;
A recovery current source for supplying a constant current;
A capacitor connected to the recovery current source and the second bipolar transistor;
An automatic gain control circuit comprising: a variable gain amplifier that changes a gain according to a charge / discharge voltage of the capacitor. 6. The automatic gain control circuit according to claim 5, wherein the input of the comparison unit is an input of the variable gain amplifier. 6. The automatic gain control circuit according to claim 5, wherein the input of the comparison unit is an output of the variable gain amplifier. A detector that detects a phase difference and a frequency difference between an input signal and a feedback signal and outputs first and second output signals indicating the detection result;
A first bipolar transistor,
A second bipolar transistor cascode-connected to the first bipolar transistor at a first connection point and having a constant bias voltage applied to the base that is not lower than a threshold voltage;
A pair of differential output signals indicating the comparison result by comparing the first output signal with a signal obtained by inverting the first output signal, the base of the first bipolar transistor, and the first connection point A first comparison unit that outputs to
A third bipolar transistor,
A fourth bipolar transistor cascode connected to the third bipolar transistor at a second connection point and having a constant bias voltage applied to the base that is not lower than a threshold voltage;
A pair of differential output signals indicating the comparison result by comparing the second output signal with a signal obtained by inverting the second output signal, the base of the third bipolar transistor, and the second connection point A second comparison unit that outputs
A capacitor connected to a connection point of the second and fourth bipolar transistors;
A voltage control oscillator that outputs a periodic signal of a frequency corresponding to the charge and discharge voltage of the capacitor;
A frequency divider that divides the periodic signal and feeds it back to the detector as the feedback signal.

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