JPS6181057A - 0-pi phase modulator - Google Patents
0-pi phase modulatorInfo
- Publication number
- JPS6181057A JPS6181057A JP20468284A JP20468284A JPS6181057A JP S6181057 A JPS6181057 A JP S6181057A JP 20468284 A JP20468284 A JP 20468284A JP 20468284 A JP20468284 A JP 20468284A JP S6181057 A JPS6181057 A JP S6181057A
- Authority
- JP
- Japan
- Prior art keywords
- microwave
- diode
- phase
- wavelength
- series
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/20—Modulator circuits; Transmitter circuits
- H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
- H04L27/2053—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases
- H04L27/206—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers
- H04L27/2064—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers using microwave technology
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/20—Modulator circuits; Transmitter circuits
- H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
- H04L27/2035—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using a single or unspecified number of carriers
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
Abstract
Description
【発明の詳細な説明】
〔発明の技術分野〕
本発明はマイクロ波の〇−π位相変調器に関するもので
ある。DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a microwave 0-π phase modulator.
マイクロ波を制御信号により位相変調し、位相が180
度異なるマイクロ波被変調信号を得る0−π位相変調器
として第3図に示すようなものが提案されている。The phase of the microwave is modulated by a control signal, and the phase is 180
As a 0-π phase modulator for obtaining microwave modulated signals of different degrees, the one shown in FIG. 3 has been proposed.
すなわち、マイクロ波入力端子1から入力したマイクロ
波をバイアス状態によりオン、オフし、かつ極性が互い
に異なるように接続された2つのダイオード2.3に導
き、バイアス供給端子4に加える制御電圧により、2つ
のダイオード2.3をチョークコイル9.10を介して
互いに異なる向きにバイアスする。すると、一方のダイ
オードがオンのとき他方はオフとなり、マイクロ波入力
端子1から加えられたマイクロ波はダイオード2と3の
いずれか一方の出力端子に出力される。そこで、ダイオ
ード2の出力端子から構成される装置クロ波を直流除去
用コンデンサ11および移相遅れ約90度の移相器5を
介して結合度が3dB、結合方向が90度のハイブリッ
ド結合回路6の入力端千人に入力する。That is, the microwave input from the microwave input terminal 1 is turned on and off depending on the bias state, and guided to two diodes 2.3 connected so that the polarities are different from each other, and by the control voltage applied to the bias supply terminal 4, The two diodes 2.3 are biased in different directions via choke coils 9.10. Then, when one diode is on, the other is off, and the microwave applied from the microwave input terminal 1 is output to the output terminal of one of the diodes 2 and 3. Therefore, a hybrid coupling circuit 6 with a coupling degree of 3 dB and a coupling direction of 90 degrees passes the device chromatic wave consisting of the output terminal of the diode 2 through a DC removal capacitor 11 and a phase shifter 5 with a phase shift delay of about 90 degrees. The input end is input to 1,000 people.
一方、ダイオード3の出力端子から送出されるマイクロ
波は直流除去用コンデンサ12を介してハイブレッド結
合回路6のもう一方の入力端子Bに入力する。On the other hand, the microwave sent out from the output terminal of the diode 3 is input to the other input terminal B of the high breadth coupling circuit 6 via the DC removal capacitor 12.
ここで、ダイオード2,3を理想的なスイッチ素子とし
、またマイクロ波入力端子1のマイクロ波入射波をV、
入力端子1−8間の線路長を無視するものとすると、ダ
イオード2.3がオン状態のときのA点、B点での波形
?a、?bは、−Jωl
となる。ただし? == V 4 で■は入射波の
振゛−幅、ωはその角周波数、lは時間である。また、
1 ハイブリッド結合回路6の出力端子7へ出
力される波形は以下のように表されることはよく知られ
ている。Here, the diodes 2 and 3 are assumed to be ideal switching elements, and the microwave incident wave at the microwave input terminal 1 is set to V,
Assuming that the line length between input terminals 1 and 8 is ignored, what are the waveforms at points A and B when diode 2.3 is in the on state? a,? b becomes -Jωl. however? == V 4 where ■ is the amplitude of the incident wave, ω is its angular frequency, and l is the time. Also,
1 It is well known that the waveform output to the output terminal 7 of the hybrid coupling circuit 6 is expressed as follows.
従って、第(1)式を第(2)式に代入すると、出力波
形は、
イオード3を通過して出力端子7に出力されるマイクロ
波信号は位相が180度異なものとなる。よって、ダイ
オード2,3のバイアス供給端子4に 1制御
信号を加えてダイオードのオン、オフを切替えてやれば
、制御信号に応じて位、相が180度異なる位相波変調
(1号を得ろことができる。なお、第3図において、8
はチョークコイル、13は終端抵抗である。Therefore, when equation (1) is substituted into equation (2), the output waveforms of the microwave signals that pass through the diode 3 and are output to the output terminal 7 are different in phase by 180 degrees. Therefore, if you apply the 1 control signal to the bias supply terminal 4 of the diodes 2 and 3 to switch the diodes on and off, you can obtain phase wave modulation (1. In addition, in Figure 3, 8
is a choke coil, and 13 is a terminating resistor.
ところが、上述した回路では、ダイオードを使用したマ
イクロ波スイッチ回路がハイブリッド結合回路6の入力
端千人に移相遅れが約90度の移相器5を介して接続さ
れ、入力端子Bには直接接続されている。このため、A
点とB点からダイオード側をみた規格化インピーダンス
をそれぞれZA。However, in the circuit described above, the microwave switch circuit using a diode is connected to the input terminal of the hybrid coupling circuit 6 via the phase shifter 5 with a phase shift delay of about 90 degrees, and the microwave switch circuit using the diode is connected directly to the input terminal B. It is connected. For this reason, A
ZA is the normalized impedance when looking at the diode side from point and point B, respectively.
ZBとすると、ダイオード2がオフのとき、0点よりダ
イオード側を見たインピーダンスは開放となり、移相器
5を介したA点でのインピーダンスZAK短絡となる。Assuming ZB, when the diode 2 is off, the impedance viewed from the 0 point to the diode side is open, and the impedance ZAK at the point A via the phase shifter 5 is short-circuited.
他方、ダイオード3がオフのときのD点よりダイオード
側を見たインピーダンスは開放となり、またB−D間の
線路長を無視するとB点でのインピーダンスZBは開放
となる。On the other hand, when the diode 3 is off, the impedance seen from point D to the diode side is open, and if the line length between B and D is ignored, the impedance ZB at point B is open.
このこと(ま、ハイブリッド結合回路6の入力側のアイ
ソレーション端子の終端条件がダイオード2゜3のオン
、オフにより大きく異なってくることを意味する。This means that the termination conditions of the isolation terminal on the input side of the hybrid coupling circuit 6 vary greatly depending on whether the diode 2.3 is on or off.
ハイブリッド結合回路6は所望の結合度を得るために各
入出力抱子と外部回路との間で整合がとれていることが
必要である。しかるに、前述したようにハイブリッド結
合回路6の片側の入力端子が開放または短絡となった場
合、ハイブリッド結合回路6においては入出力端子間の
結合度が所望の値にならず、第(2)式の関係、ひいて
は(3)式の関係式を満足することができなくなり、2
つの径路のマイクロ波の通過位相差が180度より異な
る値をとるようになり、さらに2つの径路を経て出力さ
れる信号の振幅もその値が互いに異なってしまうという
問題点があった。′
〔発明の目的〕
本発明は上記欠点を除去し、位相偏差および振幅偏差の
小さいマイクロ波位相被変調信号を得ることができる0
−π位相変調器を提供することを目的とする。The hybrid coupling circuit 6 requires matching between each input/output connector and the external circuit in order to obtain a desired degree of coupling. However, as described above, if one input terminal of the hybrid coupling circuit 6 is open or short-circuited, the degree of coupling between the input and output terminals in the hybrid coupling circuit 6 does not reach the desired value, and the equation (2) , and by extension, the relational expression (3) cannot be satisfied, and 2
There is a problem in that the passing phase difference of the microwaves in the two paths becomes different from each other by more than 180 degrees, and the amplitudes of the signals outputted through the two paths also have different values. [Objective of the Invention] The present invention eliminates the above drawbacks and provides a microwave phase modulated signal with small phase deviation and amplitude deviation.
The present invention aims to provide a −π phase modulator.
本発明は、マイクロ波スイッチ回路の出力に入力マイク
ロ波のほぼ1/4波長の伝送線路および定在波比改善用
の抵抗を直列に挿入することにより、上記目的を達成し
ている。The present invention achieves the above object by inserting in series a transmission line of approximately 1/4 wavelength of the input microwave and a resistor for improving the standing wave ratio to the output of the microwave switch circuit.
第1図は本発明の一実施例を示す回路図であり、第3図
と同一部分は同一記号で表している。第1図において、
第3図の従来回路と異なるのは、ダイオード2.3の出
力側に入力マイクロ波のほぼ1/4波長の伝送線路14
.11−直列に接続し、さらにこの伝送線路14 、1
5に定在波比改善用の抵抗16 、17を直列に接続し
ていることである。FIG. 1 is a circuit diagram showing one embodiment of the present invention, and the same parts as in FIG. 3 are represented by the same symbols. In Figure 1,
What is different from the conventional circuit shown in Fig. 3 is that a transmission line 14 with approximately 1/4 wavelength of the input microwave is installed on the output side of the diode 2.3.
.. 11 - connected in series and further this transmission line 14 , 1
5 and resistors 16 and 17 for improving the standing wave ratio are connected in series.
この構成によれば、ダイオード2がオンで、ダイオード
3がオフのときの高周波等価回路は第2図に示すような
ものとなる。但し、第2図において、抵抗16 、17
は等しい値のものを使用し、伝送線路の特性インピーダ
ンスで規格化したときの規格化インピーダンスをRとし
て表している。According to this configuration, the high frequency equivalent circuit when diode 2 is on and diode 3 is off is as shown in FIG. However, in FIG. 2, resistors 16 and 17
are of equal value, and the normalized impedance when normalized by the characteristic impedance of the transmission line is expressed as R.
この等何回路から明らかなように、抵抗17は先端開放
の1/4波長伝送線路15に接続され、また抵抗17の
接続点からダイオード側を見たインピーダンスは短絡と
なっている。このため、B点からできる。As is clear from these circuits, the resistor 17 is connected to the 1/4 wavelength transmission line 15 with an open end, and the impedance seen from the connection point of the resistor 17 to the diode side is short-circuited. Therefore, it can be done from point B.
以上の説明から明らかなように、本発明によれば、マイ
クロ波スイッチ回路の出力に入力マイクロ波のほぼ17
4波長の伝送線路および定在波比改善用の抵抗を直列に
挿入したため、ハイブリッド結合回路の入力側の一方の
アイソレーション端子の反射係数が改善され、他方のア
イソレーション端子の整合状態による影響を受けにくく
なり、位相偏差および撮幅偏差の小さいマイクロ波被変
調信号を得ることができる。As is clear from the above description, according to the present invention, approximately 17% of the input microwave is applied to the output of the microwave switch circuit.
By inserting a four-wavelength transmission line and a resistor for improving the standing wave ratio in series, the reflection coefficient of one isolation terminal on the input side of the hybrid coupling circuit is improved, and the influence of the matching state of the other isolation terminal is reduced. Therefore, it is possible to obtain a microwave modulated signal with small phase deviation and width deviation.
第1図は本発明の一実施例を示す回路図、第2図は第1
図の高周波等価回路、第3図は従来の〇−π位相変調器
を示す回路図である。
1・・・マイクロ波入力端子、2,3・・・ダイオード
、4・・・バイアス供給端子、5・・・移相器、6・・
・ハイブリッド結合回路、7・・・マイクロ波出力端子
、13・・・終端抵抗、14 、15・・・伝送線路、
16 、17・・・抵抗。Fig. 1 is a circuit diagram showing one embodiment of the present invention, and Fig. 2 is a circuit diagram showing an embodiment of the present invention.
The high frequency equivalent circuit shown in FIG. 3 is a circuit diagram showing a conventional 0-π phase modulator. 1... Microwave input terminal, 2, 3... Diode, 4... Bias supply terminal, 5... Phase shifter, 6...
・Hybrid coupling circuit, 7...Microwave output terminal, 13...Terminal resistor, 14, 15...Transmission line,
16, 17...resistance.
Claims (1)
路には第1のマイクロ波スイッチ回路、入力マイクロ波
のほぼ1/4波長の伝送線路、定在波比改善用の抵抗、
位相遅れが約90度または約180度の移相器を順に直
列接続し、第2の径路には前記第1のマイクロ波スイッ
チ回路と相反的に動作する第2のマイクロ波スイッチ回
路、入力マイクロ波のほぼ1/4波長の伝送線路、定在
波比改善用の抵抗を順に直列接続し、これら第1および
第2の径路から出力されるマイクロ波を位相差が180
度になるようにハイブリッド結合回路で合成し、このハ
イブリッド結合回路から位相が180度異なるマイクロ
波位相被変調信号を出力することを特徴とする0−π位
相変調器。The input microwave is branched into two paths, and the first path includes a first microwave switch circuit, a transmission line of approximately 1/4 wavelength of the input microwave, a resistor for improving the standing wave ratio,
Phase shifters having a phase delay of about 90 degrees or about 180 degrees are connected in series, and the second path includes a second microwave switch circuit that operates reciprocally with the first microwave switch circuit, and an input microwave. A transmission line of approximately 1/4 wavelength of the wave and a resistor for improving the standing wave ratio are connected in series, and the microwaves output from these first and second paths have a phase difference of 180 degrees.
1. A 0-π phase modulator, characterized in that a microwave phase modulated signal is synthesized by a hybrid coupling circuit so as to have a phase difference of 180 degrees, and the hybrid coupling circuit outputs a microwave phase modulated signal having a phase difference of 180 degrees.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP20468284A JPS6181057A (en) | 1984-09-28 | 1984-09-28 | 0-pi phase modulator |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP20468284A JPS6181057A (en) | 1984-09-28 | 1984-09-28 | 0-pi phase modulator |
Publications (1)
Publication Number | Publication Date |
---|---|
JPS6181057A true JPS6181057A (en) | 1986-04-24 |
Family
ID=16494556
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP20468284A Pending JPS6181057A (en) | 1984-09-28 | 1984-09-28 | 0-pi phase modulator |
Country Status (1)
Country | Link |
---|---|
JP (1) | JPS6181057A (en) |
Cited By (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
FR2636793A1 (en) * | 1988-09-19 | 1990-03-23 | Alcatel Transmission | HYPERFREQUENCY MODULATOR WITH TWO STAGES OF PHASE: 0, PI WITH VERY LOW LOSSES |
JPH0391341A (en) * | 1989-09-04 | 1991-04-16 | Oki Electric Ind Co Ltd | Phase modulator |
US5161206A (en) * | 1990-10-03 | 1992-11-03 | Telefonaktiebolaget L M Ericsson | Method of linearizing a transmission function of a modulator arrangement, and a linearized modulator |
US5363230A (en) * | 1991-12-20 | 1994-11-08 | Telefonaktiebolaget L M Ericsson | Method of linearizing the transmission function of modulator |
-
1984
- 1984-09-28 JP JP20468284A patent/JPS6181057A/en active Pending
Cited By (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
FR2636793A1 (en) * | 1988-09-19 | 1990-03-23 | Alcatel Transmission | HYPERFREQUENCY MODULATOR WITH TWO STAGES OF PHASE: 0, PI WITH VERY LOW LOSSES |
US4965866A (en) * | 1988-09-19 | 1990-10-23 | Alcatel N.V. | Very low loss microwave modulator having two phase states O, π |
JPH0391341A (en) * | 1989-09-04 | 1991-04-16 | Oki Electric Ind Co Ltd | Phase modulator |
US5161206A (en) * | 1990-10-03 | 1992-11-03 | Telefonaktiebolaget L M Ericsson | Method of linearizing a transmission function of a modulator arrangement, and a linearized modulator |
US5363230A (en) * | 1991-12-20 | 1994-11-08 | Telefonaktiebolaget L M Ericsson | Method of linearizing the transmission function of modulator |
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