JPH11308152A - Spread spectrum signal analysis device/method - Google Patents

Spread spectrum signal analysis device/method

Info

Publication number
JPH11308152A
JPH11308152A JP11026804A JP2680499A JPH11308152A JP H11308152 A JPH11308152 A JP H11308152A JP 11026804 A JP11026804 A JP 11026804A JP 2680499 A JP2680499 A JP 2680499A JP H11308152 A JPH11308152 A JP H11308152A
Authority
JP
Japan
Prior art keywords
frequency
spread spectrum
signal
spectrum signal
spread
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Pending
Application number
JP11026804A
Other languages
Japanese (ja)
Inventor
Juichi Nakada
寿一 中田
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Advantest Corp
Original Assignee
Advantest Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Advantest Corp filed Critical Advantest Corp
Priority to JP11026804A priority Critical patent/JPH11308152A/en
Publication of JPH11308152A publication Critical patent/JPH11308152A/en
Pending legal-status Critical Current

Links

Abstract

PROBLEM TO BE SOLVED: To provide a spread spectrum signal analysis device which can detect a shift of the carrier frequency of a reception signal from an official carrier frequency without executing a processing for changing the frequency and can detect the timing shift of a spreading code. SOLUTION: An input spread spectrum signal is orthogonally converted 16 and the common-mode component output Zre and orthogonal component output Zim are AD-converted 23 and 24. Zre and Zim are inputted to the series connection of a delay unit whose delay quantity is a sample period and corresponding delay output is complex-multiplied (201i). The respective outputs are converted by N point complex FET. The absolute value of respective coefficients is squared and it is displayed 32 with a lateral axis as a frequency and a vertical axis as time. A carrier frequency error is obtained by the frequency of the largest peak and timing shift by time.

Description

【発明の詳細な説明】DETAILED DESCRIPTION OF THE INVENTION

【0001】[0001]

【発明の属する技術分野】この発明は例えばスペクトラ
ムの拡散通信における信号の周期タイミングや搬送周波
数の公称周波数に対する誤差の測定や、スペクトラム拡
散信号の復調に利用されるスペクトラム拡散信号の解析
装置に関する。
BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to, for example, an apparatus for measuring an error of a signal relative to a nominal frequency of a carrier frequency or a carrier frequency in spread spectrum communication and an apparatus for analyzing a spread spectrum signal used for demodulation of a spread spectrum signal.

【0002】[0002]

【従来の技術】CDMA(符号分割多重接続)の無線通
信においてスペクトラム拡散信号を受信する場合、その
受信信号の拡散符号(PN符号、Walsh符号、Go
ld符号など)と自局の拡散符号信号とを同期させる必
要がある。この同期は入力信号と自局の拡散符号信号と
の相関を求め、その相関値が、時間軸上でピークとなる
点を求める方法がある。
2. Description of the Related Art When a spread spectrum signal is received in CDMA (code division multiple access) wireless communication, a spread code (PN code, Walsh code, Go code) of the received signal is received.
ld code) and its own spread code signal. For this synchronization, there is a method in which a correlation between an input signal and a spread code signal of the own station is obtained, and a point where the correlation value becomes a peak on a time axis.

【0003】一方無線通信においては、公称搬送波周波
数に対して、受信搬送周波数に誤差が生じる。この誤差
はドップラシフトによるものや、送信機内部の基準周波
数源の周波数誤差などによる。測定器では、中心周波数
の設定間違いや、公称周波数が正確にわからない場合が
ある。このような搬送波周波数誤差があると、相関値の
ピークが出現しないなど、時間方向の同期が困難にな
る。そこである周波数範囲を区切って順番に受信搬送波
周波数を変えて同期位置を探すことが行われている。い
ま直交変換により得られた複素信号Z(i)(iはサンプ
ル番号であり時間を表す)は、搬送波周波数がΔωだけ
ずれていた場合、その周波数誤差を補正し、Z(i)×e
xp(−jΔωi)となる。この補正された信号と、拡
散符号信号Rの複素共役R*との積の絶対値の2乗、つま
り相関が計算される。
On the other hand, in wireless communication, an error occurs in a received carrier frequency with respect to a nominal carrier frequency. This error is due to a Doppler shift or a frequency error of a reference frequency source inside the transmitter. In a measuring instrument, there are cases where the center frequency is incorrectly set and the nominal frequency is not accurately known. If there is such a carrier frequency error, synchronization in the time direction becomes difficult, for example, a peak of a correlation value does not appear. Therefore, a search for a synchronization position is performed by sequentially changing the received carrier frequency while dividing a certain frequency range. Now, the complex signal Z (i) (i is a sample number and represents time) obtained by orthogonal transformation corrects the frequency error when the carrier frequency is shifted by Δω, and Z (i) × e
xp (−jΔωi). The square of the absolute value of the product of the corrected signal and the complex conjugate R * of the spread code signal R, that is, the correlation is calculated.

【0004】 |Σi=0 L〔Z(i)×exp(−jΔωi) 〕×R*(i)|2 … (1) この相関値の絶対値の2乗値の時間推移、つまり相関曲
線のピークが同期位置となる。相関曲線C(m) は次式で
与えられる。 C(m)=|Σi=0 L〔Z(m+i)×exp(−jΔω・(m+i)〕×R*(i)|
2 周波数シフトした相関曲線は以下の通りである。
| Σ i = 0 L [Z (i) × exp (−jΔωi)] × R * (i) | 2 (1) Time transition of the square of the absolute value of the correlation value, that is, a correlation curve Is the synchronization position. The correlation curve C (m) is given by the following equation. C (m) = | Σ i = 0 L [Z (m + i) × exp (−jΔω ・ (m + i)] × R * (i) |
The correlation curve shifted by two frequencies is as follows.

【0005】C-(N/2-1)(m)=|Σi=0 L〔Z(m+i)×exp(−j
・(−(N/2−1))・Δω・(m+i)〕×R*(i)|2 ……………… C-1(m)=|Σi=0 L〔Z(m+i)×exp(−j・−Δω・(m+i))〕
×R*(i)|2 C0(m)=|Σi=0 L〔Z(m+i)×exp(−j・ (m+i))〕×R*(i) |
2 C1(m)=|Σi=0 L〔Z(m+i)×exp(−j・Δω・(m+i))〕×R*
(i)|2 C2(m)=|Σi=0 L〔Z(m+i)×exp(−j・2Δω・(m+i))〕×R
*(i)|2 ……………… CN/2(m)=|Σi=0 L〔Z(m+i)×exp(−j・(N/2))・Δω・(m
+i) 〕×R*(i)|2 Nは拡散符号の1周期(符号長) 搬送波周波数誤差のために、Δωずつずらしながら相関
曲線を求め、相関曲線のピークの最大のピークを持つ周
波数を探す。
C- (N / 2-1) (m) = | Σ i = 0 L [Z (m + i) × exp (−j
・ (− (N / 2−1)) ・ Δω ・ (m + i)] × R * (i) | 2 ……………… C -1 (m) = | Σ i = 0 L [Z ( m + i) × exp (−j ・ −Δω ・ (m + i)))
× R * (i) | 2 C 0 (m) = | Σ i = 0 L [Z (m + i) × exp (−j ・ (m + i))] × R * (i) |
2 C 1 (m) = | Σ i = 0 L [Z (m + i) × exp (−j ・ Δω ・ (m + i))] × R *
(i) | 2 C 2 (m) = | Σ i = 0 L [Z (m + i) × exp (−j ・ 2Δω ・ (m + i))] × R
* (i) | 2 ……………… C N / 2 (m) = | Σ i = 0 L [Z (m + i) × exp (−j · (N / 2)) · Δω · (m
+ i)] × R * (i) | 2 N is one period of the spreading code (code length) Due to the carrier frequency error, the correlation curve is obtained while shifting by Δω, and the frequency having the maximum peak of the correlation curve is obtained. Search for

【0006】以上のことは例えば科学技術出版社発行
横山著「スペクトル拡散通信システム」333〜337
頁に詳しく述べられている。
The above is, for example, published by Science and Technology Publishing Company
Yokoyama, "Spread Spectrum Communication System", 333-337
It is detailed on the page.

【0007】[0007]

【発明が解決しようとする課題】周波数範囲を順番に変
更させて、その周波数範囲内で拡散符号を半チップずつ
シフトさせる時間同期動作を行い、しきい値以上のピー
ク値が得られないと、周波数範囲を変更して同様のこと
を行うため、周波数を順番に変更させるハードウェアを
必要とし、また、ソフトウェア処理により求める場合
は、信号の周波数変更処理のための時間が必要であっ
た。つまり、ハードウェア規模が大きくなるか、又は処
理時間が比較的長い欠点があった。
A time synchronization operation is performed in which the frequency range is changed in order and the spreading code is shifted by half a chip within the frequency range. In order to perform the same thing by changing the frequency range, hardware for changing the frequency in order is required, and when it is determined by software processing, time is required for the signal frequency change processing. That is, there is a disadvantage that the hardware scale becomes large or the processing time is relatively long.

【0008】この発明の目的は周波数を変更する処理を
行うことなく、受信信号の搬送波周波数の公称搬送波周
波数に対するずれを検出することができ、また拡散符号
の夕イミングずれを検出することができるスペクトラム
拡散信号解析装置を提供することにある。
An object of the present invention is to provide a spectrum capable of detecting a deviation of a carrier frequency of a received signal from a nominal carrier frequency without performing a process of changing a frequency, and detecting an evening deviation of a spread code. An object of the present invention is to provide a spread signal analyzer.

【0009】[0009]

【課題を解決するための手段】この発明によれば、入カ
スペクトラム拡散信号は直交変換され、その直交変換さ
れた信号は拡散符号信号の複素共役との積が求められ、
その求めた積のデータ系列が離散的フーリエ変換され、
そのフーリエ変換された各係数値の絶対値の二乗値が求
められ、その二乗値が、拡散符号信号の1周期内におけ
る最大となる拡散符号信号の基準時間に対するずれ(タ
イミング)と、対応係数の周波数が搬送波周波数誤差と
して求まる。
According to the present invention, an input spread spectrum signal is orthogonally transformed, and the product of the orthogonally transformed signal and a complex conjugate of a spread code signal is obtained.
The data series of the obtained product is subjected to discrete Fourier transform,
The square value of the absolute value of each of the Fourier-transformed coefficient values is obtained, and the square value is the difference (timing) of the maximum value in one cycle of the spread code signal with respect to the reference time of the spread code signal and the corresponding coefficient. The frequency is determined as the carrier frequency error.

【0010】つまり式(1)において、Σの内部で積の変
換を行い、また絶対値内部でmは固定であるから式(1)
は次式(2)で表わせる。 C(m) =|Σi=0 L〔Z(m+i)×R*(i)〕×exp(−j・Δω・i) ×exp(−j・Δω・m)|2 =|Σi=0 L{〔Z(m+i)×R*(i)〕×exp(−j・Δω・i)}|2 … (2) この式(2)は(Z(m+i)×R*) を離散フーリエ変
換する式であり、よって先に述べたこの発明における入
力の直交変換された信号Z (m+i) と拡散符号系列の
複素共役R*との積を各サンプルごとに求め、その結果の
データ系列に対して、離散的フーリエ変換を行い、その
結果の各係数の絶対値の二乗がmでの相関値となる。離
散的フーリエ変換を各周波数に対して求めれば、C
f(m)を求めることができる。離散的フーリエ変換に
FFT (高速フーリエ変換) を用いれば、離散的な各周
波数に対してCf(m)が求まり、m,m+1,m+2,
・・・について同様の操作を繰り返せば、離散的な周波数
(周波数誤差)と離散的な時間(タイミングずれ) との
二次元の相関データが得られる。
That is, in equation (1), product conversion is performed inside Σ, and m is fixed inside the absolute value.
Can be expressed by the following equation (2). C (m) = | Σ i = 0 L [Z (m + i) × R * (i)] × exp (−j ・ Δω ・ i) × exp (−j ・ Δω ・ m) | 2 = | Σ i = 0 L {[Z (m + i) × R * (i)] × exp (−j · Δω · i)} | 2 (2) This equation (2) is expressed as (Z (m + i) × R * ) Is a discrete Fourier transform, and the product of the input orthogonally transformed signal Z (m + i) of the present invention and the complex conjugate R * of the spreading code sequence is obtained for each sample. Is subjected to a discrete Fourier transform, and the square of the absolute value of each coefficient is the correlation value at m. If a discrete Fourier transform is obtained for each frequency, C
f (m) can be obtained. If FFT (fast Fourier transform) is used for the discrete Fourier transform, C f (m) is obtained for each discrete frequency, and m, m + 1, m + 2,
..., two-dimensional correlation data between a discrete frequency (frequency error) and a discrete time (timing shift) can be obtained.

【0011】[0011]

【発明の実施の形態】図1にこの発明の実施例を示す。
アンテナ11で受信されたスペクトラム拡散信号は、周
波数混合器12で、局部発振器13より周波数変換さ
れ、その中間周波数成分が帯域通過フィルタ14で取出
され、その中間周波数信号は分配器15で2分配されて
直交変換器16で局部発振器17の信号により直交変換
される。つまり2分配された一方は周波数混合器18
で、局部発振器17の出力が90゜位相シフトされた信
号と周波数混合され、また周波数混合器19で2分配さ
れた他方は局部発振器17の出力と周波数混合され、同
波数混合器18,19の各出力はそれぞれ低域通過フィ
ルタ21,22で低域(ベースバンド)成分が取出され
て、AD変換器23,24でそれぞれデジタル系列とさ
れ、遅延・複素乗算部30へ供給される。
FIG. 1 shows an embodiment of the present invention.
The spread spectrum signal received by the antenna 11 is frequency-converted by a local oscillator 13 by a frequency mixer 12, its intermediate frequency component is extracted by a band-pass filter 14, and the intermediate frequency signal is split into two by a splitter 15. Then, the orthogonal transform is performed by the signal of the local oscillator 17 by the orthogonal transformer 16. In other words, one of the two distributions is the frequency mixer 18
Then, the output of the local oscillator 17 is frequency-mixed with the signal whose phase has been shifted by 90 °, and the other of the two divided by the frequency mixer 19 is frequency-mixed with the output of the local oscillator 17. From each output, low-pass (baseband) components are taken out by low-pass filters 21 and 22, converted into digital sequences by AD converters 23 and 24, and supplied to a delay / complex multiplier 30.

【0012】遅延・複素乗算部30は図2に示すように
同相成分のデジタル系列(AD変換器23の出力)は、
それぞれサンプリング周期Tsの遅延量のN個の遅延器
251〜25Nの直列接続へ入力され (Nは拡散符号の符
号長即ち1周期) 、また直交成分のデジタル系列は、そ
れぞれサンプリング周期の遅延量のN個の遅延器261
〜26Nの直列接続へ入力される。同様にTsの遅延量
のN個の遅延器261〜26Nの直列接続へ、直交成分の
デジタル系列 (AD変換器24の出力) が入力される。
As shown in FIG. 2, the delay / complex multiplication unit 30 converts the digital series of the in-phase component (the output of the AD converter 23) into
Each is input to a serial connection of N delay units 25 1 to 25 N with the delay amount of the sampling period Ts (N is the code length of the spread code, that is, one period). Quantity of N delayers 26 1
2626 N is connected to the series connection. Similarly, a digital series of quadrature components (the output of the AD converter 24) is input to a series connection of N delay units 26 1 to 26 N having a delay amount of Ts.

【0013】遅延器25i(i=1,・・・,N) の出力Zr
e(m+N−i)と対応する遅延器26iの出力Zim
(m+N−i)とがそれぞれ複素乗算手段20iで複素乗
算される。複素乗算手段20iでは遅延器25iの出力
Zre(m+N−i)と拡散符号列R* (N−i)の実部R
re(N−i)の虚部Rim(N−i)との各積がそれぞれ
乗算器27rei,28reiで求められ、また遅延器
26iの出力Zim(m+N−i)とR*(N−i)の虚部R
im(N−i)と実部Rre(N−i)の各積がそれぞれ乗
算器27imi,28imiで求められ、乗算器27r
eiと27imiの各出力の和が加算器29rei で
求められ、複素乗算結果の実部Dre(N−i)が出力さ
れ、乗算器28reiと28imiの各出力の和が加算
器29imiで求められ、複素乗算結果の虚部Dim
(N−i)が出力される。
The output Zr of the delay unit 25i (i = 1,..., N)
e (m + N−i) and output Zim of the delay unit 26i corresponding to
(m + N−i) are complex-multiplied by complex multiplying means 20i. In the complex multiplying means 20i, the output Zre (m + Ni) of the delay unit 25i and the real part R of the spread code string R * (Ni) are obtained.
Each product of re (N-i) and the imaginary part Rim (N-i) is obtained by multipliers 27 rei and 28 rei, respectively, and outputs Zim (m + N-i) and R * (N-i) of the delay unit 26 i. The imaginary part R of
The respective products of im (N−i) and real part Rre (N−i) are obtained by multipliers 27imi and 28imi, respectively, and multiplier 27r
The sum of the outputs of ei and 27imi is obtained by the adder 29rei, the real part Dre (N−i) of the result of the complex multiplication is output, and the sum of the outputs of the multipliers 28rei and 28imi is obtained by the adder 29imi. Imaginary part Dim of complex multiplication result
(Ni) is output.

【0014】これらN点の複素乗算手段201〜20N
実部出力Dre(0)〜Dre(N−1)、虚部出力Dim
(0)〜Dim(N−1)がN点複素FFT演算部31に
入力されて高速フーリエ変換(FFT変換)が行われ
る。その高速フーリ工変換部31の出力は絶対値二乗演
算部33で各係数値の絶対値が二乗されて表示部32に
供給表示される。
[0014] The real part output Dre complex multiplier means 20 1 to 20 N of these N points (0) ~Dre (N-1 ), the imaginary part output Dim
(0) to Dim (N-1) are input to the N-point complex FFT operation unit 31, and fast Fourier transform (FFT transform) is performed. The output of the fast Fourier transform unit 31 is squared with the absolute value of each coefficient value in the absolute value square operation unit 33 and is supplied to the display unit 32 for display.

【0015】つまり入力複素信号Ziのサンプル系列と
拡散符号信号の複素共役R*との対応時点の複数乗算結果
のデータ系列のN点のデータの時刻mでのFFT変換結
果が、例えば図3Aのmに示すように得られ、次の時刻
m+1におけるFFT変換結果が図3Aのm+1に示す
ように得られ、以下同様に得られる。つまり時刻(位
相)を1サンプリング周期TsずらしたFFT変換結果
が次々と得られる。
That is, the result of the FFT transformation at time m of the data at N points of the data sequence of the multiple multiplication result at the corresponding time point between the sample sequence of the input complex signal Zi and the complex conjugate R * of the spread code signal is shown in FIG. m, and the result of the FFT transform at the next time m + 1 is obtained as shown at m + 1 in FIG. 3A, and so on. In other words, FFT conversion results in which the time (phase) is shifted by one sampling period Ts are obtained one after another.

【0016】これら出力の絶対値の二乗演算結果を、周
波数fを横軸とし、レベルを縦軸としたものを、図3B
に示すように、横軸と縦軸とに直角な軸を時間(m)軸
として3次元表示し、あるいは図3Cに示すように、横
軸を周波数、縦軸を時間(m)としてレベルを輝度又は
カラー表示してもよい。図3Bの表示でレベルのピーク
値が最も大きい点(f,m)の周波数fが搬送波周波数
誤差であり、時間mが拡散符号のタイミング時間(位
相)ずれである。図3Cの表示では最も輝度が高い点
(f,m)、又は輝度と色との対応ずけで最も高い輝度
と対応する色の点(f,m)から搬送波周波数誤差とタ
イミング時間とを知ることができる。図3Cにおいて同
一ハッチング(同一番号)の部分は同一色であり、番号
が異なれば表示色も異なることを示し、番号46の点が
最も輝度が高い点(f,m)であり、この付近では輝度
(色)が急に変化しており、この位置を直ちに読み取る
ことができる。
FIG. 3B shows the result of squaring the absolute values of these outputs with the frequency f as the horizontal axis and the level as the vertical axis.
As shown in FIG. 3, the axis perpendicular to the horizontal axis and the vertical axis is displayed three-dimensionally with the time (m) axis, or as shown in FIG. 3C, the horizontal axis is frequency, and the vertical axis is time (m), and the level is shown. Brightness or color display may be used. In the display of FIG. 3B, the frequency f at the point (f, m) where the level peak value is the largest is the carrier frequency error, and the time m is the timing time (phase) shift of the spread code. In the display of FIG. 3C, the carrier frequency error and the timing time are known from the point (f, m) having the highest luminance, or the point (f, m) of the color corresponding to the highest luminance due to the correspondence between the luminance and the color. be able to. In FIG. 3C, the same hatched portions (same numbers) have the same color, and different numbers indicate different display colors. The point 46 is the point (f, m) having the highest luminance. The luminance (color) changes suddenly, and this position can be read immediately.

【0017】サンプリング周波数1/Tsを、拡散符号
信号のチップ周波数1/Tcの整数J倍とした場合は、
拡散符号信号のデータはJサンプル置きのデータでよ
く、残りのJ−1サンプルの拡散符号のデータは0にな
る。よってJ=2の場合の直交変換の周期成分Zre
(m+N−i)と拡散符号信号の実部Rre(N−i)との
乗算は図4Aに示すようになり、1つおき、つまりRr
e(N−2)、Rre(N−4)、Rre(N−6)・・・・・・
Rre(0) と対応する入力周期成分との積を求めれば
よい。他の成分の乗算も同様に1つ置きでよい。
When the sampling frequency 1 / Ts is set to an integer J times the chip frequency 1 / Tc of the spread code signal,
The data of the spread code signal may be data every J samples, and the data of the spread code of the remaining J-1 samples is zero. Therefore, the periodic component Zre of the orthogonal transform when J = 2
The multiplication of (m + N−i) and the real part Rre (N−i) of the spreading code signal is as shown in FIG.
e (N-2), Rre (N-4), Rre (N-6) ...
The product of Rre (0) and the corresponding input periodic component may be obtained. Similarly, the multiplication of other components may be every other.

【0018】拡散符号1チップは通常、1と0で表わさ
れる2値データであり、この場合、拡散符号の信号とし
ては±1が用いられる。従って、Rre(N−i)は+1
か−1の値のみを取るから、J=2の場合は、図4Bに
示すように、サンプルの1つ置きに出力を出すが、その
1つ置き出力は更に1つ置きに、直交変換出力の4つ置
き同相成分Zre(m+N−2)、Zre(m+N−6)・
・・・・・がそのまま乗算結果となり、他の1つ置き、
つまり4つ置きの同相成分が符号反転されて−Zre
(m+N−4)、−Zre(m+N−8)、・・・・・・
として乗算結果が得られる。他の成分の乗算も同様に処
理される。
One chip of the spread code is usually binary data represented by 1 and 0. In this case, ± 1 is used as a signal of the spread code. Therefore, Rre (Ni) is +1
Since only the value of -1 is taken, in the case of J = 2, as shown in FIG. 4B, an output is output for every other sample, and every other output is an orthogonal transform output. Every four in-phase components Zre (m + N-2), Zre (m + N-6).
... is the multiplication result as it is, and every other one is
In other words, every fourth in-phase component is sign-inverted and −Zre
(m + N-4), -Zre (m + N-8), ...
To obtain a multiplication result. Multiplication of other components is handled in a similar manner.

【0019】この場合は複素乗算手段201〜20Nは図
5に示すようになる。図5において図2と対応する部分
に同一符号を付けてある。この場合のサンプリング周期
Tsは、拡散符号のチップ周期Tcの1/2である。図
2中の複素乗算手段201〜20N中の偶数番目のものは
省略され、奇数番目201,203,20s,・・・が残
され、これらは各乗算器27rei,28rei,27
imi,28imiが省略され、奇数番目の加算器29
rei,29imiが残され、これら残された加算器に
おいて、拡散符号信号の同相成分Rreの0,+1,
0,−1,0,+1,・・・・・・,−1と、直交成分Rimi
の0 ,+1,0,+1,0,−1,・・・,−1に応じて、
遅延器252i-1の出力Zre(m+N−2(2i−1))
は、Rre,Rimが+1で同極のまま、−1で逆極
(符号反転)されて、遅延器262i-1のZim (m+N
−2(2i−1))は、Rre,Rimが+1で符号
(極性)がそのまま、−1で逆極(符号反転)されて、
加算器29re(2i−1)、29im(2i−1)
で、それぞれ互いに加算されて出力Dre(2i−
1)、Dim(2i−1)N/2点の複素FFT部へ供
給されることになる。
[0019] The complex multiplier means if 20 1 to 20 N are as shown in FIG. 5, parts corresponding to those in FIG. 2 are denoted by the same reference numerals. The sampling period Ts in this case is 1 / of the chip period Tc of the spreading code. Those of the even-numbered complex multiplication means 20 1 in to 20 N in FIG. 2 is omitted, the odd-numbered 20 1, 20 3, 20s, · · · are left, they each multiplier 27rei, 28rei, 27
imi and 28imi are omitted, and the odd-numbered adder 29
rei, 29 imi are left, and in these remaining adders, 0, + 1,0 of the in-phase component Rre of the spread code signal
0, -1, 0, +1, ..., -1 and the orthogonal component Rimi
According to 0, +1, 0, +1, 0, -1,.
Output Zre (m + N-2 (2i-1)) of delay unit 25 2i-1
Is inverted (sign inverted) at -1 while Rre and Rim remain the same at +1 and Zim (m + N) of the delay unit 26 2i-1.
-2 (2i-1)) is obtained when Rre and Rim are +1 and the sign (polarity) is unchanged, and -1 is reversed (sign inverted).
Adder 29re (2i-1), 29im (2i-1)
And the outputs Dre (2i-
1), Dim (2i-1) N / 2 points are supplied to the complex FFT unit.

【0020】直交変換手役16において、分配器15の
出力Asin(ωt+θ)と、周波数混合器18では余弦
波信号cos(πn/2)、つまり1,0,−1,0,
1,・・・・・・の系列を乗算し、周波数混合器19では正弦
波信号sin(πn/2)、つまり0,1,0,−1,
0,・・・・・・ の系列を乗算して、演算を頗る簡単なものと
することができる。これら0,1,−1の系列の周期は
1/f(2πf=ω) である。
In the orthogonal transformation unit 16, the output Asin (ωt + θ) of the distributor 15 and the cosine wave signal cos (πn / 2), that is, 1,0, −1,0,
Are multiplied by a series of 1,..., And the frequency mixer 19 sinusoidal signal sin (πn / 2), that is, 0, 1, 0, −1,
By multiplying the sequence of 0,..., The operation can be made very simple. The period of the series of 0, 1, and -1 is 1 / f (2πf = ω).

【0021】また帯域通過フィルタ14の出力をAD変
換器でデジタル系列とした後、直交変換を行ってもよ
い。上述において、図6Aに示すように、AD変換器2
3,24の出力をメモリ35に記憶し、そのメモリ35
の記憶データを演算処理部36で読出して、図1,図4,
図5で述べた複素乗算処理と、FFT処理をCPUやD
SPなどを用いたソフトウェア処理により行ってもよ
い。この場合図6Bに示すように、直交変換もデジタル
処理してメモリ35に格納してもよい。更には図6Cに
示すように、帯域通過フィルタ14の出力を直ちにデジ
タルデータ系列に変換してメモリ35に格納し、その後
の処理をすべてソフトウェア処理によって行ってもでき
る。
After the output of the band-pass filter 14 is converted into a digital sequence by an AD converter, orthogonal transformation may be performed. In the above description, as shown in FIG.
3 and 24 are stored in a memory 35, and the memory 35
Is read out by the arithmetic processing unit 36, and the data shown in FIGS.
The complex multiplication process and the FFT process described in FIG.
It may be performed by software processing using SP or the like. In this case, as shown in FIG. 6B, the orthogonal transform may be digitally processed and stored in the memory 35. Further, as shown in FIG. 6C, the output of the band-pass filter 14 may be immediately converted into a digital data sequence and stored in the memory 35, and all subsequent processing may be performed by software processing.

【0022】図6A,Bにおける演算処理部36におけ
る処理の流れの例を図7に示す。まずi=0に初期化し
(S1)、メモリ35のアドレスiに格納されているサン
プルデータから順次i+N−1まで読出す(S2)、そ
の各1サンプルデータの読出しごとに、そのサンプルZ
(m) と拡散符号の複素共役R*との複素乗算を行い(S
3)、その乗算結果をバッファに1次格納し(S4)、N
個のサンプルデータに対する乗算結果が得られると、こ
れらに対し、N点複素FFT演算を行い(S5)、その
演算結果の各周波数成分である係数の絶対値の二乗を求
め(S6)、データを表示か、又は格納かを判断し、つま
り、iを拡散符号の1周期相当分更新したかを調べ (S
7)、また一周期のデータを取得していなければ、格納
と判断して、S6の演算結果をメモリに格納しておき、
iを+1してステップS2に戻る(S8)。
FIG. 7 shows an example of the processing flow in the arithmetic processing section 36 in FIGS. 6A and 6B. First, initialize to i = 0
(S1) The sample data stored at the address i of the memory 35 is sequentially read out from the sample data to i + N-1 (S2). Each time one sample data is read, the sample Z is read.
(m) and the complex conjugate R * of the spreading code are subjected to complex multiplication (S
3) The result of the multiplication is primarily stored in a buffer (S4), and N
When the multiplication results for the sample data are obtained, an N-point complex FFT operation is performed on them (S5), and the square of the absolute value of the coefficient, which is each frequency component of the operation result, is obtained (S6). It is determined whether the data is displayed or stored, that is, it is checked whether or not i has been updated by one period of the spreading code (S
7) If one cycle of data has not been obtained, it is determined that the data is to be stored, and the calculation result of S6 is stored in the memory.
i is incremented by 1, and the process returns to step S2 (S8).

【0023】ステップS7で1周期分の所要データ量が
得られ、表示と判断されると、それまでに格納したステ
ップS6の演算結果中の最大ピークを求め(S9)、その
対応周波数を搬送波周波数誤差として、またタイミング
時間を位相ずれとしてそれぞれ表示する(S10)。ス
テップS9で図3B又はCに示したように表示させ、そ
の表示を操作員が見て、最大ピークを探し、そのピーク
にマーカを移動させて、その周波数誤差と、タイミング
時間を読取るようにしてもよい。
In step S7, the required data amount for one cycle is obtained, and when it is determined that the display is to be performed, the maximum peak in the calculation result of step S6 stored so far is obtained (S9), and the corresponding frequency is set to the carrier frequency. The error and the timing time are displayed as a phase shift (S10). In step S9, as shown in FIG. 3B or C, the operator looks at the display, searches for the maximum peak, moves the marker to the peak, and reads the frequency error and the timing time. Is also good.

【0024】[0024]

【発明の効果】以上述べたようにこの発明によれば、複
素FFT演算をしてその絶対値の二乗を求めることによ
り、局部発振周波数を順次変化させる必要がなく、時間
周期処理、つまり拡散符号信号と、直交変換されたデジ
タル信号の相対位相を順次ずらすことにより、拡散符号
の1周期で、搬送波周波数の誤差とタイミング時間とを
求めることができ、従来において、各周波数範囲ごとに
時間同期処理を行う場合と比較して、処理時間が著しく
短縮され、また周波数を順次変更させるためのハードウ
ェアが不要となり、ハードウェア規模も小さくなる。
As described above, according to the present invention, it is not necessary to sequentially change the local oscillation frequency by performing the complex FFT operation and obtaining the square of its absolute value. By sequentially shifting the relative phase between the signal and the orthogonally transformed digital signal, the error of the carrier frequency and the timing time can be obtained in one cycle of the spread code. Conventionally, time synchronization processing is performed for each frequency range. As compared with the case of performing the above, the processing time is remarkably reduced, hardware for sequentially changing the frequency is not required, and the hardware scale is reduced.

【0025】更に、図4,図5について示したように、
サンプリング周波数を、チップ周波数の整数J倍とする
場合、Jサンプルごとに処理を行えばよく、処理量が少
なくなる。ただし、その処理周期はチップ周期より大と
することはできない、最大でチップ周期ごとに処理すれ
ばよい。また直交変換を局部余弦波は1,0,−1,
0,1,・・・・・・ の系列とし、局部正弦波は0,1,
0,−1,0,・・・・・・ の系列とすることにより処理を単
純化することができる。
Further, as shown in FIGS. 4 and 5,
When the sampling frequency is set to an integer J times the chip frequency, the processing may be performed for each J sample, and the processing amount is reduced. However, the processing cycle cannot be longer than the chip cycle, and processing may be performed at a maximum for each chip cycle. In addition, the orthogonal cosine wave is transformed into 1, 0, -1,
.., And the local sine wave is 0, 1,
Processing can be simplified by using a sequence of 0, -1, 0,....

【図面の簡単な説明】[Brief description of the drawings]

【図1】この発明の装置の実施例を示す機能構成図。FIG. 1 is a functional configuration diagram showing an embodiment of an apparatus of the present invention.

【図2】図1中の遅延・複素乗算部30の具体例を示す
図。
FIG. 2 is a diagram showing a specific example of a delay / complex multiplication unit 30 in FIG. 1;

【図3】この発明の装置の各種表示例を示す図。FIG. 3 is a diagram showing various display examples of the device of the present invention.

【図4】Aはサンプルレートをチップレートの2倍にし
た場合の複素乗算における処理例の一部を示す図、Bは
図4Aに対し拡散符号が2値の場合の処理例の一部を示
す図である。
4A is a diagram illustrating a part of a processing example in complex multiplication when the sample rate is twice the chip rate, and FIG. 4B is a diagram illustrating a part of a processing example in the case where the spreading code is binary with respect to FIG. 4A. FIG.

【図5】図4Bの方法をハードウェアで構成した例を示
す図。
FIG. 5 is a diagram showing an example in which the method of FIG. 4B is configured by hardware.

【図6】この発明における処理手順の各種例を示す流れ
図。
FIG. 6 is a flowchart showing various examples of a processing procedure in the present invention.

【図7】図6Bに示した処理手順中の演算処理部36に
おける処理手順の例を示す流れ図。
FIG. 7 is a flowchart showing an example of a processing procedure in an arithmetic processing unit during the processing procedure shown in FIG. 6B.

Claims (12)

【特許請求の範囲】[Claims] 【請求項1】 入力スペクトラム拡散信号を直交変換す
る手段と、 前記直交変換された信号と、拡散符号信号の複素共役と
の積を求める手段と、 前記求めた積のデータ系列を離散的フーリエ変換する手
段と、 前記フーリエ変換された各係数値の絶対値の二乗値を求
める手段と、 を具備することを特徴とするスペクトラム拡散信号解析
装置。
A means for orthogonally transforming an input spread spectrum signal; a means for calculating a product of the orthogonally transformed signal and a complex conjugate of a spread code signal; a discrete Fourier transform of a data sequence of the obtained product And a means for calculating a square value of an absolute value of each of the Fourier-transformed coefficient values.
【請求項2】 前記各二乗値が、前記拡散符号信号の1
周期内における最大となる前記拡散符号信号の基準時間
に対するずれと、対応係数の周波数とを求めるピーク探
索手段をさらに備えている請求項1に記載のスペクトラ
ム拡散信号解析装置。
2. The method according to claim 1, wherein each square value is one of the spread code signals.
2. The spread spectrum signal analyzing apparatus according to claim 1, further comprising a peak searching means for obtaining a maximum shift within a cycle of the spread code signal from a reference time and a frequency of a corresponding coefficient.
【請求項3】 前記ピーク探索手段は前記二乗値を直交
軸の一方を周波数軸とし、他方を時間軸として表示する
表示手段であることを特徴とする請求項1又は2に記載
のスペクトラム拡散信号解析装置。
3. The spread spectrum signal according to claim 1, wherein said peak search means is display means for displaying said square value on one of an orthogonal axis as a frequency axis and the other on a time axis. Analysis device.
【請求項4】 前記表示手段が、前記二乗値のレベルを
前記直交軸の双方に垂直な軸を用いて3次元表示する請
求項3に記載のスペクトラム拡散信号解析装置。
4. The spread spectrum signal analyzer according to claim 3, wherein the display means displays the level of the square value three-dimensionally using an axis perpendicular to both of the orthogonal axes.
【請求項5】 前記表示手段が、前記二乗値のレベルを
輝度レベルによって表示する請求項3に記載のスペクト
ラム拡散信号解析装置。
5. The spread spectrum signal analyzing apparatus according to claim 3, wherein the display means displays the level of the square value by a luminance level.
【請求項6】 前記表示手段が、前記二乗値のレベル
を、当該二乗値の各レベルにそれぞれ対応する色によっ
て表示する請求項3に記載のスペクトラム拡散信号解析
装置。
6. The spread spectrum signal analyzing apparatus according to claim 3, wherein the display means displays the level of the square value in a color corresponding to each level of the square value.
【請求項7】 入カスペクトラム拡散信号をデジタルデ
ータ糸列に変換するAD変換工程と、 前記デジタルデータ系列を直交変換してメモリに一時格
納する直交変換工程と、 前記直交変換されたデータ系列と、拡散符号信号の複素
共役との各サンプルごとの複素乗算を行う乗算工程と、 前記複素乗算結果の拡散符号の周期と対応するサンプル
分について離散的フーリエ変換を行うフーリエ変換工程
と、 前記離散的フーリエ変換結果を各周波数成分ごとに絶対
値の二乗を演算する二乗演算工程と、 前記直交変換されたデータ系列と、前記拡散符号信号と
の相対位相をそのチップ周期以内で順次ずらして前記乗
算工程、前記フーリエ変換工程、前記二乗演算工程を、
前記相対位相のずれの和がほぼ拡散符号の1周期となる
まで繰り返す工程と、 前記二乗演算工程の演算結果から、前記拡散符号の入力
信号のそれに対するタイミングずれと、前記入力信号の
搬送波周波数誤差とを求める探索工程と、 を具備するスペクトラム拡散信号解析方法。
7. An AD conversion step of converting an input spread spectrum signal into a digital data string, an orthogonal conversion step of orthogonally converting the digital data sequence and temporarily storing the digital data sequence in a memory, A multiplication step of performing a complex multiplication for each sample with a complex conjugate of the spread code signal; a Fourier transform step of performing a discrete Fourier transform on a sample corresponding to the period of the spread code of the complex multiplication result; A square calculation step of calculating a square of an absolute value of a Fourier transform result for each frequency component; and a multiplication step of sequentially shifting a relative phase between the orthogonally transformed data sequence and the spread code signal within the chip period. , The Fourier transform step, the square operation step,
Repeating the step until the sum of the relative phase shifts is substantially equal to one cycle of the spreading code; and calculating the timing shift of the input signal of the spread code with respect to that of the input signal and the carrier frequency error of the input signal from the calculation result of the square calculation step And a search step for: determining a spread spectrum signal.
【請求項8】 前記AD変換工程におけるサンプリング
周波数を、前記拡散符号信号のチップ周波数の整数J倍
とし、前記乗算工程を、Jサンプルごとに実行すること
を特徴とする請求項7記載のスペクトラム拡散信号解析
方法。
8. The spread spectrum apparatus according to claim 7, wherein a sampling frequency in said AD conversion step is set to an integer J times a chip frequency of said spread code signal, and said multiplication step is performed for each J samples. Signal analysis method.
【請求項9】 前記拡散符号信号は2値の何れかをとる
系列であり、前記乗算工程は、拡散符号信号の2値の何
れかに応じて、直交変換されたデータの符号を変換する
か、変換することなく加算演算を行う工程であることを
特徴とする請求項7又は8記載のスペクトラム拡散信号
解析方法。
9. The spread code signal is a sequence that takes one of two values, and the multiplying step converts the code of the orthogonally transformed data according to one of the two values of the spread code signal. 9. The spread spectrum signal analyzing method according to claim 7, wherein the adding step is performed without conversion.
【請求項10】 前記探索工程は、直交軸の一方を周波
数軸、他方を時間(相対位相)軸として前記二乗値を表
示し、その最大のピークを探し、対応周波数と、時間と
を求める工程であることを特徴とする請求項7乃至9の
何れかに記載のスペクトラム拡散信号解析方法。
10. The searching step is a step of displaying the squared value with one of the orthogonal axes as a frequency axis and the other as a time (relative phase) axis, searching for the maximum peak, and finding a corresponding frequency and time. The spread spectrum signal analysis method according to any one of claims 7 to 9, wherein
【請求項11】 前記二乗値のレベルを輝度レベルによ
って表示する請求項10に記載のスペクトラム拡散信号
解析方法。
11. The spread spectrum signal analysis method according to claim 10, wherein the level of the square value is displayed by a luminance level.
【請求項12】 前記二乗値のレベルを、当該二乗値の
各レベルにそれぞれ対応する色によって表示する請求項
10に記載のスペクトラム拡散信号解析方法。
12. The spread spectrum signal analyzing method according to claim 10, wherein the level of the square value is displayed by a color corresponding to each level of the square value.
JP11026804A 1998-02-19 1999-02-04 Spread spectrum signal analysis device/method Pending JPH11308152A (en)

Priority Applications (1)

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Application Number Priority Date Filing Date Title
JP3710598 1998-02-19
JP10-37105 1998-02-19
JP11026804A JPH11308152A (en) 1998-02-19 1999-02-04 Spread spectrum signal analysis device/method

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Publication Number Publication Date
JPH11308152A true JPH11308152A (en) 1999-11-05

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JP2017069905A (en) * 2015-10-02 2017-04-06 アンリツ株式会社 Mobile terminal test apparatus, and local oscillation frequency detection method therefor
CN115499036A (en) * 2022-11-14 2022-12-20 北京航空航天大学合肥创新研究院(北京航空航天大学合肥研究生院) Parallel capturing method and storage medium for broadband spread spectrum signal

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US6518741B1 (en) 1999-11-16 2003-02-11 Anritsu Corporation Modulation analyzing apparatus with balance/imbalance converter
US8891414B2 (en) 2000-12-15 2014-11-18 Adaptix, Inc. Multi-carrier communications with adaptive cluster configuration and switching
US8934375B2 (en) 2000-12-15 2015-01-13 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US8934445B2 (en) 2000-12-15 2015-01-13 Adaptix, Inc. Multi-carrier communications with adaptive cluster configuration and switching
US8958386B2 (en) 2000-12-15 2015-02-17 Adaptix, Inc. Multi-carrier communications with adaptive cluster configuration and switching
US8964719B2 (en) 2000-12-15 2015-02-24 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US9191138B2 (en) 2000-12-15 2015-11-17 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US9203553B1 (en) 2000-12-15 2015-12-01 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US9210708B1 (en) 2000-12-15 2015-12-08 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US9219572B2 (en) 2000-12-15 2015-12-22 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
US9344211B2 (en) 2000-12-15 2016-05-17 Adaptix, Inc. OFDMA with adaptive subcarrier-cluster configuration and selective loading
JP2017069905A (en) * 2015-10-02 2017-04-06 アンリツ株式会社 Mobile terminal test apparatus, and local oscillation frequency detection method therefor
CN115499036A (en) * 2022-11-14 2022-12-20 北京航空航天大学合肥创新研究院(北京航空航天大学合肥研究生院) Parallel capturing method and storage medium for broadband spread spectrum signal
CN115499036B (en) * 2022-11-14 2023-02-24 北京航空航天大学合肥创新研究院(北京航空航天大学合肥研究生院) Parallel capturing method and storage medium for broadband spread spectrum signal

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