JPH08317639A - Ringing choke converter of synchronous rectification system - Google Patents
Ringing choke converter of synchronous rectification systemInfo
- Publication number
- JPH08317639A JPH08317639A JP7156632A JP15663295A JPH08317639A JP H08317639 A JPH08317639 A JP H08317639A JP 7156632 A JP7156632 A JP 7156632A JP 15663295 A JP15663295 A JP 15663295A JP H08317639 A JPH08317639 A JP H08317639A
- Authority
- JP
- Japan
- Prior art keywords
- mosfet
- current
- voltage
- diode
- saturable inductor
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Classifications
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Landscapes
- Dc-Dc Converters (AREA)
Abstract
Description
【0001】[0001]
【産業上の利用分野】本発明は電源装置の1つの方式て
あるリンギングチョークコンバータに関する。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a ringing choke converter which is one type of power supply device.
【0002】[0002]
【従来の技術】一般的なリンギングチョークコンバータ
は、入出力電圧が一定の条件の下では、オン期間とオフ
期間の比は一定で、出力電流の変化に対して発振周期を
変えることにより出力電圧が一定に保たれている。出力
電流が小さければ発振周期も短くなり、従ってオン期間
もオフ期間も各々短くなる。2. Description of the Related Art A general ringing choke converter has a constant ratio of an on period and an off period under a constant input / output voltage condition, and the output voltage is changed by changing an oscillation cycle with respect to a change in output current. Is kept constant. If the output current is small, the oscillation cycle becomes short, and therefore the ON period and the OFF period become short.
【0003】出力電流が最大値からゼロまで変化する負
荷条件ではオン期間が広い範囲に渡って変化するが、出
力電流がゼロに近づくに従って短くなり制御不能となり
やすい。そのため間欠発振や過電圧発生の問題を起こす
ことがある。Under load conditions in which the output current changes from the maximum value to zero, the ON period changes over a wide range, but as the output current approaches zero, it becomes shorter and control becomes more likely to be out of control. Therefore, problems such as intermittent oscillation and overvoltage may occur.
【0004】そこで本出願人は先に2次巻線に逆方向に
電流を流すことができるリンギングチョークコンバータ
を提供した(特願平6−294051)。図3はこの回
路を示すものである。Therefore, the present applicant previously provided a ringing choke converter capable of passing a current in the reverse direction in the secondary winding (Japanese Patent Application No. 6-294051). FIG. 3 shows this circuit.
【0005】[0005]
【発明が解決しようとする課題】図3において巻線11
cには出力電圧に比例する電圧が発生し、この電圧がM
OSFET14のスレッショルド電圧を越すと、MOS
FETは2次巻線11bによるフライバック電流が放出
した後も引き続きオン状態を保ち、逆にコンデンサ13
の電圧が2次巻線11bに加わる。そのため2次巻線1
1bにはフライバック電流の方向と反対向きの電流が流
れる。この電流はトランス11を逆方向に励磁する電流
となる。The winding 11 shown in FIG.
A voltage proportional to the output voltage is generated in c, and this voltage is M
When the threshold voltage of OSFET14 is exceeded, the MOS
The FET continues to be in the ON state even after the flyback current from the secondary winding 11b is discharged, and conversely the capacitor 13
Is applied to the secondary winding 11b. Therefore, the secondary winding 1
A current flows in the direction 1b in the direction opposite to the direction of the flyback current. This current becomes a current for exciting the transformer 11 in the opposite direction.
【0006】2次巻線11bを流れる励磁電流によって
トランス11に蓄積される励磁エネルギーは1次巻線1
1aに直列に接続されているスイッチング素子21が次
のサイクルでターンオンしたときに1次巻線11aを通
るフライバック電流となり、これがコンデンサ20に充
電エネルギーとして戻るので損失はほとんどない。The exciting energy accumulated in the transformer 11 by the exciting current flowing through the secondary winding 11b is the primary winding 1
When the switching element 21 connected in series to 1a turns on in the next cycle, it becomes a flyback current passing through the primary winding 11a, and this flyback current returns to the capacitor 20 as charging energy, so there is almost no loss.
【0007】また、2次巻線11bに流れる励磁電流は
補助巻線11cの電圧がMOSFET14のスレッショ
ルド電圧より小さくなるまで続く。そのため、出力電圧
はこのスレッショルド電圧に巻線11bと11cの巻線
比をかけた値にほぼ等しくなる。The exciting current flowing through the secondary winding 11b continues until the voltage of the auxiliary winding 11c becomes smaller than the threshold voltage of the MOSFET 14. Therefore, the output voltage becomes substantially equal to the threshold voltage multiplied by the winding ratio of the windings 11b and 11c.
【0008】このように図3に示した回路において、出
力電圧はMOSFET14のスレッショルド電圧によっ
て決まる。そのため1次巻線11aに直列に接続されて
いるMOSFET21の制御回路22は1次巻線を流れ
る励磁電流の最大値またはMOSFET21の最大オン
期間を制限するだけで良く出力電圧検出値によって帰還
制御機能を持つ必要はない。As described above, in the circuit shown in FIG. 3, the output voltage is determined by the threshold voltage of the MOSFET 14. Therefore, the control circuit 22 of the MOSFET 21 connected in series to the primary winding 11a only needs to limit the maximum value of the exciting current flowing through the primary winding or the maximum ON period of the MOSFET 21, and the feedback control function can be performed by the output voltage detection value. You don't have to have.
【0009】しかし、MOSFETがスレッショルド電
圧のごく近くでオン状態を保つためMOSFETのオン
抵抗を十分小さくすることができない。However, since the MOSFET maintains the ON state in the vicinity of the threshold voltage, the ON resistance of the MOSFET cannot be made sufficiently small.
【0010】そこで本発明は、補助巻線11cからMO
SFETのゲートに加わるパルスの幅を制御することに
より、出力電圧を一定に保つと同時にMOSFETのゲ
ートに加わるパルスの振幅をスレッショルド電圧より十
分高めに設定することを可能にしMOSFETのオン期
間の抵抗値を小さくし効率を改善することを目的として
いる。Therefore, according to the present invention, the auxiliary winding 11c is connected to the MO
By controlling the width of the pulse applied to the gate of the SFET, it is possible to keep the output voltage constant and at the same time set the amplitude of the pulse applied to the gate of the MOSFET sufficiently higher than the threshold voltage. Is aimed at improving efficiency.
【0011】[0011]
【課題を解決するための手段】上記の目的を達成するた
め、請求項1記載の発明において、補助巻線から2次側
整流ダイオードに並列接続されているMOSFETのゲ
ート・ソース間に加わるパルスの幅を抵抗と可飽和イン
ダクタとリセット信号制御回路によって制御する。In order to achieve the above object, in the invention according to claim 1, the pulse applied between the gate and source of the MOSFET connected in parallel with the secondary side rectifying diode from the auxiliary winding. The width is controlled by a resistor, a saturable inductor and a reset signal control circuit.
【0012】[0012]
【作用】請求項1の発明において可飽和インダクタが飽
和するまでの期間がMOSFETのオン期間にほぼ等し
く、またこのオン期間の間に2次巻線にはフライバック
電流と励磁電流が流れる。According to the first aspect of the present invention, the period until the saturable inductor is saturated is substantially equal to the ON period of the MOSFET, and the flyback current and the exciting current flow in the secondary winding during this ON period.
【0013】可飽和インダクタに補助巻線から加わる電
圧は可飽和インダクタに直列に接続されているダイオー
ドによって一方向だけであるため、この可飽和インダク
タが一度飽和すると、他の回路によってリセット信号が
加えられない限り、ほぼ短絡に近い状態を維持し、補助
巻線にパルス電圧が発生しても、抵抗によって降圧し、
MOSFETのゲート・ソース間にスレッショルド電圧
を越える電圧は加わらず、MOSFETはオン状態にな
り得ない。Since the voltage applied to the saturable inductor from the auxiliary winding is in only one direction by the diode connected in series with the saturable inductor, once the saturable inductor saturates, the reset signal is applied by another circuit. As long as it is not, it keeps a state close to a short circuit, and even if a pulse voltage is generated in the auxiliary winding, it is stepped down by a resistor,
The voltage exceeding the threshold voltage is not applied between the gate and the source of the MOSFET, and the MOSFET cannot be turned on.
【0014】補助巻線に直列に接続されている抵抗は、
可飽和インダクタが飽和した直後にわずかな期間ではあ
るが補助巻線から可飽和インダクタに流れる突入電流を
制限する。The resistor connected in series with the auxiliary winding is
Immediately after the saturable inductor is saturated, it limits the inrush current flowing from the auxiliary winding to the saturable inductor for a short period.
【0015】次にこの可飽和インダクタに他の回路によ
ってリセット信号が加えられると、MOSFETもリセ
ット信号の強さに応じた期間だけオン状態を保つ。Next, when a reset signal is applied to this saturable inductor by another circuit, the MOSFET also remains in the ON state for a period corresponding to the strength of the reset signal.
【0016】リセット信号の強さによってMOSFET
のオン期間が制御されるので、リセット信号の強さを出
力電圧検出値に応じて変えることにより出力電圧を一定
に保つことができる。Depending on the strength of the reset signal, the MOSFET
Since the ON period is controlled, the output voltage can be kept constant by changing the strength of the reset signal according to the output voltage detection value.
【0017】請求項2の発明において、MOSFETの
オン抵抗が十分小さく、2次巻線の電流によるドロップ
電圧が整流ダイオードの順方向電圧より小さければ、こ
の整流ダイオードを除くことができる。そして、過負荷
時または短絡時において、補助巻線の電圧が下がった状
態でMOSFETがオン状態にならない場合はMOSF
ETの寄生ダイオードが整流の働きをする。In the invention of claim 2, if the on-resistance of the MOSFET is sufficiently small and the drop voltage due to the current in the secondary winding is smaller than the forward voltage of the rectifying diode, this rectifying diode can be excluded. If the MOSFET does not turn on when the voltage of the auxiliary winding drops during overload or short circuit, MOSF
The ET parasitic diode acts as a rectifier.
【0018】[0018]
【実施例】図1は請求項1記載の発明の実施例を示す回
路図である。従来の例を示す図3の回路と同一または同
等な部分には同一の符号を与えた。FIG. 1 is a circuit diagram showing an embodiment of the invention described in claim 1. The same reference numerals are given to the same or equivalent portions as those of the circuit of FIG. 3 showing a conventional example.
【0019】図2は請求項2記載の発明の実施例を示す
回路図である。従来の例を示す図3の回路と同一または
同等な部分には同一の符号を与えた。FIG. 2 is a circuit diagram showing an embodiment of the invention described in claim 2. The same reference numerals are given to the same or equivalent portions as those of the circuit of FIG. 3 showing a conventional example.
【0020】図4は図1に示した回路の2次巻線11b
両端の電圧波形とMOSFET14のドレイン電流とダ
イオード12の電流の和の電流波形を同じ時間軸で測定
したものである。FIG. 4 shows the secondary winding 11b of the circuit shown in FIG.
The current waveforms of the voltage waveforms at both ends, the drain current of the MOSFET 14 and the current of the diode 12 are measured on the same time axis.
【0021】図1の回路において、リセット信号制御回
路15は出力電圧を検出し、その値が設定値より高い場
合はダイオード15fより出力される電流を大きくし、
またその値が設定値より低い場合はダイオード15fよ
り出力される電流を小さくする。ダイオード15fより
出力される電流は補助巻線11cの電圧がMOSFET
14を逆バイアスする方向に変わったときに可飽和イン
ダクタ17を通り、抵抗16及び補助巻線11cと2次
巻線11bを通って流れる。この電流が可飽和インダク
タ17をリセットする。すなわち、出力電圧が設定値よ
り高くなるとリセット電流が大きくなる。In the circuit of FIG. 1, the reset signal control circuit 15 detects the output voltage, and when the value is higher than the set value, increases the current output from the diode 15f,
When the value is lower than the set value, the current output from the diode 15f is reduced. As for the current output from the diode 15f, the voltage of the auxiliary winding 11c is MOSFET.
When the direction of reverse biasing 14 is changed, the current flows through the saturable inductor 17, the resistor 16, the auxiliary winding 11c, and the secondary winding 11b. This current resets the saturable inductor 17. That is, when the output voltage becomes higher than the set value, the reset current becomes large.
【0022】このリセット電流が大きいと、補助巻線1
1cの電圧がMOSFET14を順バイアスする方向に
変わったときに、可飽和インダクタ17が補助巻線11
cの電圧によって飽和に達するまでの時間が長くなり、
MOSFET17のオン期間も長くなる。When this reset current is large, the auxiliary winding 1
When the voltage of 1c changes to forward bias the MOSFET 14, the saturable inductor 17 causes the auxiliary winding 11 to move.
The voltage of c increases the time to reach saturation,
The ON period of the MOSFET 17 also becomes long.
【0023】MOSFET17のオン期間が長くなった
分だけ2次巻線11bを逆方向に流れる励磁電流が大き
くなる。この電流はコンデンサ13の放電によってまか
なわれるため出力電圧は下がる。このようにして出力電
圧は一定に保たれる。The exciting current flowing in the reverse direction through the secondary winding 11b increases as the ON period of the MOSFET 17 increases. Since this current is supplied by the discharge of the capacitor 13, the output voltage drops. In this way the output voltage is kept constant.
【0024】図4に示したMOSFET14のドレイン
電流波形の正の部分はフライバック電流であり、負の部
分は励磁電流である。出力電流がゼロとき、これら2つ
の電流の面積はほぼ半々になり、出力電流が最大のとき
は、励磁電流の面積がほぼゼロになる。The positive part of the drain current waveform of the MOSFET 14 shown in FIG. 4 is the flyback current, and the negative part is the exciting current. When the output current is zero, the areas of these two currents are almost half and when the output current is maximum, the area of the exciting current is almost zero.
【0025】フライバック電流と励磁電流の面積の合計
は出力電流がゼロから最大値まで変化する間は一定であ
り、各々の期間の合計も一定である。従って発振の周期
もほぼ一定となる。The total area of the flyback current and the exciting current is constant while the output current changes from zero to the maximum value, and the total of each period is also constant. Therefore, the oscillation cycle is also almost constant.
【0026】発振の周期がほぼ一定であるという点は一
般的なリンギングチョークコンバータと異なる。The fact that the oscillation period is almost constant is different from the general ringing choke converter.
【0027】図2の回路において、2次巻線に接続され
る整流回路はMOSFET17だけである。MOSFE
T17のオン抵抗が十分小さければ、コンバータの効率
を上げることが可能である。In the circuit of FIG. 2, the MOSFET 17 is the only rectifier circuit connected to the secondary winding. MOSFE
If the on resistance of T17 is sufficiently small, the efficiency of the converter can be increased.
【0028】また図2の回路において、可飽和インダク
タ17に供給するリセット信号を検出電圧と異なる出力
電圧より得ているが、このような応用は検出電圧が比較
的高いときに、リセット電流による電力損失を節約する
ときに有効である。Further, in the circuit of FIG. 2, the reset signal supplied to the saturable inductor 17 is obtained from the output voltage different from the detection voltage. In such an application, when the detection voltage is relatively high, the power by the reset current is increased. It is effective in saving loss.
【0029】[0029]
【発明の効果】以上のようにこの発明によれば、発振周
波数がほぼ一定となる動作の安定した、かつ効率の高い
コンバータが簡素な構成でできた。As described above, according to the present invention, a converter having a stable operation and a high efficiency, in which the oscillation frequency is substantially constant, can be formed with a simple structure.
【図1】請求項1記載の発明の実施例である。FIG. 1 is an embodiment of the invention described in claim 1.
【図2】請求項2記載の発明の実施例である。FIG. 2 is an embodiment of the invention described in claim 2;
【図3】従来の回路図である。FIG. 3 is a conventional circuit diagram.
【図4】図1に示した回路の動作波形である。FIG. 4 is an operation waveform of the circuit shown in FIG.
11 トランス 12 ダイオード 13 コンデンサ 14 MOSFET 15 リセット信号制御回路 16 抵抗 17 可飽和インダクタ 18 ダイオード 20 コンデンサ 21 MOSFET 22 ゲート制御回路 23 ダイオード 24 コンデンサ 11a 1次巻線 11b 2次巻線 11c 補助巻線 11d 正帰還巻線 11e 第2の2次巻線 15a 電圧検出用IC 15b 抵抗 15c 抵抗 15d 抵抗 15e 抵抗 15f ダイオード 15g トランジスタ 11 Transformer 12 Diode 13 Capacitor 14 MOSFET 15 Reset Signal Control Circuit 16 Resistor 17 Saturable Inductor 18 Diode 20 Capacitor 21 MOSFET 22 Gate Control Circuit 23 Diode 24 Capacitor 11a Primary Winding 11b Secondary Winding 11c Auxiliary Winding 11d Positive Feedback Winding 11e Second secondary winding 15a Voltage detection IC 15b Resistor 15c Resistor 15d Resistor 15e Resistor 15f Diode 15g Transistor
Claims (2)
と、前記トランスの2次巻線に直列に接続されたダイオ
ードを備えたリンギンクチョークコンバータにおいて、
前記ダイオードにMOSFETを並列接続し、前記トラ
ンスに補助巻線を付加し端子を前記MOSFETのゲー
トとソースに各々接続し、前記補助巻線と前記MOSF
ETのゲートとソースを結ぶ回路に抵抗を直列に挿入
し、前記MOSFETのゲート・ソース間に可飽和イン
ダクタとダイオードからなる直列回路を接続し、かつ出
力電圧検出値に応じたリセット信号を前記可飽和インダ
クタに供給するリセット信号制御回路を接続したことを
特徴とする同期整流方式のリンギングチョークコンバー
タ。1. A ringing choke converter comprising a transformer having a primary winding and a secondary winding, and a diode connected in series to the secondary winding of the transformer,
A MOSFET is connected in parallel to the diode, an auxiliary winding is added to the transformer, terminals are connected to the gate and the source of the MOSFET, respectively, and the auxiliary winding and the MOSF are connected.
A resistor is inserted in series in the circuit connecting the gate and source of the ET, a series circuit composed of a saturable inductor and a diode is connected between the gate and source of the MOSFET, and the reset signal corresponding to the output voltage detection value is applied to the circuit. A ringing choke converter of a synchronous rectification type, characterized in that a reset signal control circuit for supplying to a saturation inductor is connected.
ードを削除したことを特徴とする請求項1記載の同期整
流方式のリンギングチョークコンバータ。2. A synchronous rectification type ringing choke converter according to claim 1, wherein a diode connected in series to said secondary winding is deleted.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP15663295A JP3427280B2 (en) | 1995-05-19 | 1995-05-19 | Ringing choke converter with synchronous control |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP15663295A JP3427280B2 (en) | 1995-05-19 | 1995-05-19 | Ringing choke converter with synchronous control |
Publications (2)
Publication Number | Publication Date |
---|---|
JPH08317639A true JPH08317639A (en) | 1996-11-29 |
JP3427280B2 JP3427280B2 (en) | 2003-07-14 |
Family
ID=15631931
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP15663295A Expired - Fee Related JP3427280B2 (en) | 1995-05-19 | 1995-05-19 | Ringing choke converter with synchronous control |
Country Status (1)
Country | Link |
---|---|
JP (1) | JP3427280B2 (en) |
Cited By (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP1120896A2 (en) * | 2000-01-28 | 2001-08-01 | Densei-Lambda K.K. | Resonant power converter |
EP1195883A3 (en) * | 2000-10-06 | 2004-09-29 | Salcomp OY | Control circuit for rectification |
WO2005034325A1 (en) * | 2003-09-30 | 2005-04-14 | Sanken Electric Co., Ltd. | Switching power source device |
US10754366B2 (en) | 2018-06-06 | 2020-08-25 | L3 Cincinnati Electronics Corporation | Power switching circuits having a saturable inductor |
-
1995
- 1995-05-19 JP JP15663295A patent/JP3427280B2/en not_active Expired - Fee Related
Cited By (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP1120896A2 (en) * | 2000-01-28 | 2001-08-01 | Densei-Lambda K.K. | Resonant power converter |
EP1120896A3 (en) * | 2000-01-28 | 2002-01-23 | Densei-Lambda K.K. | Resonant power converter |
US6483721B2 (en) | 2000-01-28 | 2002-11-19 | Densei-Lambda K.K. | Resonant power converter |
EP1195883A3 (en) * | 2000-10-06 | 2004-09-29 | Salcomp OY | Control circuit for rectification |
WO2005034325A1 (en) * | 2003-09-30 | 2005-04-14 | Sanken Electric Co., Ltd. | Switching power source device |
US7372710B2 (en) | 2003-09-30 | 2008-05-13 | Sanken Electric Co., Ltd. | Switching power source device of the type capable of controlling power loss in generating output voltage from a secondary winding of a transformer |
US10754366B2 (en) | 2018-06-06 | 2020-08-25 | L3 Cincinnati Electronics Corporation | Power switching circuits having a saturable inductor |
Also Published As
Publication number | Publication date |
---|---|
JP3427280B2 (en) | 2003-07-14 |
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