JP6494869B1 - Radar equipment - Google Patents

Abstract

  In the plurality of range Doppler data generated by the range Doppler data generation unit (6), a phase difference corresponding to the frequency difference between the plurality of transmission signals is calculated based on the distance error to the target caused by the relative speed with the target. A phase correction unit (7) for correcting each is provided, and the target detection unit (8) detects the target from a plurality of range Doppler data whose phase difference is corrected by the phase correction unit (7).

Description

  The present invention relates to a radar apparatus that detects a target.

A MIMO (Multiple-Input Multiple-Output) radar technique is known as a technique in which a radar apparatus forms a beam and detects a target.
In the MIMO radar technology, after a plurality of antennas transmit signals orthogonal to each other toward a target, the plurality of antennas each receive a reflected signal that is a signal reflected by the target.
The MIMO radar technique is a technique for demodulating a plurality of received reflected signals, forming a beam by combining a plurality of demodulated signals, and detecting a target from the formed beam.

As signals orthogonal to each other, signals having different frequencies can be considered.
A method in which the radar apparatus uses signals orthogonal to each other is called a frequency division method, and the following Patent Document 1 discloses a basic configuration of the frequency division method.
In the frequency division method, since the frequency differs between the transmission signals, it is necessary to correct the phase of each demodulated signal according to the distance to the target.
In the radar apparatus, as a method of increasing the resolution of the distance to the target, a method of performing frequency modulation on each transmission signal is known.
As a method of performing frequency modulation on each transmission signal, there are an FMCW (Frequency Modulation Continuous Wave) method, an LFM (Linear Frequency Modulation) pulse compression method, and the like.

JP 2016-136116 A

When the frequency division method used in the conventional radar apparatus and the FMCW method or the LFM pulse compression method are used in combination, if the target moves, a distance shift occurs according to the relative speed with respect to the target. The distance shift corresponds to a distance error to the target.
If a distance shift occurs, an error is included in the distance to the target, so even if phase correction of each demodulated signal is performed according to the distance to the target, it cannot be corrected correctly. There has been a problem that target detection accuracy deteriorates.

  The present invention has been made in order to solve the above-described problems. A radar capable of suppressing deterioration in detection accuracy of a target even when a method for increasing the resolution of a distance to a target and a frequency division method are used in combination. The object is to obtain a device.

A radar apparatus according to the present invention includes a transmission signal generation unit that generates a plurality of transmission signals having different frequencies and modulated frequencies, and a plurality of transmission antennas at different positions, and generates a transmission signal. A signal transmission unit that radiates each of the plurality of transmission signals generated by the unit to the space in a synchronized state from the transmission antenna, and a signal that is reflected by the target of each of the plurality of transmission signals radiated by the signal transmission unit A signal receiving unit that receives each reflected signal and a range Doppler that generates range Doppler data that is distance and velocity axis data by Fourier transforming each of the plurality of reflected signals received by the signal receiving unit. In a plurality of range Doppler data generated by the data generation unit and the range Doppler data generation unit, A phase correction unit that corrects a phase difference corresponding to a frequency difference between a plurality of transmission signals based on a distance error to a target caused by speed versus a plurality of range Dopplers in which the phase difference is corrected by the phase correction unit It has a MIMO beamformer that corrects the spatial angle difference determined by the target angle, transmit antenna position, and receive antenna position with respect to data, and detects the target from the beam formed by the MIMO beamformer. And a target detection unit.

  According to the present invention, in the plurality of range Doppler data generated by the range Doppler data generation unit, it corresponds to the frequency difference between the plurality of transmission signals based on the distance error to the target caused by the relative speed with the target. A phase correction unit that corrects each phase difference is provided, and the target detection unit is configured to detect a target from a plurality of range Doppler data whose phase difference is corrected by the phase correction unit. Even if the enhancement method and the frequency division method are used in combination, there is an effect that deterioration in target detection accuracy can be suppressed.

It is a block diagram which shows the radar apparatus by Embodiment 1 of this invention. It is a hardware block diagram which shows the hardware structure of the signal processing part 5 of the radar apparatus by Embodiment 1 of this invention. It is a hardware block diagram of a computer in case the signal processor 5 is realized by software or firmware. It is a flowchart which shows the operation | movement content of the radar apparatus containing the process sequence in case the signal processing part 5 is implement | achieved by software or firmware. It is explanatory drawing which shows the some transmission signal produced | generated by the transmission signal production | generation part 1. FIG. It is explanatory drawing which shows the coordinate system of space. It is a block diagram which shows the radar apparatus by Embodiment 2 of this invention. It is explanatory drawing which shows the some transmission signal produced | generated by the transmission signal production | generation part.

  Hereinafter, in order to explain the present invention in more detail, modes for carrying out the present invention will be described with reference to the accompanying drawings.

Embodiment 1 FIG.
1 is a block diagram showing a radar apparatus according to Embodiment 1 of the present invention.
FIG. 2 is a hardware configuration diagram showing a hardware configuration of the signal processing unit 5 of the radar apparatus according to Embodiment 1 of the present invention.
1 and 2, the transmission signal generation unit 1 generates a plurality of transmission signals having different frequencies and modulated frequencies, and sends the plurality of transmission signals to the signal transmission unit 2 and the reception processing unit 4. Output.

The signal transmission unit 2 includes N (N is an integer of 2 or more) transmission antennas 2-0 to 2- (N-1), and each of a plurality of transmission signals output from the transmission signal generation unit 1 To the space.
The transmission antennas 2-0 to 2- (N-1) may be element antennas such as a half-wave dipole antenna, or may be subarray antennas configured with several element antennas.
The signal receiving unit 3 includes M (M is an integer of 2 or more) receiving antennas 3-0 to 3- (M-1), and each of a plurality of transmission signals radiated by the signal transmitting unit 2 is provided. A reflected signal that is a signal reflected by the target is received, and a plurality of reflected signals are output to the reception processing unit 4.
Receiving antennas 3-0 to 3- (M-1) may be element antennas such as a half-wave dipole antenna, or may be subarray antennas composed of several element antennas.
FIG. 1 shows an example in which the radar apparatus includes a signal transmission unit 2 and a signal reception unit 3, but by including a transmission / reception antenna that combines radiation of a plurality of transmission signals and reception of a plurality of reflected signals, The signal transmitter 2 and the signal receiver 3 may be integrated.

The reception processing unit 4 includes a frequency conversion unit 4a and an analog-digital converter (hereinafter referred to as “A / D converter”) 4b.
The frequency conversion unit 4 a converts the frequencies of the plurality of reflected signals output from the signal receiving unit 3 using the plurality of transmission signals output from the transmission signal generation unit 1, respectively.
The A / D converter 4b converts each of the plurality of reflected signals whose frequencies have been converted by the frequency converter 4a from analog signals to digital signals, and outputs the converted digital signals to the signal processor 5 as received signals. To do.

The signal processing unit 5 includes a range Doppler data generation unit 6, a phase correction unit 7, and a target detection unit 8.
The range Doppler data generation unit 6 is realized by, for example, a range Doppler data generation circuit 11 illustrated in FIG.
The range Doppler data generation unit 6 generates range Doppler data that is distance and velocity axis data by performing Fourier transform on each of the plurality of reception signals output from the A / D converter 4b of the reception processing unit 4. Then, a process of outputting a plurality of range Doppler data to the phase correction unit 7 is performed.

The phase correction unit 7 is realized by, for example, the phase correction circuit 12 illustrated in FIG.
The phase correction unit 7 corresponds to the frequency difference between the plurality of transmission signals based on the distance error to the target caused by the relative speed with the target in the plurality of range Doppler data output from the range Doppler data generation unit 6. The phase difference to be corrected is corrected, and a process of outputting a plurality of range Doppler data after the phase difference correction to the target detection unit 8 is performed.

The target detection unit 8 includes a MIMO beam forming unit 9 and a target detection processing unit 10, and performs processing for detecting a target from a plurality of range Doppler data after phase difference correction output from the phase correction unit 7.
The MIMO beam forming unit 9 is realized by, for example, the beam forming circuit 13 shown in FIG.
The MIMO beam forming unit 9 multiplies a plurality of range Doppler data after phase difference correction output from the phase correction unit 7 by a coefficient for correcting a spatial phase difference described later, thereby forming a MIMO beam. To implement.
The target detection processing unit 10 is realized by, for example, a target detection processing circuit 14 shown in FIG.
The target detection processing unit 10 performs a process of detecting a target by applying, for example, a CFAR (Constant False Alarm Ratio) process to the MIMO beam formed by the MIMO beam forming unit 9.

In FIG. 1, each of the range Doppler data generation unit 6, the phase correction unit 7, the MIMO beam forming unit 9, and the target detection processing unit 10, which are components of the signal processing unit 5 in the radar apparatus, is dedicated as shown in FIG. 2. It is assumed that it is realized with hardware. That is, what is realized by the range Doppler data generation circuit 11, the phase correction circuit 12, the beam forming circuit 13, and the target detection processing circuit 14 is assumed.
Here, the range Doppler data generation circuit 11, the phase correction circuit 12, the beam forming circuit 13, and the target detection processing circuit 14 are, for example, a single circuit, a composite circuit, a programmed processor, a parallel programmed processor, an ASIC (Application). A specific integrated circuit (FPGA), a field-programmable gate array (FPGA), or a combination thereof is applicable.

The components of the signal processing unit 5 are not limited to those realized by dedicated hardware, and the signal processing unit 5 may be realized by software, firmware, or a combination of software and firmware. Good.
Software or firmware is stored as a program in the memory of a computer. The computer means hardware that executes a program, for example, a CPU (Central Processing Unit), a central processing unit, a processing unit, a processing unit, a microprocessor, a microcomputer, a processor, or a DSP (Digital Signal Processor) To do.
FIG. 3 is a hardware configuration diagram of a computer when the signal processing unit 5 is realized by software or firmware.

When the signal processing unit 5 is realized by software or firmware, a program for causing the computer to execute the processing procedures of the range Doppler data generation circuit 11, the phase correction circuit 12, the beam forming circuit 13, and the target detection processing circuit 14 is stored in the memory. 32, and the processor 31 of the computer may execute the program stored in the memory 32.
FIG. 4 is a flowchart showing the operation content of the radar apparatus including a processing procedure when the signal processing unit 5 is realized by software or firmware.

Next, the operation will be described.
In the first embodiment, an example will be described in which the FMCW method is used as a method for increasing the resolution of the distance to the target, and the FMCW method and the frequency division method are used in combination.
The transmission signal generation unit 1 generates a plurality of transmission signals whose frequencies are different from each other and whose frequencies are modulated (step ST1 in FIG. 4). The plurality of transmission signals are FMCW signals that are frequency-modulated by the FMCW method.
The transmission signal generation unit 1 outputs the generated transmission signals to the signal transmission unit 2 and the reception processing unit 4.

Here, FIG. 5 is an explanatory diagram showing a plurality of transmission signals generated by the transmission signal generation unit 1.
In FIG. 5, the horizontal axis represents time t and the vertical axis represents frequency f.
f _c is the minimum frequency of the transmission signal output from the transmission signal generation unit 1 to the transmission antenna 2-0 of the signal transmission unit 2, and f _off is the transmission antenna 2-n (n = 0, n from the transmission signal generation unit 1). 1,..., N-2) and the difference between the frequency of the transmission signal output from the transmission signal generator 1 to the transmission antenna 2- (n + 1) (hereinafter referred to as the offset frequency). Called).
T _WRI is the sweep time that is the chirp width, and B is the sweep bandwidth.
In FIG. 5, sweeps corresponding to the respective sweep periods are distinguished as 0th sweep, 1st sweep,..., (L−1) sweep.

The transmission antennas 2-0 to 2- (N-1) of the signal transmission unit 2 radiate each of the N transmission signals output from the transmission signal generation unit 1 into space (step ST2 in FIG. 4).
The N transmission signals radiated to the space from the transmission antennas 2-0 to 2- (N-1) are reflected by the target and then returned to the radar apparatus as reflected signals.
The receiving antennas 3-0 to 3- (M-1) of the signal receiving unit 3 receive each of the N reflected signals that are signals reflected by the target, and send the N reflected signals to the reception processing unit 4. Output (step ST3 in FIG. 4).
Since the number of transmission signals is N and the number of reception antennas 3-0 to 3- (M-1) is M, N × M reflected signals are received and N × M reflected signals are received. Is output to the reception processing unit 4.

The frequency conversion unit 4a of the reception processing unit 4 uses the N transmission signals output from the transmission signal generation unit 1 to change the frequencies of the N × M reflected signals output from the signal reception unit 3, respectively. Conversion is performed (step ST4 in FIG. 4).
For example, paying attention to the reflected signal received by the receiving antenna 3-m (m = 0, 1,..., M−1), the frequency conversion unit 4a distributes the reflected signal to N signals and distributes the reflected signal. Of the N signals, the n (n = 0, 1,..., N−1) -th signal is transmitted from the transmission signal generator 1 to the transmission antenna 2-n (n = 0, 1,... .., N-1) is multiplied by the output transmission signal to convert the frequency of the nth signal.

式（１）において、ｃは光速である。
ただし、非特許文献１では、送信信号と反射信号の周波数差であるビート周波数が示されているが、後段の位相補正部７の説明のため、式（１）では、送信信号の周波数による位相項も含めて記載している。また、送信アンテナ２−ｎ、受信アンテナ３−ｍ及び目標との角度による空間位相差については、後述する式（４）に含めるものとして、式（１）には含めていない。
［非特許文献１］W.L.Melvin,他, PRINCIPLES OF MODERN RADAR;Vol.3RADAR APPLICATIONS,Scitech,2014.For example, 0 if the distance to the target at a sweep th relative velocity v _D between the target with R, 0 sweep th frequency-converted corresponding to a transmission signal which is radiated into space from the transmission antenna 2-n The reflection signal s _{Rx, n, 0} (t) is expressed by the following equation (1) in consideration of the following non-patent document 1.

In the formula (1), c is the speed of light.
However, in Non-Patent Document 1, the beat frequency that is the frequency difference between the transmission signal and the reflected signal is shown. However, for the description of the phase correction unit 7 in the subsequent stage, in Equation (1), the phase depending on the frequency of the transmission signal is shown. Including the section. In addition, the spatial phase difference depending on the angle with the transmitting antenna 2-n, the receiving antenna 3-m, and the target is included in Expression (4) described later, but is not included in Expression (1).
[Non-Patent Document 1] WLMelvin, et al., PRINCIPLES OF MODERN RADAR; Vol.3RADAR APPLICATIONS, Scitech, 2014.

式（２）において、スイープ間での目標の距離移動は、距離分解能と比べて小さいため、無視できるものとしている。In the l (l = 0, 1,..., L-1) sweep, the distance to the target is Rv _D 1T _WRI, and thus the reflected signal s _{Rx, n} after frequency conversion in the l sweep _{, L} (t) is expressed as the following equation (2).

In equation (2), the target distance movement between sweeps is small compared to the distance resolution, so that it can be ignored.

The A / D converter 4b of the reception processing unit 4 converts each of the plurality of reflected signals whose frequencies are converted by the frequency conversion unit 4a from analog signals to digital signals, and uses the converted digital signals as reception signals. 5 (step ST5 in FIG. 4).
Here, when the sampling period of the A / D converter 4b is δt and t = kδt (k = 0 to K−1) is substituted into the equation (2), the received signal s _{Rx, n, l} at the l-th sweep is obtained. [K, l] is expressed as the following equation (3).

式（４）において、Ａ＋Ｂが、空間位相差となる。The spatial phase difference due to the transmission antenna 2-n, the reception antenna 3-m, and the target angle is considered in the spatial coordinate system shown in FIG. FIG. 6 is an explanatory diagram showing a coordinate system of space.
Assuming that the unit direction vector of the target direction is i, the position vector of the transmission antenna 2-n is d _n ^Tx , and the position vector of the reception antenna 3-m is d _m ^Rx , the transmission antenna 2-n, the reception antenna 3-m, and the target The spatial phase difference depending on the angle is expressed as the following equation (4).

In Expression (4), A + B is a spatial phase difference.

この実施の形態１では、式（５）のように表される空間位相差を含む受信信号ｓ_Ｒｘ［ｎ，ｍ，ｋ，ｌ］が、受信処理部４のＡ／Ｄ変換器４ｂから信号処理部５に出力される。By multiplying equation (4) by equation (3), a received signal s _Rx [n, m, k, l] including a spatial phase difference is obtained. The received signal s _Rx [n, m, k, l] including the spatial phase difference is expressed as the following equation (5).

In the first embodiment, the received signal s _Rx [n, m, k, l] including the spatial phase difference expressed by the equation (5) is received from the A / D converter 4b of the reception processing unit 4. It is output to the processing unit 5.

The range Doppler data generation unit 6 of the signal processing unit 5 performs Fourier transform on the reception signal s _Rx [n, m, k, l] output from the A / D converter 4b of the reception processing unit 4 to obtain the distance and Range Doppler data as velocity axis data is generated, and the range Doppler data is output to the phase correction unit 7 (step ST6 in FIG. 4).
That is, the range Doppler data generation unit 6 uses the received signal s _Rx [n, m, k, l] output from the A / D converter 4b of the reception processing unit 4 in the k direction (within the sweep) ) To obtain the distance dimension data.
Further, the range Doppler data generation unit 6 obtains velocity dimension data by Fourier-transforming the received signal Fourier-transformed in the k-direction in the l-direction (between sweeps) shown in Equation (5).
As a result, range Doppler data, which is distance and velocity axis data, is obtained.

Hereinafter, the range Doppler data generation processing by the range Doppler data generation unit 6 will be described in detail.
Expression (5) is expressed as the following Expression (6) using variables.

式（８）において、ｋ_ｆは、レンジビンを示す番号、ｌ_ｆは、ドップラビンを示す番号である。
また、Ｋ_ＦＦＴは、ｋ方向のフーリエ変換を行う際にゼロパッティングしたＦＦＴ点数、Ｌ_ＦＦＴは、ｌ方向のフーリエ変換を行う際にゼロパッティングしたＦＦＴ点数である。ＵＰチャープの場合、ビート周波数は負の周波数となる。
ここでは、目標までの距離が遠くなるほど、レンジビンを示す番号ｋ_ｆが増加すると定義し、ｌは、ドップラ周波数の正負も考慮して、ｌ＝−Ｌ／２〜Ｌ／２−１としている。When Expression (6) is Fourier-transformed in the k direction and Fourier-transformed in the l direction, the following Expression (8) is obtained.

In the formula (8), _{k f} is a number indicating the range bin, _{l f} is a number indicating the Doppler bin.
Further, K _FFT is the number of FFT points zero-padded when performing k-direction Fourier transform, and L _FFT is the number of FFT points zero-padded when performing l-direction Fourier transform. In the case of UP chirp, the beat frequency is a negative frequency.
Here, as the distance to the target is far, defined as number k _f indicating the range bin is increased, l, even positive and negative Doppler frequency in consideration, and the l = -L / 2~L / 2-1.

When Expression (7) is substituted into Expression (8), the range Doppler data s _b [n, m, k _f , l _f ] is expressed as the following Expression (9).

The phase correction unit 7 of the signal processing unit 5 shifts the distance in the range Doppler data s _b [n, m, k _f , l _f ] output from the range Doppler data generation unit 6 according to the relative speed with the target. Based on the above, the phase difference corresponding to the offset frequency f _off is corrected (step ST7 in FIG. 4).
The phase correction unit 7 outputs the range Doppler data after the phase difference correction to the target detection unit 8.

Hereinafter, the phase difference correction processing for the range Doppler data by the phase correction unit 7 will be specifically described.
The MIMO beam forming unit 9 at the subsequent stage forms a MIMO beam by multiplying the range Doppler data after the phase difference correction by the phase correction unit 7 by a coefficient for correcting the spatial phase difference.
However, in the equation (9) indicating the range Doppler data s _b [n, m, k _f , l _f ], the first exp term is N transmission antennas 2-0 to 2- (N−1). This indicates that the phases of the N transmission signals are different because the frequencies of the N transmission signals radiated from are different from each other by the offset frequency f _off .
Therefore, after a transmission signal is radiated from each of the N transmission antennas 2-0 to 2- (N-1), a phase error occurs between a plurality of reflected signals that are transmission signals reflected by the target. Therefore, a plurality of reflected signals cannot be stacked coherently.
Therefore, it is necessary to correct the phase difference corresponding to the offset frequency f _off in the range Doppler data s _b [n, m, k _f , l _f ] output from the range Doppler data generation unit 6.

式（１０）において、Ｒは、目標までの距離である。
目標までの距離Ｒは、事前に求めることができないため、レンジビンｋ_ｆ毎に位相を補正する。
式（９）より、目標までの距離Ｒと、ビート周波数ビンとの関係は、以下の式（１１）で表される。

The phase difference corresponding to the offset frequency f _off is obtained by multiplying the range Doppler data s _b [n, m, k _f , l _f ] shown in the equation (9) by the correction coefficient w shown in the following equation (10). It can be corrected.

In equation (10), R is the distance to the target.
The distance to the target R is can not be determined in advance, to correct the phase for each range bin k _f.
From the equation (9), the relationship between the distance R to the target and the beat frequency bin is expressed by the following equation (11).

式（１２）を式（１１）に代入し、その代入結果を式（１０）に代入することで、ドップラ周波数毎に補正係数ｗを補正する。
以下の式（１３）は、補正後の補正係数ｗ［ｎ，ｍ，ｋ_ｆ，ｌ_ｆ］を表している。

However, as can be seen from equation (11), the distance R to the target, since that depends on the relative velocity v _D of the target, it is necessary to change the correction coefficients w for every Doppler frequency.
From the equation (9), the relationship between the target relative speed v _D and the Doppler bin l _f is expressed by the following equation (12).

By substituting equation (12) into equation (11) and substituting the substitution result into equation (10), the correction coefficient w is corrected for each Doppler frequency.
The following equation (13) represents a corrected correction coefficient w [n, m, k _f , l _f ].

The phase correction unit 7 uses the range Doppler data s _b [n, m, k _f , output from the range Doppler data generation unit 6 using the correction coefficient w [n, m, k _f , l _f ] shown in Expression (13). By multiplying l _f ], the phase difference corresponding to the offset frequency f _off is corrected.
Thereby, even if a distance shift occurs according to the relative speed v _D with respect to the target, the phase difference corresponding to the offset frequency f _off can be accurately corrected.

式（１４）及び式（１５）において、ｉ_ｂは、ビーム指向方向の単位方向ベクトルである。
φ_ｂは、ビーム指向方向の方位角、θ_ｂは、ビーム指向方向の仰角である。The MIMO beam forming unit 9 of the target detection unit 8 uses a weight represented by the following equation (14) that is a coefficient for correcting the spatial phase difference for the range Doppler data after the phase difference correction output from the phase correction unit 7. By multiplying a _MIMO , a MIMO beam s _MIMO [k _f , l _f , θ _b , φ _b ] shown in the following equation (15) is formed (step ST8 in FIG. 4).

In Expressions (14) and (15), i _b is a unit direction vector in the beam pointing direction.
φ _b is an azimuth angle in the beam pointing direction, and θ _b is an elevation angle in the beam pointing direction.

The target detection processing unit 10 of the target detection unit 8 applies, for example, CFAR processing to the MIMO beam s _MIMO [k _f , l _f , θ _b , φ _b ] formed by the MIMO beam forming unit 9. Then, the process of detecting the target is performed (step ST9 in FIG. 4).
The CFAR process is disclosed in Non-Patent Document 2 below, for example.
[Non-Patent Document 2] Chen, VC, "Time-Frequency transforms for Radar Imaging and Signal Analysis", ISBN-10: 1580532888, 1 January 2002.

  As apparent from the above, according to the first embodiment, in the plurality of range Doppler data generated by the range Doppler data generation unit 6, a plurality of distance Doppler data are generated based on the distance error to the target caused by the relative speed with the target. The phase correction unit 7 for correcting the phase difference corresponding to the frequency difference between the transmission signals is provided, and the target detection unit 8 detects the target from a plurality of range Doppler data whose phase difference is corrected by the phase correction unit 7. Therefore, even if the method for increasing the resolution of the distance to the target and the frequency division method are used in combination, it is possible to suppress the deterioration of the detection accuracy of the target.

In this Embodiment 1, the example which uses a FMCW system is shown as a system which raises the resolution of the distance to a target.
However, the present invention is not limited to this, and for example, an FMICW (Frequency Modulated Interrupted Continuous Wave) method may be used.
The FMICW method is a method of pulsing an FMCW signal that has been frequency-modulated by the FMCW method and receiving a reflected signal of a pulse reflected by a target during a period in which no pulse is emitted from the transmission antenna 2-n. .
Even when the FMICW method is used, the processing content of the signal processing unit 5 does not change. The FMICW method is disclosed in Non-Patent Document 3, for example.
[Non-Patent Document 3] Naka Isao, et al., Ship detection experiment using short-wave surface radar, IEICE Transactions B Vol.J82-B, No.3, pp.461-168, 1999.

Embodiment 2. FIG.
In the first embodiment, an example using the FMCW method is shown as a method for increasing the resolution of the distance to the target.
In the second embodiment, an example in which the LFM pulse compression method is used as a method for increasing the resolution of the distance to the target will be described.

FIG. 7 is a block diagram showing a radar apparatus according to Embodiment 2 of the present invention.
In FIG. 7, the same reference numerals as those in FIG.
The transmission signal generation unit 21 generates a plurality of transmission signals whose frequencies are different from each other and whose frequencies are modulated, and outputs the plurality of transmission signals to the signal transmission unit 2 and the reception processing unit 4.
The plurality of transmission signals are LFM pulse signals that are frequency-modulated by the LFM pulse compression method.

FIG. 8 is an explanatory diagram showing a plurality of transmission signals generated by the transmission signal generation unit 21.
In FIG. 8, the horizontal axis represents time t, and the vertical axis represents frequency f.
f _c is the minimum frequency of the transmission signal output from the transmission signal generation unit 21 to the transmission antenna 2-0 of the signal transmission unit 2, and f _off is the transmission antenna 2-n (n = 0, 1,..., N−2) is a difference (offset frequency) between the frequency of the transmission signal output to the transmission antenna 2- (n + 1) from the transmission signal generation unit 21. .
T _P is the pulse _{width, T WRI} the pulse repetition period, B is a sweep bandwidth.
In FIG. 8, each pulse is distinguished as the 0th pulse,..., (L-1) pulse.

The signal processing unit 5 includes a range Doppler data generation unit 22, a phase correction unit 7, and a target detection unit 8.
The range Doppler data generation unit 22 is realized by, for example, the range Doppler data generation circuit 11 illustrated in FIG.
The range Doppler data generation unit 22 Fourier-transforms the plurality of reception signals output from the A / D converter 4b of the reception processing unit 4 within the pulse repetition period _TWRI , and the result of Fourier transform on the frequency axis The received signal is pulse-compressed by multiplying a reference function that is a complex conjugate in the frequency domain of the transmission signal.
The range Doppler data generation unit 22 obtains distance dimension data by performing inverse Fourier transform on the pulse-compressed reception signal.
Further, the range Doppler data generation unit 22 obtains velocity dimension data by Fourier-transforming the distance dimension data between pulse repetition periods T _WRI to obtain range Doppler data.
Also in the LFM pulse compression method, the distance shift by Doppler is not different from that in the FMCW method, so that the phase correction can be performed by the processing of the phase correction unit 7 shown in the first embodiment.

  Even in the second embodiment, similarly to the first embodiment, even when the method for increasing the resolution of the distance to the target and the frequency division method are used in combination, it is possible to suppress the deterioration of the target detection accuracy.

In the second embodiment, an example in which the LFM pulse compression method is used as a method for increasing the resolution of the distance to the target is shown.
However, the present invention is not limited to this. For example, an NLFM (NoLiner Frequency Modulation) pulse compression method in which frequency modulation is nonlinear may be used.
Even when the NLFM method is used, the processing content of the signal processing unit 5 does not change.

  In the present invention, within the scope of the invention, any combination of the embodiments, or any modification of any component in each embodiment, or omission of any component in each embodiment is possible. .

  The present invention is suitable for a radar apparatus that detects a target.

  DESCRIPTION OF SYMBOLS 1 Transmission signal production | generation part, 2 Signal transmission part, 2-0-2- (N-1) Transmission antenna, 3 Signal reception part, 3-0-3- (M-1) Reception antenna, 4 Reception processing part, 4a Frequency conversion unit, 4b A / D converter, 5 signal processing unit, 6 range Doppler data generation unit, 7 phase correction unit, 8 target detection unit, 9 MIMO beam forming unit, 10 target detection processing unit, 11 range Doppler data generation Circuit, 12 Phase correction circuit, 13 Beam forming circuit, 14 Target detection processing circuit, 21 Transmission signal generation unit, 22 Range Doppler data generation unit, 31 Processor, 32 Memory

Claims (3)

A transmission signal generating unit that generates a plurality of transmission signals having different frequencies and modulated in frequency;
A plurality of transmission antennas at different positions, and a signal transmission unit that radiates each of the plurality of transmission signals generated by the transmission signal generation unit into space in a synchronized state from the transmission antenna; and
A signal receiving unit that receives a reflected signal, each of which is a signal reflected by a target, of each of the plurality of transmission signals radiated by the signal transmitting unit;
A range Doppler data generation unit that generates range Doppler data that is data of a distance and a speed axis by Fourier-transforming each of the plurality of reflected signals received by the signal receiving unit;
In a plurality of range Doppler data generated by the range Doppler data generation unit, a phase difference corresponding to a frequency difference between the plurality of transmission signals based on a distance error to the target caused by a relative speed with the target A phase correction unit for correcting
A MIMO beam forming unit that performs beam forming by correcting a spatial phase difference determined by a target angle, a transmitting antenna position, and a receiving antenna position with respect to a plurality of range Doppler data whose phase difference is corrected by the phase correcting unit. And a target detection unit for detecting the target from the beam formed by the MIMO beam forming unit.   The transmission signal generation unit generates a signal whose frequency is modulated by the FMCW (Frequency Modulation Continuous Wave) method or the FMICW (Frequency Modulated Continuous Wave) method as the transmission signal. The radar apparatus described.   The transmission signal generation unit generates a pulse signal whose frequency is modulated by an LFM (Liner Frequency Modulation) pulse compression method or an NLFM (NoLiner Frequency Modulation) pulse compression method as the transmission signal. Item 2. The radar device according to item 1.

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