JP5645902B2 - Synchronous motor control method and apparatus - Google Patents

Synchronous motor control method and apparatus Download PDF

Info

Publication number
JP5645902B2
JP5645902B2 JP2012235246A JP2012235246A JP5645902B2 JP 5645902 B2 JP5645902 B2 JP 5645902B2 JP 2012235246 A JP2012235246 A JP 2012235246A JP 2012235246 A JP2012235246 A JP 2012235246A JP 5645902 B2 JP5645902 B2 JP 5645902B2
Authority
JP
Japan
Prior art keywords
value
command value
phase error
phase
axis
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
JP2012235246A
Other languages
Japanese (ja)
Other versions
JP2014087190A (en
Inventor
大輔 豊田
大輔 豊田
清史 越智
清史 越智
Original Assignee
株式会社日本製鋼所
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 株式会社日本製鋼所 filed Critical 株式会社日本製鋼所
Priority to JP2012235246A priority Critical patent/JP5645902B2/en
Publication of JP2014087190A publication Critical patent/JP2014087190A/en
Application granted granted Critical
Publication of JP5645902B2 publication Critical patent/JP5645902B2/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Description

  The present invention relates to a synchronous motor control method and apparatus, and more particularly, to a synchronous motor control method and apparatus that can estimate the rotor magnetic pole position of a synchronous motor that has stopped rotating with high accuracy and in a short time.

Conventionally, the minute voltage change amounts vhd and vhq are given to the dc axis and the dq axis to measure current change rates ΔIdc and ΔIqc on the dc axis and dq axis, and the measured minute voltage change amounts vhd and vhq are measured. Based on ΔIdc and ΔIqc, the phase error Δθ between the estimated magnetic pole position and the actual magnetic pole position is calculated, or the minute voltage change amount vhd is given to the dc axis to measure the current change rate ΔIqc on the dq axis. There is known an AC motor drive system that calculates a phase error Δθ between an estimated magnetic pole position and an actual magnetic pole position based on a voltage change amount vhd and a measured current change rate ΔIqc (see, for example, Patent Documents 1 and 2). Note that the names / symbols of the dc axis to current change rate Δθ are the names / symbols used in Patent Documents 1 and 2, and do not necessarily match the names / symbols in the present specification.
On the other hand, applying a sinusoidal current command value id1 * to the d-axis and measuring the current value Iq ′ on the q′-axis is repeated while changing the magnetic pole position estimated value θ ′, and the current value Iq ′ = 0. A synchronous motor magnetic pole position estimation method, an electric motor control device, and an electric vehicle that obtain an estimated magnetic pole position estimated value θ ′ as an actual magnetic pole position are known (see, for example, Patent Document 3. Note that d-axis to magnetic pole position. The name / symbol of the estimated value θ ′ is the name / symbol used in Patent Document 3, and does not necessarily match the name / symbol in the present specification.

JP 2002-78391 A JP 2002-78392 A JP 2003-143894 A

In the above Patent Documents 1 and 2, the current change rate is measured. However, the current change rate has an error due to noise caused by switching of a PWM (Pulse Width Modulation) circuit, so that the magnetic pole position estimation accuracy is lowered.
On the other hand, in Patent Document 3, the current value Iq ′ is repeatedly measured by changing the magnetic pole position estimated value θ ′ until the current value Iq ′ = 0 on the q ′ axis converges. For this reason, there is a problem that the average time required for convergence becomes long.
SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide a synchronous motor control method and apparatus capable of estimating the rotor magnetic pole position of a synchronous motor whose rotation is stopped with high accuracy and in a short time.

In a first aspect, the present invention provides a high-frequency current command value or a high-frequency voltage command value to a synchronous motor that has stopped rotating, and a phase error θ that is the phase of the rotor magnetic pole (P) with respect to the stator U-phase position (dc0). The minimum value of the peak-to-peak value of the d-axis current measurement value or the d-axis voltage measurement value when the phase error command value θ * for commanding the phase of the dc axis with respect to the stator U phase position (dc0) is changed. The maximum value is stored in advance, the d-axis current measurement value or the d-axis when the high-frequency current command value or the high-frequency voltage command value is given to the synchronous motor whose rotation is stopped and the predetermined value is given as the phase error command value θ * The peak-to-peak value of the voltage measurement value is measured, two phase error candidate values (θa, θb) of the phase error θ are calculated from the measurement value, the minimum value, and the maximum value, and then the synchronous power while the rotation is stopped From the change of the d-axis current measurement value or the d-axis voltage measurement value when a high frequency current command value or a high frequency voltage command value is given to the machine and a value obtained by slightly changing the predetermined value as the phase error command value θ * is given. A synchronous motor control method is provided in which one of two candidate values (θa, θb) is selected as a phase error θ.
In the synchronous motor control method according to the first aspect, since the influence of the switching noise of the PWM circuit is small as compared with the conventional technique for measuring the current change rate, high estimation accuracy can be obtained. Further, immediately before starting the synchronous motor, the phase error θ can be estimated by measuring the peak-to-peak value of the d-axis current measurement value only twice, so that the synchronous motor can be started quickly.
Since the minimum value and the maximum value are obtained without changing the rotor angle, the implementation is easy.

In a second aspect, the present invention provides a phase error command value θ * that gives a high-frequency current command value or a high-frequency voltage command value to a synchronous motor that has stopped rotating and commands the phase of the dc axis with respect to the stator U-phase position (dc0). = Minimum of peak-to-peak value of d-axis current measurement value or d-axis voltage measurement value when a constant value is given and the phase error θ which is the phase of the rotor magnetic pole (P) with respect to the stator U-phase position (dc0) is changed Value and maximum value are stored in advance, a measured value of the d-axis current when a high-frequency current command value or a high-frequency voltage command value is given to the synchronous motor whose rotation is stopped and a predetermined value is given as the phase error command value θ *, or The peak-to-peak value of the d-axis voltage measurement value is measured, and two phase error candidate values (θa, θb) of the phase error θ are calculated from the measurement value and the minimum value and the maximum value, and then the rotation is stopped. same From the change in the d-axis current measurement value or the d-axis voltage measurement value when a high frequency current command value or a high frequency voltage command value is given to the motor and a value obtained by slightly changing the predetermined value as the phase error command value θ * is given. A synchronous motor control method is provided in which one of two candidate values (θa, θb) is selected as a phase error θ.
In the synchronous motor control method according to the second aspect, since the influence of the switching noise of the PWM circuit is small as compared with the conventional technique for measuring the current change rate, high estimation accuracy can be obtained. Further, immediately before starting the synchronous motor, the phase error θ can be estimated by measuring the peak-to-peak value of the d-axis current measurement value only twice, so that the synchronous motor can be started quickly.

In a third aspect, the present invention relates to a high-frequency command value generating means (2) for giving a high-frequency current command value (ih *) or a high-frequency voltage command value to the synchronous motor (M) whose rotation is stopped, and the high-frequency current command value. (Ih *) or current control means (6) for outputting a d-axis voltage command value (vd *) and a q-axis voltage command value (vq *) based on the high-frequency voltage command, and the d-axis voltage command value (vd *) ) And the q-axis voltage command value (vq *), dq / UVW conversion means (7) for outputting a three-phase voltage command value (vU *, vV *, vW *), and the three-phase voltage command value (vU) *, VV *, vW *) based on three-phase power supply means (8) for supplying three-phase power to the synchronous motor (M), and current detection for detecting the three-phase motor current (iU ', iV') Based on the means (9, 10) and the three-phase motor current (iU ′, iV ′) UVW / dq conversion means (11) for outputting a flow measurement value (id '), and a stator U phase position (dc0) of a synchronous motor (M) whose rotation is stopped based on the d-axis current measurement value (id') A phase error estimating means (12) for obtaining a phase error θ which is a phase of the rotor magnetic pole with respect to a minimum value (idhmin) of a peak-to-peak value (idh) of a d-axis current measurement value (id ′) with respect to a change in the phase error θ ) And a memory (13) storing a maximum value (idhmax), and during the rotation stop, the high frequency command value generating means (2) outputs a high frequency current command value (ih *) or a high frequency voltage command value. The phase error estimation means (12) outputs a phase error command value θ for instructing the dq / UVW conversion means (7) the phase of the dc axis with respect to the stator U phase position (dc0) of the synchronous motor (M). * As The dq / UVW conversion means (7) determines three values based on the d-axis voltage command value (vd *), the q-axis voltage command value (vq *), and the phase error command value (θ *). The phase voltage command value (vU *, vV *, vW *) is output, and the phase error estimating means (12) outputs the d-axis current measurement value (idc ′) or the peak-to-peak value (idh) of the d-axis voltage measurement value. And two phase error candidate values (θa, θb) of the phase error θ are calculated from the measured value and the minimum value (idhmin) and the maximum value (idhmax) stored in the memory (13), Next, the high frequency command value generating means (2) outputs a high frequency current command value or a high frequency voltage command value to the synchronous motor (M) whose rotation is stopped, and the phase error estimating means (12) is the dq / UVW conversion means ( 7) As a phase error command value θ * A value obtained by slightly changing the predetermined value is given, and the dq / UVW conversion means (7) is configured to output the d-axis voltage command value (vd *), the q-axis voltage command value (vq *), and the phase error command value ( The three-phase voltage command value (vU *, vV *, vW *) is output based on θ *), and the phase error estimating means (12) is a d-axis current measurement value (idc ′) or a d-axis voltage measurement value. A synchronous motor control device (100) is provided in which one of the two candidate values (θa, θb) is selected as a phase error θ from the change.
In the synchronous motor control device according to the third aspect, since the influence of the switching noise of the PWM circuit is small as compared with the conventional technique for measuring the current change rate, high estimation accuracy can be obtained. Further, immediately before starting the synchronous motor, the phase error θ can be estimated by measuring the peak-to-peak value of the d-axis current measurement value only twice, so that the synchronous motor can be started quickly.

In a fourth aspect, the present invention relates to the synchronous motor control device (100) according to the third aspect, wherein the high-frequency command value generating means (2) is a high-frequency current command value (ih *) or a high-frequency signal while rotation is stopped. A voltage command value is output, the phase error estimating means (12) changes the phase error command value θ *, and the dq / UVW conversion means (7) is configured to output the d-axis voltage command value (vd *) and the q A three-phase voltage command value (vU *, vV *, vW *) is output based on the shaft voltage command value (vq *) and the phase error command value (θ *), and the phase error estimation means (12) The peak-to-peak value (idh) of the d-axis current measurement value (idc ′) or the d-axis voltage measurement value is measured, and the minimum from the change in the peak-to-peak value (idh) with respect to the change in the phase error command value θ * Value (idhmin) and maximum value (idh) max), and the memory (13) stores the minimum value (idhmin) and the maximum value (idhmax), thereby providing a synchronous motor control device (100).
The synchronous motor control device according to the fourth aspect is easy to implement because the minimum value and the maximum value are obtained without changing the rotor angle.

  According to the synchronous motor control method and apparatus of the present invention, the rotor magnetic pole position of the synchronous motor whose rotation is stopped can be accurately estimated in a short time.

1 is an explanatory diagram illustrating a configuration of a synchronous motor control device according to a first embodiment. It is explanatory drawing of phase error (theta). It is an illustration figure of the waveform which gave d-axis current command value idc * to the reluctance motor in rotation stop, and observed d-axis current measurement value idc '. FIG. 6 is a characteristic curve diagram in which a change in peak-to-peak value idh of a d-axis current measurement value idc ′ with respect to a change in phase error θ is plotted. It is an illustration figure of the waveform which gave d-axis current command value idc * to the permanent-magnet motor in which rotation was stopped, and observed d-axis current measurement value idc '. It is a flowchart which shows the peak-to-peak maximum value minimum value measurement process which concerns on Example 1. FIG. FIG. 5 is a conceptual diagram showing a relationship between a dc axis (broken arrow) corresponding to the characteristic curve of FIG. 4 and a dc axis (solid arrow) corresponding to the phase error command value θ *. FIG. 6 is a characteristic curve diagram plotting changes in peak-to-peak value idh of d-axis current measurement value idc ′ with respect to changes in phase error command value θ *. FIG. 5 is a conceptual diagram showing a relationship between a dc axis (broken arrow) corresponding to the characteristic curve of FIG. 4 and a dc axis (solid arrow) corresponding to the phase error command value θ * = θ * min + 90 °. It is a flowchart which shows the rotor magnetic pole position estimation process which concerns on Example 1. FIG. It is explanatory drawing which shows two magnetic pole error candidate values (theta) a and (theta) b obtained from the measured value idh1 of the peak-to-peak value idh of d-axis current measured value idc '. It is explanatory drawing which shows the measured value idh2 of the peak-to-peak value idh of the d-axis current measured value idc 'obtained when the phase error command value θ * is changed by ψ.

  Hereinafter, the present invention will be described in more detail with reference to embodiments shown in the drawings. Note that the present invention is not limited thereby.

Example 1
FIG. 1 is a configuration explanatory view showing a synchronous motor control device 100.
The synchronous motor control device 100 includes a vector controller 1 that outputs a d-axis current control value id0 * and a q-axis current command value iq * for performing speed control and torque control of the synchronous motor M having saliency, The high-frequency current command value generator 2 that outputs a high-frequency current command value ih * for detecting the magnetic pole position of the rotor of the synchronous motor M that is stopped, and the high-frequency current command value ih * is added to the d-axis current control value id0 * An adder 3 that outputs a d-axis current command value id *, a differencer 4 that outputs a d-axis current difference value that is a difference between the d-axis current command value id * and the d-axis current measurement value id ′, q A differencer 5 that outputs a q-axis current difference value that is a difference between the axis current command value iq * and the q-axis current measurement value iq ′, and a d-axis voltage command value based on the d-axis current difference value and the q-axis current difference value. Current that outputs vd * and q-axis voltage command value vq * Controller 6 and dq / UVW conversion that outputs three-phase voltage command values vU *, vV *, and vW * based on d-axis voltage command value vd *, q-axis voltage command value vq *, and phase error command value θ * 7, a PWM inverter 8 that supplies three-phase power to the synchronous motor M based on the three-phase voltage command values vU *, vV *, vW *, and a current detector 9 for detecting the U-phase motor current iU ′. A current detector 10 for detecting the V-phase motor current iV ′, a d-axis current measurement value id ′ and a q-axis current measurement value iq ′ based on the U-phase motor current iU ′ and the V-phase motor current iV ′. And a phase error estimator 12 for obtaining a phase error θ between the stator U phase position of the synchronous motor M and the magnetic pole position of the rotor that are stopped from rotation based on the measured d-axis current value id ′. , Peak-to-peak value id of d-axis current measurement value id ′ with respect to change in phase error θ When the rotor 13 of the synchronous motor M is of a permanent magnet type, the polarity of the magnetic pole closest to the stator U phase position of the synchronous motor M that has stopped rotating is discriminated from the memory 13 that stores the maximum value idhmax and the minimum value idhmin. The polarity discriminator 14 is provided.

FIG. 2 is an explanatory diagram of the phase error θ. In addition, (a)-(c) shows a 4-pole reluctance motor, (A)-(D) shows a 4-pole permanent magnet motor.
As shown in FIG. 2A, a state where the stator U-phase position dc0 of the four-pole reluctance motor that is stopped in rotation and the position of the magnetic pole P of the rotor coincide is defined as a phase error θ = 0 °.
As shown in FIG. 2 (b), the phase error θ = 45 is a state in which the stator U phase position dc0 of the four-pole reluctance motor being stopped and the position of the magnetic pole P of the rotor are mechanically shifted by 22.5 °. °.
As shown in FIG. 2 (c), the phase error θ = 90 ° is a state where the stator U phase position dc0 of the four-pole reluctance motor being stopped rotating and the position of the magnetic pole P of the rotor are mechanically shifted by 45 °. To do.

As shown in FIG. 2A, a state in which the stator U-phase position dc0 of the four-pole permanent magnet electric motor whose rotation is stopped coincides with the position of the N-pole N of the rotor is defined as a phase error θ = 0 °.
As shown in FIG. 2B, the phase error θ is a state in which the stator U-phase position dc0 of the four-pole permanent magnet electric motor whose rotation is stopped and the position of the rotor N-pole N are mechanically shifted by 22.5 °. = 45 °.
As shown in FIG. 2C, a phase error θ = 90 is a state in which the stator U-phase position dc0 of the four-pole permanent magnet motor that is stopped from rotating and the position of the rotor N-pole N are mechanically shifted by 45 °. °.
As shown in FIG. 2D, a state in which the stator U-phase position dc0 of the four-pole permanent magnet motor whose rotation is stopped and the position of the S-pole S of the rotor coincide with each other is defined as a phase error θ = 180 °.

The solid line arrow shown in FIG. 2 represents the phase of the dc axis of the estimated coordinate system of the rotor magnetic pole position.
When the phase error command value θ * = 0 ° is given from the phase error estimator 12, the dq / UVW converter 7 matches the dc axis with the stator U phase position dc0.

As shown in FIG. 2A, when the phase error θ = 0 °, the phase error command value θ * = 0 ° is given, the d-axis current control value id0 * = 0, and the q-axis current command When a value iq * = 0 and a sine wave having a period of 3 ms and a peak-to-peak value of 10 A is given as the high-frequency current command value ih *, a d-axis current measurement value idc ′ as shown in FIG. 3A is observed. . In addition, in order to show that it is d axis | shaft current measured value id 'in a rotation stop, the symbol is set to idc'.
As shown in FIG. 2B, when the same condition is given when the phase error θ = 45 °, a d-axis current measurement value idc ′ as shown in FIG. 3B is observed. .
As shown in FIG. 2C, when the same condition is given when the phase error θ = 90 °, a d-axis current measurement value idc ′ as shown in FIG. 3C is observed. .

FIG. 4 is a characteristic curve diagram in which the change of the peak-to-peak value idh of the d-axis current measurement value idc ′ with respect to the change of the phase error θ is plotted.
This characteristic curve is expressed by the following equation as can be seen from FIG.
idh = (idhmax + idhmin) / 2 − {(idhmax−idhmin) / 2} cos (2 · θ)
The maximum value idhmax and the minimum value idhmin are different values depending on the high-frequency current command value ih *, the winding resistance and inductance of the synchronous motor M, and the like.

The characteristic curves in FIG. 4 are the same even in the case of the four-pole permanent magnet motor shown in FIGS.
However, a state in which the stator U-phase position dc0 of the four-pole permanent magnet motor being stopped and the position of the N-pole N of the rotor coincide with each other is defined as a phase error θ = 0 °, and the stator U-phase position dc0 and the S-pole of the rotor Since the phase error θ = 180 ° is the state in which the positions of S match, the range is 0 ° ≦ θ <360 °.
On the other hand, in the case of a 4-pole reluctance motor, since there is no distinction of polarity, the range is 0 ° ≦ θ <180 °.

Further, as shown in FIG. 5, the waveform of the d-axis current measurement value idc ′ becomes asymmetric between positive and negative according to the phase error θ.
That is, as shown in FIG. 5A, when 0 ° ≦ θ <180 °, the d-axis current measurement value idc ′ becomes a negative value during the positive period Tp in which the d-axis current measurement value idc ′ is a positive value. The negative period Tn becomes longer.
On the other hand, as shown in FIG. 5B, when 180 ° ≦ θ <360 °, the positive period Tp is longer than the negative period Tn.

FIG. 6 is a flowchart showing a peak-to-peak maximum value minimum value measurement process. This process is started with the d-axis current control value id0 * = 0 and the q-axis current command value iq * = 0.
In step H1, a constant high-frequency current command value ih * is output from the high-frequency current command value generator 2, a phase error command value θ * = 0 ° is output from the phase error estimator 12, and a d-axis current measurement value is output. The peak-to-peak value idh of idc ′ is measured.

  As shown in FIG. 7A, the dc axis corresponding to the characteristic curve in FIG. 4 (the broken line arrow in FIG. 7 coincides with the d axis) and the phase error command value θ * = 0 °. The phase difference θ ′ of the corresponding dc axis (solid arrow) becomes, for example, θ ′ = θ * −θ = 0 ° −62 ° = −62 ° if the actual phase error θ = 62 °. Therefore, the measured peak-to-peak value idh is the peak-to-peak value at θ = −62 ° = 118 ° in FIG. This corresponds to the point a shown in FIG.

  Subsequently, the peak-to-peak value idh is measured by increasing the phase-error command value θ * from the phase-error estimator 12 by δ1 (for example, 15 °) and measuring the peak-to-peak value idh of the d-axis current measurement value idc ′. Until the minimum value idhmin is found, the phase error command value θ * min when the minimum value idhmin is obtained is obtained.

  As shown in FIG. 7B, when the dc axis in the characteristic curve shown in FIG. 4 (the broken line arrow in FIG. 7 coincides with the d axis) and the phase error command value θ * = 45 °. For example, if the phase error θ = 62 °, the phase difference θ ′ of the dc axis (solid arrow) is θ ′ = θ * −θ = 45 ° −62 ° = −17 °. Therefore, the measured peak-to-peak value idh is the peak-to-peak value at θ = −17 ° = 163 ° in FIG. This corresponds to the point b shown in FIG.

  As shown in FIG. 7C, when the dc axis in the characteristic curve shown in FIG. 4 (the broken line arrow in FIG. 7 coincides with the d axis) and the phase error command value θ * = 90 °. For example, if the phase error θ = 62 °, the phase difference θ ′ of the dc axis (solid arrow) becomes θ ′ = θ * −θ = 90 ° −62 ° = 28 °. Therefore, the measured peak-to-peak value idh is the peak-to-peak value at θ = 28 ° in FIG. This corresponds to the point c shown in FIG.

  If the measured value of the peak-to-peak value idh changes as shown by the curve shown in FIG. 8A, the phase error command value can be obtained by repeating the measurement from the phase error command value θ * = 0 ° to about 90 °. θ * min = 60 ° is found. For example, if δ1 = 15 ° and θ * = 0 ° to about 90 ° is repeated, the number of times the measurement is repeated in step H1 is about 7 times. Further, the resolution of the phase error command value θ * min found in step H1 is δ1.

  Returning to FIG. 6, in step H2, a constant high-frequency current command value ih * is output, and the phase error command value θ * is set to δ2 within a predetermined range R1 centered on the phase error command value θ * = θ * min. When the peak-to-peak value idh of the d-axis current measurement value idc ′ is measured by changing (for example, 3 °) each time until the minimum value idhmin of the peak-to-peak value idh is found, the minimum value idhmin is obtained. A phase error command value θ * min is obtained. Note that δ2 is a value smaller than δ1, and R1 is an integral multiple of δ2.

  As shown in FIG. 8B, for example, if the phase error command value found in step H1 is θ * min = 60 °, R1 = 24 °, δ2 = 3 °, the number of times the measurement is repeated in step H2 is 9 Times. By repeating this, the phase error command value θ * min = 63 ° is found. The resolution of the phase error command value θ * min found in step H2 is δ2.

  Returning to FIG. 6, in step H3, a constant high-frequency current command value ih * is output, and the phase error command value θ * is set to δ3 within a predetermined range R2 centered on the phase error command value θ * = θ * min. Measurement of the peak-to-peak value idh of the measured d-axis current value idc ′ is repeated until the minimum value idhmin of the peak-to-peak value idh is found to obtain the minimum value idhmin and the phase error command value θ * min. . Note that δ3 is a value smaller than δ2, and R2 is an integral multiple of δ3.

  As shown in FIG. 8C, for example, if the phase error command value found in step H2 is θ * min = 63 °, R2 = 10 °, and δ3 = 1 °, the number of times the measurement is repeated in step H3 is 11. Times. By repeating this, the phase error command value θ * min = 62 ° is found. This phase error command value θ * min = 62 ° is the phase error θ. The resolution of the phase error command value θ * min obtained in step H3 is δ3.

  Returning to FIG. 6, in step H4, a constant high-frequency current command value ih * is output, a phase error command value θ * = θ * min + 90 ° is output, and the peak-to-peak value of the d-axis current measurement value idc ′ is output. idh is measured, and the measured value is set as the maximum value idhmax of the peak-to-peak value idh.

  As shown in FIG. 9, since the phase error command value θ * min obtained in step H3 is the phase error θ, the dc axis (solid arrow) in the phase obtained by adding 90 ° to the phase error θ is the characteristic shown in FIG. This corresponds to a phase error θ = 90 ° in the curve. Therefore, the peak-to-peak value idh measured here is the maximum value idhmax.

In step H5, the phase error estimator 12 stores the minimum value idhmin and the maximum value idhmax in the memory 13.
The memory 13 stores in advance whether the synchronous motor M is a reluctance motor or a permanent magnet motor.

FIG. 10 is a flowchart showing the rotor magnetic pole position estimation processing. This process is started with the d-axis current control value id0 * = 0 and the q-axis current command value iq * = 0.
In step H11, a constant high-frequency current command value ih * is output from the high-frequency current command value generator 2, a phase error command value θ * = 0 ° is output from the phase error estimator 12, and a d-axis current measurement value is output. The peak-to-peak value idh, positive value period Tp, and negative value period Tn of idc ′ are measured. The peak-to-peak value idh measured in step H11 is set to idh1.

In Step H12, the phase error candidate values θa (<90 °) and θb (≧ 90 °) are obtained by the following equations.

  As can be seen from FIG. 11, two phase error candidate values θa (<90 °) and θb (≧ 90 °) are calculated for one measured peak-to-peak value idh. For example, θa = 62 ° and θb = 118 °.

Returning to FIG. 10, in step H13, a constant high-frequency current command value ih * is output from the high-frequency current command value generator 2, and a phase error command value θ * = − ψ is output from the phase error estimator 12. The peak-to-peak value idh of the d-axis current measurement value idc ′ is measured. It is assumed that the peak-to-peak value idh = idh2 measured in step H13.
Note that ψ is, for example, the maximum resolution δ3 when the minimum value idhmin and the maximum value idhmax are obtained.

In step H14, it is determined whether or not idh2> idh1, and if idh2> idh1, the process proceeds to step H15, and if idh2 ≦ idh1, the process proceeds to step H16.
As shown in FIG. 12, setting the phase error command value θ * = − ψ in step H13 means that the apparent phase error θ is increased by ψ, so if the true phase error θ is θa, idh2> If idh1 and the true phase error θ is θb, then idh2 ≦ idh1.

  In step H15, the phase error θ is set to θa. Then, the process proceeds to Step H17.

  In step H16, the phase error θ = θb. Then, the process proceeds to Step H17.

  In step H17, it is determined whether the synchronous motor M is a reluctance motor or a permanent magnet motor. If it is a reluctance motor, the process is terminated, and if it is a permanent magnet motor, the process proceeds to step H18.

  In step H18, the positive value period Tp and the negative value period Tn of the d-axis current measurement value idc ′ are compared. If Tn> Tp, the process is ended because FIG. Since it is the case of (b) of 5, it progresses to step H19.

  In step H19, the phase error θ = θ + 180 ° is set, and the process is terminated.

According to the synchronous motor control apparatus 100 according to the first embodiment, the following effects can be obtained.
(1) Since the influence of the switching noise of the PWM inverter 8 is small as compared with the conventional technique for measuring the current change rate, high estimation accuracy can be obtained.
(2) Immediately before starting the synchronous motor M, the phase error θ can be estimated by measuring the peak-to-peak value idh of the d-axis current measured value idc ′ only twice, so that the synchronous motor M can be started quickly.
(3) Even when the characteristics of the synchronous motor M change due to long-term use or the like, the estimation accuracy can be maintained by performing the peak-to-peak maximum value / minimum value measurement process again.
(4) Since only the minimum value idhmin and the maximum value idhmax of the peak-to-peak value idh of the d-axis current measurement value idc ′ need only be obtained and stored in advance, the amount of use of the memory 13 can be reduced.
(5) In an existing synchronous motor control device that performs vector control by measuring the d-axis current measurement value idc ′, it is not necessary to add hardware, and therefore it can be easily implemented.

Next, the present invention is demonstrated by a mathematical model.
The voltage equation of the synchronous motor is expressed by the following equation.
In the following equation, vd is a d-axis voltage, vq is a q-axis voltage, id is a d-axis current, iq is a q-axis current, R is an armature winding resistance, p is a differential operator, Ld is a d-axis inductance, and Lq is The q-axis inductance, ω is the electrical angular rotational speed of the synchronous motor, and φ is the field linkage magnetic flux.

Since the voltage equation of the synchronous motor that is not rotating is ω = 0, the following equation is obtained.

The rotational coordinate system dq axis of the synchronous motor (= the coordinate system in which the position where the rotor magnetic pole is actually located is the d axis) and the estimated coordinate system dcqc axis (= the stator d-axis position coincides with the stator U phase position dc0). When there is a phase error θ in the estimated coordinate system), the relationship between the dq axis and the dcqc axis is as follows.

Therefore, the relationship between the voltages vd, vq on the dq axis and the voltages vdc, vqc on the dcqc axis is as follows.

The relationship between the currents id and iq on the dq axis and the currents idc and iqc on the dcqc axis is also expressed by the following equation.

Substituting these into equation (2) gives the following equation.

Here, there is a relationship of the following equation.

When the right side of Equation (7) is multiplied by the right side of Equation (6) and the left side of Equation (7) is multiplied by the left side of Equation (6), the following equation is obtained.
When La = (Ld + Lq) / 2 and Lb = (Ld−Lq) / 2 in the following equation, the following equation is obtained.

Further, when the voltage vdc is replaced with the voltage command value vdc *, the voltage vqc is replaced with the voltage command value vqc *, the current idc is replaced with the current command value idc *, and the current idc is replaced with the current command value idc *, the following equation is obtained.

When simply considering proportional control of the difference between the command value and the detected value, the following equation is established.
In the following equation, Kd is a d-axis current proportional gain, and Kq is a q-axis current proportional gain.

Substituting Equation (11) into Equation (10) gives the following equation.

Here, the following equation holds.

If Δ = La 2 −Lb 2 , the following equation is obtained. Assuming a synchronous motor with saliency, Δ ≠ 0.

Furthermore, when the qc-axis current command value iqc * = 0, the relationship between the dc-axis current measurement value idc ′ and the qc-axis current measurement value iqc ′ with respect to the dc-axis current command value idc * is as follows.

  As can be seen from the equation (15), the dc-axis current measurement value idc ′ and the qc-axis current measurement value iqc ′ vary depending on the inductances Ld and Lq depending on the magnetic flux position and the phase error θ. The phase error θ can be estimated by measuring the dc axis current measurement value idc ′ generated by *.

-Example 2-
From the equation (10), the relationship between the d-axis voltage command value vdc * and the dc-axis current measurement value idc ′ is expressed by the following equation.

Further, assuming that the qc-axis voltage command value vqc * = 0, the relationship between the dc-axis current measurement value idc ′ and the qc-axis current measurement value iqc ′ with respect to the dc-axis voltage command value vdc * is as follows.

  As can be seen from the equation (17), the phase error θ can be estimated by measuring the dc axis current measurement value idc ′ generated by the dc axis voltage command value vdc *.

-Example 3-
Similarly to the second embodiment, the phase error θ can also be estimated by measuring the dc axis voltage measurement value vdc ′ generated by the dc axis voltage command value vdc *.

Example 4
The high-frequency current command value ih * may be a rectangular wave.

-Example 5
When the high-frequency current command value ih * or the high-frequency voltage command value is given to the synchronous motor M that is stopped from rotation, the phase error command value θ * = 0 ° is given, and the phase error θ is changed by changing the rotor angle. The minimum value idhmin and the maximum value idhmax of the peak-to-peak value of the axis current measurement value idc ′ or the d-axis voltage measurement value may be obtained and stored in the memory 13 in advance.

  The synchronous motor control method and apparatus of the present invention can be used in, for example, an injection molding apparatus.

DESCRIPTION OF SYMBOLS 1 Vector controller 2 High frequency electric current command value generator 3 Adder 4,5 Difference 6 Current controller 7 dq / UVW converter 8 PWM inverter 9,10 Current detector 11 UVW / dq converter 12 Phase error estimator 13 Memory 14 Polarity discriminator 100 Synchronous motor controller M Synchronous motor

Claims (4)

  1.   A high frequency current command value or a high frequency voltage command value is given to the synchronous motor whose rotation is stopped, the phase error θ which is the phase of the rotor magnetic pole (P) with respect to the stator U phase position (dc0) is made constant, and the stator U phase position (dc0) The minimum and maximum values of the peak-to-peak value of the d-axis current measurement value or d-axis voltage measurement value when the phase error command value θ * that commands the phase of the dc axis with respect to is changed are stored in advance. Measure the peak-to-peak value of the d-axis current measurement value or d-axis voltage measurement value when a high-frequency current command value or high-frequency voltage command value is given to the stopped synchronous motor and a predetermined value is given as the phase error command value θ * Then, two phase error candidate values (θa, θb) of the phase error θ are calculated from the measured value and the minimum value and the maximum value, and then there is a high-frequency current command value in the synchronous motor that has stopped rotating. Indicates the two candidate values (θa) based on the change in the d-axis current measurement value or the d-axis voltage measurement value when a high frequency voltage command value is given and a value obtained by slightly changing the predetermined value as the phase error command value θ * is given. , Θb) is selected as the phase error θ, and the synchronous motor control method is characterized.
  2.   A high-frequency current command value or a high-frequency voltage command value is given to the synchronous motor whose rotation is stopped, and a phase error command value θ * = constant value is given to command the phase of the dc axis with respect to the stator U-phase position (dc0), and the stator U-phase position The minimum and maximum values of the peak-to-peak value of the d-axis current measurement value or d-axis voltage measurement value when the phase error θ, which is the phase of the rotor magnetic pole (P) with respect to (dc0), is stored in advance. The peak-to-peak value of the d-axis current measurement value or the d-axis voltage measurement value when the high-frequency current command value or the high-frequency voltage command value is given to the synchronous motor that has stopped rotating and the predetermined value is given as the phase error command value θ * And two phase error candidate values (θa, θb) of the phase error θ are calculated from the measured value and the minimum value and the maximum value, and then the high frequency current command value is supplied to the synchronous motor whose rotation is stopped. Alternatively, the two candidate values (from the change in the d-axis current measurement value or the d-axis voltage measurement value when a high-frequency voltage command value is given and a value obtained by slightly changing the predetermined value as the phase error command value θ * is given. A method of controlling a synchronous motor, wherein one of θa and θb) is selected to be a phase error θ.
  3. Based on the high frequency command value generating means (2) for giving the high frequency current command value (ih *) or the high frequency voltage command value to the synchronous motor (M) whose rotation is stopped, and the high frequency current command value (ih *) or the high frequency voltage command. Current control means (6) for outputting the d-axis voltage command value (vd *) and the q-axis voltage command value (vq *), the d-axis voltage command value (vd *) and the q-axis voltage command value (vq *) ) Based on the three-phase voltage command values (vU *, vV *, vW *) and the dq / UVW conversion means (7) that outputs the three-phase voltage command values (vU *, vV *, vW *). Three-phase power supply means (8) for supplying three-phase power to the synchronous motor (M), current detection means (9, 10) for detecting the three-phase motor current (iU ′, iV ′), Outputs d-axis current measurement value (id ') based on phase motor current (iU', iV ') The phase error θ which is the phase of the rotor magnetic pole with respect to the stator U phase position (dc0) of the synchronous motor (M) whose rotation is stopped based on the UVW / dq conversion means (11) and the d-axis current measurement value (id ′) And a phase error estimating means (12) for obtaining a minimum value (idhmin) and a maximum value (idhmax) of a peak-to-peak value (idh) of a d-axis current measurement value (id ′) with respect to a change in the phase error θ. A memory (13) having
    While the rotation is stopped, the high frequency command value generating means (2) outputs a high frequency current command value (ih *) or a high frequency voltage command value, and the phase error estimating means (12) is the dq / UVW conversion means (7). Is given a predetermined value as a phase error command value θ * for commanding the phase of the dc axis with respect to the stator U phase position (dc0) of the synchronous motor (M), and the dq / UVW conversion means (7) Based on the command value (vd *), the q-axis voltage command value (vq *), and the phase error command value (θ *), a three-phase voltage command value (vU *, vV *, vW *) is output. The phase error estimating means (12) measures the d-axis current measurement value (idc ′) or the peak-to-peak value (idh) of the d-axis voltage measurement value and stores the measurement value and the memory (13). Minimum value (idhmin) and maximum value (idhm) Since the x) 2 single phase error candidate value of the phase error θ is calculated (θa, θb),
    Next, the high-frequency command value generating means (2) outputs a high-frequency current command value or a high-frequency voltage command value to the synchronous motor (M) whose rotation is stopped, and the phase error estimating means (12) is the dq / UVW conversion means. A value obtained by slightly changing the predetermined value as a phase error command value θ * is given to (7), and the dq / UVW conversion means (7) is configured to output the d-axis voltage command value (vd *) and the q-axis voltage. A three-phase voltage command value (vU *, vV *, vW *) is output based on the command value (vq *) and the phase error command value (θ *), and the phase error estimating means (12) A synchronous motor control device characterized by measuring a current measurement value (idc ′) or a d-axis voltage measurement value and selecting one of the two candidate values (θa, θb) from the change as a phase error θ ( 100).
  4. In the synchronous motor control device (100) according to claim 3,
    While the rotation is stopped, the high frequency command value generation means (2) outputs a high frequency current command value (ih *) or a high frequency voltage command value, and the phase error estimation means (12) changes the phase error command value θ *. The dq / UVW conversion means (7) is configured to generate a three-phase voltage based on the d-axis voltage command value (vd *), the q-axis voltage command value (vq *), and the phase error command value (θ *). The command value (vU *, vV *, vW *) is output, and the phase error estimating means (12) measures the d-axis current measurement value (idc ′) or the peak-to-peak value (idh) of the d-axis voltage measurement value. And obtaining the minimum value (idhmin) and the maximum value (idhmax) from the change of the peak-to-peak value (idh) with respect to the change of the phase error command value θ *,
    The synchronous motor control device (100), wherein the memory (13) stores the minimum value (idhmin) and the maximum value (idhmax).
JP2012235246A 2012-10-25 2012-10-25 Synchronous motor control method and apparatus Active JP5645902B2 (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
JP2012235246A JP5645902B2 (en) 2012-10-25 2012-10-25 Synchronous motor control method and apparatus

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
JP2012235246A JP5645902B2 (en) 2012-10-25 2012-10-25 Synchronous motor control method and apparatus

Publications (2)

Publication Number Publication Date
JP2014087190A JP2014087190A (en) 2014-05-12
JP5645902B2 true JP5645902B2 (en) 2014-12-24

Family

ID=50789775

Family Applications (1)

Application Number Title Priority Date Filing Date
JP2012235246A Active JP5645902B2 (en) 2012-10-25 2012-10-25 Synchronous motor control method and apparatus

Country Status (1)

Country Link
JP (1) JP5645902B2 (en)

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP3401155B2 (en) * 1997-02-14 2003-04-28 株式会社日立製作所 Synchronous motor control device and electric vehicle
JP2001008486A (en) * 1999-06-18 2001-01-12 Hitachi Building Systems Co Ltd Controller for permanent magnet synchronous motor
JP3832443B2 (en) * 2003-03-28 2006-10-11 株式会社日立製作所 AC motor control device
JP5652217B2 (en) * 2011-01-18 2015-01-14 ダイキン工業株式会社 Power converter

Also Published As

Publication number Publication date
JP2014087190A (en) 2014-05-12

Similar Documents

Publication Publication Date Title
US8159161B2 (en) Motor control device
Liu et al. Improved sensorless control of permanent-magnet synchronous machine based on third-harmonic back EMF
TWI499198B (en) Motor control apparatus and motor control method
US8174220B2 (en) Apparatus for controlling permanent-magnet rotary electric machine
US9438153B2 (en) Rotary electric machine control device
DE102006047206B9 (en) A method of estimating a magnetic pole position in an engine and apparatus for controlling the motor based on the estimated position
DE60024222T2 (en) Method for estimating the rotor position of a synchronous motor, method for controlling a sensorless synchronous motor and a controller for a synchronous motor
US9531313B2 (en) Apparatus for controlling controlled variable of rotary machine to command value
KR100423715B1 (en) Synchronous motor control device and method
Yoon et al. Sensorless control for induction machines based on square-wave voltage injection
JP3502040B2 (en) Brushless DC motor constant detection device, brushless DC motor control device, and brushless DC motor constant detection program
JP5130031B2 (en) Position sensorless control device for permanent magnet motor
Paulus et al. Sensorless field-oriented control for permanent magnet synchronous machines with an arbitrary injection scheme and direct angle calculation
KR100747941B1 (en) Method and apparatus for controling position sensorless motor
JP5291184B2 (en) AC rotating machine control device
JP5652664B2 (en) Rotating electrical machine control device
CN107078674B (en) Control device for inverter and motor driven systems
US7872435B2 (en) Motor control apparatus
JP3755424B2 (en) AC motor drive control device
US9112436B2 (en) System for controlling controlled variable of rotary machine
JP4677852B2 (en) Vector controller for permanent magnet synchronous motor
Liu et al. Sensorless control strategy by square-waveform high-frequency pulsating signal injection into stationary reference frame
US7932692B2 (en) Control system for rotary electric machine with salient structure
JP4754417B2 (en) Control device for permanent magnet type rotating electrical machine
JP3783695B2 (en) Motor control device

Legal Events

Date Code Title Description
A131 Notification of reasons for refusal

Free format text: JAPANESE INTERMEDIATE CODE: A131

Effective date: 20140826

A977 Report on retrieval

Free format text: JAPANESE INTERMEDIATE CODE: A971007

Effective date: 20140828

A521 Written amendment

Free format text: JAPANESE INTERMEDIATE CODE: A523

Effective date: 20141014

TRDD Decision of grant or rejection written
A01 Written decision to grant a patent or to grant a registration (utility model)

Free format text: JAPANESE INTERMEDIATE CODE: A01

Effective date: 20141104

A61 First payment of annual fees (during grant procedure)

Free format text: JAPANESE INTERMEDIATE CODE: A61

Effective date: 20141104

R150 Certificate of patent or registration of utility model

Ref document number: 5645902

Country of ref document: JP

Free format text: JAPANESE INTERMEDIATE CODE: R150

R250 Receipt of annual fees

Free format text: JAPANESE INTERMEDIATE CODE: R250

R250 Receipt of annual fees

Free format text: JAPANESE INTERMEDIATE CODE: R250