JP5230682B2 - Control device for synchronous motor - Google Patents

Description

  The present invention relates to a synchronous motor control device that controls the driving of a synchronous motor used in, for example, an electric vehicle drive device and an electric power steering device.

  As a method for driving the synchronous motor, AC voltage driving using a PWM (Pulse Width Modulation) method is known. As the PWM method, there are an asynchronous PWM method and a synchronous PWM method. The asynchronous PWM system is a system in which the period of the carrier wave generated by the time counter is fixed, and the period of the carrier wave and the period of the command AC wave calculated by the arithmetic unit are not synchronized.

  On the other hand, the synchronous PWM method is a method in which the cycle of the carrier wave and the cycle of the command AC wave are synchronized. Therefore, in the synchronous PWM method, the number of carrier waves included in one command AC wave is constant, and the cycle of the carrier wave generated by the phase counter is different from the fixed cycle of the carrier wave in the asynchronous PWM method, and the rotational speed of the synchronous motor It becomes variable according to.

  When the synchronous motor is driven by the asynchronous PWM method, as the rotational speed of the synchronous motor increases, the carrier wave cycle and the AC voltage cycle become closer, and the carrier wave cycle and the command AC wave cycle are not synchronized. There was a risk that the waveform collapsed. Therefore, there has been a problem that harmonic components are increased and the motor efficiency is lowered.

  Therefore, in order to solve the above problem, when the synchronous motor is controlled at high speed, the harmonic component is suppressed by synchronizing the period of the carrier wave and the period of the command AC wave using the synchronous PWM method. Thus, a method for suppressing a decrease in motor efficiency has been proposed.

  By the way, in the case where the synchronous motor is driven with an AC voltage using the PWM method, when the sample value control is executed, the update of the AC voltage command value used for the PWM calculation is discrete with respect to the voltage of the continuous command AC wave. It becomes the target. In general, the AC voltage command value is updated at peaks and troughs of the carrier wave so that the update interval is equal. By executing PWM calculation on the updated AC voltage command value and generating the PWM voltage, a current corresponding to the average voltage of the PWM voltage flows to the synchronous motor that is the RL load.

  Here, the average voltage of the PWM voltage is set equal to the average value of the voltage of the continuous command AC wave within the PWM calculation period corresponding to the half cycle of the carrier wave (hereinafter referred to as “carrier half cycle”). Therefore, the value is equal to the voltage value of the command AC wave at the center of the PWM calculation period, so that the AC voltage command value set in the sample value control is the voltage value of the command AC wave at the start of the PWM calculation. A PWM response delay corresponding to a time delay of 0.5 times the half cycle of the carrier wave occurs.

  In addition, when executing the sample value control, a calculation delay of a certain time occurs until the AC voltage command value is calculated by the calculation device. That is, a delay time is generated by adding a time corresponding to 0.5 carrier half-cycle due to the PWM response delay and a certain time due to the calculation delay.

  Even within such a delay time including a PWM response delay and a calculation delay, the rotor of the synchronous motor rotates and the phase changes. Therefore, an AC voltage cannot be applied to the synchronous motor at an appropriate phase, and the motor There was a problem that efficiency and torque decreased.

  Therefore, in order to solve the above problem, in a synchronous motor control apparatus in which the operation delay is equal to the carrier half-cycle (that is, the delay time is 1.5 carrier half-cycle), the phase delay due to the PWM response delay and the operation delay Has been proposed (see Non-Patent Document 1, for example).

  That is, in Non-Patent Document 1, the phase for 1.5 carrier half cycle is added as a delay compensation amount to the phase of the rotor of the synchronous motor, and this is used as the phase when calculating the AC voltage command value by the arithmetic unit. A method of compensating for the phase delay by using the value after addition is shown.

Takeshi Oda and three other authors, “Estimated Error Factors of Magnetic Position Sensorless PM Motors and Countermeasures”, Proceedings of the 2008 IEEJ Industrial Application Conference, Vol. I-273-I-276

However, the prior art has the following problems.
In the synchronous motor control device of Non-Patent Document 1, the period of the carrier wave generated by the time counter is fixed, and the period of the carrier wave is synchronized with the calculation period of the AC voltage command value, that is, the synchronous motor is operated by asynchronous PWM. The phase delay is compensated on the assumption that the system is driven.

  Here, as described above, in the synchronous PWM method, the cycle of the carrier wave and the cycle of the command AC wave are synchronized, and the number of carrier waves included in one command AC wave is constant. Therefore, since the cycle of the carrier wave changes in accordance with the AC voltage cycle, the carrier wave cycle and the calculation cycle cannot be synchronized.

  Therefore, the phase lag compensation method shown in Non-Patent Document 1 cannot be applied when the synchronous motor is driven by the synchronous PWM method. Therefore, when the synchronous motor rotates at high speed, the phase lag cannot be compensated, and an AC voltage cannot be applied to the synchronous motor at an appropriate phase, which causes a problem that motor efficiency and torque are reduced.

  The present invention has been made to solve the above-described problems, and is a case where the period of the carrier wave becomes variable due to the driving of the synchronous motor by the synchronous PWM method, and cannot be synchronized with the calculation period. Another object of the present invention is to obtain a control device for a synchronous motor that can prevent a reduction in motor efficiency and torque by applying an AC voltage to the synchronous motor at an appropriate phase.

The synchronous motor control device according to the present invention is a synchronous motor control device that controls the driving of the synchronous motor, based on the torque command, the current flowing through the stator of the synchronous motor, and the phase of the rotor of the synchronous motor, A voltage command calculation means for calculating an AC voltage command for the synchronous motor, a PWM inverter for generating an AC voltage for driving the synchronous motor in an asynchronous PWM method or a synchronous PWM method based on the AC voltage command, and an operation state of the synchronous motor Based on the switching means for switching between the asynchronous PWM system and the synchronous PWM system, the voltage command calculation means and the PWM inverter in each of the case where the synchronous motor is driven by the asynchronous PWM system and the case where the synchronous motor is driven by the synchronous PWM system Of the synchronous motor rotor due to computation delay and PWM response delay respectively Comprising phase compensation means for compensating the delay, the voltage command calculation means includes a first coordinate transformation means for coordinate transformation the current flowing through the stator of the synchronous motor in dq current, the torque command, dq current and rotor phase And a second coordinate conversion means for converting the dq voltage command value into an AC voltage command. The phase compensation means is a synchronous PWM. When the synchronous motor is driven by this method, the phase for the calculation delay and the phase change from the previous value of the dq voltage command value are added to the current AC voltage phase, and the AC voltage phase after the addition and the carrier wave When generating a PWM pulse over a half-cycle of carrier wave from peak to valley or from valley to peak, compare the PWM calculation start phase group which is the peak and valley of the carrier, The phase obtained by adding the phase corresponding to the PWM response delay to the PWM calculation start phase following the current voltage phase is defined as the AC voltage phase after phase compensation, and the voltage command calculation means The AC voltage command is calculated using the phase.

According to the control device for a synchronous motor according to the present invention, the phase compensation means includes the voltage command calculation means and the PWM in each of the case where the synchronous motor is driven by the asynchronous PWM method and the case where the synchronous motor is driven by the synchronous PWM method. Compensating for the phase delay of the synchronous motor rotor caused by the calculation delay and PWM response delay respectively generated by the inverter, the voltage command calculation means uses the phase of the rotor phase-compensated by the phase compensation means to output an AC voltage command. calculate.
Therefore, even when the synchronous motor is driven by the synchronous PWM method, the period of the carrier wave becomes variable and synchronization with the calculation period cannot be achieved, the AC motor is applied by applying an AC voltage to the synchronous motor with an appropriate phase. A reduction in efficiency and torque can be prevented.

It is a block block diagram which shows the control apparatus of the synchronous motor which concerns on Embodiment 1 of this invention with a synchronous motor. In the synchronous motor control apparatus according to Embodiment 1 of the present invention, it is an explanatory diagram showing a phase compensation process when the synchronous motor is driven by an asynchronous PWM method. In the synchronous motor control apparatus according to Embodiment 1 of the present invention, it is an explanatory diagram showing phase compensation processing when the synchronous motor is driven by a synchronous PWM method. (A), (b) is explanatory drawing which shows the effect by the phase compensation process which concerns on Embodiment 1 of this invention.

Hereinafter, preferred embodiments of a control apparatus for a synchronous motor according to the present invention will be described with reference to the drawings. In the drawings, the same or corresponding parts will be described with the same reference numerals.
In the following embodiments, a case will be described in which the synchronous motor control device is a magnetic pole position sensorless control device.

Embodiment 1 FIG.
FIG. 1 is a block configuration diagram showing a synchronous motor control device 1 according to Embodiment 1 of the present invention together with a synchronous motor 2. In FIG. 1, a synchronous motor control device 1 controls the driving of a synchronous motor 2. The synchronous motor 2 has a stator 3 and a rotor 4.

  The control apparatus 1 for the synchronous motor includes a field current control means 11, a subtractor 12, a PI compensator 13, a subtractor 14, a dq voltage command calculation means 15 (voltage command calculation means, dq voltage command calculation means), an integration 16, asynchronous PWM / synchronous PWM switching means 17, phase compensation means 18, dq-3 phase coordinate conversion means 19 (voltage command calculation means, second coordinate conversion means), PWM inverter 20, 3 phase-dq coordinate conversion means 21 (First coordinate conversion means), total current calculation means 22 and frequency compensation means 23 are provided.

  Since the synchronous motor control device 1 is a magnetic pole position sensorless control device as described above, it cannot know the dq axes. Therefore, the synchronous motor 2 is energized using the virtual dq axis set on the control side, and the total magnetic flux of the rotor magnetic flux and the armature reaction magnetic flux is maintained at a constant value. Use matching magnetic flux vector control. Thereby, the voltage can be controlled without losing sight of the magnetic pole position.

The field current control unit 11 controls the field current of the rotor 4 based on the input field current command and the detected field current value, and generates a field magnetic flux from the rotor 4.
The subtracter 12 subtracts the total current calculated by the total current calculation means 22 from the input torque command, and outputs a torque deviation. That is, the control apparatus 1 for a synchronous motor has a torque feedback configuration in which torque control is performed by using a torque deviation as a control operation amount. The function of the total current calculation means 22 will be described later.

The PI compensator 13 calculates a rotational speed command value ω * based on the torque deviation from the subtractor 12 and outputs the rotational speed command value ω *.
The subtracter 14 subtracts the frequency compensation amount fc calculated by the frequency compensation means 23 from the rotational speed command value ω * from the PI compensator 13 and outputs the rotational speed command value ωc * after frequency compensation. The function of the frequency compensation means 23 will be described later.

  The dq voltage command calculation means 15 includes the input field magnetic flux, the rotational speed command value ωc * after frequency compensation from the subtractor 14, the d-axis current Id and q-axis from the three-phase-dq coordinate conversion means 21. Based on the current Iq, the d-axis voltage command Vd * and the q-axis voltage command Vq * are calculated and output. In the first embodiment, dq voltage command values (d-axis voltage command Vd * and q-axis voltage command Vq *) are calculated based on an algorithm of magnetic flux vector control which is one of sensorless controls. Based on this, the dq voltage command value may be calculated.

  The integrator 16 calculates the phase θ of the rotor 4 based on the rotation speed command value ωc * after frequency compensation from the subtractor 14 and outputs the phase θ of the rotor 4. In other words, the integrator 16 is a magnetic pole position calculation means for calculating the phase θ of the rotor 4. Specifically, the integrator 16 calculates the phase θ of the rotor 4 by adding the phase advance for the calculation period to the previous phase of the rotor 4. This phase advance is obtained by multiplying the calculation cycle by the rotational speed command value ωc * after frequency compensation.

  The asynchronous PWM / synchronous PWM switching means 17 drives the synchronous motor 2 by the asynchronous PWM method or synchronizes by the synchronous PWM method based on the rotational speed command value ωc * (operation state) after frequency compensation from the subtractor 14. It switches whether the electric motor 2 is driven. Specifically, the asynchronous PWM / synchronous PWM switching means 17 selects the asynchronous PWM method when the rotational speed command value ωc * after frequency compensation is, for example, 100,000 rpm or less, and selects the asynchronous PWM method from 100,000 rpm. If it is higher, the PWM method is switched, such as selecting the synchronous PWM method.

  In the switching of the PWM method by the asynchronous PWM / synchronous PWM switching means 17, when the rotational speed command value ωc * after frequency compensation fluctuates around the above 100,000 rpm, the switching of the PWM method is performed a plurality of times. May become unstable. Therefore, the asynchronous PWM / synchronous PWM switching means 17 has a hysteresis (hysteresis means) with respect to the set rotational speed, or after the rotational speed command value ωc * after frequency compensation reaches the set rotational speed. The timer means for switching the PWM system after a certain time has elapsed.

Here, it is conceivable that the hysteresis has a hysteresis of about several percent (for example, ± 2%) with respect to the set rotation speed. The asynchronous PWM / synchronous PWM switching means 17 may have both hysteresis and timer means.
As a result, even if the rotational speed command value ωc * after frequency compensation fluctuates before and after the set rotational speed, the switching of the PWM system is prevented from being executed continuously, and the control stability is improved. Can be improved.

  The phase compensation means 18 is based on the rotational speed command value ωc * after frequency compensation from the subtractor 14, the phase θ of the rotor 4 from the integrator 16, and the PWM method switched by the asynchronous PWM / synchronous PWM switching means 17. When the synchronous motor 2 is driven by the asynchronous PWM method and when the synchronous motor 2 is driven by the synchronous PWM method, the dq voltage command calculating means 15, the dq-3 phase coordinate converting means 19 and the PWM inverter 20 are used. Compensate for the phase delay of the rotor 4 due to the calculation delay, phase change delay and PWM response delay that occur, and output the rotor phase θc after phase compensation. The detailed function of the phase compensation means 18 will be described later.

  The dq-3 phase coordinate conversion unit 19 is based on the d-axis voltage command Vd * and the q-axis voltage command Vq * from the dq voltage command calculation unit 15 and the rotor phase θc after phase compensation from the phase compensation unit 18. The U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * are calculated and output. The U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * are represented by the following equations (1) to (3), respectively.

    Vu * = Vd * · cos θc−Vq * · sin θc (1)

    Vv * = Vd * · cos (θc + 240 °) −Vq * · sin (θc + 240 °) (2)

    Vw * = − (Vu * + Vv *) (3)

  The PWM inverter 20 is switched by the asynchronous PWM / synchronous PWM switching means 17 from the U-phase voltage command Vu *, the V-phase voltage command Vv * and the W-phase voltage command Vw * from the dq-3 phase coordinate conversion means 19. The AC voltage for driving the synchronous motor 2 is generated based on the asynchronous PWM method or the synchronous PWM method.

  Specifically, when the synchronous motor 2 is driven by the asynchronous PWM method, the PWM inverter 20 generates a PWM pulse using a carrier wave having a constant period based on a time counter and a three-phase AC voltage command value. Further, when the synchronous motor 2 is driven by the synchronous PWM method, the PWM inverter 20 generates a PWM pulse using a carrier wave based on a phase counter that counts a three-phase AC voltage phase and a three-phase AC voltage command value. .

  Here, the time counter first prepares a count upper limit value set based on the carrier half-cycle and the time count width. Subsequently, the count value is incremented from 0 for each count time width, and after the count value reaches the count upper limit value, the count value is decremented to 0. By repeating this process, a carrier wave having a desired period can be generated.

  Further, the phase counter first prepares a count upper limit value set based on the phase shift amount and the phase count width that advance in the carrier half-cycle. Subsequently, the count value is incremented from 0. At this time, the count value is incremented each time the voltage phase advances while referring to the AC voltage phase. Next, after the count value reaches the count upper limit value, the count value is decremented to zero. Also, if the dq voltage command value is updated and there is a phase change, the count value is changed based on the value obtained by adding the phase change amount due to the update of the dq voltage command value to the referenced AC voltage phase To do.

  Based on the phase θ of the rotor 4 from the integrator 16, the three-phase-dq coordinate conversion means 21 converts the dq current value (d-axis current Id and q-axis current from the three-phase current flowing in the stator 3 of the synchronous motor 2. Iq) is calculated and output. In the first embodiment, the U-phase current Iu and the V-phase current Iv are detected from the three-phase currents, and the dq current value is calculated based on these detected currents. However, the present invention is not limited to this. The phase current to be detected may be any two of the three phases. In addition to the following, the d-axis current Id and the q-axis current Iq calculated by the three-phase-dq coordinate conversion means 21 are used for calculation of a torque command.

The total current calculation means 22 calculates the square root of the square sum of the d-axis current Id and the q-axis current Iq from the three-phase-dq coordinate conversion means 21, and outputs the total current.
The frequency compensation means 23 calculates and outputs a frequency compensation amount fc for compensating the frequency of the rotational speed command value ω * from the PI compensator 13 based on the q-axis current Iq from the three-phase-dq coordinate conversion means 21. To do. The frequency compensation amount fc is expressed by the following equation (4).

fc = (k _hpf × ω _hpf ) × s / (s + ω _hpf ) × Iq (4)

In addition, the definition of the variable indicates that k _hpf is a frequency compensation gain, ω _hpf is a high-pass filter pass frequency, and s is a Laplace operator.

Next, the detailed function of the phase compensation means 18 will be described.
First, the phase compensation processing of the phase compensation means 18 when the synchronous motor 2 is driven by the asynchronous PWM method will be described with reference to FIG.

In the first embodiment, similarly to the non-patent document 1 described above, the period of the carrier wave generated by the time counter is fixed, and the period of the carrier wave is synchronized with the calculation period of the AC voltage command value. Yes.
As shown in FIG. 2, when the calculation time of the three-phase AC voltage command value is the carrier half-cycle Thc and the three-phase AC voltage command value is updated in the peak and valley of the carrier wave, A phase delay of ωc * · Thc occurs with the calculation delay.

  Further, the average voltage of the PWM voltage needs to be equal to the average value of the voltage of the continuous command AC wave in the PWM calculation period corresponding to the half cycle of the carrier wave, and the value is in the center of the PWM calculation period. It is equal to the voltage value of the command AC wave. Therefore, if the AC voltage command value set in the sample value control is the voltage value of the command AC wave at the start of PWM calculation, a PWM response delay corresponding to a time delay of 0.5 times the half cycle of the carrier wave occurs. Therefore, in order to compensate for this PWM response delay, a phase addition of 0.5ωc * · Thc is required.

  Therefore, the phase compensation means 18 adds 1.5ωc * · Thc, which is the phase sum of the calculation delay and the PWM response delay, to the phase θ of the rotor 4 from the integrator 16, and the rotor phase θc after phase compensation. Is calculated and output as a phase for coordinate conversion in the dq-3 phase coordinate conversion means 19.

  Next, the phase compensation processing of the phase compensation means 18 when the synchronous motor 2 is driven by the synchronous PWM method will be described with reference to FIG.

  In FIG. 3, when the dq voltage command value is updated, the phase compensation means 18 first refers to the current three-phase AC voltage phase φ from the phase counter, and the three-phase AC voltage command value is calculated for the calculation time Tcal. The phase delay and the phase change δ generated when the dq voltage command value is updated from the previous values Vd * and Vq * to Vd_n * and Vq_n *, respectively, are added. The phase change δ is expressed by the following equation (5).

    δ = arctan (−Vd_n * / Vq_n *) − arctan (−Vd * / Vq *) (5)

Further, the three-phase AC voltage phase φ_n after the addition is expressed by the following equation (6). Thereby, the phase at the time of generating an AC voltage command value can be calculated.
φ_n = φ + ω * · Tcal + δ (6)

  Subsequently, the phase compensation means 18 compares the added three-phase AC voltage phase φ_n with the PWM calculation start phase group. Here, as shown in FIG. 2, the PWM calculation start phase group is generated when, for example, a PWM pulse is generated over a half period of the carrier wave from the peak to the valley of the carrier or from the valley to the peak. The phase is determined by the number of carrier waves entering one AC voltage cycle. Assuming that the number of carriers included in one AC voltage cycle is N, the PWM calculation start phase group φi is expressed by the following expression (7) using an integer value i of 0 to 2N−1.

    φi = 180i / N + 90 / N [deg] (7)

The PWM calculation start phase group need not be limited to the peaks and valleys of the carrier wave, and may be phases shifted from the peaks and valleys.
Further, when the synchronous motor 2 is driven by the synchronous PWM method, the phase corresponding to the PWM response delay described above is a phase corresponding to 0.5 times the half cycle of the carrier wave regardless of the PWM calculation start phase group. That is, when the number of carrier waves entering one cycle of the AC voltage is N, the phase of the PWM response delay is expressed as 90 / N [deg], and it can be seen that it does not depend on the rotational speed of the synchronous motor 2.

  The phase compensation means 18 compares the three-phase AC voltage phase φ_n after addition with the PWM calculation start phase group, selects the next PWM calculation start phase, and the phase 90 / N corresponding to the PWM response delay is selected as the phase. Is the phase of the three-phase AC voltage command value. Therefore, the rotor phase θc after phase compensation is expressed by the following equation (8) using the selected PWM calculation start phase φi (i = 0, 1,..., 2N).

    θc = φi + 90 / N-arctan (−Vd_n * / Vq_n *) (8)

  In FIG. 3, φ3 is the PWM calculation start phase, and PWM calculation is executed between φ3 and φ4. Note that FIG. 3 shows a case where the phase change δ is a negative value, but even if the phase change δ is 0 and a positive value, phase compensation can be executed by the same method. it can.

  FIG. 4 is an explanatory diagram showing the effect of the phase compensation processing according to Embodiment 1 of the present invention. FIG. 4 shows the effect of phase compensation when the torque proportional to the q-axis current Iq is controlled with the d-axis current Id = 0. Further, the d′-q coordinate is the coordinate when the dq coordinate synchronized with the phase of the rotor 4 advances by the phase φd corresponding to the calculation delay and the PWM response delay.

  The synchronous motor control apparatus 1 according to the first embodiment of the present invention is not only for driving the synchronous motor 2 by the asynchronous PWM method but also for the case of driving the synchronous motor 2 by the synchronous PWM method. An effect can be obtained. That is, as shown in FIG. 4A, when phase compensation is not executed, a shift from the q′-axis in the direction of the q-axis current Iqcom, that is, a shift in AC phase occurs, so that the q′-axis current decreases. Torque decreases. On the other hand, as shown in FIG. 4B, when phase compensation is executed, Iqcom is obtained as the q′-axis current, so that a reduction in torque can be suppressed.

As described above, according to the first embodiment, the phase compensation means includes the voltage command calculation means and the PWM in each of the case where the synchronous motor is driven by the asynchronous PWM method and the case where the synchronous motor is driven by the synchronous PWM method. Compensating for the phase delay of the synchronous motor rotor caused by the calculation delay and PWM response delay respectively generated by the inverter, the voltage command calculation means uses the phase of the rotor phase-compensated by the phase compensation means to output an AC voltage command. calculate.
Further, when the synchronous motor is driven by the synchronous PWM method, the phase compensation means calculates the phase of the calculation delay and the phase change generated when the dq voltage command is coordinate-converted into the AC voltage command. Add the phase to the phase, compare the AC voltage phase after the addition with the PWM calculation start phase group, and add the phase corresponding to the PWM response delay to the PWM calculation start phase next to the AC voltage phase after the addition. The compensated AC voltage phase.
Therefore, even when the synchronous motor is driven by the synchronous PWM method, the period of the carrier wave becomes variable and synchronization with the calculation period cannot be achieved, the AC motor is applied by applying an AC voltage to the synchronous motor with an appropriate phase. A reduction in efficiency and torque can be prevented.

  In the first embodiment, the case where the synchronous motor control device 1 is a magnetic pole position sensorless control device has been described. However, a control device using a magnetic pole position detection sensor may be used. By using the magnetic pole position detection sensor for detecting the magnetic pole position, the integrator 16 shown in FIG. 1 is not necessary, and by using the differentiator, the actual rotational speed of the synchronous motor 2 can be calculated. In addition, closed loop control of the motor rotation speed can be executed by feeding back the actual rotation speed.

  In the first embodiment, the configuration is shown in which the vector control is performed using the dq coordinate system. However, the configuration may be such that the AC voltage itself is controlled without using the dq coordinate system. In this case, it is not necessary to consider the phase change in the update of the dq voltage command value, so that the phase compensation process when driving the synchronous motor 2 by the synchronous PWM method becomes simple.

In the first embodiment, the number of alternating phases is three, but the number of alternating phases is not limited to this.
In the first embodiment, the field magnetic flux is generated by the field current control. However, the field magnetic flux may be generated by a rotor using a permanent magnet. By using a permanent magnet for the rotor, a field current is not required, so that the motor efficiency can be improved.

  In the first embodiment, the switching rotational speed in the asynchronous PWM / synchronous PWM switching means 17 is 100,000 rpm and the hysteresis is ± 2%. However, these values are the performance of the microcomputer and the performance of the synchronous motor. Therefore, a different value is set for each applied system.

  In the first embodiment, the effect of the phase compensation process in the control for setting the d-axis current Id = 0 in FIG. 4 has been described. However, the control apparatus 1 for the synchronous motor according to the first embodiment of the present invention is as follows. The present invention is not limited to a specific control algorithm, and can be applied to various other control algorithms.

  DESCRIPTION OF SYMBOLS 1 Control apparatus of synchronous motor, 2 Synchronous motor, 3 Stator, 4 Rotor, 11 Field current control means, 12 Subtractor, 13 PI compensator, 14 Subtractor, 15 dq voltage command calculating means, 16 Integrator, 17 asynchronous PWM / synchronous PWM switching means, 18 phase compensation means, 19 dq-3 phase coordinate conversion means, 20 PWM inverter, 21 3 phase-dq coordinate conversion means, 22 total current calculation means, 23 frequency compensation means.

Claims (3)

A control device for a synchronous motor that controls driving of the synchronous motor,
Voltage command calculation means for calculating an AC voltage command for the synchronous motor based on a torque command, a current flowing through the stator of the synchronous motor and a phase of the rotor of the synchronous motor;
A PWM inverter that generates an AC voltage for driving the synchronous motor in an asynchronous PWM method or a synchronous PWM method based on the AC voltage command;
Switching means for switching between the asynchronous PWM method and the synchronous PWM method based on the operating state of the synchronous motor;
In each of the case where the synchronous motor is driven by the asynchronous PWM method and the case where the synchronous motor is driven by the synchronous PWM method, calculation delays and PWM response delays respectively generated by the voltage command calculation means and the PWM inverter are caused. and a phase compensation means for compensating the phase lag of the rotor of the synchronous motor,
The voltage command calculation means is
First coordinate conversion means for converting the current flowing in the stator of the synchronous motor into a dq current;
Dq voltage command calculation means for calculating a dq voltage command value for the synchronous motor based on the torque command, the dq current and the phase of the rotor;
Second coordinate conversion means for converting the dq voltage command value into the AC voltage command,
When the synchronous motor is driven by the synchronous PWM method, the phase compensation means adds the phase for the calculation delay and the phase change from the previous value of the dq voltage command value to the current AC voltage phase. In the case where the PWM pulse is generated by taking the AC voltage phase after the addition and the half-cycle of the carrier wave from the peak to the valley of the carrier or from the valley to the peak, the PWM that is the peak and valley of the carrier. Comparing the calculation start phase group, the phase obtained by adding the phase corresponding to the PWM response delay to the PWM calculation start phase following the AC voltage phase after the addition is defined as the AC voltage phase after phase compensation,
The control apparatus for a synchronous motor, wherein the voltage command calculation means calculates the AC voltage command using the phase of the rotor phase-compensated by the phase compensation means.   2. The synchronous motor control device according to claim 1, wherein the operation state of the synchronous motor is a rotation speed of the synchronous motor. The synchronous motor control device according to claim 1, wherein the switching unit includes at least one of a hysteresis unit and a timer unit when switching between the asynchronous PWM method and the synchronous PWM method.

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