JP4552576B2 - Magnetic pole position estimation method, magnetic pole position estimation apparatus, inverter control method, and inverter control apparatus - Google Patents

Magnetic pole position estimation method, magnetic pole position estimation apparatus, inverter control method, and inverter control apparatus Download PDF

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JP4552576B2
JP4552576B2 JP2004274972A JP2004274972A JP4552576B2 JP 4552576 B2 JP4552576 B2 JP 4552576B2 JP 2004274972 A JP2004274972 A JP 2004274972A JP 2004274972 A JP2004274972 A JP 2004274972A JP 4552576 B2 JP4552576 B2 JP 4552576B2
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magnetic pole
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estimated value
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inductance
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広之 山井
守満 関本
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ダイキン工業株式会社
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  The present invention relates to a technique for estimating a magnetic pole position of a motor.

  When operating the motor, it is necessary to know the magnetic pole position of the motor so that problems such as reverse rotation of the motor do not occur. In order to know the magnetic pole position, for example, there is a method using a position sensor. The method using the position sensor has a problem that it is expensive and the position sensor cannot be attached when the motor is provided in the high-temperature and high-pressure part.

  For this reason, conventionally, a method for estimating the magnetic pole position of a motor without using a position sensor has been considered. For example, there are a method using the saliency of the motor and a method using the induced voltage of the motor. Many of these methods estimate the magnetic pole position of the motor by solving the voltage equation of the motor. In this case, preset device constants are substituted into the voltage equation.

  Non-patent documents 1 to 5 are shown below as related documents.

Chen Zhen, 3 others, "Sensorless position estimation method and stability of salient pole type brushless DC motor", 1998 IEEJ National Conference on Industrial Applications, No. 59, 1998, p. 179-182 Masayuki Kamimae, two others, "Online Parameter Identification of PMSM Drive System", IEEJ Semiconductor Power Conversion Study Group Material, February 2003, SPC-03-3, p. 13-18 Kazunori Yamada and two others, "Position sensorless speed control method of salient pole type PM motor including low speed region", IEEJ Semiconductor Power Conversion Study Group Material, January 1997, SPC-97-13, p75-82 Kiyoshi Sakamoto and two others, "Position Sensorless Control of IPM Motors by Direct Estimation of Axis Error", IEEJ Semiconductor Power Conversion / Industry Power Electrical Application Joint Study Group, November 2000, SPC-00-67, p73 -76 Shigeya Tanimoto, "Brushless DC motor for air conditioner / compressor", '94 Motor Technology Symposium (hosted by Japan Management Association), 1994, C5-1-1 to C5-1-16

  In the method of estimating the magnetic pole position of the motor without using the position sensor, an error that occurs in the device constant is affected, and an error also occurs in the magnetic pole position estimated from the voltage equation.

  In the case of mass production of motors, it is difficult to make the dimensions and shapes of the motors the same, resulting in variations in equipment constants. In addition, depending on the operating condition of the motor, the device constant changes due to the influence of magnetic saturation. It is conceivable to measure the device constants accurately in order to reduce the errors generated in these device constants. For example, Non-Patent Document 2 proposes a method for identifying device constants using an observer.

  However, the above-described method is intended to identify device constants and is not intended to reduce an error that occurs in the magnetic pole position when the magnetic pole position is estimated. For this reason, when the magnetic pole position is estimated using the device constant identified by the above method, it cannot be said that the error generated in the magnetic pole position is reduced.

  The present invention has been made in view of the above-described circumstances, and an object of the present invention is to efficiently reduce errors generated in the magnetic pole position when estimating the magnetic pole position of the motor.

A magnetic pole position estimation method according to claim 1 of the present invention is a method for estimating a magnetic pole position of a motor, wherein the current (i α , i β ; i d ^ , i q ^ ) flowing in the motor and / or the A first estimation value (θ ^ ef1 ) of the magnetic pole position based on the motor voltage (v α , v β ; v d ^ , v q ^ ) and the motor inductance (L ^ q ) Performing an estimation method (1A), performing a second estimation method (1B; 21B) for obtaining a second estimated value (θ ^ e2 ) of the magnetic pole position without depending on the inductance, and the first A step (12) of obtaining an error estimated value (Δθ ^ e ) as a difference between the estimated value of the error and the second estimated value; updating the inductance based on the estimated value of the error; and Using the first Performing an estimation method.

A magnetic pole position estimation method according to a second aspect of the present invention is the magnetic pole position estimation method according to the first aspect, wherein in the first estimation method (1A), the first estimated value (θ ^ ef1 ) The pulsating component with respect to the rotation of the motor is removed.

A magnetic pole position estimation method according to a third aspect of the present invention is the magnetic pole position estimation method according to the first or second aspect, wherein in the second estimation method (21B), no current is supplied to the motor. The second estimated value (θ ^ e2 ) is obtained based on the voltage (v α , v β ).

A magnetic pole position estimation method according to claim 4 of the present invention is the magnetic pole position estimation method according to claim 1 or 2, wherein in the second estimation method, the second estimated value (θ ^ e2 ) is used. A predetermined magnetic pole position for each rotation of the motor is obtained.

According to a fifth aspect of the present invention, there is provided a magnetic pole position estimation method for estimating a magnetic pole position of a motor, wherein the current (i α , i β ; i d ^ , i q ^ ) flowing in the motor and / or the A first estimation value (θ ^ ef1 ) of the magnetic pole position based on the motor voltage (v α , v β ; v d ^ , v q ^ ) and the motor inductance (L ^ q ) An error of the magnetic pole position based on the step (1A) of performing the estimation method and the harmonic component of the voltage in a state where the harmonic current ( id ^ h , iq ^ h ) is passed through the motor A step (6) of performing a second estimation method for obtaining an estimated value (Δθ ^ e ), updating the inductance based on the estimated value of the error, and using the updated inductance, the first estimation Performing the method.

A magnetic pole position estimation method according to a sixth aspect of the present invention is the magnetic pole position estimation method according to the fifth aspect, wherein in the second estimation method (6), the first estimated value (θ ^ ef1 ). And a first coordinate conversion step of performing rotational coordinate conversion on the voltage (v α , v β ) of the motor based on the phase angle (θ h ) of the harmonic current (i d ^ h , i q ^ h ). (601), a first filtering step (602) for filtering the result of the first coordinate transformation step in the vicinity of the angular frequency (ω h ) of the harmonic current, and the angular frequency of the harmonic current. Based on the result of the first filtering step, a second coordinate transformation step (603) for performing rotational coordinate transformation, and a low frequency of the result of the second coordinate transformation step (v δh ; v δh , v γh ) component (v δh0; v δh0, v γh0) , said high And; (606 605) and the second filtering step (604) for filtering at the angular frequency near the wave current, the second estimate of the error based on the result of the filtering step ([Delta] [theta] ^ e) determining a Is done.

According to a seventh aspect of the present invention, in the inverter control method, the first estimated value (θ ^ ef1 ) obtained by the magnetic pole position estimating method according to any one of the first to sixth aspects is used as the magnetic pole position. Based on this, an inverter that drives the motor is controlled.

According to an eighth aspect of the present invention, there is provided a magnetic pole position estimating apparatus for estimating a magnetic pole position of a motor, wherein the current (i α , i β ; i d ^ , i q ^ ) flowing in the motor and / or the A first estimation value (θ ^ ef1 ) of the magnetic pole position based on the motor voltage (v α , v β ; v d ^ , v q ^ ) and the motor inductance (L ^ q ) An estimation means (1A), a second estimation means (1B; 21B) for obtaining a second estimated value (θ ^ e2 ) of the magnetic pole position without depending on the inductance, the first estimated value and the first Error estimation means (12) for obtaining an error estimation value (Δθ ^ e ) as a difference between the two estimation values, and an inductance calculation unit (5) for updating the inductance based on the error estimation value.

A magnetic pole position estimation apparatus according to a ninth aspect of the present invention is the magnetic pole position estimation apparatus according to the eighth aspect, wherein the first estimation means (1A) is based on the first estimated value (θ ^ ef1 ). The filter (3) which removes the pulsation component with respect to rotation of the said motor is provided.

A magnetic pole position estimation apparatus according to a tenth aspect of the present invention is the magnetic pole position estimation apparatus according to the eighth or ninth aspect, wherein the second estimation means (21B) does not pass a current to the motor. The second estimated value (θ ^ e2 ) is obtained based on the voltage (v α , v β ).

A magnetic pole position estimation apparatus according to an eleventh aspect of the present invention is the magnetic pole position estimation apparatus according to the eighth or ninth aspect, wherein the second estimation means is the second estimated value (θ ^ e2 ). A predetermined magnetic pole position for each rotation of the motor is obtained.

A magnetic pole position estimation apparatus according to a twelfth aspect of the present invention is an apparatus for estimating a magnetic pole position of a motor, wherein the current (i α , i β ; i d ^ , i q ^ ) flowing in the motor and / or the A first estimation value (θ ^ ef1 ) of the magnetic pole position based on the motor voltage (v α , v β ; v d ^ , v q ^ ) and the motor inductance (L ^ q ) An estimated value of the magnetic pole position error based on the estimation means (1A) and the harmonic component of the voltage in a state where harmonic currents ( id ^ h , iq ^ h ) are passed through the motor. 2nd estimation means (6) which calculates | requires ((DELTA) (theta) ^ e ), and the inductance calculation part (5) which updates the said inductance based on the estimated value of the said error.

A magnetic pole position estimation apparatus according to a thirteenth aspect of the present invention is the magnetic pole position estimation apparatus according to the twelfth aspect, wherein the second estimation method (6) includes the first estimated value (θ ^ ef1 ) and Based on the phase angle (θ h ) of the harmonic current (i d ^ h , i q ^ h ), a first coordinate conversion unit that performs rotational coordinate conversion on the motor voltages (v α , v β ) ( 601), a bandpass filter (602) for filtering the output of the first coordinate transformation unit in the vicinity of the angular frequency (ω h ) of the harmonic current, and the angular frequency of the harmonic current, the second coordinate conversion unit for rotating coordinate transformation to the output of the band-pass filter and (603), the output of the second coordinate conversion unit (v δh; v δh, v γh) of the low frequency components (v δh0; v δh0, the v γh0), with the angular frequency near the harmonic current And a; (606 605) the wave to the low-pass filter (604), the estimated value of said error on the basis of the output of the low-pass filter ([Delta] [theta] ^ e) calculation unit for obtaining the.

An inverter control device according to a fourteenth aspect of the present invention provides the first estimated value (θ ^ ef1 ) obtained by the magnetic pole position estimating device according to any one of the eighth to thirteenth aspects as the magnetic pole position. Based on this, an inverter that drives the motor is controlled.

  According to the magnetic pole position estimation method according to claims 1 and 5 of the present invention or the magnetic pole position estimation apparatus according to claims 8 and 12, a response is obtained by calculating and determining the magnetic pole position for each sampling of current and / or voltage. In addition, it is possible to efficiently obtain an accurate magnetic pole position.

  According to the magnetic pole position estimation method according to the second aspect of the present invention or the magnetic pole position estimation device according to the ninth aspect, it is possible to suppress the influence of current pulsation on the first estimated value of the magnetic pole position.

  According to the magnetic pole position estimation method according to the third and fourth aspects of the present invention or the magnetic pole position estimation device according to the tenth and eleventh aspects, the second estimated value of the magnetic pole position can be obtained without depending on the inductance.

  According to the magnetic pole position estimation method according to the sixth aspect of the present invention or the magnetic pole position estimation apparatus according to the thirteenth aspect, an estimated value of the magnetic pole position error can be obtained.

  According to the inverter control method of the seventh aspect of the present invention or the inverter control device of the fourteenth aspect of the present invention, the voltage to be applied to the motor is generated based on the magnetic pole position that is accurately obtained. Therefore, the motor can be accurately controlled using the inverter.

First embodiment.
A block diagram showing a magnetic pole position estimation method according to the present embodiment is shown in FIG. The relationship between the magnetic pole position and the coordinate axis is shown in FIG. In FIG. 2, the α-β axis orthogonal to the rotation axis of the magnetic pole as the origin 100 is fixed. As a rotation coordinate corresponding to the rotation of the magnetic pole, a coordinate composed of a d-axis from the origin 100 to the N-pole direction of the magnetic pole, for example, and a q-axis perpendicular to the d-axis is employed. And the magnetic pole position, based on the positive part of the example α-axis, the angle theta e formed by the d-axis direction around the opposite watch is employed. The d ^ -q ^ axis is an estimated axis, and the d ^ axis forms an angle θ ^ e corresponding to the estimated value of the position in the direction opposite to the clock with the positive part of the α axis as a reference. In the following, α, β, d, q, d ^, q ^ are used as subscripts for variables such as current and voltage in order to indicate that they are components in the axial direction.

The block diagram (FIG. 1) showing the magnetic pole position estimation method includes a first estimation unit 1A, a second estimation unit 1B, a sample / hold unit 4, an integration unit 5, and an adder / subtractor 12. The first estimating means 1A and the second estimating means 1B obtain a first estimated value θ ^ ef1 and a second estimated value θ ^ e2 of the magnetic pole position, respectively. The first estimation means 1A includes an original estimation unit 2 and a filter 3. The original estimation unit 2 obtains the original estimated value θ ^ e1 of the magnetic pole position. The filter 3 filters the original estimated value θ ^ e1 to obtain a first estimated value θ ^ ef1 . The sample / hold unit 4 includes sample / hold elements 4a and 4b, and performs sample / hold on the original estimated value θ ^ e1 and the first estimated value θ ^ ef1 at timings described later.

The adder / subtracter 12 subtracts the output of the sample / hold element 4b from the output of the sample / hold element 4a to obtain an error estimated value Δθ ^ e . Therefore, the adder / subtractor 12 can be grasped as error estimation means. The integration unit 5 integrates the estimated value Δθ ^ e of the error and multiplies it by a predetermined coefficient to obtain the estimated value L ^ q of the q-axis inductance. Therefore, the integrating unit 5 can be grasped as an inductance calculating unit.

Since the second estimated value θ ^ e2 is obtained as the output of the sample / hold element 4a, it can be understood that the second estimating means 1B includes the original estimating unit 2 and the sample / hold element 4a. That is, the original estimation unit 2 is shared between the first estimation unit 1A and the second estimation unit 1B, and the sample / hold element 4a is shared between the sample / hold unit 4 and the second estimation unit 1B.

As a first step, a first estimation value θ ^ ef1 of the magnetic pole position is obtained by the first estimation means 1A. First, the original estimation unit 2 obtains an original estimated value θ ^ e1 based on the currents i α and i β flowing through the motor, the motor voltages v α and v β and the estimated value L ^ q of the motor q-axis inductance. The currents i α and i β are the α component and β component of the current with respect to the fixed α-β axis. The same applies to the voltages v α and v β . These can be measured using known techniques. When the original estimation unit 2 first obtains the original estimated value θ ^ e1 , the rated value can be adopted as the initial estimated value L ^ q0 of the q-axis inductance estimated value L ^ q .

FIG. 3 is a graph showing the behavior of the original estimated value θ ^ e1 . When the true values θ ~ e of the magnetic pole positions are repeatedly changed (operated) linearly between 0 and 2π, if the currents i d and i q are pulsated for reasons described later, the original estimated value θ ^ e1 Also pulsates. This is because the estimated value L ^ q of the q-axis inductance is different from the true values L to q . Therefore, the filter 3 obtains the first estimated value θ ^ ef1 of the magnetic pole position by filtering the pulsation component of the estimated value θ ^ e1 with respect to the rotation of the motor.

Filtering is performed by executing the blocks 3 a, 3 b, 3 c included in the filter 3 in that order and feeding back to the adder / subtractor 11 along the feedback path 21. The adder / subtractor 11 subtracts the first estimated value θ ^ ef1 from the original estimated value θ ^ e1 . The block 3a converts the output of the adder / subtractor 11 from -π to + π. This is because the standard of the difference between the two is not 2nπ (n is an integer other than 0) but 0. The block 3b gives a straight line having the same inclination as the change of the true values θ to e as an offset by applying the Z transformation expressed by the equation (1) to the output of the block 3a. Further, the pulsation component is removed to suppress the influence of the pulsation of the currents i d and i q . Block 3c converts the estimated value obtained in block 3b from 0 to + 2π. This is because the original estimated value θ ^ e1 is generally obtained in the range of 0 to + 2π.

As a result, the first estimated value θ ^ ef1 of the magnetic pole position is output as a sawtooth wave (FIG. 3B) having a constant inclination. The above method for removing the pulsating component is an example, and other methods can be employed.

As a second step, a second estimated value θ ^ e2 of the magnetic pole position is obtained by the sample / hold element 4a. In the sample / hold element 4a, the original estimated value θ ^ e1 when the current (illustrated by i d and i q in FIG. 3) becomes 0 is extracted, and the value is adopted as the second estimated value θ ^ e2. (FIG. 3). For reasons that will be described later, the estimated value θ ^ e2 does not depend on the estimated value L ^ q of the q-axis inductance of the motor. That is, the estimated value θ ^ e2 is not affected by the error of the q-axis inductance L q and is considered to be equal to the true value θ ~ e when the current is 0. When the situation where the current becomes 0 is obtained in this way, the currents i d and i q pulsate as described above.

As a third step, an error Δθ ^ e is obtained by the adder / subtractor 12. First, in the sample / hold element 4b, a first estimated value θ ^ ef1 when the current becomes 0 is extracted, and this value is indicated as θ ^ ef1 , 0 (FIG. 3). The adder / subtracter 12 subtracts the first estimated value θ ^ ef1,0 when the current is 0 from the second estimated value θ ^ e2 to obtain an error Δθ ^ e when the current is 0 (FIG. 3). .

As a fourth step, the estimated value L ^ q of the q-axis inductance of the motor is updated. The updated estimated value L ^ q of the q-axis inductance is obtained by the integrating unit 5. That is, the estimated value L ^ q of the q-axis inductance is obtained by setting the initial value L ^ q0 and the Z conversion expressed by the equation (2) acting on the error Δθ ^ e . Here, k represents a gain of integration.

The error Δθ ^ e when the current is 0 is caused by the difference between the estimated value L ^ q of the q-axis inductance and the true values L to q, and the estimated value L ^ q of the q-axis inductance depends on the current value. Therefore, the updated estimated value L ^ q of the q-axis inductance approaches the true value L to q . Then, by repeating the first to fourth steps using the updated q-axis inductance estimated value L ^ q , the first estimated value θ ^ ef1 can be brought close to the true value θ ~ e . FIG. 4 shows how the estimated value L ^ q of the q-axis inductance approaches the true values L to q for each update.

The first to fourth iteration of step (q-axis estimated values L ^ q inductance) is, for example, the difference between the estimated value L ^ q of the before and after updating the q-axis inductance and the determined absolute value, the absolute value It is possible to adopt a method of performing the process until it becomes smaller than a certain value, for example, an allowable value as an error.

As a method for obtaining the original estimated value θ ^ e1 described in the first step, for example, a method of calculating using Equation (3) can be employed.

Equation (3) represents a voltage equation on a stationary coordinate indicated by the α-β axis (FIG. 2). Here, R a is an armature resistance, L d is a d-axis inductance, θ e is a magnetic pole position, p is a time differential operator, and φ a is a field main magnetic flux.

If λ α and λ β are defined by equation (4), the magnetic pole position θ e can be represented by equation (5), and equation (3) can be transformed into equation (6). By substituting this into the equation (5), the magnetic pole position θ e can be obtained. Such a technique is introduced in Non-Patent Document 1, for example.

As can be seen from the equation (6), λ α and λ β are determined using the measurable quantities i α , i β , v α , v β and the armature resistance R a and the q-axis inductance L q which are the device constants. The magnetic pole position θ e obtained by Expression (5) also depends on the armature resistance R a and the q-axis inductance L q . However, in a somewhat high speed operation, the voltages v α and v β are large, and R a i α and R a i β can be ignored. Therefore, the magnetic pole position θ e depends only on the q-axis inductance L q as a device constant. If therefore the estimated value L ^ q of the q-axis inductance is known, by adopting it as the q-axis inductance L q of formula (6), the original estimate the magnetic pole position theta e of formula (5) theta ^ e1 Can be adopted as.

If the estimated value L ^ q of the q-axis inductance is larger than the true value L ~ q , the original estimated value θ ^ e1 is estimated to be smaller (i.e., delayed) than the true value θ ~ e. Δθ ^ e is a positive value. In addition, when the estimated value L ^ q of the q-axis inductance is smaller than the true value L to q , the original estimated value θ ^ e1 is estimated to be larger (that is, advanced) than the true value θ to e . The error Δθ ^ e becomes a negative value.

Therefore, in order to approximate the estimated value L ^ q of the q-axis inductance to the true value L ~ q is by increasing the reverse sign to the sign of the estimated value L ^ q error [Delta] [theta] ^ e of the q-axis inductance Good. Therefore, the integration gain k (see Expression (2)) of the integration unit 5 is set to be negative.

As a method for obtaining the original estimated value θ ^ e1 , for example, a method of calculating using the equation (7) can also be adopted. Expression (7) represents a voltage equation on the rotation coordinate indicated by the d ^ -q ^ axis (FIG. 2) which is an estimated axis. Here, θ err represents an axis error of the estimated axis (d ^ −q ^ axis) with the dq axis, ω represents an angular frequency of the dq axis, and ω ^ represents an angular frequency of the estimated axis.

Assuming that the angular frequency and current are constant, an approximation that ignores the term including the differential operator p is applied to Equation (7). Equation (8) is obtained by solving Equation (7) under this approximation with respect to the axis error θ err . The angular frequency ω ^ of the estimated axis is adjusted using, for example, PLL control so that the axis error θ err becomes zero. Then, the estimated value θ ^ e1 of the magnetic pole position is obtained by integrating with respect to time using the adjusted angular frequency ω ^ of the estimated axis. Such a technique is introduced in Non-Patent Document 4, for example.

In the same manner as that for determining the original estimate theta ^ e1 using equation (3) to (6), the term including armature resistance R a is negligible, the estimate L ^ q of the q-axis inductance is known If there is, by adopting this as the q-axis inductance L q of the equation (8), the magnetic pole position θ e obtained by the method can be adopted as the original estimated value θ ^ e1 .

In any of the above methods for determining the original estimate theta ^ e1, current is 0, that (i α, i β) relative to the = (0,0) (d ^ -q ^ axis (i d ^, i q ^) if = (0,0)), the axis error theta err obtained from equation (derived from 5) the magnetic pole position theta e or formula (8) does not depend on the q-axis inductance L q. That is, since the second estimated value θ ^ e2 of the magnetic pole position obtained according to these methods does not depend on the estimated value L ^ q of the q-axis inductance, the true value θ ~ e of the magnetic pole position when the current is zero. It is thought that.

As a method of obtaining the second estimated value θ ^ e2 of the magnetic pole position, a method of calculating based on the voltage when the current is 0 may be employed. This method uses the voltage equation of equation (3). Equation (3) can be transformed into Equation (9). Here, L 0 = (L d + L q ) / 2 and L 1 = (L d −L q ) / 2. By setting the current and the time derivative of the current to 0 (that is, (i α , i β ) = (0, 0), p (i α , i β ) = (0, 0)), equation (9) is (10) Therefore, the magnetic pole position θ e is expressed by the equation (11) only by the voltages v α and v β .

The magnetic pole position θ e (formula (11)) does not depend on the q-axis inductance L q of the motor. That is, the magnetic pole position θ e is not affected by the error of the q-axis inductance L q and is considered to be equal to the true value θ to e when the current and its derivative are zero. Therefore, the magnetic pole position θ e obtained from Equation (11) can be adopted as the second estimated value θ ^ e2 .

A block diagram when this method is used is shown in FIG. Blocks corresponding to FIG. 1 are given the same reference numerals. The second estimating means 21B has an arc tangent part 210, which obtains tan −1 (−v α / v β ) from the input of the voltages v α and v β and outputs it to the sample / hold element 4a. In this case, the sample / holds 4a and 4b perform the sample / hold operation when the current and the time differentiation of the current are zero. Then, as the outputs of the sample / hold elements 4a and 4b, the second estimated value θ ^ e2 and the first estimated value θ ^ ef1,00 when the current and current time derivatives are 0 are obtained. The second estimating means 21B shares the sample / hold element 4a with the sample / hold unit 4.

As a method for obtaining the second estimated value θ ^ e2 , a 120-degree energization method can be adopted particularly when the motor is a three-phase motor. The three-phase voltages are each out of phase by 120 degrees. In this method, the voltage applied to each of the three phases is switched every time the phase changes by 60 degrees. When rotating the motor in one direction, an energization period in which the phase continuously changes by 120 degrees and a non-energization period in which the phase continuously changes by 60 degrees are repeated in each phase.

  When the voltage is measured, the applied voltage and the induced voltage are overlapped and detected during the energization period, but only the induced voltage is detected during the non-energization period. Therefore, the magnetic pole position can be estimated by processing the induced voltage detected during the non-energization period. Such a technique is introduced in Non-Patent Document 5, for example.

As a method for obtaining the second estimated value θ ^ e2 , a sensor may be employed. Unlike a position sensor that constantly detects the magnetic pole position, this sensor only needs to be able to accurately detect the position and time at which the magnetic pole has passed through a certain point. For example, a so-called Z signal indicating the reference position of the rotary encoder can be employed. The position and time are detected only once (that is, every rotation) during one rotation of the magnetic pole, and this is adopted as the second estimated value θ ^ e2 .

Thus, according to the present embodiment, the first estimated value θ ^ ef1 of the magnetic pole position and the second estimated value θ ^ e2 of the magnetic pole position are obtained, and the error estimated value Δθ ^ e is obtained as a difference between them. Ask. Then, by updating the estimated value L ^ q of the q-axis inductance by using the estimated value [Delta] [theta] ^ e of the error, it is possible to reduce the estimated value [Delta] [theta] ^ e of the error. Therefore, by calculating the magnetic pole position for each sampling of voltage and current, it is possible to obtain an accurate magnetic pole position with good responsiveness.

Second embodiment.
A block diagram showing the magnetic pole position estimation method according to the present embodiment is shown in FIG. This block diagram includes a first estimation unit 1A, an error estimation unit 6 as a second estimation unit, and an integration unit 5. The first estimating means 1A obtains a first estimated value θ ^ ef1 of the magnetic pole position. The first estimation means 1A includes an original estimation unit 2 and a filter 3. The original estimation unit 2 obtains the original estimated value θ ^ e1 of the magnetic pole position. The filter 3 filters the original estimated value θ ^ e1 to obtain a first estimated value θ ^ ef1 . The functions and operations described in the first embodiment can be adopted as these functions and operations.

The error estimating means 6 obtains an error estimated value Δθ ^ e . The integration unit 5 integrates the estimated value Δθ ^ e of the error and multiplies it by a predetermined coefficient to obtain the estimated value L ^ q of the q-axis inductance.

As a first step, a first estimated value θ ^ ef1 is obtained by the first estimating means 1A. The method according to this step is the same as the method according to the first step described in the first embodiment. Then, for example, a calculation method using the voltage equation represented by the formula (3) or the formula (7) can be adopted.

As a second step, an error estimated value Δθ ^ e is obtained by the error estimating means 6. As a method for obtaining the error estimated value Δθ ^ e , for example, a method of injecting a high frequency current into the motor can be employed.

For example, the voltage equation (9) on the stationary coordinates indicated by the α-β axis (a modification of the equation (3)) is used. Expression (12) can be adopted for the high-frequency currents i d ^ h and i q ^ h to be injected into the motor. Here, I h is the amplitude of the high-frequency current. The high-frequency currents i d ^ h and i q ^ h have a constant angular frequency ω h on the rotation coordinate d ^ -q ^ axis, and the phase angle θ h of the high-frequency current is ω h t (t represents time. )be equivalent to.

The equation (12) is coordinate-transformed on the α-β axis and substituted into the equation (9), and this is transformed onto the rotational coordinate d ^ -q ^ axis, thereby appearing on the d ^ axis and q ^ axis. The high-frequency voltages v d ^ h and v q ^ h are expressed by Expression (13). The first estimated value θ ^ ef1 is adopted as the magnetic pole position θ ^ e .

Since the change in (θ e −θ ^ e ) is considered to be small compared to the change in θ h , the high-frequency voltages v d ^ h , v q ^ h (Equation (13)) are converted into a bandpass with an angular frequency ω h . By passing the filter (BPF), the fourth term of Expression (13) is removed. Then, the equation (14) can be obtained by performing coordinate transformation on the coordinates (γ-δ axis) rotating at the angular frequency ω h . Since the change in 2θ e −2θ ^ e is also considered to be smaller than the changes in θ h and θ e , the high-frequency voltages v γh and v δh represented by the equation (14) are applied to the low-pass filter (LPF). By passing, the third term of the equation (14) can be obtained. Such a technique is introduced in Non-Patent Document 3, for example.

Two methods are conceivable for obtaining the error estimated value Δθ ^ e from the third term of the equation (14). FIG. 7 is a block diagram illustrating the configuration of the error estimation means 6 when these methods are employed.

In the configuration illustrated in FIG. 7A, the error estimation unit 6 includes coordinate conversion units 601 and 603, filters 602 and 604, and a divider 605. The coordinate conversion unit 601 converts the observed voltages v α and v β into rotational coordinates d ^ -q ^ axes. Filter 602 functions as a band-pass filter corner frequency omega h, giving the first to third terms of equation (13) to the coordinate transformation unit 603. The coordinate conversion unit 603 performs coordinate conversion based on each frequency ω h and gives a high frequency voltage v δh to the filter 604. The filter 604 functions as a low-pass filter, and outputs only the third term of the high-frequency voltage v δh in Expression (14) as the filtered voltage v δh0 .

Since the filtered voltage v δh0 has a coefficient L 1 I hh −2ω + ω ^), in order to remove this, the divider 605 converts the filtered voltage v δh0 to the divisor Q = (L ^ d − L ^ q) performs a dividing operation in the I h (ω h -ω ^) , and output with a the quotient as an estimated value Δθ ^ e of error. This relationship is expressed in Equation (15). Here, an approximation is introduced that each frequency ω is almost equal to its estimated value ω ^ and the estimated error value Δθ ^ e is small.

The coefficient m in the equation (15) is the values of L ^ d , L ^ q , I h , ω h , ω ^ that are device constants included in the filtered voltage v δh 0 , and those input to the divider 605. In consideration of an error occurring between the device constant value and the estimated value Δθ ^ e of the error. That is, the coefficient m is 1 when there is no error, and the coefficient m is a value other than 1 in other cases.

In the repetition of the first to third steps to be described later, m · Δθ ^ e acts so as to decrease. Therefore, even when the coefficient m is a value other than 1, the operation performed by the divider 605 is the coefficient m. It is effective as in the case where is 1. Therefore, the calculation result of the divider 605 is a device constant L ^ d, L ^ q, I h, ω h, the value of omega ^ does not affect.

The function of the divider 605 can be incorporated into the function of the integrator 5. In this case, the error estimation means 6 outputs the filtered voltage vδh0, and the gain k of the integrator 5 is set as shown in the equation (16). Here, K represents a gain, and K <0.

In the configuration illustrated in FIG. 7B, the error estimation unit 6 includes coordinate conversion units 601 and 603, filters 602 and 604, and an arctangent unit 606. Coordinate conversion is performed by the coordinate conversion unit 603 in the same manner as the configuration shown in FIG. 7A, but not only the high-frequency voltage v δh but also the high-frequency voltage v γh is given to the low-pass filter 604. The low-pass filter 604 outputs the filtered voltages v γh0 and v δh0 of the high-frequency voltages v γh and v δh , respectively.

The arc tangent unit 606 performs an operation of dividing the arc tangent value of the post-filtering voltage ratio (v δh0 / v γh0 ) by 2. In view of the equation (17), this result is an error estimation value Δθ ^ e . This is a device constant from the outside to the arctangent unit 606 in a manner L ^ d, L ^ q, I h, ω h, ω ^ it is not necessary to enter, thus the device constant in the calculation of the inverse tangent unit 606 It is not affected by errors.

As a third step, the estimated value L ^ q of the q-axis inductance of the motor is updated. This updating method is the same as the method for updating the estimated value L ^ q of the q-axis inductance according to the fourth step described in the first embodiment.

Since the estimated value Δθ ^ e of the error is caused by the difference between the estimated value L ^ q of the q-axis inductance and the true value L ~ q , the updated estimated value L ^ q of the q-axis inductance is the true value L ~ approaches q . Then, by repeating the first to third steps using the updated q-axis inductance estimated value L ^ q , the first estimated value θ ^ ef1 can be brought close to the true value θ ~ e .

The first to third steps are repeated until, for example, the difference between the estimated value L ^ q of the q-axis inductance before and after the update and the absolute value thereof are obtained until the absolute value becomes smaller than a certain value, for example, an allowable value as an error. Can be used.

Above high frequency current minute, that is, when I h is small, the filtering adopted in the first step (filter 3) is not necessary. A block diagram corresponding to this is shown in FIG. The estimated value θ ^ e1 of the magnetic pole position is adopted as the magnetic pole position θ ^ e .

Thus, according to the present embodiment, the first estimated value θ ^ ef1 (or θ ^ e ) of the magnetic pole position and the estimated value Δθ ^ e of the error of the magnetic pole position are obtained. Then, by updating the estimated value L ^ q of the inductance using the estimated value Δθ ^ e of the error, the estimated value of the error can be reduced. Therefore, by calculating the magnetic pole position for each sampling of voltage and current, it is possible to obtain an accurate magnetic pole position with good responsiveness.

In any of the above-described embodiments, once the estimated inductance value L ^ q is updated during the first operation to obtain an estimated inductance value that has only an acceptable error, Based on the estimated value of the inductance, an accurate magnetic pole position can be obtained only by the first estimating means 1A.

The first estimating means 1A uses the first current i α , i β ; i d ^ , i q ^ or the motor voltage v α , v β ; v d ^ , v q ^ The estimated value θ ^ ef1 may be obtained.

In any of the above-described embodiments, the first estimated value θ ^ ef1 obtained by the magnetic pole position estimation technique described therein is adopted as the magnetic pole position, and based on this, the inverter that drives the motor is controlled. Also good. At this time, you may control an inverter, for example using a control apparatus.

  According to such an inverter control technique, a voltage to be applied to the motor is generated based on the magnetic pole position obtained accurately. Therefore, the motor can be accurately controlled by the inverter.

It is a block diagram which shows the magnetic pole position estimation method demonstrated by 1st Embodiment. It is a figure which shows the magnetic pole position and coordinate system which are demonstrated by 1st Embodiment. It is a figure which shows the relationship between time and a magnetic pole position demonstrated by 1st Embodiment. It is a figure which shows the update of the inductance demonstrated by 1st Embodiment. It is a block diagram which shows the magnetic pole position estimation method demonstrated by 1st Embodiment. It is a block diagram which shows the magnetic pole position estimation method demonstrated by 2nd Embodiment. It is a block diagram which shows the error estimation means demonstrated by 2nd Embodiment. It is a block diagram which shows the magnetic pole position estimation method demonstrated by 2nd Embodiment.

Explanation of symbols

i α , i β ; i d ^ , i q ^ current v α , v β ; v d ^ , v q ^ voltage L ^ q inductance estimated value θ ^ ef1 first estimated value of magnetic pole position θ ^ e2 magnetic pole Second estimate of position
Δθ ^ e Estimated error
i d ^ h , i q ^ h high-frequency current θ h phase angle of high-frequency current ω h each frequency of high-frequency current 1A first estimation means 1B, 21B second estimation means 3 filter 5 integration section (inductance calculation section)
6 Error estimation means 12 Adder / Subtracter (Error estimation means)
601 and 603 coordinate conversion unit 602 filter (band pass filter)
604 filter (low-pass filter)
605 Divider 606 Inverse tangent

Claims (14)

  1. A method for estimating the magnetic pole position of a motor,
    Current (i α , i β ; i d ^ , i q ^ ) and / or motor voltage (v α , v β ; v d ^ , v q ^ ) flowing through the motor and inductance (L ^ (1A) performing a first estimation method for obtaining a first estimated value (θ ^ ef1 ) of the magnetic pole position based on q );
    Performing a second estimation method (1B; 21B) for obtaining a second estimated value (θ ^ e2 ) of the magnetic pole position without depending on the inductance;
    Obtaining an error estimate (Δθ ^ e ) as a difference between the first estimate and the second estimate;
    A magnetic pole position estimation method comprising: updating the inductance based on the estimated value of the error, and performing the first estimation method using the updated inductance.
  2. The magnetic pole position estimation method according to claim 1, wherein, in the first estimation method (1A), a pulsation component of the first estimation value (θ ^ ef1 ) with respect to rotation of the motor is removed.
  3. In the second estimation method (21B), the second estimated value (θ ^ e2 ) is obtained based on the voltages (v α , v β ) when no current is passed through the motor. The magnetic pole position estimation method according to claim 2.
  4. 3. The magnetic pole position estimation method according to claim 1, wherein in the second estimation method, a predetermined magnetic pole position for each rotation of the motor is obtained as the second estimated value (θ ^ e2 ).
  5. A method for estimating the magnetic pole position of a motor,
    Current (i α , i β ; i d ^ , i q ^ ) and / or motor voltage (v α , v β ; v d ^ , v q ^ ) flowing through the motor and inductance (L ^ (1A) performing a first estimation method for obtaining a first estimated value (θ ^ ef1 ) of the magnetic pole position based on q );
    An estimated value (Δθ ^ e ) of the magnetic pole position error is obtained based on the harmonic component of the voltage in a state where the harmonic current ( id ^ h , i q ^ h ) is supplied to the motor. Performing a second estimation method (6);
    A magnetic pole position estimation method comprising: updating the inductance based on the estimated value of the error, and performing the first estimation method using the updated inductance.
  6. In the second estimation method (6),
    Based on the first estimated value (θ ^ ef1 ) and the phase angle (θ h ) of the harmonic current ( id ^ h , i q ^ h ), the voltage (v α , v β ) of the motor is determined. A first coordinate conversion step (601) for performing rotational coordinate conversion;
    A first filtering step (602) for filtering the result of the first coordinate transformation step in the vicinity of the angular frequency (ω h ) of the harmonic current;
    A second coordinate transformation step (603) for performing a rotational coordinate transformation on the result of the first filtering step based on the angular frequency of the harmonic current;
    The second coordinate conversion step results (v δh; v δh, v γh) low-frequency component of (v δh0; v δh0, v γh0) a second for filtering by the angular frequency near the harmonic current A filtering step (604);
    The magnetic pole position estimation method according to claim 5, wherein a step (605; 606) of obtaining the estimated value (Δθ ^ e ) of the error based on the result of the second filtering step is performed.
  7. The first estimated value (θ ^ ef1 ) obtained by the magnetic pole position estimation method according to any one of claims 1 to 6 is adopted as the magnetic pole position, and the motor is driven based on the first estimated value (θ ^ ef1 ). An inverter control method for controlling an inverter.
  8. An apparatus for estimating the magnetic pole position of a motor,
    Current (i α , i β ; i d ^ , i q ^ ) and / or motor voltage (v α , v β ; v d ^ , v q ^ ) flowing through the motor and inductance (L ^ q )) first estimation means (1A) for obtaining a first estimated value (θ ^ ef1 ) of the magnetic pole position;
    Second estimating means (1B; 21B) for obtaining a second estimated value (θ ^ e2 ) of the magnetic pole position without depending on the inductance;
    Error estimation means (12) for obtaining an error estimated value (Δθ ^ e ) as a difference between the first estimated value and the second estimated value;
    A magnetic pole position estimation apparatus comprising: an inductance calculation unit (5) that updates the inductance based on the estimated value of the error.
  9. The magnetic pole position estimation apparatus according to claim 8, wherein the first estimation means (1A) includes a filter (3) for removing a pulsation component with respect to rotation of the motor from the first estimated value (θ ^ ef1 ).
  10. The second estimation means (21B) obtains the second estimated value (θ ^ e2 ) based on the voltages (v α , v β ) at a time when no current is passed through the motor. The magnetic pole position estimation apparatus according to claim 9.
  11. The magnetic pole position estimation apparatus according to claim 8 or 9, wherein the second estimation means obtains a predetermined magnetic pole position for each rotation of the motor as the second estimated value (θ ^ e2 ).
  12. An apparatus for estimating the magnetic pole position of a motor,
    Current (i α , i β ; i d ^ , i q ^ ) and / or motor voltage (v α , v β ; v d ^ , v q ^ ) flowing through the motor and inductance (L ^ q )) first estimation means (1A) for obtaining a first estimated value (θ ^ ef1 ) of the magnetic pole position;
    An estimated value (Δθ ^ e ) of the magnetic pole position error is obtained based on the harmonic component of the voltage in a state where the harmonic current (i d ^ h , i q ^ h ) is supplied to the motor. Second estimation means (6);
    A magnetic pole position estimation apparatus comprising: an inductance calculation unit (5) that updates the inductance based on the estimated value of the error.
  13. The second estimation method (6) is:
    Based on the first estimated value (θ ^ ef1 ) and the phase angle (θ h ) of the harmonic current ( id ^ h , i q ^ h ), the voltage (v α , v β ) of the motor is determined. A first coordinate conversion unit (601) that performs rotational coordinate conversion;
    A bandpass filter (602) for filtering the output of the first coordinate transformation unit in the vicinity of the angular frequency (ω h ) of the harmonic current;
    A second coordinate transformation unit (603) that performs rotational coordinate transformation on the output of the bandpass filter based on the angular frequency of the harmonic current;
    The output of the second coordinate conversion unit (v δh; v δh, v γh) low-frequency component of; low-pass filter a (v δh0 v δh0, v γh0 ), is filtered by the angular frequency near the harmonic current ( 604),
    The magnetic pole position estimation apparatus of Claim 12 provided with the calculating part (605; 606) which calculates | requires the estimated value ((DELTA) (theta) ^ e ) of the said error based on the output of the said low-pass filter.
  14. The first estimated value (θ ^ ef1 ) obtained by the magnetic pole position estimating device according to any one of claims 8 to 13 is adopted as the magnetic pole position, and the motor is driven based on the first estimated value (θ ^ ef1 ). An inverter control device that controls the inverter.
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