JP4542844B2 - 2 transformer type DC-DC converter - Google Patents

2 transformer type DC-DC converter Download PDF

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JP4542844B2
JP4542844B2 JP2004207597A JP2004207597A JP4542844B2 JP 4542844 B2 JP4542844 B2 JP 4542844B2 JP 2004207597 A JP2004207597 A JP 2004207597A JP 2004207597 A JP2004207597 A JP 2004207597A JP 4542844 B2 JP4542844 B2 JP 4542844B2
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voltage
converter
current
circuit
main switch
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JP2005051994A (en
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剛 山下
宏治 川崎
恵二 重岡
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株式会社デンソー
株式会社日本自動車部品総合研究所
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Description

The present invention relates to an improvement of 2-trans-DC-DC converter.

  Transformer DC-DC converters using a transformer are widely used because the input and output can be completely electrically isolated. In this transformer type DC-DC converter, a two transformer type DC-DC converter having two transformers is known.

  The following Patent Document 1 proposes a two-transformer DC-DC converter that essentially connects a normal one-transform DC-DC converter in parallel and performs a complementary operation. In this two-transformer DC-DC converter, since the two transformers alternately output current, it is possible to reduce the ripple with only a small-capacity smoothing capacitor without using the output side choke coil.

  Patent Document 2 below proposes a two-transform DC-DC converter of the type shown in FIG. In this two-transformer DC-DC converter, the primary coils W2, W5 of the transformers T1, T2 are fed from the input DC power supply 2 through the switching element Q1. A clamp circuit in which a capacitor C2 and a switching element Q2 are connected in series is connected in parallel with the primary coils W2 and W5. D is a parasitic (built-in) diode of the switching elements Q1 and Q2 which are MOS transistors. The voltages of the secondary coils W3 and W6 of the transformers T1 and T2 are alternately rectified and output by the normal synchronous rectifier circuit 100. The switching elements Q1 to Q4 are PWM controlled to make the output voltage Vo constant. The switching elements Q1 and Q2 are alternately (complementarily) intermittently connected. The operation of the DC-DC converter of FIG. 9 will be briefly described below.

(Mode 1)
When the switching element Q1 is turned on, an input DC voltage is applied to the primary coils W2 and W5. It is assumed that the switching element Q2 is turned off. The current i1 flows from the input terminal to the primary coils W2 and W5, and the capacitor C1 is discharged. The currents flowing through the primary coils W2 and W5 increase with time due to the inductances of the primary coils W2 and W5, and the secondary coils W3 and W6 generate a voltage with a positive terminal on the dot side. A current i3 is output from the next coil W6, and magnetic energy is accumulated in the core of the transformer T1.

(Mode 2)
Next, when the switching element Q1 is turned off, the potential at the connection point 40 rapidly rises due to the energy stored in the transformer T1, and the capacitor C2 is charged through the parasitic diode D of the switching element Q2 in order to extinguish the energy stored in the transformer T1.

(Mode 3)
Next, when the switching element Q2 is turned on, the capacitor C2 is charged better through the switching element Q2 by the energy stored in the transformer T1. After completion of this operation, the capacitor C2 causes the capacitor C2 to pass a current in the discharge direction through the switching element Q2 to the primary coils W5 and W2, and the capacitor C2 is discharged. This current increases with time, a positive voltage is generated at the non-dot-side terminals of the primary coils W2 and W5, the current is output from the primary coil W3 when the switching element Q4 is turned on, and the transformer T2 has magnetic energy. Accumulated.

(Mode 4)
Next, when the switching element Q2 is turned off, the potential at the connection point 40 is suddenly lowered by the stored energy of the transformer T2, and the primary coil W5 is connected to the input terminal, the parasitic diode D of the switching element Q1, in order to extinguish the stored energy of the transformer T2. A current is passed through the capacitor C1, and the capacitor C1 is charged.

(Mode 1)
When the switching element Q1 is turned on, the capacitor C1 is further charged by the energy stored in the transformer T2. After the end of this operation, the operation cycle ends and returns to the beginning.
JP 2003-102175 A USP 5291382

  However, the two-transform DC-DC converter disclosed in the above publication has a problem that a large number of switching elements are required and current ripple is large.

  Further, the two-transform DC-DC converter shown in FIG. 9 has a problem that a ripple component of the input current is large because a reverse current flows from the transformer to the input DC power supply 2 side.

  For this reason, in order to reduce the current change of the input DC power supply 2, a large-capacitance capacitor has to be employed as the input-side smoothing capacitor C1 connected in parallel with the input DC power supply 2. However, a high-withstand-voltage and large-capacitance capacitor is large and expensive, leading to an increase in size and cost of the device. In addition, shielding of electromagnetic wave noise radiated from a line connecting the input DC power supply 2 and the DC-DC converter has become an important problem. Furthermore, the fact that the ripple component of the current is large has the disadvantage that the effective value of the input current becomes large, so that loss and heat generation increase. Furthermore, since the transformer is demagnetized by a direct current component, the size of the transformer is increased.

The present invention has been made in view of the above problems, and can be reduced in size and weight and the number of switching elements can be reduced. In addition, the input side smoothing capacitor can be reduced in capacity with excellent efficiency and a small input current ripple component. it you are its object to provide a 2-trans-DC-DC converter capable.

  In the description of the claims and the disclosure of the present invention, the reference numerals given to the constituent elements of the invention are merely for facilitating understanding, and are not limited to the constituent elements of the embodiment having the corresponding reference numerals. Of course.

2 trans DC-DC converter of the present invention includes a transformer T1 composed of a primary coil W1, W2 and secondary coil W3 Prefecture, two transformers and transformer T2 consisting of primary coil W4, W5 and secondary coil W6 Prefecture The coils W1 and W4 are connected in series to form a first coil pair, and the coils W2 and W5 are connected in series to form a second coil pair, and a first DC power supply voltage is used. The AC / DC converting circuit 11 connected to the one voltage system 1000 so as to be able to transmit / receive power, and the AC / DC converting circuit 21 connected so as to be able to transmit / receive power to the second voltage system 2000 operated with a DC power supply voltage different from that of the first voltage system 1000. And a controller for controlling power transmission between the first voltage system 1000 and the second voltage system 2000, and one end of the first coil pair is connected to the first voltage system. Is connected to one end of the unified 1000, the AC-DC converter circuit 11 includes a main switch Q1 which connects the other ends of said first coil pair of the first voltage line 1000,
A capacitor C1 that connects the other end of the first voltage system 1000 and one end of the second coil pair, one end is one end of the second coil pair, and the other end is the other end of the first coil pair and a second one. A clamp circuit which is connected to the other end of the coil pair and bypasses the current flowing through the main switch Q1 when the main switch Q1 is off, and the controller turns off the main switch Q1 to turn off the capacitor C1. A charging mode for charging and a discharging mode in which the main switch Q1 is turned on to discharge the capacitor C1 are repeatedly performed in a predetermined cycle.

  That is, the two-transform DC-DC converter of the present invention is the same as the conventional two-transform DC-DC converter shown in FIG. 9, except that a direct-current coil body (primary coils W1, W4) is added. Employs a circuit configuration that feeds power to a connection point between the main switch Q1 and the AC coil bodies (primary coils W2, W5) through the DC coil bodies (primary coils W1, W4). In addition to the PWM control, various known pulse control methods can be adopted for the control of each switch.

  As the clamp circuit, various known circuits that absorb the surge voltage when the main switch Q1 is off can be used. As the main switch Q1, a MOS transistor capable of bidirectional energization is suitable. The AC / DC converter circuit referred to in the present specification converts input AC power (which may include a DC power component) into DC power, or input DC power (which includes an AC power component). Is also a circuit that converts AC power into AC power.

  As the AC / DC converter circuit 21, a normal synchronous rectifier circuit or a diode rectifier circuit can be used, and power transmission from the second voltage system 2000 to the first voltage system 1000 can be performed by operating the synchronous rectifier circuit as an inverter.

  In this way, as will be described below, a two-transformer DC− capable of reducing the size and weight of the smoothing capacitor, reducing the number of switching elements, improving the efficiency, and reducing the ripple component of the input current and output current, as will be described below. A DC converter can be realized.

  In the first aspect, the clamp circuit includes a capacitor C2 and a sub switch Q2 connected in series, and the controller turns off the main switch Q1, turns on the sub switch Q2, and charges the capacitor C1. And a discharge mode in which the main switch Q1 is turned on and the sub switch Q2 is turned off to discharge the capacitor C1. In this way, the clamp power can be effectively reused.

  In the aspect 2, the first coil pair is interrupted from the first voltage system 1000 to the AC / DC conversion circuit 11 side when the first voltage system 1000 is the power transmission side and the second voltage system 2000 is the power reception side. Without passing current.

  In this way, current backflow from the DC-DC converter to the first voltage system 1000 that is the input DC power supply, which was a drawback of the conventional two-transform DC-DC converter, can be significantly reduced. The ripple component of the input current can be satisfactorily reduced without using the smoothing capacitor. Therefore, in the two-transform DC-DC converter of the present invention, it is not necessary or necessary to provide a current smoothing circuit for smoothing the input current of the AC / DC converter circuit 11 between the first voltage system 1000 and the AC / DC converter circuit 11. Even if this is the case, it can be greatly reduced in size.

  In aspect 3, the AC / DC converter circuit 21 has one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W6. A switching element Q4 having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W3, and the controller includes the main switch Q1 One of the switching elements Q3 and Q4 is synchronously turned on, and the sub switch Q2 and the other of the switching elements Q3 and Q4 are synchronously turned on.

  In this way, the AC / DC converter circuit 21 can be configured by a so-called synchronous rectifier circuit, so that loss can be reduced. Note that the term “synchronization on” as used herein includes not only the case where both are turned on simultaneously but also the case where the other is turned on after a predetermined time has elapsed after one of them is turned on. Also, power transmission from the second voltage system 2000 to the first voltage system 1000 is possible. Note that one of the switching elements Q3 and Q4 may be constituted by a diode as a two-terminal switching element instead of a transistor as a three-terminal switching element.

  In the aspect 4, the controller restricts the change of the on-duty ratio of the output switches Q1 and Q2 within a predetermined range, thereby setting the ripple component of the output current of the synchronous rectifier circuit below a predetermined value level.

  In this way, the ripple component of the output current can be reduced without using a choke coil.

  In the aspect 5, the first voltage system 1000 is the high voltage side, and the second voltage system 2000 is the low voltage side. Thereby, step-down transmission from the high-voltage first voltage system 1000 to the low-voltage second voltage system 2000 or step-up transmission from the low-voltage second voltage system 2000 to the high-voltage first voltage system 10 is performed. be able to.

  In the aspect 6, the first voltage system 1000 is set to a lower voltage than the second voltage system 2000. Thereby, step-up power transmission from the low-voltage first voltage system 1000 to the high-voltage second voltage system 2000 or step-down power transmission from the high-voltage second voltage system 2000 to the low-voltage first voltage system 10 is performed. be able to.

  In the aspect 7, the AC / DC converter circuit 11 is connected to a DC power source or an electric load forming the first voltage system 1000 without a current smoothing choke coil element. Thereby, a physique and a weight can be reduced, maintaining the ripple component of input current in an allowable range.

  In the aspect 8, the AC / DC converter circuit 21 is connected to a DC power source or an electric load forming the second voltage system 2000 without a current smoothing choke coil element. Thereby, a physique and a weight can be reduced, maintaining the ripple component of output current in a tolerance.

  In the aspect 9, the controller controls the first voltage system 1000 from the first voltage system 1000 to the second voltage system 2000 that is operated with different DC power supply voltages by changing the on-duty ratio of the main switch Q1 that performs PWM control. The mode is switched between the power transmission mode and the second power transmission mode in which power is transmitted from the second voltage system 2000 to the first voltage system 1000.

  As a result, power is transmitted bidirectionally using a point that does not require a choke coil. As a result, the convenience of the two-transform DC-DC converter can be improved without complicating the configuration.

  In particular, in this embodiment, the AC / DC conversion circuit 11 is connected to a DC power source or an electric load forming the first voltage system 1000 without a current smoothing choke coil element, and the AC / DC conversion circuit 21 is connected to the second voltage system 2000. It is suitable for the case where it is connected to a DC power source or an electric load that constitutes the above without passing through a choke coil element for current smoothing.

  That is, in the two-transform DC-DC converter of the present invention, the ripple component contained in the synchronously rectified output current is very small. This eliminates the need for a ripple removing choke coil. For this reason, when power is transmitted in the reverse direction, a switching surge voltage is not generated in the choke coil when power is transmitted back from the AC / DC converter circuit 21 side, which is a synchronous rectifier circuit, to the AC / DC converter circuit 11 side. A circuit device for attenuation or prevention can be omitted.

  Note that the change of the power transmission direction can be performed only by changing the duty ratio of the switching element. On the other hand, when bidirectional power transmission is performed by a conventional transformer type DC-DC converter, when reverse power transmission is performed from the original rectification side to the original inverter side, the output ripple of the conventional DC-DC converter is large. Therefore, it is difficult to omit the choke coil for removing the ripple from the current output from the rectifier circuit. For this reason, when the original rectifying circuit is operated as an inverter during reverse power transmission, a switching surge voltage is generated in the choke coil. Therefore, a circuit for attenuating or suppressing the switching surge voltage must be added. It is difficult to avoid complication of the system, and it is difficult to realize bidirectional power transmission of the conventional transformer type DC-DC converter.

  In aspect 10, in the first power transmission mode, the controller increases the on-duty ratio of the main switch Q1 when the voltage of the second voltage system 2000 is lower than a predetermined target voltage, and decreases when the voltage is higher. In the second power transmission mode, the on-duty ratio of the main switch Q1 decreases when the voltage of the first voltage system 1000 is lower than a predetermined target voltage, and increases when it is higher. In this way, output voltage control in bidirectional power transmission can be easily performed.

  In the aspect 11, the first voltage system 1000 includes a smoothing capacitor that is connected in parallel to the first voltage system 1000 and exchanges AC power with the AC / DC converter circuit 11, and the first voltage system 1000 includes a DC power source and a power switch connected in series with each other. The controller performs power transmission from the second voltage system 2000 to the first voltage system 1000 to precharge the smoothing capacitor, and then closes the power switch.

  In this way, even when power is transmitted from the second voltage system 2000 to the second voltage system 2000 through the two-transform DC-DC converter from the state in which the power switch of the first voltage system 1000 is open, the first voltage from the second voltage system 2000 in advance. Since the power switch of the first voltage system 1000 can be turned on after the smoothing capacitor of the system 1000 is precharged, the inrush current is prevented from flowing from the DC power source of the first voltage system 1000 to the smoothing capacitor when the power switch is turned on. It is possible to protect the smoothing capacitor and the power supply system.

  In the aspect 12, the controller gradually increases the voltage applied to the smoothing capacitor in the precharging of the smoothing capacitor. Thereby, further protection of the smoothing capacitor can be realized.

  In aspect 13, it has a current sensor which detects the current of the main switch Q1, and the controller determines a decrease in the load current in the synchronous rectifier circuit based on the magnitude of the detected current of the current sensor at a predetermined timing. As a result, the synchronous rectification operation of the synchronous rectification circuit is stopped.

  In the synchronous rectification circuit, when the load current decreases and becomes no load or close to it, the voltage induced in the secondary coils W3 and W6 and the battery voltage of the second voltage system 2000 become substantially equal. However, as the switching elements Q3 and Q4 constituting the synchronous rectifier circuit, MOS transistors that can be energized in both directions are usually employed, and thus deviates from the original purpose of power transmission from the first voltage system 1000 to the second voltage system 2000. The situation of reverse power transmission from the second voltage system 2000 to the first voltage system 1000 is likely to occur. To prevent this, synchronous rectification is interrupted under conditions where the secondary current is very small. This interruption of synchronous rectification is also preferable from the viewpoint of reducing the drive loss of the element.

  For this purpose, it is conceivable to detect whether the synchronous rectification is interrupted by detecting the output current of the synchronous rectification circuit. However, in the step-down DC-DC converter, the secondary side current, which is larger than the primary side, is considered. The resistance value of the resistance element used for detection needs to be extremely small, and there is a problem that it is difficult to manufacture a current detection resistance element with high accuracy, and power loss and heat generation of the current detection resistance also become problems.

  Of course, the interruption of the synchronous rectification when the load current is still large causes an increase in rectification loss. Therefore, the interruption of the synchronous rectification is preferably performed when the load current becomes sufficiently small. For this purpose, it is conceivable to interrupt the synchronous rectification by detecting the current reverse flow of the switching elements Q3 and Q4 of the synchronous rectification circuit. However, there are some loads as the second voltage system 2000 where the current reverse flow is not desirable. Therefore, it is desired to reliably determine whether or not to perform synchronous rectification without causing a backflow in the second voltage system 2000. Further, as described above, in the step-down DC-DC converter, power loss and heat generation become a problem in current detection on the transformer secondary side with a large current.

  This mode is made from the above viewpoint. Based on the current value of the primary side of the transformer, that is, the inverter side, in particular, the main switch Q1, it is estimated that the secondary side current is substantially zero, and the appropriateness of the synchronous rectification is determined. . In detection by suitable CT (current transformer), the detection signal can be made independent of the potential of the primary circuit of the transformer, that is, the AC / DC converter circuit 11, so that signal processing can be performed independently of the potential of the first voltage system 1000. it can. Further, the step-down DC-DC converter can reduce the power loss of CT as a current sensor or a current detection resistor. In addition, it is not necessary to manufacture a current detection resistor having a low resistance value with high accuracy, and the manufacture is facilitated. The restart after the interruption of the synchronous rectification on the transformer secondary side may be performed based on the change in the current of the main switch Q1 to increase the load current flowing through the diodes of the switching elements Q3 and Q4.

  In the fourteenth aspect, the controller determines whether or not the synchronous rectification operation should be stopped based on the magnitude of the current of the main switch Q1 after a predetermined time has elapsed since the main switch Q1 was commanded to be turned on.

  In the two-transform DC-DC converter of the present invention, the current flowing through the main switch Q1 during the ON period of the main switch Q1 is the sum of a load current component that is a DC component and an excitation current component that is an AC component. The excitation current component here refers to a current that flows through the primary coil of the transformer when there is no load. As the load current increases, the timing at which the current of the main switch Q1 crosses the current value line indicating a predetermined threshold current close to or close to the current 0 changes. Therefore, by determining the magnitude of the current of the main switch Q1 after a predetermined time after the main switch Q1 is turned on, in other words, by comparing the current of the main switch Q1 with the current value line after the predetermined time. It can be determined whether or not the current of the switching elements Q3 and Q4, that is, the load current of the synchronous rectifier circuit is substantially zero.

  Note that, in this determination method, which is earlier, the timing at which the current value of the main switch Q1 crosses the current value line is earlier than a predetermined timing (the load current is set to approximately 0). Is synonymous with judging. The timing here is preferably set easily with reference to the time from the on-command of the main switch Q1.

  In the aspect 15, the predetermined time is set longer than the energization time of the transient resonance current that flows in the reverse direction to the main switch Q1 immediately after the main switch Q1 is turned on.

  In this way, it is possible to prevent the current of the main switch Q1 from oscillating due to the LC resonance in the current when the main switch Q1 is intermittent, thereby preventing the erroneous determination of the synchronous rectification stop. That is, even if the main switch Q1 of the AC / DC conversion circuit 11 as the inverter of the resonance type DC-DC converter is turned on and a transient LC resonance current flows in the reverse direction to the main switch Q1, this is reversed in the synchronous rectifier circuit. There is no problem of erroneously stopping the synchronous rectifier circuit when confused with the generation of current.

  In the aspect 16, the controller is configured to transmit the power from the first voltage system 1000 to the second voltage system 2000 as the DC power supply voltage applied from the first voltage system 1000 to the AC / DC converter circuit 11 increases. The maximum value of the on-duty ratio of the main switch Q1 is decreased.

  In this way, when power is transmitted from the first voltage system 1000 to the second voltage system 2000, when the input DC voltage from the first voltage system 1000 to the AC / DC converter circuit 11 as an inverter increases rapidly due to surge voltage superposition or the like. A simple circuit reliably prevents a sudden increase in the voltage applied to the main switch Q1 and ensures a wide control range of the on-duty ratio when the input DC voltage is low, so output voltage control and ripple reduction control High adaptability to can be taken.

  In the aspect 17, when power is transmitted from the second voltage system 2000 to the first voltage system 1000, the on-duty of the main switch Q1 increases as the DC power supply voltage applied from the second voltage system 2000 to the AC / DC converter circuit 21 increases. Increase the maximum value of the ratio.

  In this case, when power is transmitted from the second voltage system 2000 to the first voltage system 1000, the input DC voltage from the second voltage system 2000 to the AC / DC converter circuit 21 as an inverter increases rapidly due to superposition of a surge voltage or the like. A simple circuit reliably prevents a sudden increase in the voltage applied to the main switch Q1 and ensures a wide control range of the on-duty ratio when the input DC voltage is low, so output voltage control and ripple reduction High adaptability to control can be achieved.

In the aspect 18 , the controller closes the power switch after the precharge to the smoothing capacitor is completed, and performs power transmission from the first voltage system 1000 to the second voltage system 2000. In this way, even when power is transmitted from the second voltage system 2000 to the second voltage system 2000 through the two-transform DC-DC converter from the state in which the power switch of the first voltage system 1000 is open, the first voltage from the second voltage system 2000 in advance. Since the power switch of the first voltage system 1000 can be turned on after the smoothing capacitor of the system 1000 is precharged, the inrush current is prevented from flowing from the DC power source of the first voltage system 1000 to the smoothing capacitor when the power switch is turned on. Thus, the smoothing capacitor and the DC power supply of the first voltage system 1000 can be protected.

In embodiments 19, wherein the controller is characterized in that for regulating the maximum value Dmax of the on-duty ratio D of the main switch Q1 below a predetermined limit value decreases as before Kijika flow supply voltage increases.

In this way, even when the input DC voltage from the first voltage system 1000 to the AC / DC converter circuit 11 as the inverter suddenly increases due to superposition of the surge voltage or the like, the applied voltage to the main switch Q1 is rapidly increased by a simple circuit. It can be surely prevented. Furthermore, when the input DC voltage is low, a wide control range of the on-duty ratio can be ensured, so that high adaptability to output voltage control and ripple reduction control can be achieved.
In the aspect 20, the AC / DC converter circuit 21 includes a first rectifier diode having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W6. And a second rectifier diode having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W3. In this way, the AC / DC converter circuit 21 can be configured by a so-called diode rectifier circuit, so that loss can be reduced.

  The preferred embodiment of the DC-DC converter of the present invention will be specifically described with reference to the following examples. The present invention is not limited to the following embodiments, and it is obvious that each component can be replaced with one or more known components having the same main function.

(Overall circuit configuration)
The DC-DC converter of Example 1 will be described with reference to the circuit diagram shown in FIG. The DC-DC converter 1 is a unidirectional step-down converter, and is disposed between a high-voltage input DC power supply (high-voltage battery) 2 and a low-voltage load 3. The DC-DC converter 1 may be a unidirectional boost converter, and the load 3 may be a direct current power source.

  The DC-DC converter 1 includes transformers T1, T2, switching elements Q1, Q2, Q3, Q4, capacitors C1, C2, C3, and a controller 4.

  The controller 4 controls on / off of the switching elements Q1 to Q4. In this embodiment, the controller 4 reads the output voltage of the DC-DC converter 1 in order to feedback control the output voltage of the DC-DC converter 1 to a set value. The on-duty ratio of the switching element Q1 is PWM-controlled based on the deviation between the output voltage and the set value. The carrier frequency in PWM control is set to several tens to several hundreds kHz as in the normal case, but is preferably set as high as possible as long as problems due to increased loss and electromagnetic noise are allowed.

  Each of the switching elements Q1 to Q4 is a MOS transistor as shown in FIG. 1, but may be replaced with a well-known configuration in which a junction diode and another transistor such as an IGBT are connected in parallel.

  The transformer T1 has primary coils W1, W2 and a secondary coil W3, and the transformer T2 has primary coils W4, W5 and a secondary coil W6. The capacitor C3 is a well-known output-side smoothing capacitor connected in parallel with the load 3 to reduce ripples. Switching elements Q3 and Q4 constitute a rectifier circuit. Instead of the output-side smoothing circuit composed of the capacitor C3, an output-side smoothing circuit may be configured by the choke coil and the capacitor C3 in the same manner as a normal output-side smoothing circuit. Switching elements Q1 to Q4 are configured by N-channel MOS transistors.

  10 is a connection point between the positive end of the input DC power source 2 and the primary coil W1, 20 is a connection point between the negative end of the input DC power source 2 and the main switch Q1 and the capacitor C1, and 30 is a connection point between the capacitor C1 and the capacitor C2. A connection point 40 with the primary coil W2 is a connection point with the main switch Q1, the sub switch Q2, the primary coil W4, and the primary coil W5.

  Coils W1, W2, W4, and W5, capacitors C1 and C2, and switching elements Q1 and Q2 are also referred to as input side circuits, and coils W3 and W6, capacitor C3, and switching elements Q3 and Q4 are also referred to as output side circuits. .

(Input side circuit)
Hereinafter, the input side circuit will be described.

  In this embodiment, if the normally set dead time is ignored, the main switch Q1 and the sub switch Q2 operate alternately. Therefore, the circuit current flow other than the current passing through the parasitic diode D of the MOS transistor is the main switch Q1. It is easy to think based on the main switch circuit formed when the switch is turned on and the sub switch circuit formed when the sub switch Q2 is turned on. The main switch circuit includes a connection point 10, primary coils W1, W4, a main switch Q1, a connection point 20, a first circuit portion connecting the input DC power supply 2, a connection point 20, a capacitor C1, coils W2, W5, and a main switch. Q1 and a second circuit portion connecting the connection points 20. The sub switch circuit includes a connection point 10, a coil W1, W4, a connection point 40, a sub switch Q2, a capacitor C2, a connection point 30, a capacitor C1, a connection point 20, and a third circuit unit connecting the input DC power source 2. 40, the sub switch Q2, the capacitor C2, the coils W2, W5, and the fourth circuit portion connecting the connection point 40. Further, a capacitor C1 charging circuit is formed from the input DC power supply 2 to the input DC power supply 2 through the primary coils W1, W4, the primary coils W2, W5, and the capacitor C1.

(Output side circuit)
Hereinafter, the output side circuit will be described.

The output side circuit of this embodiment is the same synchronous rectifier circuit as the output side circuit of the conventional two transformer type DC-DC converter shown in FIG. Since the output switches Q3 and Q4 operate in principle, they are divided into a fifth circuit portion formed when the output switch Q3 is turned on and a sixth circuit portion formed when the output switch Q4 is turned on. Can do. The fifth circuit unit is a circuit connecting the connection point 60, the output switch Q3, the coil W6, the capacitor C3, and the connection point 60. The sixth circuit unit is the connection point 60, the output switch Q4, the coil W3, the capacitor C3, and the connection. A circuit connecting the points 60. Either or both of the output switches Q3 and Q4 may be replaced with a diode. Output switch Q3 has almost the same operating state as the main switch Q1, the output switch Q 4 are of the same operating state as a sub-switch Q2. The operation timing of the switching elements Q1 to Q4 is shown in FIG. 2, and the voltage change of the coils W1 to W6 is schematically shown in FIG. In the state transition of switching elements Q1 and Q2, a dead time is provided at the time of state transition of switching elements Q3 and Q4, but this is not essential.

  In this embodiment, when the number of turns of the coil W1 and the coil W2 is N1, the number of turns of the coil W3 is N2, the number of turns of the coils W4 and W5 is N3, and the number of turns of the coil W6 is N4, N1 / N2 = N3 / N4, but this turn ratio is not essential. Preferably, N1 = N3 and N2 = N4. Of course, the turn ratio can be changed.

(Overall description of operation)
Next, the operation principle of the DC-DC converter 1 will be described below.

  2 is a timing chart showing a schematic operation of the switching elements Q1 to Q4, FIG. 3 is a timing chart showing voltage waveforms of the coils, FIG. 4 is a timing chart showing the waveform of the current i1, and FIG. 5 is a waveform of the current i2. FIG. 6 is a timing chart showing waveforms of the currents i3 and i4, FIG. 7 is a circuit diagram when the main switch Q1 is on (mode A), and FIG. 8 is a circuit diagram when the sub switch Q2 is on (mode B). . Note that the directions of currents during charging and discharging of the capacitors C1 and C2 are as shown in FIGS. In FIG. 6, the current i3 of the coil W6 of the transformer T2 and the current i4 of the coil W3 of the transformer T1 are shown together. In mode A, the current i4 can be regarded as almost zero, and in mode B, the current i3 can be regarded as almost zero. In the actual timing chart shown in FIG. 2, the dead time is preferably set. In practice, the switching elements Q1 to Q4 do not rise instantaneously, but naturally transition between ON and OFF at a predetermined inclination rate. As shown in FIG. 2, the switching elements Q1 and Q3 operate synchronously, the switching elements Q2 and Q4 operate synchronously, the switching elements Q1 and Q2 operate complementarily, and the switching elements Q3 and Q4 operate complementary. In mode A, switching elements Q1, Q3 are turned on, and switching elements Q2, Q4 are turned off. In mode B, switching elements Q1, Q3 are turned off, and switching elements Q2, Q4 are turned on.

The circuit operation will be described with reference to FIGS. Figure 7 is a main switch Q 1 on the state (mode A) is a sub switch Q2 off, Figure 8 shows a main switch Q1 OFF, the state is a sub switch Q2 ON (mode B). In this embodiment, since the switching elements Q1 and Q2 have the parasitic diode D, when the potential of the connection point 20 is assumed to be 0V, the connection point 40 does not drop to about −0.8V or less.

  When the circuit of this embodiment shown in FIGS. 7 and 8 is compared with the conventional circuit shown in FIG. 9, the circuit 200 is the same. That is, in the conventional circuit shown in FIG. 9, the input DC power supply (corresponding to the first voltage system 1000) 2 supplies power directly to the circuit 200. On the other hand, in FIG. 7 and FIG. 8, the input DC power supply 2 supplies the current controlled by the main switch Q1 to the circuit 200 through the primary coils W1 and W4 and also supplies them to the secondary coils W3 and W6. For this reason, in the conventional circuit shown in FIG. 9, a direct current flows through the primary coils W2 and W5 when the main switch Q1 is on. In contrast, on the primary side of the circuit 200 of this embodiment shown in FIGS. 7 and 8, only the capacitors C1 and C2 are charged and discharged, and no direct current flows through the primary coils W2 and W5.

  Hereinafter, each period (mode) of the operation of one cycle will be sequentially described. The input current from the input DC power supply 2 is referred to as i1, the charge / discharge current of the capacitor C1 is referred to as i2, and the charge / discharge current of the capacitor C2 is referred to as i3. For simplification, it is assumed that the transformers T1 and T2 each have one turn, and each magnetomotive force (ampere turn) is equal to the current value, and the magnetic resistances of the transformers T1 and T2 are equal. However, it should be considered that elements corresponding to the leakage inductance are connected in series to each coil, but this leakage inductance is ignored in the following description. In addition, each mode described below describes one cycle after sufficient time has elapsed since the DC-DC converter 1 is started.

  In the following description, the dead time is omitted, but the dead time can be set.

(Description of mode A)
A mode A in which the main switch Q1 is turned on and the sub switch Q2 is turned off will be described with reference to FIG. In the following description, since the number of turns is 1, the same sign in (n is a number) as the current is used as the sign of the magnetomotive force (ampere turn) of each coil. Accordingly, the magnetomotive force (ampere turn) of each coil has a positive or negative sign depending on the direction of magnetic flux formation.

  When the main switch Q1 is turned on, the current i1 that has passed through the primary coils W1 and W4 from the input DC power supply 2 is directly commutated from the flow toward the capacitor C1 via the primary coils W5 and W2 in the preceding mode B. Head to the connection point 20. Thereby, the input current i1 increases with time, and in the transformer T2, a magnetomotive force (ampere turn) i1 of the primary coil W4 and a magnetomotive force (ampere turn) i2 of the primary coil W5 are formed. Further, the capacitor C1, which is charged in the mode B described later and is higher than the average voltage Vin, is discharged with the current i2 through the primary coils W2, W5 and the main switch Q1. This current i2 becomes a flow that increases with time in the discharge direction.

  In this embodiment, the direction of the magnetic flux formed by the magnetomotive force (ampere turn) i1 of the primary coil W4 is made equal to the direction of the magnetic flux formed by the magnetomotive force (ampere turn) i2 of the primary coil W5. As a result, a magnetic flux φ2 corresponding to the sum (i1 + i2) of these magnetomotive forces (ampere turns) is formed. A secondary voltage V6 having a magnitude corresponding to the change of the magnetic flux φ2 is formed in the secondary coil W6. The winding direction of the secondary coil W6 is the direction in which the current i3 is output in mode A.

  If the load 3 is regarded as a resistance, a current i3 having a magnitude proportional to the secondary voltage V6 flows. Therefore, ideally, this current i3 becomes a substantially direct current having a predetermined amplitude during this mode period. In response to the outflow of the current i3 from the transformer T2, the current i1 flowing through the primary coil W4 and the current i2 flowing through the primary coil W5 increase. That is, since the current i1 flowing through the primary coil W4 and the current i2 flowing through the primary coil W5 correspond to the exciting current and current i3 of the transformer T2, the exciting current of the transformer T2 is subtracted from the current i2 flowing through the primary coil W5. And the current i1 flowing through the primary coil W4 are output from the secondary coil W6 as a current i3.

  Eventually, during the period in which the main switch Q1 is on, in the transformer T2, an increase in magnetomotive force (ampere turn) in the first direction due to an increase in current in the primary coil W4 and a current in the discharge direction of the primary coil W5. A secondary voltage is formed in the secondary coil W6 by the combined magnetomotive force (ampere turn), which is the sum of the increase in magnetomotive force (ampere turn) due to the increase in current, and the current i3 is output. At this time, the transformer T1 operates as a choke coil. That is, during the period in which the main switch Q1 is on, in the transformer T2, an increase in magnetic flux due to an increase in current in the primary coil W4 and an increase in magnetic flux due to an increase in current in the charging direction of the primary coil W5 that promotes the increase in magnetic flux. The current i3 is output from the secondary coil W6, and the power energy for outputting the current i3 is supplied from the primary coils W4 and W5.

(Description of mode B)
Next, the mode B in which the main switch Q1 is turned off and the sub switch Q2 is turned on will be described with reference to FIG.

  The current i1 from the input DC power supply 2 through the primary coils W1 and W4 to the connection point 40 tends to decrease due to the increase in potential at the connection point 40 due to the main switch Q1 being turned off. A current i2 'flowing from the connection point 40 through the primary coils W5 and W2 charges the capacitor C1. The current ic2 flowing from the input DC power supply 2 through the primary coils W1 and W4 flows through the capacitor C2 and charges the capacitor C1. Therefore, the current i1 is equal to the sum of i2 'and ic2, that is, the charging current i2 of the capacitor C1.

  At this time, the magnetic flux in a certain direction of the transformer T1 decreases due to the decrease of the current i1 of the primary coil W1. At this time, the current change in the primary coil W2, that is, the change from the discharge current i2 shown in FIG. 7 to the charge current i2 ′ shown in FIG. 8 is directed to the direction of the transformer T1 due to the decrease in the current i1 of the primary coil W1. The direction is to promote the reduction of the magnetic flux.

  As a result, the secondary coil W3 outputs the current i4 due to the change (decrease from increase) of the current i1 in the primary coil W1 and the current change (i2 ′ in the opposite direction from i2) in the primary coil W2. A voltage V3 is generated. This voltage is a substantially DC voltage having a magnitude proportional to the rate of change of magnetic flux due to the current change. The winding direction of secondary coil W3 is the direction in which current i4 is output in mode B.

  If the load 3 is a resistance, a current i4 proportional to the magnitude of the voltage V4 flows. Ideally, this current i4 becomes a substantially direct current having a predetermined amplitude during this mode period. In response to the outflow of the current i4 from the transformer T1, the current i1 flowing through the primary coil W1 and the current i2 ′ flowing through the primary coil W2 increase, so that the exciting current of the transformer T1 from the current i2 ′ flowing through the primary coil W2 And the current i1 flowing through the primary coil W1 are output as the current i4 from the secondary coil W3.

  Eventually, during the period in which the main switch Q1 is off, in the transformer T1, it is caused by a decrease in magnetomotive force (ampere turn) due to a decrease in the current of the primary coil W1, and an increase in current in the charging direction of the capacitor C1 of the primary coil W2. A secondary voltage is formed in the secondary coil W3 by the combined magnetomotive force (ampere turn), which is the sum of the increase in magnetic force (ampere turn), and the current i4 is output. At this time, the transformer T2 operates as a choke coil. That is, during the period in which the main switch Q1 is off, the transformer T1 has a magnetic flux reduction due to the current reduction of the primary coil W1, and a magnetic flux reduction due to a current change in the discharge direction of the primary coil W2 that promotes the magnetic flux reduction. The current i4 is output from the secondary coil W3, and the power for outputting the current i4 is supplied from the primary coils W1 and W2. Capacitor C2 and sub switch Q2 are essentially a clamp circuit that prevents the generation of surge voltage when main switch Q1 is off.

(Explanation of power flow)
The flow of power in the DC-DC converter 1 will be described with reference to FIG.

  In FIG. 10, 1000 is an input voltage system (first voltage system) corresponding to the input DC power supply 2, 11, 21 are AC / DC converter circuits, and 2000 is an output voltage system (second voltage system) corresponding to the load 3 including the battery. It is. FIG. 10 shows a case where power is transmitted from the first voltage system 1000 to the second voltage system 2000, and the AC / DC conversion circuit 11 converts DC power to AC power for transmission from the first voltage system 1000 to the transformers T1 and T2, and AC / DC The conversion circuit 21 converts AC power into DC power for transmission from the transformers T1 and T2 to the second voltage system 2000. Coils W2 and W5 are connected in series to form a second coil pair, and coils W1 and W4 are connected in series to form a first coil pair.

  As shown in FIG. 1, the AC / DC converter circuit 11 includes a main switch Q1, a capacitor C1, a capacitor C2, and a sub switch Q2, and has a function of flowing an alternating current through the second coil pair W2 and W5. Further, the AC / DC converter circuit 11 has a function of flowing a switching current including a direct current component to the coils W1 and W4 through the main switch Q1.

  The first coil pair is formed by connecting in series a coil W1 that is electromagnetically coupled to the coils W2 and W3 by the transformer T1, and a coil W4 that is electromagnetically coupled to the coils W5 and W6 by the transformer T2. The AC / DC converter circuit 11 is connected to the lower side of the first voltage system 1000 through the main switch Q1. As a result, a DC current flows from the input DC power supply 2 to the first coil pair at the same time as the main switch Q1 is turned on to discharge the capacitor C1. For this reason, the input current i1 flowing through the first coil pair can flow both when the capacitor C1 is charged and when the capacitor C1 is discharged, and the ripple component of the input current i1 is reduced. The ripple component here means the carrier frequency of the main switch Q1 or its harmonic component, but does not mean slow current fluctuation or surge current.

  More specifically, the second coil pair outputs power from the coil W5 during the capacitor C1 discharging period when the main switch Q1 is turned on, and outputs power from the coil W2 during the capacitor C1 charging period when the main switch Q1 is turned off.

  The capacitor C1 is charged by charging the capacitor C1 through the first voltage system 1000 through a first coil pair and a second coil pair, which will be described later, from the first voltage system 1000 when the main switch Q1 is turned off. That is, power is transmitted from the coil W2 of the second coil pair to the coil W3 when the capacitor C1 is charged, and power is transmitted from the coil W5 of the second coil pair to the coil W6 when the capacitor C1 is discharged when the main switch Q1 is turned on. After all, in charging of the capacitor C1 in the mode B, the AC / DC converter circuit 11 outputs the power through the primary coil W5 through the transformer T2, and charges the capacitor C1, and the energy charged in the capacitor C1 in the next mode A is primary. It outputs as electric power through the coil W2.

  Further, the increase in the input current i1 when the main switch Q1 is turned on accompanied by the discharge of the capacitor C1 causes the accumulation of magnetic energy in the coil W1 constituting the first coil pair, the choke coil action, and the power from the coil W4 to the coil W6. Enable transmission. Conversely, a decrease in the input current i1 when the main switch Q1 is off accompanied by charging of the capacitor C1 results in power transmission from the coil W1 to the coil W3, accumulation of magnetic energy in the coil W4, and choke coil action. Thereby, the ripple component of the input current i1 is further reduced. Eventually, the power flow is as shown in FIG. That is, during the period in which the main switch Q1 is turned on and the capacitor C1 is discharged, energy P5 is transmitted from the coil W5, energy P4 is transmitted from the coil W4 to the coil W6, and the coils W1 and W2 store magnetic energy as choke coils. . In the period in which the main switch Q1 is turned off and the capacitor C1 is charged, energy P1 is transmitted from the coil W1, energy P2 is transmitted from the coil W2 to the coil W3 side, and the coils W4 and W5 store magnetic energy as choke coils. It is clear that these stored magnetic energies are used effectively later.

(Example effect)
In the DC-DC converter 1 of this embodiment described above, even if the ON period and the OFF period of the main switch Q1 are changed, the current i3 + i4 of the DC-DC converter 1 can be regarded as a substantially continuous DC current without being interrupted. it can. Further, the input current i1 from the input DC power supply 2 to the DC-DC converter 1 always has a current waveform flowing into the DC-DC converter 1 from the input DC power supply 2 side, and there is almost no reverse flow period as in the conventional case. There is no need to install a smoothing capacitor in parallel with the DC power supply 2.

  Some of the transmitted power is transmitted from the primary coils W1, W4 to the secondary coils W3, W6 by the ripple of the current i1, but most of the transmitted power is the mode B of the current flowing through the primary coils W2, W5. Since power is transmitted by reversing the mode A, large power can be output to the secondary side even though the ripple component of the input current is small. In addition, the capacitor C1 can significantly reduce the capacitance as compared with a smoothing capacitor that is conventionally connected in parallel with the input DC power supply 2. Further, since the transformer T1 in the mode A and the transformer T2 in the mode B act as choke coils, the current changes of the primary coils W1 and W4, that is, the input current of the DC-DC converter is smoothed, and the secondary coil The current, that is, the output current of the DC-DC converter can be smoothed.

(Modification 1)
Of course, the dead time may be set.

  If a dead time is provided between the turn-off of the sub switch Q2 and the subsequent turn-on of the main switch Q1, an LC circuit of the parasitic capacitance between both ends of the main switch Q1 and the leakage inductance of the transformer is formed by turning off the sub switch Q2. Then, a current flows resonantly to the main switch Q1 through this parasitic capacitance, so that soft switching is realized if the main switch Q1 is turned on at a timing when the voltage drop or current of the main switch Q1 becomes 0 or in the vicinity thereof. Can do.

  In addition, if a dead time is provided between the main switch Q1 being turned off and the subsequent sub switch Q2 being turned on, the surge voltage caused by the main switch Q1 being turned off is commutated to the capacitor C2 through the parasitic diode of the sub switch Q2. Can be prevented from occurring.

(Modification 2)
The ratio between the number of turns of the primary coils W1 and W4 and the number of turns of the primary coils W2 and W5 may be 1 or a value other than 1.

(Modification 3)
The ratio between the leakage inductance and the excitation inductance of the transformers T1 and T2 can be variously set according to the application. As will be described later, the ripple component of the input current and the ripple component of the output current are reduced in a wide duty range. It is preferable to set.

(Modification 4)
In this embodiment, the transformers T1 and T2 are configured by a transformer having a gap type core in which a predetermined gap is formed in the closed magnetic circuit, thereby preventing magnetic saturation due to a direct current component from being easily generated. However, it is not essential to have a gapped core.

(Modification 5)
In this embodiment, the step-down DC-DC converter has been described. However, it is possible to obtain a step-up DC-DC converter by changing the turn ratio of the primary coil and the secondary coil of the transformers T1 and T2. Is natural.

(Deformation mode 6)
In this embodiment, the output switches Q3 and Q4 are complementarily operated to perform synchronous rectification. However, one or both of the output switches Q3 and Q4 may be replaced with a rectifier diode. Further, the positions of the coils W3 and W6 and the output switches Q3 and Q4 and the rectifier diode instead thereof may be exchanged.

(Deformation mode 7)
In the first embodiment, a circuit using the main switch Q1, the sub switch Q2, the capacitor C1, and the capacitor C2 as an inverter circuit that converts input DC power into AC power and supplies it to the transformers T1 and T2 receives power from the power source. Although the input side circuit is a synchronous rectifier circuit including the output switches Q3 and Q4 and the capacitor C3, the output side circuit is configured to supply power to the power supply, but conversely, it may be a unidirectional DC-DC converter that transmits power.

(Deformation mode 8)
The core of the transformer T1 and the core of the transformer T2 may be a merged core having a common magnetic path. Alternatively, the core of the transformer T1 and the core of the transformer T2 may be juxtaposed, and a coil may be wound around a pillar portion having one core and a pillar portion of another core. In this way, the primary coils W2 and W5 can be the same coil, and the primary coils W1 and W4 can also be the same coil.

  A DC-DC converter 1 according to a second embodiment will be described with reference to FIG.

  In this embodiment, the range of change of the on-duty ratio of the main switch Q1 is limited to a predetermined range X (here, 40 to 60%) centering on 50% in applications that do not require a wide range of on-duty ratio changes. is there. Thereby, as shown in FIG. 11, the ripple component of output current can be suppressed to less than a predetermined value. The ripple component of the output current io has a characteristic (see FIG. 13) that continuously changes depending on the on-duty ratio (on-duty ratio) of the main switch Q1, and the minimum point of the on-duty ratio at this time is various circuit constants, In particular, it is influenced by the magnetic resistance of the transformers T1 and T2. When the electromagnetic characteristics of the transformer T1 and the transformer T2 are equal, the ripple component of the output current io becomes 0 at an on-duty ratio of 50%, and the output current can be flattened.

  A DC-DC converter 1 of Example 3 will be described with reference to FIG.

  In this embodiment, the unidirectional DC-DC converter of the first embodiment is changed to a bidirectional DC-DC converter. Reference numeral 300 denotes a primary side circuit of the transformers T1 and T2, and corresponds to a primary side circuit portion of the circuit 200 shown in FIG. Reference numeral 400 denotes a synchronous rectifier circuit, which corresponds to the secondary circuit portion of the circuit 200 shown in FIG. 7, and includes output switches Q3 and Q4 and a capacitor C3. The primary side circuit 300 corresponds to the AC / DC converting circuit 11 referred to in the present invention, and the secondary side circuit 400 is a synchronous rectifier circuit and corresponds to the AC / DC converting circuit 21 referred to in the present invention. Reference numeral 500 denotes a controller, 600 denotes a driver for PWM control of the switching elements Q1 and Q2, and 700 denotes a driver for PWM control of the switching elements Q3 and Q4.

  The controller 500 reads the output voltage of the synchronous rectifier circuit 400 when power is transmitted from the power source 2, that is, the first voltage system 1000 side in the present invention, to the power source 3, that is, the second voltage system 2000 side, and stores the output voltage in advance. The on-duty ratio of the main switch Q1 is increased when the output voltage is smaller than the target voltage, and the main switch Q1 is turned on when the output voltage is higher than the target voltage. Control to reduce the duty ratio. By this feedback control, power can be transmitted from the power source 2 to the power source 3 to converge the voltage of the power source 3 to the target voltage.

  Next, when the controller 500 transmits power from the power source 3 side to the power source 2 side, the input voltage of the circuit 300 (the input DC voltage Vin in the first embodiment) is read, and this input voltage and a pre-stored target voltage are read. Based on the comparison result, the on-duty ratio of the main switch Q1 is decreased when the input voltage is lower than the target voltage, and the on-duty ratio of the main switch Q1 is increased when the input voltage is higher than the target voltage. Take control. By this feedback control, power can be transmitted from the power source 3 to the power source 2 to converge the voltage of the power source 2 to the target voltage.

  A DC-DC converter 1 of Embodiment 4 will be described with reference to FIG.

  In this embodiment, in the DC-DC converter of the first embodiment, the output switch Q3 is turned on by a predetermined time Δt1 before the main switch Q1 is turned on. Similarly, the output switch Q4 is turned on for a predetermined time Δt2 after the sub switch Q2 is turned on. It is an early one. In this way, since the output switches Q3 and Q4 are short-circuited at the time of switching, the surge voltage can be suppressed without using a snubber circuit due to this short-circuit current, and power that has been wasted until now can be reduced. Effective recovery can be achieved electromagnetically by the transformers T1 and T2. The predetermined time for turning on the output switch Q3 earlier than the on of the main switch Q1 may be equal to or different from the predetermined time for turning on the output switch Q4 earlier than the on of the sub switch Q2.

  A DC-DC converter 1 according to a fifth embodiment will be described below with reference to FIG.

  This embodiment is characterized in that the AC / DC conversion circuit 11 including the main switch Q1, the sub switch Q2, and the capacitors C1 and C2 shown in FIG. 1 is used on both sides of the transformers T1 and T2. That is, in FIG. 14, the AC / DC conversion circuits 11 and 11 'are each composed of the AC / DC conversion circuit 11 including the main switch Q1, the sub switch Q2, and the capacitors C1 and C2 shown in FIG.

  However, the main switch Q1 of the AC / DC conversion circuit 11 ′ shown in FIG. 14 and the main switch Q1 of the AC / DC conversion circuit 11 are operated in opposite phases, and the sub switch Q2 of the AC / DC conversion circuit 11 ′ shown in FIG. The circuit 11 is operated in reverse phase with respect to the sub switch Q2.

  A DC-DC converter 1 according to a sixth embodiment will be described with reference to FIGS. 15 and 16. This embodiment is a modification of the bidirectional DC-DC converter of Embodiment 3 shown in FIG. 12, FIG. 15 is a circuit diagram of the bidirectional DC-DC converter of this embodiment, and FIG. 16 is its timing chart. It is.

  In this embodiment, in the DC-DC converter of Embodiment 1 shown in FIG. 1, the load 3 as the second voltage system 2000 is changed to the low voltage battery 31, and the input DC power supply 2 as the first voltage system 1000 is replaced with the power supply. The switch 32, the high voltage battery 33, and the inverter 34 are changed. In other words, this embodiment is obtained by adding an inverter 34 and a power switch 32 to the high voltage battery 2 in the bidirectional DC-DC converter of the second embodiment shown in FIG. The power switch 32 is connected in series with the high voltage battery 33 at the high end of the high voltage battery 33 and is controlled by the controller 4. Therefore, the AC / DC conversion circuit 11 of the DC-DC converter 1 is supplied with power from the high voltage battery 33 through the power switch 32. The inverter 34 is a three-phase inverter that drives and controls a high-voltage three-phase AC motor 35 for generating driving power of the automobile, and a smoothing capacitor Cin is connected between a pair of DC input terminals.

  Hereinafter, the operation of the DC-DC converter 1 will be described with reference to a timing chart shown in FIG.

  If the power switch 32 is open when the controller 4 receives a start request for the DC-DC converter 1 from an external vehicle ECU, the DC-DC converter 1 is first turned on before the power switch 32 configured by a relay is turned on. The operation is performed to precharge the smoothing capacitor Cin. In this precharge operation, the on-duty ratio of the switching element Q1 is gradually decreased from a predetermined value.

  In the DC-DC converter 1, as the on-duty ratio of the switching element Q1 increases, the secondary voltage generated in the coil W6 increases, and as a result, the output current to the load 3 as the second voltage system 2000 increases. . Further, as the on-duty ratio of the switching element Q1 decreases, the secondary voltage generated in the coil W6 decreases, and as a result, the output current to the load 3 as the second voltage system 2000 decreases. Further, when the on-duty ratio of the switching element Q1 is reduced, the secondary voltage generated in the coil W6 is reduced, and as a result, power is supplied from the second voltage system 2000 to the first voltage system 1000 side through the DC-DC converter 1. Done. The secondary voltage generated in the coil W3 has an increase / decrease change characteristic in the opposite direction to the secondary voltage generated in the coil W6.

  That is, in the state where the power switch 32 is open, the on-duty ratio of the switching element Q1 is gradually reduced, and thereby the voltage applied by the AC / DC converter circuit 11 to the smoothing capacitor Cin gradually increases. After performing this precharge operation for a predetermined period or until the stored voltage of the smoothing capacitor Cin reaches a predetermined value close to the voltage of the high voltage battery 33, the power switch 32 is turned on. Then, the voltage of the high voltage battery 33 is applied to the AC / DC converter circuit 11, and power is transmitted from the AC / DC converter circuit 11 to the low voltage battery 31 through the DC-DC converter 1. Thereby, when the power switch 32 is turned on, an inrush current can be prevented from flowing from the high voltage battery 33 into the smoothing capacitor Cin of the inverter 34.

  The precharge operation of the smoothing capacitor Cin can be used not only for power transmission from the high voltage battery 33 to the low voltage battery 31 but also for driving the traveling motor 35 from a state where the power switch 32 is open. That is, it is preferable that the power switch 32 is always turned on. In this way, it is possible to always prevent the inrush current from flowing through the smoothing capacitor Cin when the power switch 32 is turned on.

  Control of the precharge operation when the power switch 32 is turned on will be described with reference to the flowchart shown in FIG. First, the presence / absence of a turn-on command for the power switch 32 is detected (S100). If the turn-on command is given, a precharge operation for gradually decreasing the on-duty ratio D of the switching element Q1 is performed (S102). When the time elapses (S104), it is determined that the precharge operation is completed, and the power switch 32 is turned on (S106). This precharge control can naturally be processed by a hardware circuit having an equivalent function instead of the software processing shown in FIG.

  A DC-DC converter 1 of Example 7 will be described with reference to FIG. In this embodiment, the input voltage from the high voltage battery 2 as the first voltage system 1000 to the AC / DC converter circuit 11 of the DC-DC converter 1 rises rapidly, and the feedback control described in the third embodiment cannot catch up. Considering this, it is made to prevent an excessive voltage from being applied to the switching element Q1 or the like.

  More specifically, in the third embodiment shown in FIG. 12, in the step-down power transmission operation from the power source 2 (first voltage system 1000) to the power source 3 (second voltage system 2000), the secondary side circuit corresponding to the AC / DC converter circuit 21. When the output voltage of 400 exceeds the set value, the on-duty ratio of the switching element Q1 is decreased, and when it is smaller than the set value, the feedback control is performed to increase the on-duty ratio of the switching element Q1. Thereby, even if the voltage of the power supply 2, that is, the first voltage system 1000 shown in FIG. 1 increases, the output voltage of the synchronous rectifier circuit 400 can be converged to a predetermined set value.

  However, in FIG. 12, when the output voltage of the power source 2, that is, the input voltage of the primary side circuit 300, that is, the AC / DC conversion circuit 11 of the DC-DC converter 1, changes abruptly, there is a possibility that this feedback control cannot follow.

  In the circuit of FIG. 1, the voltage Vds between the source and the drain of the switching element Q1 is Vin / when the output voltage of the power source 2, that is, the input voltage of the AC / DC converter circuit 11, is Vin and the on-duty ratio D of the switching element Q1. It is approximately equal to (1-D). That is, the maximum value Vdsmax (also referred to as Vsmax) of the source-drain voltage Vds of the switching element Q1 exceeds Vin / (1-Dmax) when the on-duty ratio D of the switching element Q1 at that time is Dmax. There is nothing. Therefore, if the maximum value Dmax of the on-duty ratio D is decreased as the input voltage Vin increases, the maximum value Vdsmax can be prevented from exceeding a constant value, that is, the breakdown voltage Vdsth between the source and drain of the switching element Q1. Can do.

  The operation for setting the maximum value Dmax of the on-duty ratio D will be described with reference to the flowchart shown in FIG. First, the input voltage Vin is read (S200), and the maximum value Dmax of the on-duty ratio D corresponding to the input voltage Vin is determined from a table indicating the relationship between the input voltage Vin stored in advance and the maximum value Dmax of the on-duty ratio D. Extraction is performed (S202), and the current on-duty ratio D command is read (S204). Next, it is determined whether or not the current on-duty ratio D is larger than the maximum value Dmax of the on-duty ratio D (S206). If larger, the on-duty ratio D is set to the maximum value Dmax and AC / DC conversion is performed. Otherwise, the current on-duty ratio D is output to the AC / DC conversion circuits 11 and 21 (S210). The relationship between the input voltage Vin stored in the table and the maximum value Dmax of the on-duty ratio D is shown as a line L1 in FIG. Note that the line L2 in FIG. 19 is the maximum value of the on-duty for suppressing the switching element Q1 within the breakdown voltage.

  In this way, it is possible to reliably prevent breakdown of the switching element Q1 due to a sudden rise in the input voltage Vin. Further, when the input voltage Vin is small, the maximum value Dmax of the on-duty ratio D can be set large, so that the degree of freedom in control can be increased.

  Note that the processing in FIG. 18 is performed by the controller 500 shown in FIG. The PWM control command determined by the controller 500 or input to the controller 500 from the outside is subjected to the maximum value limitation as shown in FIG. 18, and then the primary side circuit 300, that is, the AC / DC conversion circuit 11 through the drivers 600 and 700, and The output is output to the secondary side circuit (synchronous rectifier circuit) 400 that operates in synchronization, that is, the AC / DC converter circuit 21, to drive the switching elements Q1 to Q4. The processing shown in FIG. 18 is preferably performed as fast and frequently as possible. Of course, it can be processed by a hardware circuit having an equivalent function instead of the software processing shown in FIG.

  20 and 21 show waveforms of the source-drain voltage of the switching element Q1 when the maximum value Dmax of the on-duty ratio D of this embodiment is limited according to the input voltage Vin and when it is not limited. FIG. 20 shows a case where no restriction is made, and FIG. 21 shows a case where restriction is made.

  A DC-DC converter 1 of Example 8 will be described with reference to FIG. FIG. 22 shows a configuration in which a current sensor 71 is connected in series with the switching element Q1 in FIG. Various known sensors can be used as the current sensor. This embodiment is characterized in that the current detected by the current detection sensor 71 is used to control the operation and stop of synchronous rectification by the AC / DC conversion circuit 21 which is a synchronous rectification circuit.

  An example in which the synchronous rectification stop control is performed by software executed by the controller 4 will be described with reference to a flowchart shown in FIG. This subroutine is started when the main switch Q1 is turned on, that is, when the operation command Vgq1 of the main switch Q1 becomes high level.

  First, when the routine starts, the built-in timer starts counting (S300), and the current iq1 of the switching element Q1 is read from the current sensor 71 (S302). Next, it is checked whether or not the count value of the timer has reached a predetermined threshold time Tth set in advance (S304). If not reached, the process returns to S302, and if it has reached, the detected current iq1 is set in advance. It is determined whether or not it is larger than the predetermined current threshold value ith in the forward (positive) direction (S306). If it is larger, it is determined that no reverse flow of the synchronous rectifier circuit has occurred. Therefore, the switching elements Q3 and Q4 of the AC / DC converter circuit 21 are turned off, and the synchronous rectification is interrupted (S308). Even if the load current output from the AC / DC converter circuit 21, which is a synchronous rectifier circuit, to the second voltage system 2000 is 0, the predetermined current threshold value ith is at least as much as the load of the transformer Q1. In consideration of occurrence as a component, it is preferable to adopt a value larger in the forward direction by a predetermined value than the current 0 level of the main switch Q1. Therefore, the threshold time Tth is preferably set to a time until the detected current of the main switch Q1 coincides with the current threshold value ith when there is no load.

(Modification)
The inverter, that is, the AC / DC converter circuit 11, may be stopped for a predetermined time simultaneously with the switching elements Q3 and Q4 being turned off.

(Modification)
In the above embodiment, the current value of the main switch Q1 after a predetermined time after the main switch Q1 is turned on (the time until the current value of the main switch Q1 at the load current 0 becomes the current threshold value ith after the main switch Q1 is turned on) is The load current level, that is, the possibility of occurrence of a reverse current was determined by determining whether it is larger or smaller in the forward direction than a predetermined current threshold value ith.

  Instead, a subroutine shown in the flowchart shown in FIG. 24 may be adopted. In this subroutine, first, the count of the built-in timer is started simultaneously with the start of the routine (S300), and the current iq1 of the switching element Q1 is read from the current sensor 71 (S302). Next, it is determined whether or not the current iq1 has reached a predetermined current threshold value isth (S314). When the current iq1 is reached, the timer is stopped and the count value at that time, that is, the elapsed time Tc from the main switch Q1 is turned on Compared with a predetermined threshold time Tth (S316), if the elapsed time Tc is equal to or less than the threshold time Tth, it is determined that no reverse flow of the synchronous rectifier circuit has occurred, and the routine is terminated. If it is longer than the time Tth, it is determined that there is a possibility of backflow in the synchronous rectifier circuit, the switching elements Q3 and Q4 of the AC / DC converter circuit 21 are turned off, and the synchronous rectification is interrupted (S318).

  In this embodiment, for example, a hardware circuit having an equivalent function can be adopted instead of the software processing shown in FIGS.

  FIG. 25 shows the current waveform of the main switch Q1 and the gate voltage waveform of the main switch Q1 at various load currents.

  In FIG. 25, iq1M of the current iq1 of the main switch Q1 is an exciting current component, iq1L is a predetermined load current component of the current iq1 of the main switch Q1, and the sum of both is the current iq1 of the main switch Q1. It becomes. The excitation current component iq1M is such that the current value of the main switch Q1 first flows in the reverse direction from the start of turning on of the main switch Q1 when there is no load, and this reverse current gradually decreases at a substantially constant current change rate. After that, a waveform in which the forward current gradually increases until the main switch Q1 is turned off is obtained. This is because the current flowing through the main switch Q1 when the main switch Q1 is turned on leaves the first voltage system 1000 and returns to the first voltage system 1000 through the coils W1, W4 and the main switch Q1, and the positive electrode of the capacitor C1. This is because it becomes the sum of the capacitor supply components of the current components that return from the coils W2 and W5, the main switch Q1, and the negative electrode of the capacitor C1. When the load current increases, the current component that flows in the reverse direction from the start of turning on the main switch Q1 by the amount of the load current gradually decreases to zero at a substantially constant current change rate. Thereafter, the forward current has a waveform that gradually increases until the main switch Q1 is turned off. Therefore, if the current iq1 of the main switch Q1 exceeds a certain current threshold value isth and becomes positive, the load current is large, and if it is late, the load current is small. If the current threshold value isth is set to a negative value, it is theoretically possible to determine the magnitude of the load current, that is, the possibility of reverse current generation immediately after the main switch Q1 is turned on. However, when soft switching of the main switch Q1 is used, the current waveform sinks negatively immediately after the main switch Q1 is turned on as shown in FIG. 25 due to the influence of LC resonance of the circuit. The determination is preferably set after a predetermined time from which the main switch Q1 is turned on.

(Other effects)
According to each embodiment described above, the following effects can be obtained.

  (1) First, the primary ampere turns of the transformers T1 and T2, that is, the reversal of the direction of the magnetic flux, the main switch Q1 and the sub switch Q2 are turned on and off to change the direction in which the input current i1 flows. And the primary coils W2 and W5 are switched so that the current direction to the capacitors C1 and C2 is reversed, so that the input current i1 itself is generated from the input DC power supply 2 to the primary coils of the transformers T1 and T2 in all modes. As a result, the ripple component of the current of the input current i1 can be greatly reduced as compared with the two-transform DC-DC converter shown in FIG. 9, which is necessary in FIG. The large-capacity smoothing capacitor C1, which had to be connected in parallel with the input DC power supply 2, may be omitted or miniaturized. it can. The importance of reducing the ripple component of the input current and output current in the DC-DC converter is well known, and electromagnetic noise can be reduced and the smoothing capacitor can be downsized or omitted.

  (2) Since the output switches Q3 and Q4 alternately output during the on period and the off period of the main switch Q1, the output current can be constantly output, and the ripple component of the output current can be greatly reduced. As a result, the output side choke coil can be omitted. In addition, the withstand voltage of the rectifying element can be reduced as compared with a type in which the output between the neutral point of one transformer and both ends is full-wave rectified by two rectifying elements.

  (3) Next, since the direct current excitation current component of the transformer T2, that is, the direct current component included in the sum of the current changes of the primary coils W4 and W5, becomes small, the direct current magnetization and magnetic saturation of the transformer T2 can be suppressed. The gap width in the magnetic circuit (core) can be greatly shortened, and the core of the transformer T2 can be downsized accordingly.

  (4) The ripple component of the output current can be adjusted, and the ripple component of the output current can be minimized within a suitable use range.

  (5) Energy efficiency can be improved by reducing input current and output current ripples, realizing soft switching, and reducing copper loss in transformers and the like.

  (6) By turning on the output switches Q3 and Q4 early, it is possible to effectively recover power while suppressing surges.

  (7) The power transmission direction can be switched by a control circuit having a simple configuration.

1 is a circuit diagram showing a two-transform DC-DC converter of Example 1. FIG. 2 is a timing chart of the DC-DC converter 1 in FIG. 1. It is each part voltage waveform of the DC-DC converter 1 of FIG. It is a timing chart which shows the waveform of input current i1. It is a timing chart which shows the waveform of the charging / discharging electric current i2 of the capacitor | condenser C1. It is a timing chart which shows an output current waveform. FIG. 3 is a circuit diagram illustrating an operation in mode A in the two-transform DC-DC converter according to the first embodiment. FIG. 3 is a circuit diagram illustrating an operation in mode B in the two-transformer DC-DC converter according to the first embodiment. It is a circuit diagram which shows an example of the conventional 2 transformer type DC-DC converter. FIG. 3 is a block diagram illustrating a power flow of the DC-DC converter according to the first embodiment. FIG. 6 is a circuit diagram illustrating a second embodiment. FIG. 6 is a circuit diagram showing Example 3. 10 is a characteristic diagram showing Example 4. FIG. FIG. 10 is a circuit diagram showing Example 5. FIG. 9 is a circuit diagram showing Example 6. 10 is a timing chart showing the operation of the sixth embodiment. 10 is a flowchart showing the operation of the sixth embodiment. 18 is a flowchart showing the operation of the seventh embodiment. It is a figure which shows the input voltage and ONTY ratio of Example 7. FIG. 10 is a test waveform diagram (not yet implemented) showing the effect of Example 7. FIG. 10 is a test waveform diagram (implementation) showing the effect of Example 7; FIG. 10 is a circuit diagram showing Example 8. 19 is a flowchart showing the operation of the eighth embodiment. 19 is a flowchart showing the operation of the eighth embodiment. It is a timing chart which shows the current waveform in the load state of a main switch.

Explanation of symbols

1 2 transformer type DC-DC converter 2 input power supply 3 load system 4 controller 10 first voltage system 11 AC / DC conversion circuit 20 second voltage system 21 AC / DC conversion circuit T1 transformer T2 transformer W1 transformer T1 primary coil W2 transformer T1 primary coil W3 secondary coil of transformer T1 W4 primary coil of transformer T2 W2 primary coil of transformer T2 W3 secondary coil of transformer T2 Q1 main switch (switching element)
Q2 Sub switch (switching element)
C1 capacitor C2 capacitor

Claims (21)

  1. A transformer T1 composed of a primary coil W1, W2 and secondary coil W3 Prefecture, which has two transformers and transformer T2 consisting of primary coil W4, W5 and secondary coil W6 Prefecture, coil W1, W4 is connected in series A magnetic circuit constituting a first coil pair, wherein the coils W2, W5 are connected in series to form a second coil pair;
    An AC / DC converter circuit 11 connected to the first voltage system 1000 operated at a predetermined DC power supply voltage so as to be able to exchange power;
    An AC / DC conversion circuit 21 connected to a second voltage system 2000 operated at a DC power supply voltage different from that of the first voltage system 1000 so as to be able to transmit and receive power;
    A controller for controlling power transmission between the first voltage system 1000 and the second voltage system 2000;
    With
    One end of the first coil pair is
    Connected to one end of the first voltage system 1000;
    The AC / DC converter circuit 11 includes:
    A main switch Q1 for connecting the other end of the first voltage system 1000 and the other end of the first coil pair;
    A capacitor C1 connecting the other end of the first voltage system 1000 and one end of the second coil pair;
    One end is connected to one end of the second coil pair, the other end is connected to the other end of the first coil pair and the other end of the second coil pair, and the current flowing through the main switch Q1 when the main switch Q1 is off is A bypass clamp circuit,
    Have
    The controller is
    A two-transform DC, wherein a charging mode in which the main switch Q1 is turned off and the capacitor C1 is charged and a discharging mode in which the main switch Q1 is turned on and the capacitor C1 is discharged are repeatedly performed at a predetermined cycle. DC converter.
  2. The two-transform DC-DC converter according to claim 1,
    The clamp circuit is
    It is composed of a capacitor C2 and a sub switch Q2 connected in series,
    The controller is
    A charging mode in which the main switch Q1 is turned off and the sub switch Q2 is turned on to charge the capacitor C1, and a discharge mode in which the main switch Q1 is turned on and the sub switch Q2 is turned off to discharge the capacitor C1. Are implemented alternately. A two-transform DC-DC converter.
  3. The two-transform DC-DC converter according to claim 1 or 2,
    The first coil pair is:
    When the first voltage system 1000 is the power transmission side and the second voltage system 2000 is the power reception side, current flows in the same direction from the first voltage system 1000 to the AC / DC converter circuit 11 without interruption. 2 transformer type DC-DC converter.
  4. In the 2 transformer type DC-DC converter in any one of Claims 1 thru | or 3,
    The AC / DC converting circuit 21 includes:
    A switching element Q3 having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W6;
    A switching element Q4 having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W3;
    With
    The controller is
    A two-transform DC-DC converter, wherein the main switch Q1 and one of the switching elements Q3 and Q4 are synchronously turned on, and the sub switch Q2 and the other of the switching elements Q3 and Q4 are synchronously turned on.
  5. In the 2 transformer type DC-DC converter in any one of Claims 1 thru | or 4,
    The controller is
    The two-transformer DC− is characterized in that the ripple component of the output current of the synchronous rectifier circuit is made less than a predetermined value level by restricting the change of the on-duty ratio of the output switches Q1, Q2 within a predetermined range. DC converter.
  6. In the 2 transformer type DC-DC converter in any one of Claims 1 thru | or 5,
    The first voltage system 1000 includes:
    A two-transform DC-DC converter characterized in that the voltage is higher than that of the second voltage system 2000.
  7. In the 2 transformer type DC-DC converter in any one of Claims 1 thru | or 5,
    The first voltage system 1000 includes:
    A two-transform DC-DC converter characterized in that the voltage is lower than that of the second voltage system 2000.
  8. In the 2 transformer type DC-DC converter in any one of Claims 1 thru | or 7,
    The AC / DC converter circuit 11 includes:
    2. A two-transformer DC-DC converter characterized in that it is connected to a DC power source or an electric load constituting the first voltage system 1000 without a current smoothing choke coil element.
  9. The two-transform DC-DC converter according to any one of claims 1 to 8,
    The AC / DC converting circuit 21 includes:
    2. A two-transformer DC-DC converter, characterized in that it is connected to a DC power source or an electric load constituting the second voltage system 2000 without passing through a choke coil element for current smoothing.
  10. The two-transform DC-DC converter according to any one of claims 1 to 9,
    The controller is
    A first power transmission mode from the first voltage system 1000 to the second voltage system 2000 operated at different DC power supply voltages by changing the on-duty ratio of the main switch Q1 that performs PWM control, and the second voltage A two-transform DC-DC converter characterized by switching to a second power transmission mode in which power is transmitted from the system 2000 to the first voltage system 1000.
  11. The two-transform DC-DC converter according to claim 10,
    The controller is
    In the first power transmission mode, the on-duty ratio of the main switch Q1 increases when the voltage of the second voltage system 2000 is lower than a predetermined target voltage, and decreases when the voltage is higher.
    In the second power transmission mode, the on-duty ratio of the main switch Q1 decreases when the voltage of the first voltage system 1000 is lower than a predetermined target voltage, and increases when the voltage is high. Type DC-DC converter.
  12. The two-transform DC-DC converter according to claim 10,
    A smoothing capacitor that is connected in parallel with the first voltage system 1000 and exchanges AC power with the AC / DC converter circuit 11;
    The first voltage system 1000 includes a DC power source and a power switch connected in series with each other,
    The controller performs power transmission from the second voltage system 2000 to the first voltage system 1000 to precharge the smoothing capacitor, and then closes the power switch.
  13. The two-transform DC-DC converter according to claim 12,
    The controller is
    A two-transform DC-DC converter characterized by gradually increasing the voltage applied to the smoothing capacitor in the precharging of the smoothing capacitor.
  14. The two-transform DC-DC converter according to any one of claims 4 to 13,
    A current sensor for detecting the current of the main switch Q1,
    The controller is
    A two-transform DC-DC circuit characterized in that the synchronous rectification operation of the synchronous rectifier circuit is stopped by determining a decrease in load current in the synchronous rectifier circuit based on the magnitude of the current detected by the current sensor at a predetermined timing. DC converter.
  15. The two-transform DC-DC converter according to claim 14,
    The controller is
    A two-transform DC-- characterized in that it is determined whether or not the synchronous rectification operation should be stopped based on the magnitude of the current of the main switch Q1 after a lapse of a predetermined time since the command to turn on the main switch Q1. DC converter.
  16. The two-transform DC-DC converter according to claim 15,
    The predetermined time is
    A two-transform DC-DC converter characterized in that it is set to be longer than the energizing time of a transient resonance current that flows in the reverse direction to the main switch Q1 immediately after the main switch Q1 is turned on.
  17. The two-transform DC-DC converter according to any one of claims 2 to 16,
    The controller is
    When power is transmitted from the first voltage system 1000 to the second voltage system 2000, the on-duty ratio of the main switch Q1 increases as the DC power supply voltage applied from the first voltage system 1000 to the AC / DC converter circuit 11 increases. 2 transformer type DC-DC converter characterized by reducing the maximum value of.
  18. The two-transform DC-DC converter according to any one of claims 2 to 16,
    When power is transmitted from the second voltage system 2000 to the first voltage system 1000, the on-duty ratio of the main switch Q1 increases as the DC power supply voltage applied from the second voltage system 2000 to the AC / DC converter circuit 21 increases. 2 transformer type DC-DC converter characterized by increasing the maximum value of.
  19. The two-transform DC-DC converter according to claim 12 ,
    The controller is
    A two-transform DC-DC converter characterized in that after the precharge to the smoothing capacitor is completed, the power switch is closed and power is transmitted from the first voltage system 1000 to the second voltage system 2000.
  20. In second transformer DC-DC converter according to any one of claims 1 to 19,
    The controller is
    Before Kijika flow supply voltage increases the main on-duty ratio maximum value 2 trans DC-DC converter, characterized in that to regulate the Dmax of D of the switch Q1 below a predetermined limit value decreases as the.
  21. The two-transform DC-DC converter according to any one of claims 1 to 9 and claims 17 , 18, and 20,
    The AC / DC converting circuit 21 includes:
    A first rectifier diode having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W6;
    And a second rectifier diode having one end connected to one end of the second voltage system 2000 and the other end connected to the other end of the second voltage system 2000 through the coil W3. DC-DC converter.
JP2004207597A 2003-07-16 2004-07-14 2 transformer type DC-DC converter Active JP4542844B2 (en)

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