JP3886876B2 - Power semiconductor element drive circuit - Google Patents

Power semiconductor element drive circuit Download PDF

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Publication number
JP3886876B2
JP3886876B2 JP2002283658A JP2002283658A JP3886876B2 JP 3886876 B2 JP3886876 B2 JP 3886876B2 JP 2002283658 A JP2002283658 A JP 2002283658A JP 2002283658 A JP2002283658 A JP 2002283658A JP 3886876 B2 JP3886876 B2 JP 3886876B2
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Japan
Prior art keywords
circuit
power semiconductor
voltage
semiconductor element
gate voltage
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JP2003284318A (en
Inventor
靖 中山
健史 大井
隆一 橋戸
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三菱電機株式会社
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    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03KPULSE TECHNIQUE
    • H03K17/00Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/16Modifications for eliminating interference voltages or currents
    • H03K17/168Modifications for eliminating interference voltages or currents in composite switches
    • HELECTRICITY
    • H03BASIC ELECTRONIC CIRCUITRY
    • H03KPULSE TECHNIQUE
    • H03K17/00Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/08Modifications for protecting switching circuit against overcurrent or overvoltage
    • H03K17/081Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit
    • H03K17/0812Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the control circuit
    • H03K17/08128Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the control circuit in composite switches
    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01LSEMICONDUCTOR DEVICES; ELECTRIC SOLID STATE DEVICES NOT OTHERWISE PROVIDED FOR
    • H01L2924/00Indexing scheme for arrangements or methods for connecting or disconnecting semiconductor or solid-state bodies as covered by H01L24/00
    • H01L2924/0001Technical content checked by a classifier
    • H01L2924/0002Not covered by any one of groups H01L24/00, H01L24/00 and H01L2224/00

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a drive circuit for a power semiconductor element, and more particularly to a drive circuit for driving a power semiconductor element such as an IGBT, and driving the power semiconductor element for suppressing a surge voltage generated during switching. Regarding the circuit.
[0002]
[Prior art]
A method for overcurrent protection of an IGBT in a conventional power semiconductor device will be described. When an overcurrent flows through the IGBT, the gate potential of the IGBT and the reference voltage are compared by a comparator, and the comparator operates only when the IGBT gate-emitter voltage rises from the reference voltage to turn on the switch. As a result, the gate potential is clamped to the breakdown voltage of the Zener diode and the breakdown voltage of the diode, thereby prolonging the time until the IGBT is destroyed at the time of overcurrent. Note that a switch once turned off by the action of mono-multi is not turned on even if the gate-emitter voltage of the IGBT decreases again (see, for example, Patent Document 1).
[0003]
[Patent Document 1]
JP-A-4-165916 (page 2)
[0004]
An example of a driving circuit for a power semiconductor element having a protection device using a voltage increase between a gate and an emitter during an overcurrent in another conventional device will be described. In the conventional device, the input side is inserted between the gate of the power element inserted into the load circuit and the positive electrode of the DC power source for driving the gate, and the rise of the gate voltage of the power element due to the overcurrent is limited and caused by the overcurrent. This is an overcurrent protection device for a power element that includes a photocoupler that can output an isolated current on the input side, and blocks or limits the overcurrent of the power element using the output of the photocoupler as an overcurrent detection signal. The photocoupler includes an LED on the input side and a phototransistor on the output side (see, for example, Patent Document 2).
[0005]
In the conventional device having such a configuration, the LED is energized when the gate voltage rises during overcurrent, and the phototransistor outputs a current corresponding to the energization current of the LED to the control circuit as an overcurrent detection signal. The control circuit outputs a control signal so as to cut off or limit the energization of the power semiconductor element by the input of the detection signal. The LED also clamps the rise in the gate voltage Vge and suppresses the peak current of the overcurrent (see, for example, Patent Document 2).
[0006]
[Patent Document 2]
Japanese Patent No. 2674355 (page 3, FIG. 1)
[0007]
[Problems to be solved by the invention]
In the overcurrent detection method using the increase in the gate-emitter voltage represented by the above-mentioned Patent Document 1 and the above-mentioned Patent Document 2, the gate-emitter voltage needs to be increased to some extent. The current value that can be detected reaches several times the rated current. For this reason, when an off command is input from the outside in a state in which a current equal to or lower than the current that can be detected as an overcurrent flows even if the current is higher than the rated current of the power semiconductor element, it is normally cut off. For this reason, the surge voltage generated at the time of interrupting the current is increased, and the device may be destroyed. In particular, when the current rise rate di / dt is low, such as when a short circuit occurs at a location away from the power semiconductor element, it takes time for the gate-emitter voltage to reach the detection level. There is a problem that the probability that an off command will be input increases, and the possibility of destroying the device increases.
[0008]
The present invention has been made in order to solve such a problem. As in the above-described conventional example, overcurrent detection and protection interruption are performed at a current level lower than a large current that increases the gate-emitter voltage. An object of the present invention is to obtain a drive circuit for a possible power semiconductor element.
[0009]
[Means for Solving the Problems]
The present invention is a drive circuit for driving a power semiconductor element, wherein an input control signal is inputted from the outside and the power semiconductor element is turned on / off, and the input control signal is A sampling signal generating circuit for outputting a sampling signal at approximately the start time of the mirror period of the power semiconductor element when the input control signal is detected and instructing to turn off; and the power semiconductor element When the mirror voltage of the power semiconductor element is detected at the timing when the sampling signal is input and the mirror voltage is equal to or higher than a predetermined threshold, an overcurrent detection signal is output. The output gate voltage detection circuit is connected to the gate line of the power semiconductor element and receives the overcurrent detection signal from the gate voltage detection circuit. To turn off the semiconductor element for the electric power at normal slower rate than during a driving circuit of a power semiconductor device having a gate voltage control circuit for controlling the gate voltage of the semiconductor device for said power.
[0010]
DETAILED DESCRIPTION OF THE INVENTION
Embodiment 1 FIG.
In the present invention, in order to suppress the occurrence of a surge voltage, the gate voltage waveform at the time of current interruption can be adjusted by the current value flowing through the power semiconductor element. Therefore, the gate voltage (mirror voltage) of the semiconductor power element at the turn-off time can be detected in order to estimate the current value of the power semiconductor element. Further, the detection timing can be adjusted by the sampling signal generation circuit using the input control signal.
[0011]
FIG. 1 is a circuit diagram showing a configuration of a drive circuit for a power semiconductor element according to Embodiment 1 of the present invention. In FIG. 1, 100 is a main inverter circuit that performs on / off switching of a power semiconductor element, 7 is a sampling signal generating circuit that generates a sampling signal only when turning off with reference to a signal from an input control signal, and 8 Is connected to the gate line of the power semiconductor element, and based on the detection result of the gate voltage detection circuit described later, a gate voltage control circuit for controlling the gate voltage, 9 is connected to the gate line of the power semiconductor element, When a sampling signal is input, a gate voltage detection circuit 10 for detecting the gate voltage at that time is a power semiconductor element 10 to be driven by the drive circuit of the present invention. In this embodiment, an IGBT is used. An example will be described (hereinafter referred to as IGBT 10). 1 to 6 are provided in the main inverter 100, 1 is a power source, 2 is ground, 3 is a Pch MOSFET, 4 and 5 are resistors, and 6 is an Nch MOSFET. 1 shows an example in which the main inverter 100 includes the power source 1, the ground 2, the MOSFET 3, the resistors 4, 5, and the MOSFET 6, but the present invention is not limited to this case. As long as the main inverter 100 has the function of an inverter, any one may be used, and the present invention is not limited to the above configuration. If the logic is changed, the buffer configuration may be used.
[0012]
The operation will be described. First, when an input control signal is input to the main inverter 100, the sampling signal generation circuit 7 refers to the signal from the input control signal, and if it is turned off (such as near the start of the mirror period). A sampling signal is generated after a predetermined time. On the other hand, in the case of turn-on, no signal is generated. Thereby, it can be operated only at the time of turn-off. Next, when the sampling signal is input to the gate voltage detection circuit 9, the gate voltage detection circuit 9 detects the gate voltage (that is, the mirror voltage) at that time. When the gate voltage detection circuit 9 detects that an overcurrent is flowing through the IGBT 10, a predetermined overcurrent detection signal is output to the gate voltage control circuit 8 and feedback is applied. Thereby, the gate voltage is feedback-controlled only in the case of an overcurrent, and the IGBT 10 is slowly cut off (turned off). In addition, since the sampling signal generation circuit 7 generates the sampling signal at a time near the start of the mirror period, the gate voltage detection circuit 9 and the gate voltage control circuit 8 are turned off at high speed under normal conditions. Since the gate voltage is controlled so that the turn-off loss is small and the IGBT 10 can be shut off slowly during an overcurrent, the surge voltage generated at the turn-off can be reduced.
[0013]
Here, the gate waveform at turn-off is shown in FIG. 2, the collector voltage waveform is shown in FIG. 3, and the collector current waveform is shown in FIG.
[0014]
The operation at turn-off will be described in more detail with reference to FIGS. First, HIGH is input to the input of the main inverter 100 at time T1 in order to turn off. Therefore, since the switch 3 is turned off and the switch 6 is turned on, the electric charge stored in the gate-emitter capacitance of the IGBT 10 is discharged to the ground 2 through the resistor 5 and the switch 6. The discharge current at this time can be ignored because the on-resistance of the switch 6 is sufficiently low, is determined by the size of the resistor 5, and the gate voltage begins to decrease as shown in FIG.
[0015]
At time T2, a displacement current passing through the feedback capacitance existing between the gate collector of the IGBT 10 begins to flow from the gate of the IGBT 10. Therefore, since the discharge of the gate-emitter capacitance apparently stops, the gate voltage becomes a predetermined voltage V as shown in FIG. M It becomes constant at.
[0016]
As the time T3 approaches, the collector voltage of the IGBT 10 begins to increase as shown in FIG.
[0017]
After the time T3, the gate-emitter capacitance of the IGBT 10 is discharged again by the current determined by the resistor 5, so that the gate voltage starts to decrease as shown in FIG. During this time, the collector current is rapidly cut off as shown in FIG.
[0018]
By the way, the wiring of the main circuit such as an inverter including the IGBT 10 must always have a parasitic inductance L. S Exists. As a result, the surge voltage (V CP -V C = L S × dI C / Dt) occurs. Therefore, the collector voltage waveform is once peak voltage V as shown in FIG. CP After becoming steady value V C It becomes.
[0019]
Thereafter, the discharge of the gate-emitter capacitance of the IGBT 10 is completed at time T4, so that the gate voltage becomes 0V.
[0020]
Accordingly, when the interruption is performed when an overcurrent flows through the IGBT 10, a surge voltage larger than that in the normal state is generated, so that the breakdown voltage of the IGBT 10 is exceeded and the IGBT 10 is destroyed.
[0021]
In the present invention, the time from time T2 to time T3 (referred to as “mirror period” for convenience) and the gate voltage at that time (“mirror voltage V for convenience”). M ")". When the IGBT is turned off, the gate voltage must be (the predetermined voltage V M There is a fact that there is a mirror period (which is constant at the end of this period) and that the collector current begins to drop at the end of this period and that the mirror voltage increases depending on the magnitude of the collector current of the IGBT.
[0022]
Therefore, if the mirror voltage is detected and greater than a predetermined value, it is determined as an overcurrent, and the gate voltage is controlled so that it can be shut off slowly, so that the surge voltage generated at turn-off can be reduced. In this method, the turn-off is normally fast, and therefore the loss can be reduced. In addition, since the time until the mirror period is as fast as normal during an overcurrent, control is less likely to be delayed, and after the mirror period it is slowly interrupted, so that surge voltage generation is also suppressed, so that the IGBT is Can protect. Further, according to the present invention, the mirror voltage at the turn-off depending on the collector current is detected, and it is determined whether or not it is an overcurrent based on the value. That is, since the overcurrent detection level can be adjusted by the mirror voltage value, it is possible to detect a current lower than the conventional method of detecting the rise in the gate voltage as the overcurrent.
[0023]
As shown in FIG. 1, by providing a sampling signal generation circuit 7, the present invention can be operated only when it is turned off by referring to a signal from an input control signal. As a result, the turn-on loss does not increase because there is no effect at turn-on.
[0024]
The sampling signal generation circuit 7 has a function of adjusting the sampling time near the start of the mirror period. In response to the generated sampling signal, the gate voltage detection circuit 9 can detect the mirror voltage at that time.
[0025]
Feedback is applied to the gate voltage control circuit 8 only when the gate voltage detection circuit 9 detects that an overcurrent is flowing through the IGBT 10. As a result, the gate voltage is controlled only when overcurrent occurs, and the IGBT 10 is slowly shut off. Thereby, the surge voltage generated at the time of turn-off can be suppressed to be small, and the destruction of the IGBT 10 due to the surge voltage can be prevented.
[0026]
As described above, according to the present embodiment, since the sampling signal generation circuit 7, the gate voltage detection circuit 9, and the gate voltage control circuit 8 are provided, the gate voltage detection circuit 9 operates only when it is turned off. Turn-on loss can be reduced. In addition, the turn-off loss is small because the turn-off is normally performed at a high speed. In addition, by adjusting the mirror voltage detection level, the overcurrent detection level can be changed, so that overcurrent protection that suppresses the occurrence of surge voltage by detecting overcurrent even at a low current, not a large current as in the past. It can be carried out.
[0027]
Embodiment 2. FIG.
As a configuration example of the sampling signal generation circuit 7 of the first embodiment, there is a method using an ASIC. In this case, an arbitrary waveform can be created. For example, when an input control signal 90 (solid line) as shown by the solid line in FIG. 5 is input to the driver, the gate voltage 91 is as shown by the broken line in FIG. Since the above circuit of the present invention only needs to work at the time of turn-off, the ASIC may output a waveform as shown in FIG.
[0028]
Also in the present embodiment, the same effect as in the first embodiment can be obtained.
[0029]
Embodiment 3 FIG.
FIG. 7 shows an example of a sampling signal generation circuit 7 different from the above-described second embodiment. In FIG. 7, 11 is a resistor, 12 is a capacitance, 13 is a buffer, and 14 is a node. As shown in FIG. 7, the resistor 11 is provided on the input side of the input control signal, and the buffer 13 is provided on the gate voltage detection circuit 9 side. The resistor 11 and the buffer 13 are connected, and the capacitance 12 is connected between a node 14 provided therebetween and the ground. Since the resistor 11 and the capacitance 12 constitute a low-pass filter, when an input control signal as shown in FIG. 5 is input, the voltage 93 (broken line) at the node 14 has a distorted waveform as shown in FIG. A waveform with a gradual change in the change point) occurs. Here, the threshold voltage of the buffer 13 is set to half the input V DD By inputting the voltage generated at the node 14 to the buffer as / 2, a rectangular wave 92 as shown in FIG. 8 is obtained. The rectangular wave 92 becomes a signal delayed by a certain time from the input control signal 90 shown in FIG. 5, and this delay time can be adjusted by changing the value of the resistor 11 or the capacitance 12. In addition, this method is less expensive than using an ASIC. Since the delay time can be adjusted arbitrarily, the threshold voltage of the buffer 13 is strictly V. DD It is not necessary to be / 2.
[0030]
As described above, according to the present embodiment, the sampling signal generation circuit 7 of FIG. 1 is configured by the delay circuit using the resistor 11 and the capacitance 12, and the buffer 13, so that the cost can be reduced. it can. Further, the delay time can be arbitrarily adjusted by changing the value of the resistor 11 or the capacitance 12.
[0031]
Embodiment 4 FIG.
In the case of the circuit of the third embodiment, the delay time when changing from HIGH (high) to LOW (low) and the delay time when changing from LOW to HIGH are the same. Therefore, if the time at which the mirror period at the turn-on starts is sufficiently earlier than the time at which the mirror period at the turn-off starts, using the circuit of the third embodiment causes a problem of sampling even at the turn-on. .
[0032]
Therefore, an example of the sampling signal generation circuit 7 according to the fourth embodiment as shown in FIG. 9 is shown. In FIG. 9, a diode 15 is further connected to the circuit of the third embodiment shown in FIG. 7 and connected in the direction of FIG. 9. That is, the diode 15 is connected in parallel with the resistor 11 so that the input control signal direction is the forward direction. Since the diode 15 causes a current to flow in the forward direction, the discharge current flows through the diode 15 only when the charge is discharged from the capacitance 12, that is, when the input control signal changes from HIGH to LOW. Therefore, as compared with the case where the third embodiment discharges through the resistor 11, the present embodiment can discharge very quickly, so that there is almost no delay time during discharge. Thus, as shown in FIG. 10, a sampling signal is obtained that is delayed only when changing from LOW to HIGH, and synchronized with the input control signal when changing from HIGH to LOW.
[0033]
When this circuit is used, a sampling signal is generated with a certain time delay at the time of turn-off, and conversely, at the time of turn-on, it is turned off in synchronization with the input control signal, so that sampling is not performed. As a result, a more accurate sampling signal can be obtained as compared with Example 1.
[0034]
As described above, according to the present embodiment, as in the third embodiment, the sampling signal generation circuit 7 is configured by the delay circuit using the resistor 11 and the capacitance 12, and the buffer 13, so that the cost can be reduced. can do. Further, the delay time can be arbitrarily adjusted by changing the value of the resistor 11 or the capacitance 12.
[0035]
In the present embodiment, the sampling signal generation circuit 7 further includes a diode 15 connected in parallel to the resistor 11 so that the input control signal direction is the forward direction. When changing to LOW, it can be synchronized with the input control signal. Therefore, the drive circuit is reliably prevented from operating when the power semiconductor element is turned on.
[0036]
Embodiment 5 FIG.
FIG. 11 shows an example of the sampling signal generation circuit 7 according to the fifth embodiment. Instead of the buffer 13 of the above-described third and fourth embodiments, a reference circuit including a comparator 16 and resistors 17 and 18 is configured. In FIG. 11, 16 is a comparator, 17 and 18 are provided in the previous stage of the comparator 16, resistors for dividing the voltage of the power supply 1, and 20 is provided in the subsequent stage of the comparator 16 between the comparator 16 and the power supply 1. Resistance. Since other configurations are the same as those in the above embodiment, the description thereof is omitted here. When the input control signal is switched from LOW to HIGH at the time of turn-off, the capacitor 12 is charged. When the voltage at the node 14 input to the comparator 16 becomes higher than the voltage at the node 19 obtained by dividing the power supply 1 of the reference circuit by the resistors 17 and 18, the comparator 16 outputs a sampling signal. The output delay time may be adjusted by changing the value of the resistor 11 or the capacitance 12, or the reference circuit resistors 17 and 18 may be adjusted to adjust the voltage of the reference circuit.
[0037]
When using a buffer as in the third embodiment, there is a possibility that the delay time of the output of the sampling signal varies due to variations in the threshold value of the buffer. However, when a configuration using a comparator as in this embodiment is used. Can reduce variations in output and further prevent malfunction.
[0038]
As described above, according to the present embodiment, the sampling signal generation circuit 7 includes the reference circuits 17 and 18 that output a predetermined reference voltage set in advance, the delay circuit including the resistor 11 and the capacitance 12, and the delay. Since the comparator 16 (voltage comparator) detects whether the output voltage of the circuit is larger or smaller than the reference voltage, it is possible to reduce the variation in the delay time of the output of the sampling signal, and to further prevent malfunction. Can be prevented.
[0039]
Embodiment 6 FIG.
FIG. 12 shows an example of the sampling signal generation circuit 7 according to the sixth embodiment. In addition to the first circuit including the resistor 11, the capacitance 12, and the diode 15, the inverter 21 for inverting the input control signal, the second circuit including the resistor 22 and the capacitance 23, and the AND element 24 are included. Yes. The first circuit and the second circuit are connected in parallel, and their outputs are input to the AND element 24. The configuration of the first circuit is the same as that except for the buffer 13 of the fourth embodiment. In the second circuit configuration, a resistor 22 is connected in series to the inverter 21, and a capacitance 23 is provided between a node 25 provided between the resistor 22 and an input on one side of the AND element 24 and ground. Yes.
[0040]
The operation at turn-off is shown in FIG. When the input control signal is switched from LOW to HIGH at the time of turn-off, the voltage at the node 14 gradually increases and reaches the threshold voltage of the AND element 24. On the other hand, since the inverter 21 is inserted, the voltage at the node 25 gradually decreases and becomes lower than the threshold value of the AND element 24. If the time until the voltage at the node 25 falls below the threshold voltage of the AND element 24 is delayed with respect to the time until the voltage at the node 14 reaches the threshold voltage of the AND element 16, the AND element only during that time. 24 outputs a sampling signal.
[0041]
As described above, in this embodiment, the output period of the sampling signal can be determined by using such a circuit configuration. If the output period of the sampling signal is set shorter than the mirror period, it is possible to prevent malfunction due to noise when the current decreases after the mirror period. Although not shown in FIG. 1, if the detection signal when an overcurrent is detected is output to an external circuit, it operates near the end of the mirror period and the gate voltage control circuit 8 operates. Although it is delayed, the detection signal can be prevented from being output in spite of being normally cut off.
[0042]
Further, according to the present embodiment, the sampling signal generation circuit 7 is constituted by a delay circuit using the resistor 11 and the capacitance 12, and a circuit constituted by the inverter 21 and the resistor 22, so that the cost is reduced. Can be small. Further, the delay time can be arbitrarily adjusted by changing the value of the resistor 11 or the capacitance 12.
[0043]
Embodiment 7 FIG.
In the above-described sixth embodiment, the AND element 24 is directly input from the nodes 14 and 25. However, the comparator (FIG. 11) is used as in the fifth embodiment, and the output is input to the AND element 24. Also good. A configuration using a comparator for input to the AND element 24 is shown in FIG. In the configuration of FIG. 14, two circuits in which diodes are deleted from the circuit shown in FIG. 11 are provided in parallel, and the outputs of the circuits are input to the AND element 24. Note that the input signal may be inverted on one side of the input to the AND element 24 using the inverter 21 as shown in FIG. 12, but the terminal input to the comparator may be replaced with the reference side.
[0044]
As described above, when the comparator is used, the variation in the sampling output due to the variation in the threshold value of the AND element 24 can be reduced.
[0045]
As described above, according to the present embodiment, the sampling signal generation circuit 7 includes the reference circuits 17 and 18 that output a predetermined reference voltage set in advance, the delay circuit including the resistor 11 and the capacitance 12, A first circuit and a second circuit composed of a comparator 26 for detecting whether the output voltage of the delay circuit is larger or smaller than the reference voltage, and an AND to which outputs of these circuits are inputted Since it is composed of the element 24, it is possible to reduce variations in the delay time of the output of the sampling signal and prevent malfunction.
[0046]
Embodiment 8 FIG.
In the seventh embodiment, the comparators are used for both inputs of the AND element. However, as shown in FIG. 15, only the start point of the monitoring period is set using the comparator 16, and the output of the comparator 16 is set to the resistor 32 and the capacitance 33 for the end point. The delay circuit may be set. In FIG. 15, 31 is an inverter, 32 is a resistor, and 33 is a capacitor. The other configuration is the same as the configuration of FIG. 11 described above, and thus the description thereof is omitted here. As shown in FIG. 15, the former part has the same configuration as that obtained by deleting the diode from the circuit of FIG. 11, and the output obtained by inverting the output by the inverter 31 is input to one side of the AND element 24. The other input of the AND element 24 is input by delaying the output of the comparator 16 by a delay circuit including a resistor 32 and a capacitance 33.
[0047]
In the case of the circuit of FIG. 15, the end point varies depending on the variation of the AND element 24, but is smaller than the case where the comparator 16 is not used. Further, when the start point of the monitoring period changes, the period during which the gate voltage is high may be monitored and malfunction may occur. However, even if the end point slightly varies, the possibility of malfunctioning is small.
[0048]
Embodiment 9 FIG.
FIG. 16 shows another example of the sampling signal generation circuit 7 according to the ninth embodiment. The capacitor 12 used in the delay circuit shown in FIG. 15 and one end of the resistor 18 of the reference circuit are connected to the emitter power supply 34 instead of the ground. In general, even when the power supply voltage fluctuates, the voltage fluctuation between the power supply 1 and the emitter power supply 34 is small compared to the voltage fluctuation of the power supply 1 and is often kept constant. For this reason, when one end of the capacitor 12 and the resistor 18 of the reference circuit is connected to the emitter power supply, even when the power supply voltage fluctuates, the voltage fluctuation between the power supply 1 and the emitter power supply 34 is small, so that the monitoring period can be output in a certain time. it can. Therefore, even if the power supply voltage fluctuates, it is possible to prevent malfunction due to variations in the monitoring period. Although the emitter power supply is used here, a power supply with little voltage fluctuation using a regulator or the like may be used.
[0049]
Embodiment 10 FIG.
FIG. 17 shows another example of the sampling signal generation circuit 7 according to the tenth embodiment. When the power supply voltage of the drive circuit fluctuates, the operation time of the MOSFETs 3 and 6 fluctuates in the main inverter as shown in FIG. 1, and the time from when the input control signal is turned off until the gate voltage decreases may fluctuate. . The present invention corresponds to such a case. As shown in FIG. 17, the terminal of the capacitor 12 remains connected to the emitter power supply 34 as in FIG. 16, but one end of the resistor 18 of the reference circuit of the comparator 16 is connected to the ground 2 as in FIG. is doing. A Zener diode 35 is provided between the resistor 17 of the reference circuit of the comparator and the power supply 1. When the power supply voltage fluctuates, the input control signal and the emitter power supply 34 fluctuate, and the voltage at the node 14 that is the input to the comparator 16 fluctuates in the same manner as the power supply voltage. On the other hand, the voltage at the node 19 that is the input on the reference circuit side to the comparator 16 is different from the power supply voltage because one end of the resistor 18 is connected to the ground 2. Therefore, the timing for outputting the sampling signal changes when the power supply voltage fluctuates. The amount of change in timing can be adjusted by changing the voltage dividing ratio of the resistors 17 and 18 of the reference circuit and the value of the Zener diode 35. Therefore, even if the power supply voltage fluctuates, if the adjustment is made so that the output timing of the sampling signal changes to the same extent as the fluctuation of the time from the OFF time of the input control signal to the gate voltage drop when the power supply voltage fluctuates The time from the voltage drop to the output of the sampling signal can be made constant.
[0050]
In FIG. 17, if the values of the Zener diode 35 and the resistors 17 and 18 are adjusted (in some cases, the position of the Zener diode 35 is changed between the resistor 18 and the ground 2), the gate voltage starts to decrease with respect to fluctuations in the power supply voltage. It is also possible to set so that the period until detection is changed. When the power supply voltage decreases, the gate voltage during the mirror period decreases even if the current value is the same. Therefore, the period until the gate voltage reaches a certain mirror period becomes longer due to the influence of the inductance of the gate line. Therefore, if the detection period is set to be delayed when the power supply voltage is lowered, it is possible to prevent malfunction due to the influence thereof. Further, when the power supply voltage is lowered, the difference in ground potential with respect to the emitter is lowered, so that the mirror period becomes longer. Therefore, even if the detection period is delayed to the extent that it falls within the range of the mirror period when the power supply voltage drops, detection will not be delayed.
[0051]
As described above, according to the present embodiment, the output period of the sampling signal generation circuit 7 does not depend on the power supply voltage of the drive circuit, but is a fixed period after the gate voltage starts to decrease due to the off command of the input control signal. In this way, it is possible to prevent malfunctions due to variations in the monitoring period.
[0052]
In addition, since the output period of the sampling signal generation circuit 7 depends on the power supply voltage and the output period is delayed as the power supply voltage decreases, the time until the gate voltage reaches a certain mirror period when the power supply voltage is reduced is reduced. Even when the length becomes longer, it is possible to prevent malfunction.
[0053]
Embodiment 11 FIG.
In the above-described first to tenth embodiments, the period for detecting the gate voltage is set using the input control signal, but it can also be set directly from the gate voltage.
[0054]
FIG. 18 is a circuit diagram showing the configuration of the drive circuit for the power semiconductor device according to the eleventh embodiment of the present invention. In the eleventh embodiment, the sampling signal generating circuit 7 shown in the first to tenth embodiments is eliminated, and a timing detection circuit 200 is newly provided instead. Since other configurations are the same as those in FIG. 1, the description thereof is omitted here.
[0055]
The timing detection circuit 200 is a circuit that outputs a signal when the mirror period starts. As a result, the gate voltage detection circuit can be operated as soon as the mirror period starts, and the delay time until the input control signal is turned off can be further shortened. In the eleventh embodiment, since the timing can be directly generated from the gate voltage, no extra load is applied to the input control signal generation circuit.
[0056]
FIG. 19 shows a specific circuit example of the timing detection circuit 200 according to the eleventh embodiment of the present invention. In FIG. 19, 1 is a power supply, 36 is a capacitance, 37 is a node, 38 is a resistor, and 39 is a buffer. As shown in FIG. 19, a capacitance 36 is connected to the gate line, and a resistor 38 is connected between the capacitance 36 and the power supply 1. As shown in the figure, a buffer 39 is connected to a node 37 provided between the resistor 38 and the capacitance 36, and an output signal output from the buffer 39 is input to the gate voltage detection circuit 9.
[0057]
Capacitance 36 and resistor 38 constitute a high-pass filter. As a result, a pulse is generated at the node 37 only when the gate potential changes rapidly. When the input control signal becomes HIGH and the turn-off starts, the gate voltage starts to decrease, but the change at this time is more gradual than the change when entering the mirror period. Therefore, by adjusting the values of the capacitance 36 and the resistor 38, this circuit can accurately detect the mirror period.
[0058]
In the present embodiment, the resistor 38 is connected to the power source 1. This is because if the resistor 38 is connected to the ground, the potential of the node 37 takes a value equal to or lower than the ground at the time of turn-off. The output signal of the timing detection circuit 200 should be in the range from the ground to the power supply so that logic processing can be performed. In such a case, the signal cannot be processed.
[0059]
Conversely, by connecting to the power supply 1, the potential of the node 37 always has a voltage higher than the ground. In particular, due to a change in the gate voltage during the turn-off mirror period, the potential of the node 37 decreases via the capacitance 36 and takes the ground potential.
[0060]
As described above, the timing detection circuit 200 according to the present embodiment can output a signal when the mirror period starts.
[0061]
As described above, in the present embodiment, the timing detection circuit 200 is provided so that the gate voltage is detected directly from the gate voltage when the mirror period is entered. 9 can be operated, the delay time until the input control signal is turned off can be further shortened, and an extra load can be prevented from being applied to the input control signal generation circuit.
[0062]
As described above, according to the present embodiment, since the timing detection circuit 200, the gate voltage detection circuit 9, and the gate voltage control circuit 8 are provided, the turn-on loss can be reduced because the circuit operates only at the time of turn-off. In addition, the turn-off loss is small because the turn-off is normally performed at a high speed. In addition, by adjusting the mirror voltage detection level, the overcurrent detection level can be changed, so that overcurrent protection that suppresses the occurrence of surge voltage by detecting overcurrent even at a low current, not a large current as in the past. It can be carried out. In addition, the gate voltage detection circuit can be operated immediately after entering the mirror period, and the delay time until the turn-off from the input control signal can be further shortened. In addition, since the timing can be made directly from the gate voltage, no extra load is applied to the input control signal generation circuit.
[0063]
Furthermore, as shown in FIG. 19, the timing detection circuit 200 includes a delay circuit using a resistor 38 and a capacitance 36, and a buffer 39, so that the cost can be reduced. Further, the delay time can be arbitrarily adjusted by changing the value of the resistor 38 or the capacitance 36.
[0064]
Embodiment 12 FIG.
The gate voltage detection circuit 9 is a circuit that measures the gate voltage of the IGBT 10. The gate voltage changes from the ground 2 to the power source 1 to analog. FIG. 20 shows an example of the gate voltage detection circuit 9 according to the twelfth embodiment. In FIG. 20, reference numeral 40 denotes a voltage amplifier connected to the gate line of the IGBT 10, and 41 denotes a switch that switches based on the sampling signal output from the sampling signal generation circuit 7 or the timing detection circuit 200. In this example, the gate voltage is changed and amplified to an arbitrary level by the voltage amplifier 40. Whether or not the output signal from the voltage amplifier 40 is output by the switch 41 controlled by the sampling signal generated by the sampling signal generation circuit 7 or the timing detection circuit 200 is determined. In this example, since the gate voltage value is detected and output as an analog value, it can be applied to variable control.
[0065]
As described above, according to the present embodiment, the gate voltage detection circuit 9 is connected to the gate line of the power semiconductor element and is connected to the voltage amplifier 40 that amplifies the gate voltage to a predetermined level. And a switch 41 that switches based on a signal output from the timing detection circuit 200, the gate voltage detection circuit 9 includes a switch 41 that switches according to a signal generated from the voltage amplifier 40 and the sampling signal generation circuit 7. Since the detected information is output as an analog value, it can be applied to variable control.
[0066]
Embodiment 13 FIG.
FIG. 21 shows an example of the gate voltage detection circuit 9 according to the thirteenth embodiment. In FIG. 21, 41 is a switch that switches based on the sampling signal output from the sampling signal generation circuit 7 or the timing detection circuit 200, 42 is connected to the gate line of the IGBT 10, and the gate voltage is from a reference voltage generation adjustment circuit 43 to be described later. A voltage comparator 43 for detecting whether the reference voltage is LOW or HIGH is a reference voltage generation adjusting circuit for outputting a predetermined reference voltage set in advance. In this example, the voltage comparator 42 determines whether the gate voltage is greater than the reference voltage output from the reference voltage generation adjustment circuit 43, and outputs the output as a binary value of LOW or HIGH. Whether the output signal from the voltage comparator 42 is output by the switch 41 controlled by the sampling signal generated by the sampling signal generation circuit 7 or the timing detection circuit 200 is determined.
[0067]
As described above, the mirror voltage is uniquely determined by the current value. Therefore, if the reference voltage is set to a predetermined mirror voltage value, the output of the voltage comparator 42 can be switched when more current flows. it can.
[0068]
In this example, since the gate voltage value is detected and output as a digital value, it cannot be applied to variable control, but a circuit that is resistant to noise and hardly malfunctions can be configured.
[0069]
Embodiment 14 FIG.
FIG. 22 shows an example of the gate voltage detection circuit 9 according to the fourteenth embodiment. In FIG. 22, 1 is a power source, 2 is ground, 44 and 49 are Zener diodes, 45 and 50 are diodes, 46, 48, 51 and 55 are resistors, 47 is a MOS transistor, 52 is a bipolar transistor, 53 is a capacitance, 54 Is a buffer.
[0070]
The Zener diode 49, the resistor 48, and the MOS transistor 47 are circuits that determine whether or not to operate the detection circuit. As shown in the figure, a Zener diode 49 is connected to the sampling signal generation circuit 7 or the timing detection circuit 200, and a MOS transistor 47 is connected to the other end of the Zener diode 49. A resistor 48 is connected between a node provided between the Zener diode 49 and the MOS transistor 47 and the ground 2. When a voltage greater than the sum of the breakdown voltage of the Zener diode 49 and the threshold voltage of the MOS transistor 47 is input from the sampling signal, the detection circuit is turned on. Here, the voltage when turning on can be controlled by inserting the Zener diode 49. The resistor 48 is for discharging electric charge between the Zener diode 49 and the MOS transistor 47. Therefore, when a Zener diode is not used, these may be omitted.
[0071]
The resistor 55, the capacitance 53, and the buffer 54 are circuits for outputting detected signals. As shown in the figure, a buffer 54 is connected to the gate voltage control circuit 8, and a resistor 55 is connected between the buffer 54 and the power supply 1. A capacitance 53 is connected between a node provided between the buffer 54 and the resistor 55 and the ground. Here, since the voltage of the capacitance 53 is connected to the power source 1 through the resistor 55, it is normally HIGH. Here, the buffer 54 receives the capacitance signal and amplifies the current or voltage. Therefore, when an inverter is used instead of the buffer 54, the logic is inverted, but there is no problem because it is sufficient to construct a circuit based on such a logic configuration from the beginning.
[0072]
The zener diode 44, the diodes 45 and 50, the resistors 46 and 51, and the bipolar transistor 52 are circuits for detecting the gate voltage. As shown in the figure, a Zener diode 44 is connected to the gate line of the IGBT 10, a diode 45 is connected to the other end of the Zener diode 44, and a resistor 46 is connected to the other end of the diode 45. The aforementioned MOS transistor 47 is connected to the resistor 46. Between the node provided between the diode 45 and the resistor 46 and the node provided between the buffer 54 and the resistor 55, the diode 50, A resistor 51 and a bipolar transistor 52 are connected.
[0073]
The operation method of this embodiment will be described below. First, when the MOS transistor 47 is off, no voltage is generated between the terminals of the resistor 46. Therefore, since the bipolar transistor 52 is not turned ON, the charge of the capacitance 53 is not discharged and remains HIGH. As a result, the output signal is HIGH.
[0074]
In contrast, it is assumed that the MOS transistor 47 is turned on in response to the sampling signal. Then, if the on-resistance of the MOS transistor 47 is sufficiently low, one terminal of the resistor 46 may be considered as a ground potential. Therefore, the voltage applied to the resistor 46 is a voltage obtained by subtracting the breakdown voltage of the Zener diode 44 and the forward ON voltage of the diode 45 from the gate voltage.
[0075]
When the gate voltage is sufficiently larger than this, the voltage applied to the resistor 46 is sufficiently large, and the bipolar transistor 52 is turned on. Accordingly, since the voltage of the capacitance 53 is discharged to the ground through the bipolar transistor 52, the output signal becomes LOW.
[0076]
On the other hand, when the gate voltage is sufficiently low, no voltage is generated in the resistor. Accordingly, since the bipolar transistor 52 remains in the OFF state, the charge of the capacitance 53 is not discharged and the output signal remains HIGH.
[0077]
That is, the breakdown voltage of the Zener diode 44 is selected so that a voltage is generated in the resistor 46 at the mirror voltage at the current value to be determined as an overcurrent. Further, the sample signal generation circuit 7 or the timing detection circuit 200 is adjusted so that the gate voltage during the mirror period, that is, the mirror voltage can be detected. From these, in this example, it can be determined whether there is an overcurrent.
[0078]
Note that the collector current value of the bipolar transistor 52 in the ON state can be changed depending on the magnitude of the bias current, but this can be changed by adjusting the resistors 51 and 46.
[0079]
The diode 50 is inserted so that no current flows in the reverse direction, but may be omitted.
[0080]
The diode 45 corrects the temperature characteristic of the Zener diode 44 and may not be used when there is no temperature change.
[0081]
Further, since the buffer 54 is inserted to facilitate the driving capability and logic synthesis, the buffer 54 may be omitted.
[0082]
Embodiment 15 FIG.
FIG. 23 shows a specific circuit example of the gate voltage detection circuit 9 using a comparator as the voltage comparator 42 as shown in the thirteenth embodiment. In the figure, 56 is a comparator that compares and outputs the divided value of the gate voltage and the input from the reference voltage generation adjustment circuit 43, 57 and 58 are resistors that divide the power supply 1 and constitute the reference voltage generation adjustment circuit 43, 59 and 60 are resistors for dividing the gate voltage, 47 is connected in parallel with the resistor 60 and is operated by a sampling signal from the sampling signal generation circuit 7 or the timing detection circuit 200, 61 is a capacitor as a filter, and 62 is a power supply Reference numeral 1 denotes a resistor connected between 1 and the output of the comparator 56, and 63 denotes a capacitor as a filter.
[0083]
In the steady ON state, an ON command is input to the MOS transistor 47 from the sampling signal generation circuit 7 or the timing detection circuit 200, and the resistor 60 is short-circuited. In this embodiment, the sampling signal is directly input to the MOS transistor 47. However, as shown in the fourteenth embodiment, a Zener diode 49 and a resistor 48 may be used. Since the resistor 60 is short-circuited, the input on the gate voltage side to the comparator 56 is almost at the ground potential and is not detected because it is lower than the voltage input from the reference voltage generation adjustment circuit 43. In the mirror period during the off operation, the MOS transistor 47 is turned off by the sampling signal generation circuit 7 or the timing detection circuit 200, and the MOS transistor 47 is turned off. Thereby, the value obtained by dividing the gate voltage by the resistors 59 and 60 is input to the comparator 56. When the value is higher than the input voltage from the reference voltage generation adjustment circuit 43, the comparator 56 operates to output a detection signal. The output of the comparator 56 is connected to the power source 1 through the resistor 62, and HIGH or LOW is output by the operation of the comparator 56. Depending on the connection method on the input side of the comparator 56, LOW is output during an overcurrent, and HIGH is output during normal operation. It can be set to either output high or output LOW during an overcurrent, and is determined by the state of the gate voltage setting circuit that receives the output signal.
[0084]
Therefore, if the configuration as in this embodiment is used, an input obtained by dividing the gate voltage of the comparator 56 is supplied from the reference voltage generation adjustment circuit 43 at the mirror voltage at the current value to be determined as an overcurrent. If the resistors 57 to 60 are adjusted so as to be higher than the input, it is possible to determine whether there is an overcurrent and to output it.
[0085]
As described above, according to the present embodiment, the gate voltage detection circuit 9 is connected to the reference voltage generation adjustment circuit 43 that outputs a predetermined reference voltage set in advance, and the gate line of the power semiconductor element, A comparator 56 that detects whether the gate voltage of the power semiconductor element is larger or smaller than the reference voltage, and a switch 47 that is connected to the comparator 56 and switches based on a signal output from the timing detection circuit 200. Therefore, the gate voltage detection circuit 9 is composed of a comparator, a circuit for generating a reference voltage, and a switch that is switched by a signal generated from the sampling signal generation circuit, and the detected information is output as a digital value. Therefore, a circuit that is resistant to noise and hardly malfunctions can be configured.
[0086]
Embodiment 16 FIG.
FIG. 24 shows a specific circuit example of the gate voltage detection circuit 9 according to another form different from the fifteenth embodiment. In the present embodiment, the reference voltage generation adjusting circuit 43 and the gate voltage are compared using the bipolar transistor 64 as the voltage comparator 42 shown in the thirteenth embodiment. In the figure, 64 is a bipolar transistor, 65 is a diode for preventing reverse current flow, 66 is a resistor connected between the output of the reference voltage generation adjustment circuit 43 and the emitter of the bipolar transistor 64, and 67 is a reference voltage generation adjustment circuit 43. The resistor 68 connected between the output of the transistor and the base of the bipolar transistor 64 is connected to the bipolar transistor at one end, and connected to the MOS transistor 47, resistor 60, and capacitance 61 at the other end. Other configurations are the same as those in FIG. 23, and thus description thereof is omitted here.
[0087]
In the steady ON state, an ON command is input to the MOS transistor 47 from the sampling signal generation circuit 7 or the timing detection circuit 200, and the output signal is almost at ground potential. An off command is input to the MOS transistor 47 during the off mirror period, and the MOS transistor 47 is turned off. When the gate voltage during the mirror period is higher than the voltage of the reference voltage generation adjustment circuit 43, a current flows through the resistor 66, a voltage is generated between the base and the emitter of the bipolar transistor 64, and the bipolar transistor 64 becomes conductive. Since a current flows through the resistors 68 and 60, a voltage is generated in the resistor 60, and the voltage is output as an output signal.
[0088]
Therefore, even in the configuration as in the present embodiment, the input voltage of the gate is higher than the input from the voltage of the reference voltage generation adjustment circuit 43 at the mirror voltage at the current value to be determined as an overcurrent. If the resistors 57 and 58 are adjusted, it is possible to determine whether there is an overcurrent and to output it.
[0089]
Embodiment 17. FIG.
When the gate voltage detection circuit as shown in the above fourteenth to sixteenth embodiments is used, when the power supply voltage for driving the IGBT 10 varies, the gate voltage varies, and the current level for detecting overcurrent may vary. is there. FIG. 25 shows a simplified circuit configuration when turning off the IGBT 10. The IGBT 10 may include a balance resistor 69 inside. In some cases, an emitter power source 70 may be used separately from the power source for driving the gate to the emitter of the IGBT 10. The figure shows such a case. The gate voltage Vg viewed from the ground potential during the mirror period output to the gate voltage detection circuit 8 is the emitter voltage Ve when viewed from the ground potential, the gate-emitter voltage Vgem during the mirror period at the OFF time determined by the current, and the balance. The resistance R1 and the gate resistance R2 at the time of OFF are given as follows.
[0090]
Vg = (Ve + Vgem) R2 / (R1 + R2) (1)
[0091]
Vgem is determined by the current and does not change with the power supply voltage. However, when the voltage Ve of the emitter power supply 70 changes due to power supply voltage fluctuation, the gate voltage Vg changes. Therefore, the current level for detecting overcurrent changes.
[0092]
Therefore, when the reference voltage generation adjusting circuit 43 is used as shown in the fifteenth and sixteenth embodiments, even if the current value is the same, the reference voltage generation in which the output voltage fluctuates as the gate voltage fluctuates due to fluctuations in the power supply voltage. By using the adjustment circuit 43, it is possible to detect with the same current value even when the power supply voltage fluctuates.
[0093]
FIG. 26 shows a drive circuit for a power semiconductor device according to the seventeenth embodiment of the present invention. A specific configuration of the overcurrent detection circuit is shown. In addition to the reference voltage generation adjustment circuit 43 shown in the fifteenth and sixteenth embodiments, a Zener diode 71 is inserted between the power source 1 and the resistor 57 of the reference voltage generation adjustment circuit 43. From the equation (1), the emitter voltage Ve is generally given by Ve = Vcc-Vge from the power supply voltage Vcc and the gate-emitter voltage Vge in a steady state. Therefore, equation (1) is as follows.
[0094]
Vg = (Vcc-(Vge-Vgem)) R2 / (R1 + R2) (2)
[0095]
As a result, if the voltage of the Zener diode 71 is Vge−Vgem and the resistance ratio of the resistors 57 and 58 of the reference voltage generation adjustment circuit 43 is approximately the same as the resistance ratio of the balance resistor 69 and the gate resistor 5 in the off state, the desired current is set. The voltage of 43 of the reference voltage generation circuit, which is the same as the gate voltage in value, can be obtained, and by comparing it with a voltage comparator, even when the power supply voltage fluctuates, it can be detected with a constant current. Therefore, even when the power supply voltage fluctuates, it is possible to prevent malfunctions such as not detecting with the current value to be detected but detecting with the current value or less to be detected. When the gate voltage is further divided and compared as in the fifteenth embodiment, the resistance of the reference voltage generation adjustment circuit is set so that the voltage division ratio is the same as the value divided by the voltage division ratio in (2). do it.
[0096]
As described above, according to the present embodiment, even when the mirror voltage fluctuates due to the fluctuation of the power supply voltage of the drive circuit in the gate voltage detection circuit, the detection level for the overcurrent does not change, and a current exceeding a certain level. Since an overcurrent detection signal is output when the current flows, it is possible to prevent malfunctions such as not detecting at the current value to be detected even when the power supply voltage fluctuates, or detecting below the current value to be detected. .
[0097]
Embodiment 18 FIG.
FIG. 27 shows a specific example of the overcurrent detection circuit of the power semiconductor element drive circuit according to the eighteenth embodiment. In the fourteenth embodiment, temperature correction using a diode is described, but here, a more specific method will be described by taking the circuit shown in the sixteenth embodiment as an example. In FIG. 27, in addition to the configuration of the sixteenth embodiment (FIG. 24), a Zener diode 71 and diodes 74, 73, 72 are inserted between the resistor 57 in the reference voltage generation adjustment circuit 43 and the power supply 1. is doing. Other configurations are the same as those in FIG. The Zener diode 71 prevents the malfunction when the power supply voltage varies as described in the seventeenth embodiment. In general, the on-voltage of an element such as a Zener diode, a diode, or a bipolar transistor changes with temperature. Therefore, when an overcurrent detection circuit is configured using these elements, the current level for detecting overcurrent may change. For this reason, when an element having the opposite temperature characteristic is used, or when the voltage comparator 42 is used, the overcurrent detection level depends on the temperature by using an element having the same temperature characteristic for the gate voltage input side and the reference voltage generation adjustment circuit 43. The change can be corrected. In the circuit of FIG. 27, a diode 65 is used for the input from the gate voltage. Therefore, if the diode 72 is inserted into the reference voltage generation adjustment circuit 43, the characteristic change due to temperature can be canceled. Further, since the bipolar transistor 64 is used, the characteristic change due to temperature can be canceled by inserting the diode 73 having the same temperature characteristic into the reference voltage generation adjusting circuit 43. Further, the Zener diode 71 used in the reference voltage generation adjustment circuit 43 has a voltage variation depending on the temperature depending on the voltage used. Voltage fluctuation due to temperature may or may not increase when the temperature rises. When the voltage rises, a diode 74 that falls due to temperature rise as shown in FIG. . On the contrary, if the voltage drops, a diode having the same temperature characteristic may be inserted in series with the diode 65 on the gate voltage input side. One diode may be inserted, or a plurality of diodes may be used so that the temperature characteristics are comparable.
[0098]
Here, for explanation, one diode is used for one element. However, this is not always necessary, and it is sufficient that the temperature characteristics are the same as a whole. Further, when a diode is used, the voltage input to the bipolar transistor 64 changes by the on-voltage, so that the Zener diode 71 needs to be adjusted. Although the case where the bipolar transistor 64 is used has been described here, the temperature can be similarly corrected in other systems.
[0099]
Since the correction for the temperature is performed as described above, it is possible to prevent malfunctions such as not detecting the current value to be detected even when the temperature is changed, and detecting the current value to be detected or less.
[0100]
Also, even if the mirror voltage fluctuates due to fluctuations in the power supply voltage of the drive circuit in the gate voltage detection circuit, the detection level for overcurrent does not change, and an overcurrent detection signal is output when a certain current or more flows. Since the output is made, it is possible to prevent malfunctions such as not detecting the current value to be detected even when the power supply voltage fluctuates, or detecting the current value to be detected or less.
[0101]
Even when the temperature of the drive circuit of the gate voltage detection circuit fluctuates, the detection level for overcurrent does not change, and an overcurrent detection signal is output when a current exceeding a certain level flows. It is possible to prevent malfunctions such as not detecting with a current value to be detected even when the value of is changed, or detecting with a current value less than a desired value.
[0102]
Embodiment 19. FIG.
The gate voltage control circuit is a circuit for adjusting the gate voltage at the time of current interruption. FIG. 28 is a circuit diagram showing a specific example of the gate voltage control circuit 8 of the first and eleventh embodiments. In FIG. 28, 1 is a power supply, 2 is ground, 3 is a Pch MOSFET, 4 and 5 are resistors, 6 is an Nch MOSFET, 10 is an IGBT, 75 is a MOS transistor, 76 is a Zener diode, and 77 is a resistor. In this example, the gate voltage control circuit 8 includes a MOS transistor 75, a Zener diode 76, and a resistor 77. In addition, the structure of the codes | symbols 1-6 comprises the main inverter 100 as above-mentioned. As shown in the figure, the MOS transistor 75 is connected to the gate voltage detection circuit 9 and outputs HIGH during normal operation and LOW during overcurrent based on a detection signal from the gate voltage detection circuit 9. A Zener diode 76 is connected to the MOS transistor 75 as shown in the figure. The resistor 77 is connected to the Zener diode 76 and the gate line of the IGBT 10. At the time of turn-off, since the input control signal is HIGH, the MOS transistor 3 is turned off and the MOS transistor 6 is turned on, so that the gate charge of the IGBT 10 is discharged through the resistor 5 and the IGBT 10 is turned off. Therefore, in order to suppress the surge voltage that occurs when the power supply is suddenly turned off when overcurrent flows, for example, as much charge as the amount of charge discharged from the gate or more charge flows into the gate. The gate voltage does not drop rapidly, and the generation of surge voltage can be suppressed.
[0103]
Here, since the MOS transistor 75 is Pch, the output signal from the gate voltage detection circuit 9 is set to HIGH when normal, and is set to LOW when an overcurrent flows. In this way, since the MOS transistor 75 is turned on only during an overcurrent, a voltage obtained by subtracting the breakdown voltage and the gate voltage of the Zener diode 76 from the power source 1 is applied to the resistor 77. As a result, a charging current flows into the gate of the IGBT 10 via the resistor 77. Therefore, the gate voltage does not drop rapidly, and the generation of surge voltage can be suppressed.
[0104]
Here, when the charging current flowing through the resistor 77 is set to be smaller than the discharging current flowing through the resistor 5, the gate potential gradually decreases, so that the IGBT is slowly cut off.
[0105]
On the other hand, when it is set large, the gate potential is clamped to a certain potential determined by the breakdown voltage of the Zener diode 76, and no turn-off is performed. In this case, in the present embodiment, the output signal of the gate voltage detection circuit is always turned ON and OFF by the sample signal, so that the output signal of the gate voltage detection circuit 9 becomes HIGH until the next turn-off, and is restored to the normal state. ing. Therefore, the next turn-off operation is restored to the detectable state.
[0106]
If Nch is used in reverse for the MOS transistor 75, the same effect can be obtained simply by inverting the logic to the gate.
[0107]
As described above, according to the present embodiment, the gate voltage control circuit 8 is connected to the gate voltage detection circuit 9 and switched based on the signal from the gate voltage detection circuit 9, and the power semiconductor element. Since the Zener diode 76 connected between the gate line and the MOS transistor 75 is provided, the gate voltage control circuit 8 is switched by the output from the gate voltage detection circuit 9, a Zener diode, Since the resistor can be set arbitrarily, the gate voltage can be lowered slowly or clamped to a certain value. Therefore, since it can be turned off slowly, the generation of surge voltage can be suppressed. Further, since the delay time is the same as the normal turn-off from the time when the input control signal becomes HIGH until the mirror period, the control circuit for generating the input control signal only needs to be designed in consideration of the delay time, so that the design is simple. become.
[0108]
Embodiment 20. FIG.
FIG. 29 is a circuit diagram showing a specific example of the gate voltage control circuit 8 different from the nineteenth embodiment. In FIG. 29, 1 is a power supply, 2 is ground, 3 is a Pch MOSFET, 4 and 5 are resistors, 6 is an Nch MOSFET, 10 is an IGBT, 78 is a resistor, and 79 is a MOS transistor. In this example, the gate voltage control circuit 8 includes a MOS transistor 79 and a resistor 78. In addition, the structure of the codes | symbols 1-6 comprises the main inverter 100 as above-mentioned. As shown in the figure, the MOS transistor 79 is connected to the gate voltage detection circuit 9. A resistor 78 is connected to the gate lines of the MOS transistor 79 and the IGBT 10 as shown in the figure.
[0109]
In the present embodiment, the MOS transistor 79 is normally in a normal state. In this case, the off resistance of the main inverter is a combined resistance of the resistor 5 and the resistor 78. Further, since the Nch MOS transistor 79 becomes conductive, the output signal of the gate voltage detection circuit 9 outputs HIGH.
[0110]
When an overcurrent flows, the output signal of the gate voltage detection circuit 9 outputs LOW, so that the MOS transistor 79 is turned off. Accordingly, the off resistance of the main inverter 100 is only the resistor 5, and the resistance value is larger than that in the normal state. As a result, the discharge current from the IGBT 10 becomes smaller than normal, so the IGBT 10 is slowly cut off, and the generation of surge voltage is suppressed.
[0111]
FIG. 30 shows an IGBT turn-off waveform when no control is performed. As shown in FIG. 30, the input control signal becomes HIGH and the gate voltage starts to decrease. The IGBT collector-emitter voltage suddenly rises near the end of the mirror period. Also, since the IGBT starts to shut down, the collector current starts to decrease rapidly. A surge voltage much larger than this occurs, and for example, 439 V is generated in the case of FIG.
[0112]
On the other hand, FIG. 31 shows a turn-off waveform of the IGBT when the gate voltage of the IGBT is controlled using the circuits shown in FIGS. As shown in FIG. 31, a sampling signal is generated with a delay from the input control signal, and the voltage in the mirror period can be detected. Since an overcurrent is detected here, the MOS transistor 54 is turned off. Therefore, the off resistance of the main inverter 100 is only the resistor 5, and the discharge current starts to be suppressed as compared with the normal time. As a result, the gate voltage increases once and then slowly decreases. From FIG. 31, when the circuit of the twentieth embodiment is used, the generated surge voltage is small and is about 138V. That is, the surge voltage is suppressed to about 31%.
[0113]
Further, as another feature of this example other than the above effect, as shown in FIG. 31, the delay time is the same as the normal turn-off period from when the input control signal becomes HIGH until the mirror period. Therefore, since the control circuit for generating the input control signal only needs to be designed in consideration of the delay time, the design is simplified.
[0114]
In this example, only one set of the gate voltage detection circuit, the resistor 78, and the MOS transistor 79 is used. Of course, a number of pairs of the gate voltage detection circuit 9, the resistor 78, and the MOS transistor 79 are arranged in parallel to form a multi-stage. It is also possible to change the gate voltage by controlling the discharge current. In this case, finer control can be performed.
[0115]
As described above, in the present embodiment, the sampling signal generation circuit 7 refers to the signal from the input control signal and outputs the sampling signal only when it is turned off, so that it operates only when it is turned off. The turn-on loss can be reduced. Further, since the sampling signal generation circuit 7 generates the sampling signal at a time near the start of the mirror period, the gate voltage control circuit 8 can detect the mirror voltage, and the mirror voltage is equal to or higher than a predetermined threshold value. In such a case, it is determined that an overcurrent is flowing in the IGBT 10, and the turn-off loss is small because the turn-off is normally performed at a high speed. At the time of overcurrent, the gate voltage is controlled by the gate voltage control circuit 8 so that the IGBT 10 can be shut off slowly. Therefore, the surge voltage generated at turn-off can be reduced, and overcurrent protection can be implemented.
[0116]
The gate voltage control circuit is connected to the gate voltage detection circuit, and is switched between a MOS transistor that switches based on a signal from the gate voltage detection circuit, and a gate line of the power semiconductor element and the MOS transistor. Since the output from the resistor is connected to the off-side switch of the main inverter, the gate voltage can be lowered slowly. Therefore, since it can be turned off slowly, the generation of surge voltage can be suppressed. Further, since only the discharge current from the gate of the power semiconductor element is suppressed, the through current does not flow through the circuit for switching on / off of the power semiconductor element, and the power consumption does not increase so much. Further, since the delay time is the same as the normal turn-off from the time when the input control signal becomes HIGH until the mirror period, the control circuit for generating the input control signal only needs to be designed in consideration of the delay time, so that the design is simple. become.
[0117]
Embodiment 21. FIG.
In the above embodiment, a protection method for suppressing a surge generated at the time of interruption is described, but this method does not have a means for detecting an overcurrent when a large overcurrent such as a short-circuit current continues to flow in an on state. Therefore, the power semiconductor element cannot be protected. In order to protect the overcurrent in the on state, it is necessary to combine with other detection circuits.
[0118]
FIG. 32 shows an example in combination with another detection circuit. In the figure, reference numeral 80 denotes an overcurrent detection circuit that detects an overcurrent in the on state, and 81 is a control circuit that outputs an input control signal in response to an input command from the outside. As the overcurrent detection circuit 80, for example, a detection method using an increase in gate voltage as shown in the prior art is used. When an overcurrent flows in the ON state, the overcurrent detection circuit 80 outputs a detection signal to the control circuit 81, and the control circuit 81 receives the output from the overcurrent detection circuit 80 and shuts down the IGBT 10. Therefore, no overcurrent flows for a long time, and the device can be prevented from being destroyed. At the same time, the detection signal of the overcurrent detection circuit 80 is output to the gate voltage control circuit 8, and the gate voltage control circuit 8 cuts off at a slower speed than usual. Thereby, surge can be suppressed and destruction of the apparatus can be prevented.
[0119]
Further, even when an overcurrent occurs, the gate voltage detection circuit 9 determines that the overcurrent is detected when a cutoff command is input when the detection level is below the detection level of the overcurrent detection circuit 80, and the gate voltage detection circuit 9 The gate voltage control circuit 8 is cut off at a slower speed than usual by the detection signal from. Therefore, surge can be suppressed and destruction of the device can be prevented.
[0120]
Therefore, with this configuration, even when a large overcurrent such as a short-circuit current flows in the ON state, it is possible to protect even a relatively low overcurrent that the overcurrent detection circuit 80 cannot detect. it can.
[0121]
A driving circuit for driving the power semiconductor element, wherein an input control signal is input from the outside and the power semiconductor element is turned on / off; and the input control signal is detected. When the input control signal is an instruction to turn off, the sampling signal generating circuit for outputting the sampling signal at approximately the start time of the mirror period of the power semiconductor element or the gate line of the power semiconductor element Connected to the timing detection circuit for detecting the mirror period of the power semiconductor element and outputting a timing signal at approximately the start time of the mirror period; and connected to the gate line of the power semiconductor element and the sampling The mirror voltage of the power semiconductor element is detected at a timing when a signal or a timing signal is input, and the mirror A gate voltage detection circuit that outputs an overcurrent detection signal when the pressure exceeds a predetermined threshold, and detects the gate voltage in the on state, detects the overcurrent from the rise in the gate voltage, and turns on An overcurrent detection circuit that outputs an overcurrent detection signal, and an overcurrent detection signal from the gate voltage detection circuit or overcurrent detection circuit connected to the gate line of the power semiconductor element, or the on-time overcurrent detection signal. And a gate voltage control circuit for controlling the gate voltage of the power semiconductor element so that the power semiconductor element is turned off at a slower speed than usual. Even when an overcurrent flows, it is possible to protect even a relatively low overcurrent that cannot be detected by the overcurrent detection circuit.
[0122]
【The invention's effect】
The present invention is a drive circuit for driving a power semiconductor element, wherein an input control signal is inputted from the outside and the power semiconductor element is turned on / off, and the input control signal is A sampling signal generating circuit for outputting a sampling signal at approximately the start time of the mirror period of the power semiconductor element when the input control signal is detected and instructing to turn off; and the power semiconductor element When the mirror voltage of the power semiconductor element is detected at the timing when the sampling signal is input and the mirror voltage is equal to or higher than a predetermined threshold, an overcurrent detection signal is output. The output gate voltage detection circuit is connected to the gate line of the power semiconductor element and receives the overcurrent detection signal from the gate voltage detection circuit. Since the power semiconductor element drive circuit includes a gate voltage control circuit that controls the gate voltage of the power semiconductor element so that the power semiconductor element is turned off at a slower speed than normal, the sampling signal Due to the operation of the generation circuit, the gate voltage detection circuit and the gate voltage control circuit are operated only when they are turned off, so that the turn-on loss can be reduced. In addition, the turn-off loss is small because the turn-off is normally performed at a high speed. In addition, by adjusting the mirror voltage detection level, the overcurrent detection level can be changed, so that overcurrent protection that suppresses the occurrence of surge voltage by detecting overcurrent even at a low current, not a large current as in the past. It can be carried out.
[Brief description of the drawings]
FIG. 1 is a configuration diagram showing a configuration of a drive circuit for a power semiconductor element according to a first embodiment of the present invention.
FIG. 2 is an explanatory diagram showing a waveform of a gate voltage when the IGBT is turned off.
FIG. 3 is an explanatory diagram showing a waveform of a collector voltage when the IGBT is turned off.
FIG. 4 is an explanatory diagram showing a waveform of a collector current when the IGBT is turned off.
FIG. 5 is an explanatory diagram showing waveforms of an input control signal and a gate voltage in Embodiment 2 of the present invention.
FIG. 6 is an explanatory diagram showing sampling waveforms according to the second embodiment of the present invention.
FIG. 7 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a third embodiment of the present invention.
FIG. 8 is an explanatory diagram showing sampling waveforms according to the third embodiment of the present invention.
FIG. 9 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a fourth embodiment of the present invention.
FIG. 10 is an explanatory diagram showing sampling waveforms according to the fourth embodiment of the present invention.
FIG. 11 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a fifth embodiment of the present invention.
FIG. 12 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a sixth embodiment of the present invention.
FIG. 13 is an explanatory diagram showing sampling waveforms according to the sixth embodiment of the present invention.
FIG. 14 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a seventh embodiment of the present invention.
FIG. 15 is a configuration diagram showing a configuration of a sampling signal generation circuit according to an eighth embodiment of the present invention.
FIG. 16 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a ninth embodiment of the present invention.
FIG. 17 is a configuration diagram showing a configuration of a sampling signal generation circuit according to a tenth embodiment of the present invention.
FIG. 18 is a configuration diagram showing a configuration of a drive circuit for a power semiconductor element according to an eleventh embodiment of the present invention.
FIG. 19 is a configuration diagram showing a configuration of a timing detection circuit according to an eleventh embodiment of the present invention.
FIG. 20 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a twelfth embodiment of the present invention.
FIG. 21 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a thirteenth embodiment of the present invention.
FIG. 22 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a fourteenth embodiment of the present invention.
FIG. 23 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a fifteenth embodiment of the present invention.
FIG. 24 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a sixteenth embodiment of the present invention.
FIG. 25 is a configuration diagram showing a simplified circuit configuration when turning off an IGBT;
FIG. 26 is a configuration diagram showing a configuration of a gate voltage detection circuit according to a seventeenth embodiment of the present invention.
FIG. 27 is a configuration diagram showing a configuration of a gate voltage detection circuit according to an eighteenth embodiment of the present invention.
FIG. 28 is a configuration diagram showing a configuration of a gate voltage control circuit according to a nineteenth embodiment of the present invention.
FIG. 29 is a configuration diagram showing a configuration of a gate voltage control circuit according to a twentieth embodiment of the present invention.
FIG. 30 is an explanatory diagram showing an input control signal waveform, a collector-emitter voltage waveform, a collector current waveform, and a gate voltage waveform when the IGBT is turned off without control.
FIG. 31 is an explanatory diagram showing an input control signal waveform, a sampling signal waveform, a collector-emitter voltage waveform, a collector current waveform, and a gate voltage waveform when the IGBT is turned off without control. .
32 is a configuration diagram showing a configuration of a drive circuit for a power semiconductor device according to a twenty-first embodiment of the present invention. FIG.
[Explanation of symbols]
1 power supply, 2 ground, 3 MOSFET, 4 resistance, 5 resistance, 6 MOSFET, 7 sampling signal generation circuit, 8 gate voltage control circuit, 9 gate voltage detection circuit, 10 IGBT (power semiconductor device), 11 resistance, 12 capacitance , 13 buffer, 14 node, 15 diode, 16 comparator, 17 resistor, 18 resistor, 19 node, 20 resistor, 21 inverter, 22 resistor, 23 capacitance, 24 AND element, 25 node, 26 comparator, 27 resistor, 28 resistor, 29 nodes, 30 resistors, 31 inverters, 32 resistors, 33 capacitors, 34 emitter power supply, 35 Zener, 36 capacitances, 37 nodes, 38 resistors, 39 buffers, 40 voltage amplifiers, 41 switches, 42 voltage comparators, 43 reference voltage generation Adjustment circuit 44 Zener diode, 45 diode, 46 resistor, 47 MOS transistor, 48 resistor, 49 Zener diode, 50 diode, 51 resistor, 52 bipolar transistor, 53 capacitance, 54 buffer, 55 resistor, 56 comparator, 57 resistor, 58 resistor, 59 Resistor, 60 Resistor, 61 Capacitance, 62 Resistor, 63 Capacitance, 64 Bipolar Transistor, 65 Diode, 66 Resistor, 67 Resistor, 68 Resistor, 69 Balance Resistor, 70 Emitter Power Supply, 71 Zener Diode, 72 Diode, 73 Diode, 74 Diode 75 MOS transistor, 76 Zener diode, 77 resistor, 78 resistor, 79 MOS transistor, 80 overcurrent detection circuit, 81 control circuit, 90 input control signal 91 gate voltage, 92 buffers the output of the voltage of the 93 node, the voltage of the 94 node 14, the voltage of the 95 node 25, 100 main inverter, 200 a timing detection circuit.

Claims (16)

  1. A drive circuit for driving a power semiconductor element,
    A switching circuit that receives an input control signal from the outside and performs on / off switching of the power semiconductor element;
    A sampling signal generating circuit that outputs the sampling signal at approximately the start time of the mirror period of the power semiconductor element when the input control signal is detected and the input control signal instructs to turn off;
    When the mirror voltage of the power semiconductor element is detected at the timing when the sampling signal is input and connected to the gate line of the power semiconductor element, and the mirror voltage is equal to or higher than a predetermined threshold value, A gate voltage detection circuit that outputs an overcurrent detection signal;
    The power semiconductor element is connected to a gate line of the power semiconductor element and receives the overcurrent detection signal from the gate voltage detection circuit so as to turn off the power semiconductor element at a speed slower than normal. And a gate voltage control circuit for controlling the gate voltage of the power semiconductor element drive circuit.
  2. A drive circuit for driving a power semiconductor element,
    A switching circuit that receives an input control signal from the outside and performs on / off switching of the power semiconductor element;
    A timing detection circuit connected to the gate line of the power semiconductor element, detecting a mirror period of the power semiconductor element, and outputting a timing signal at approximately the start time of the mirror period;
    When the mirror voltage of the power semiconductor element is detected at the timing when the timing signal is input and connected to the gate line of the power semiconductor element, and the mirror voltage is equal to or higher than a predetermined threshold value, A gate voltage detection circuit that outputs an overcurrent detection signal;
    The power semiconductor is connected to the gate line of the power semiconductor element, receives the overcurrent detection signal from the gate voltage detection circuit, and turns off the power semiconductor element at a speed slower than normal. A drive circuit for a power semiconductor element, comprising: a gate voltage control circuit that controls a gate voltage of the element.
  3.   3. The drive circuit for a power semiconductor device according to claim 1, wherein the sampling signal generation circuit or the timing detection circuit includes a delay circuit having a resistor and a capacitance and a buffer.
  4. The sampling signal generator circuitry includes a reference voltage generating circuit for outputting a predetermined reference voltage set in advance, a delay circuit having a resistance and capacitance, the output voltage of the delay circuit, than the reference voltage, greater 3. The power semiconductor element drive circuit according to claim 1, further comprising a voltage comparator that detects whether the power is small.
  5. The sampling signal generator circuits are in parallel with the resistor, the power according to claim 3 or 4 the input control signal direction and further comprising a diode connected such that the forward direction Semiconductor device drive circuit.
  6. Output period of said sampling signal generating circuits are power semiconductor device according to any one of claims 1 to 5, characterized in that it is set to be shorter than the mirror period of the semiconductor device for the power Drive circuit.
  7. 7. The method according to claim 1 , wherein a time from when the gate voltage starts to decrease to when the sampling signal is output is set to be constant by an OFF command of the input control signal. A drive circuit for a power semiconductor device according to the item.
  8. Depending on the sampling signal generating circuits output period the power supply voltage, the higher the supply voltage decreases, the power semiconductor device according to any one of claims 1 to 6, characterized in that the output period delayed Driving circuit.
  9. The gate voltage detection circuit is
    A voltage amplifier connected to the gate line of the power semiconductor element and amplifying the gate voltage to a predetermined level;
    9. The switch according to claim 1, further comprising: a switch connected to the voltage amplifier and switching based on a signal output from the sampling signal generation circuit or the timing detection circuit. Drive circuit for power semiconductor element.
  10. The gate voltage detection circuit is
    A reference voltage generating circuit for outputting a predetermined reference voltage set in advance;
    A voltage comparator connected to the gate line of the power semiconductor element and detecting whether the gate voltage of the power semiconductor element is larger or smaller than the reference voltage;
    9. The switch according to claim 1, further comprising: a switch connected to the voltage comparator and switching based on a signal output from the sampling signal generation circuit or the timing detection circuit. Drive circuit for power semiconductor elements.
  11.   Even if the mirror voltage fluctuates due to fluctuations in the power supply voltage of the drive circuit, the gate voltage detection circuit outputs an overcurrent detection signal when a current exceeding a certain level flows without changing the detection level for the overcurrent. 11. The drive circuit for a power semiconductor element according to claim 1, wherein the drive circuit is a power semiconductor element drive circuit.
  12.   The gate voltage detection circuit outputs an overcurrent detection signal when a current exceeding a certain level flows without changing the detection level for the overcurrent even when the temperature of the drive circuit fluctuates. 12. A drive circuit for a power semiconductor device according to any one of 1 to 11.
  13. The gate voltage control circuit is
    A MOS transistor connected to the gate voltage detection circuit and switching based on a signal from the gate voltage detection circuit;
    13. The power semiconductor element drive circuit according to claim 1, further comprising: a Zener diode connected between the gate line of the power semiconductor element and the MOS transistor. .
  14. The gate voltage control circuit is
    A MOS transistor connected to the gate voltage detection circuit and switching based on a signal from the gate voltage detection circuit;
    A resistor connected between the gate line of the power semiconductor element and the MOS transistor;
    The drive circuit for a power semiconductor device according to any one of claims 1 to 12, wherein an output from the resistor is connected to an off-side switch of the switching circuit.
  15. A drive circuit for driving a power semiconductor element,
    A switching circuit that receives an input control signal from the outside and performs on / off switching of the power semiconductor element;
    A sampling signal generating circuit that outputs the sampling signal at approximately the start time of the mirror period of the power semiconductor element when the input control signal is detected and the input control signal instructs to turn off;
    When the mirror voltage of the power semiconductor element is detected at the timing when the sampling signal is input and connected to the gate line of the power semiconductor element, and the mirror voltage is equal to or higher than a predetermined threshold value, A gate voltage detection circuit that outputs an overcurrent detection signal;
    An overcurrent detection circuit that detects a gate voltage in an on state, detects an overcurrent based on an increase in the gate voltage, and outputs an oncurrent overcurrent detection signal;
    The power semiconductor element is connected to the gate line of the power semiconductor element and receives the overcurrent detection signal from the gate voltage detection circuit or the on-time overcurrent detection signal from the overcurrent detection circuit, And a gate voltage control circuit for controlling a gate voltage of the power semiconductor element so as to be turned off at a slower speed.
  16. A drive circuit for driving a power semiconductor element,
    A switching circuit that receives an input control signal from the outside and performs on / off switching of the power semiconductor element;
    A timing detection circuit connected to the gate line of the power semiconductor element, detecting a mirror period of the power semiconductor element, and outputting a timing signal at approximately the start time of the mirror period;
    When the mirror voltage of the power semiconductor element is detected at the timing when the timing signal is input and connected to the gate line of the power semiconductor element, and the mirror voltage is equal to or higher than a predetermined threshold value, A gate voltage detection circuit that outputs an overcurrent detection signal;
    An overcurrent detection circuit that detects a gate voltage in an on state, detects an overcurrent from an increase in the gate voltage, and outputs an overcurrent detection signal when on;
    The power semiconductor element is connected to the gate line of the power semiconductor element and receives the overcurrent detection signal from the gate voltage detection circuit or the on-time overcurrent detection signal from the overcurrent detection circuit, And a gate voltage control circuit for controlling a gate voltage of the power semiconductor element so as to be turned off at a slower speed.
JP2002283658A 2002-01-17 2002-09-27 Power semiconductor element drive circuit Expired - Fee Related JP3886876B2 (en)

Priority Applications (3)

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JP2002008596 2002-01-17
JP2002-8596 2002-01-17
JP2002283658A JP3886876B2 (en) 2002-01-17 2002-09-27 Power semiconductor element drive circuit

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JP2002283658A JP3886876B2 (en) 2002-01-17 2002-09-27 Power semiconductor element drive circuit
US10/345,388 US6967519B2 (en) 2002-01-17 2003-01-16 Drive circuit for a power semiconductor device
CNB031017118A CN1314201C (en) 2002-01-17 2003-01-17 Drive circuit of power semiconductor element
DE2003101655 DE10301655B4 (en) 2002-01-17 2003-01-17 Control circuit for a power semiconductor device

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JP3886876B2 true JP3886876B2 (en) 2007-02-28

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US6967519B2 (en) 2005-11-22
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CN1445928A (en) 2003-10-01
DE10301655A1 (en) 2003-08-07
JP2003284318A (en) 2003-10-03

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