JP2023083501A - Information processing device, optical apparatus, control method, program, and storage medium - Google Patents

Abstract

To provide an information processing device capable of suitably reducing noise superposed on an output signal from a light-receiving unit.SOLUTION: A lidar 1 includes: a scan unit 55 for radiating emission light Lo according to a pulse trigger signal PT while changing a radiation direction; an absorber 7 arranged in a prescribed radiation direction for absorbing the emission light Lo; an APD 41 for receiving return light Lr of the emission light Lo; and a DSP 16. The DSP 16 includes a synchronization obstruction estimation unit 71 for calculating estimated synchronization obstruction w, i.e. a noise signal due to the pulse trigger signal PT or the like on the basis of an output signal of the APD 41 when the radiation direction of the emission light Lo is in a direction to irradiate the absorber 7.SELECTED DRAWING: Figure 11

Description

The present invention relates to information processing technology for optical equipment using laser light.

Conventionally, as a distance measuring device using light, a pulsed light is emitted to an object to be measured, and the distance to the object is measured based on the reception timing of the pulsed light reflected by the object to be measured. are widely used. For example, in Patent Document 1, a trigger signal is generated by a control unit, pulsed light is emitted to a measurement object with the trigger signal as a starting point, and the pulsed light reflected by the measurement object is received by a light receiving unit and received. A ranging device is disclosed that generates a signal.

JP 2007-256191 A

Since the trigger signal, which defines the timing of emitting the pulsed light, is extremely large compared to the output level of the light receiving section, noise caused by high-frequency components included in the rise and fall of the trigger signal is superimposed on the output signal of the light receiving section. It causes an error if

SUMMARY OF THE INVENTION The present invention has been made to solve the problems described above, and a main object of the present invention is to provide an information processing apparatus capable of suitably reducing noise superimposed on an output signal of a light receiving section. and

The claimed invention includes an irradiation unit that irradiates a laser beam according to a first signal while changing the irradiation direction, an absorber that is arranged in a predetermined irradiation direction and absorbs the laser beam, and the laser beam. an information processing device for processing an output signal of the light receiving unit of an optical device, comprising: a light receiving unit for receiving return light of light, wherein the output when the irradiation direction of the laser light is the predetermined irradiation direction An estimator for estimating a noise signal caused by the first signal based on the signal.

A claimed invention is an optical apparatus comprising: an irradiating unit that irradiates laser light according to a first signal while changing the irradiation direction; and an irradiation unit that is arranged in a predetermined irradiation direction and absorbs the laser light. a noise signal caused by the first signal based on an absorber, a light receiving portion for receiving the return light of the laser beam, and the output signal when the irradiation direction of the laser beam is the predetermined irradiation direction; an estimating unit for estimating

Further, the invention described in the claims includes an irradiation unit that irradiates laser light according to a first signal while changing the irradiation direction, an absorber that is arranged in a predetermined irradiation direction and absorbs the laser light, A control method executed by an optical device having a light receiving unit that receives the return light of the laser light, wherein the control method is executed based on the output signal when the irradiation direction of the laser light is the predetermined irradiation direction. There is a noise estimation step of estimating a noise signal due to the first signal.

Further, the invention described in the claims includes an irradiation unit that irradiates laser light according to a first signal while changing the irradiation direction, an absorber that is arranged in a predetermined irradiation direction and absorbs the laser light, A program executed by a computer that processes an output signal of the light receiving unit of an optical device having a light receiving unit that receives the return light of the laser light, wherein the irradiation direction of the laser light is the predetermined irradiation direction. The computer functions as an estimator that estimates a noise signal caused by the first signal based on the output signal at the time.

1 shows the overall configuration of a lidar according to an embodiment; 1 shows the configuration of a transmitter and a receiver; 4 shows the configuration of a scanning optical unit; 4 shows an example of setting a register for a control signal generated by a synchronization control unit; 4 shows the temporal relationship of control signals generated by the synchronization control section; 4 is a graph showing the relationship between ADC output signals and gates; Fig. 3 shows the temporal relationship of the pulse train of the rotary encoder; Fig. 4 shows the temporal relationship between encoder pulses and segment slots in steady state; 4 shows the relationship between the pulse trigger signal and the received segment signal; It is the figure which showed roughly the arrangement|positioning of an absorber. FIG. 4 shows a block diagram of signal processing performed by the DSP in the first embodiment. FIG. 4 shows waveforms of a reference received pulse and an impulse response within one segment period; FIG. It is an example of a filtered segment. FIG. 4 is a diagram schematically showing the arrangement of reflectors; FIG. 11 shows a block diagram of signal processing performed by the DSP in the second embodiment. FIG. 4 is a diagram schematically showing the arrangement of absorbers and reflectors; The reflection component from the absorber, the synchronous interference component, and the reflection component from the reflector in the reception segment within one segment period are shown, respectively. FIG. 11 shows a block diagram of signal processing performed by the DSP in the third embodiment. FIG. 4 is a block diagram of signal processing performed by a synchronization disturbance second estimator;

According to a preferred embodiment of the present invention, an irradiation unit that irradiates a laser beam according to a first signal while changing the irradiation direction, and an absorber that is arranged in a predetermined irradiation direction and absorbs the laser beam. and a light receiving portion for receiving the return light of the laser beam, the information processing device for processing an output signal of the light receiving portion of the optical equipment, wherein the irradiation direction of the laser beam is the predetermined irradiation direction. an estimating unit for estimating a noise signal caused by the first signal based on the output signal of .

The information processing device processes an output signal from a light receiving section of an optical device comprising an irradiation section that irradiates a laser beam according to a first signal while changing the direction of irradiation, an absorber, and a light receiving section. and an estimator. The estimation unit estimates a noise signal caused by the first signal based on the output signal of the light receiving unit when the irradiation direction of the laser light is the predetermined irradiation direction in which the absorber is arranged. In this aspect, the information processing device utilizes an absorber arranged in a predetermined irradiation direction, so that the output signal of the light receiving unit when no return light received by the light receiving unit is generated or when the amount of the return light is extremely small. is intentionally generated. As a result, the information processing device can accurately estimate the noise signal superimposed on the output signal of the light receiving unit due to the high frequency component of the first signal that defines the timing at which the irradiating unit irradiates the laser light. can be done.

In one aspect of the information processing device, the information processing device further includes a section signal extraction unit that extracts a section signal for each irradiation direction of the laser beam from the output signal, and the estimation unit includes: estimating the noise signal based on at least one corresponding interval signal; According to this aspect, the information processing device can suitably estimate the noise signal caused by the first signal superimposed on each section signal based on the section signal corresponding to the irradiation direction in which the absorber is arranged. .

In another aspect of the information processing device, the information processing device further includes a subtraction section that subtracts the noise signal estimated by the estimation section from the output signal of the light receiving section. According to this aspect, the information processing device can preferably generate the output signal of the light receiving unit from which the noise signal caused by the first signal has been removed.

In another aspect of the information processing device, the absorber has a multiple reflection structure. With this aspect, the absorber can suitably achieve a low reflectance, so that the information processing device can acquire the output signal of the light receiving unit when no return light is generated or when the amount of return light is low. .

In another aspect of the information processing device, the estimation unit averages the output signal for each irradiation angle when the irradiation direction of the laser light is the predetermined irradiation direction, thereby averaging the noise signal. presume. According to this aspect, the information processing device can estimate the noise signal with high accuracy.

In another aspect of the information processing device, the estimating unit averages the output signals obtained when the irradiating unit scans a plurality of times in the predetermined irradiation direction so that the noise Estimate the signal. According to this aspect, the information processing device can suitably calculate a highly accurate estimated value of the noise signal regardless of the magnitude of the angle of the irradiation direction in which the absorber is arranged.

In another aspect of the information processing apparatus, the first signal is a pulse signal, and the irradiation section irradiates pulse light as the laser light in different irradiation directions based on the pulse signal. In this aspect, the information processing device can preferably estimate the noise signal caused by the high-frequency components included in the rise and fall of the pulse signal that defines the irradiation timing of the laser beam of the irradiation unit.

In another preferred embodiment of the present invention, the optical device includes an irradiating unit that irradiates laser light according to a first signal while changing the irradiation direction, and an irradiation unit that is arranged in a predetermined irradiation direction and absorbs the laser light. a light-receiving unit for receiving the return light of the laser light; and noise caused by the first signal based on the output signal when the irradiation direction of the laser light is the predetermined irradiation direction. an estimator for estimating the signal. According to this aspect, the optical device can accurately estimate the noise signal superimposed on the output signal of the light receiving unit due to the high frequency component of the first signal that defines the timing at which the irradiating unit irradiates the laser light. can be done.

According to another preferred embodiment of the present invention, an irradiating unit that irradiates a laser beam according to a first signal while changing the irradiation direction, and an absorption unit that is arranged in a predetermined irradiation direction and absorbs the laser beam. and a light-receiving unit that receives the return light of the laser light, the control method being executed by the optical device based on the output signal when the irradiation direction of the laser light is the predetermined irradiation direction. and a noise estimation step of estimating a noise signal caused by the first signal. By executing this control method, the optical device reduces the noise signal superimposed on the output signal of the light receiving unit due to the high frequency component of the first signal that defines the timing at which the irradiating unit irradiates the laser beam. can be estimated accurately.

According to another preferred embodiment of the present invention, an irradiating unit that irradiates a laser beam according to a first signal while changing the irradiation direction, and an absorption unit that is arranged in a predetermined irradiation direction and absorbs the laser beam. A program executed by a computer for processing an output signal of the light receiving unit of an optical device having a body and a light receiving unit that receives the return light of the laser light, wherein the irradiation direction of the laser light is the predetermined irradiation The computer functions as an estimator that estimates a noise signal caused by the first signal based on the output signal when the direction is the direction. By executing this program, the computer accurately removes the noise signal superimposed on the output signal of the light receiving unit due to the high frequency component of the first signal that defines the timing at which the irradiation unit irradiates the laser light. can be estimated. Preferably, the program is stored in a storage medium.

Preferred embodiments of the present invention will be described below with reference to the drawings.

<Basic description>
First, the basic configuration of the lidar according to the embodiment will be described.

(1) Overall Configuration FIG. 1 shows the overall configuration of a lidar according to an embodiment. The lidar 1 scans the surrounding space by appropriately controlling the emission direction of light pulses repeatedly emitted (hereinafter referred to as "scanning direction"), and observes the returned light to detect objects in the surroundings. information (such as distance, its probability of existence, reflectance, etc.). Specifically, the rider 1 emits a light pulse (hereinafter referred to as "emission light") Lo, which is reflected by an external object (target) (hereinafter referred to as "return light") Lr. generates information about the object. The lidar 1 is an example of an "optical instrument" in the present invention.

As shown in FIG. 1, the lidar 1 is roughly divided into a system CPU 5, an ASIC 10, a transmitter 30, a receiver 40, and a scanning optical section 50. The transmitter 30 repeatedly outputs laser light pulses with a width of about 5 nsec in response to the pulse trigger signal PT supplied from the ASIC 10 . A light pulse output from transmitter 30 is guided to scanning optical section 50 .

The scanning optical unit 50 emits the light pulse output by the transmitter 30 in an appropriate direction, and collects the return light Lr that has returned by being reflected or scattered when the emitted light encounters an object in space. and lead it to the receiver 40 . The scanning optical section 50 is an example of the "irradiation section" in the present invention. The receiver 40 outputs to the ASIC 10 a signal proportional to the intensity of the return light Lr. The receiver 40 is an example of the "light receiving section" in the present invention.

The ASIC 10 analyzes the output signal of the receiver 40 to estimate and output parameters related to the object in the scanning space, such as its distance. The ASIC 10 also controls the scanning optics 50 to ensure proper scanning. In addition, ASIC 10 provides transmitter 30 and receiver 40 with the high voltages required by each.

The system CPU 5 performs at least initial setting, monitoring and control of the ASIC 10 through the communication interface. Other functions vary depending on the application. In the case of the simplest lidar, the system CPU 5 merely converts the target information TI output by the ASIC 10 into an appropriate format and outputs it. The system CPU 5, for example, converts the target information TI into a highly versatile point group format, and then outputs it through the USB interface.

(2) Transmitter The transmitter 30 outputs an optical pulse with a width of about 5 nsec in response to the pulse trigger signal PT supplied from the ASIC 10 . The configuration of the transmitter 30 is shown in FIG. 2(A). Transmitter 30 includes charging resistor 31 , driver circuit 32 , capacitor 33 , charging diode 34 , laser diode (LD) 35 and CMOS switch 36 .

A pulse trigger signal PT input from the ASIC 10 drives a switch 36 such as a CMOS through a driver circuit 32 . A driver circuit 32 is inserted to drive the switch 36 at high speed. During non-asserted periods of pulse trigger signal PT, switch 36 is open and capacitor 33 within transmitter 30 is charged with the high voltage V _TX supplied by ASIC 10 . On the other hand, during the assertion period of the pulse trigger signal PT, the switch 36 is closed, and the charge stored in the capacitor 33 is discharged through the LD35. As a result, an optical pulse is output from the LD35.

(3) Receiver The receiver 40 outputs a voltage signal proportional to the intensity of the return light Lr from the object. Generally, photodetection elements such as PDs and APDs output current, so the receiver 40 converts this current into voltage (I/V conversion) and outputs it. The configuration of the receiver 40 is shown in FIG. 2(B). The receiver 40 includes an APD (Avalanche Photodiode) 41 , an I/V converter 42 , a resistor 45 , a capacitor 46 and a low pass filter (LPF) 47 . The I/V converter 42 includes a feedback resistor 43 and an operational amplifier 44 .

In this embodiment, an APD 41 is used as the photodetector. A high voltage _VRX supplied from the ASIC 10 is applied to the APD 41 as a reverse bias, and a detection current proportional to the return light Lr from the object flows. By applying a reverse bias close to the breakdown voltage of the APD 41, it is possible to obtain a high avalanche gain and detect weak return light. The final stage LPF 47 is provided for the purpose of limiting the bandwidth of the signal prior to sampling by the ADC 20 within the ASIC 10 . In this embodiment, the sampling frequency of the ADC 20 is 512 MHz, and the cutoff frequency of the LPF 47 is approximately 250 MHz.

(4) Scanning Optical Section The scanning optical section 50 emits the light pulse input from the transmitter 30 as emitted light Lo in an appropriate direction, and this emitted light Lo encounters an object in space and is reflected or scattered. As a result, the returned light Lr is guided to the receiver 40 . A configuration example of the scanning optical unit 50 is shown in FIG. The scanning optical section 50 includes a rotating mirror 61 , a collimator lens 62 , a condenser lens 64 , an optical filter 65 , a coaxial mirror 66 and a rotary encoder 67 .

A light pulse output from the LD 35 of the transmitter 30 enters the collimator lens 62 . A collimator lens 62 collimates the laser light to an appropriate divergence angle (generally on the order of 0-1°). Emitted light from the collimator lens 62 is reflected vertically downward by a small coaxial mirror 66 and enters the rotating shaft (center) of the rotating mirror 61 . The rotating mirror 61 horizontally reflects the laser light incident from vertically above and emits it into the scanning space. The rotating mirror 61 is attached to the rotating portion of the motor 54 , and the laser light reflected by the rotating mirror 61 scans the horizontal plane as the emitted light Lo as the motor 54 rotates.

The return light Lr returned to the lidar 1 by being reflected or scattered by an object existing in the scanning space is reflected vertically upward by the rotating mirror 61 and enters the optical filter 65 . In addition to the return light Lr, the background light caused by the object being illuminated by the sun or the like also enters the optical filter 65 . An optical filter 65 is provided to selectively eliminate such background light. Specifically, the optical filter 65 selectively allows only components around ±10 nm around the wavelength of the emitted light Lo (905 nm in this embodiment) to pass. If the passband of the optical filter 65 is wide, a large amount of background light will enter the subsequent receiver 40 . As a result, a large DC current component appears in the output of the APD 41 in the receiver 40, and shot noise (background light shot noise) caused by this DC component degrades SN, which is not preferable. However, if the passband is too narrow, the emitted light itself is also suppressed, which is not preferable. The condenser lens 64 collects the light that has passed through the optical filter 65 and guides it to the APD 41 of the receiver 40 .

A rotary encoder 67 is attached to the motor 54 to detect the scanning direction. The rotary encoder 67 has a turntable 68 attached to the motor rotating portion and a code detector 69 attached to the motor base. A slit representing the rotation angle of the motor 54 is engraved on the outer circumference of the rotating disk 68, and the code detector 69 reads and outputs the slit. Specific specifications of the rotary encoder 67 and motor control based on its output will be described later.

In the above configuration, the collimator lens 62 constitutes the transmission optical system 51 shown in FIG. 1, the rotating mirror 61 and the motor 54 constitute the scanning unit 55 shown in FIG. , and the rotary encoder 67 constitutes the scanning direction detector 53 in FIG.

(5) ASICs
The ASIC 10 performs timing control of emitted light pulses, AD conversion of APD output signals, and the like. The ASIC 10 also performs appropriate signal processing on the AD conversion output to estimate parameters (distance, return light intensity, etc.) related to the object, and outputs the estimation results to the outside. As shown in FIG. 1, the ASIC 10 includes a register section 11, a clock generation section 12, a synchronization control section 13, a gate extraction section 14, a reception segment memory 15, a DSP 16, and a transmitter high voltage generation section (TXHV ) 17 , a receiver high voltage generator (RXHV) 18 , a preamplifier 19 , an AD converter (ADC) 20 , and a scanning controller 21 .

Registers for communication with the system CPU 5 as an external processor are arranged in the register unit 11 . The registers provided in the register unit 11 are roughly classified into R registers that can be referenced only from the outside and W registers that can be set from the outside. The R register mainly holds status values inside the ASIC, and the system CPU 5 can monitor the internal status of the ASIC 10 by reading these values through the communication interface. On the other hand, the W register holds various parameter values referred to inside the ASIC 10 . These various parameter values can be set from the system CPU 5 through the communication interface. Note that the communication register may be implemented by a flip-flop or may be implemented as a RAM.

The clock generator 12 generates a system clock SCK and supplies it to each block within the ASIC 10 . Many blocks of the ASIC 10 operate in synchronization with the system clock SCK. In this embodiment, the system clock SCK has a frequency of 512 MHz. The system clock SCK is generated by the PLL so as to synchronize with the externally input reference clock RCK. A crystal oscillator is normally used as the source of the reference clock RCK.

TXHV 17 generates the DC high voltage (on the order of 100V) required by transmitter 30 . This high voltage is generated by boosting a low voltage (about 5V to 15V) by a DCDC converter circuit.

RXHV 18 generates the DC high voltage (on the order of 100V) required by receiver 40 . This high voltage is generated by boosting a low voltage (about 5V to 15V) by a DCDC converter circuit.

The synchronization control unit 13 generates and outputs various control signals. The synchronization control section 13 in this embodiment outputs two control signals, ie, a pulse trigger signal PT and an AD gate signal GT. A setting example of these control signals is shown in FIG. 4, and their temporal relationship is shown in FIG. As shown in FIG. 5, these control signals are generated in synchronization with time intervals (segment slots) divided at predetermined intervals. The time interval width (segment period) of the segment slot can be set by "nSeg". In this embodiment, it is assumed that "nSeg=8192" is set unless otherwise specified.

A pulse trigger signal PT is supplied to a transmitter 30 provided outside the ASIC 10 . The transmitter 30 outputs optical pulses in response to the pulse trigger signal PT. For the pulse trigger signal PT, it is possible to set a delay "dTrg" and a pulse width "wTrg" with respect to the segment slot start point. If the pulse width wTrg is too narrow, the transmitter 30 will not respond, so it is determined in consideration of the trigger response specifications of the transmitter 30 .

The AD gate signal GT is supplied to the gate extractor 14 . As will be described later, the gate extraction unit 14 extracts only the asserted period of the AD gate signal GT from the ADC output signal input from the ADC 20 and stores it in the reception segment memory 15 . For the AD gate signal GT, it is possible to set a delay time "dGate" and a gate width "wGate" with respect to the segment slot start point.

The preamplifier 19 amplifies the voltage of the analog voltage signal input from the receiver 40 installed outside the ASIC 10 and supplies it to the subsequent ADC 20 . Note that the voltage gain of the preamplifier 19 can be set by the W register.

The ADC 20 AD-converts the output signal of the preamplifier 19 and converts it into a digital series. In this embodiment, the system clock SCK is used as the sampling clock for the ADC 20, and the input signal of the ADC 20 is sampled at 512 MHz.

The gate extraction unit 14 extracts only the asserted period of the AD gate signal GT from the ADC output signal input from the ADC 20 and stores it in the reception segment memory 15 . The segment signal extracted by the gate extractor 14 is hereinafter referred to as "received segment signal RS". That is, the received segment signal RS is a real vector whose vector length is equal to the gate width wGate. The gate extractor 14 is an example of the "interval signal extractor" in the present invention.

Here, the relationship between the ADC output signal and the received segment and the setting of the gate position will be described. FIG. 6A shows segment slots. As shown in FIG. 6B, the pulse trigger signal PT is asserted with a delay of dTrg from the segment slot start point. Since dTrg=0 in the example of FIG. 6, the pulse trigger signal PT is asserted at the starting point of the segment slot. FIG. 6C shows the ADC output signal (received segment signal RS) when an object is placed at the scanning origin of the lidar. That is, FIG. 6C illustrates the received segment signal RS when the target distance (radius radius R) is 0 m. As shown, even when R=0 m, the rising edge of the received pulse is observed after the rising edge of the pulse trigger signal by the system delay _{D_SYS} . The system delay _DSYS is caused by electrical delay in the LD driver circuit in the transmitter 30, optical delay in the transmission optical system 51, optical delay in the reception optical system 52, and electrical delay in the receiver 40. Delays, conversion delays in the ADC 20, etc. are conceivable.

FIG. 6(D) illustrates the received segment signal RS when the object is placed at the radius R. FIG. In this case, the delay increases by the round trip time of the light from the scanning origin to the object as compared with FIG. 6(C). This increased delay is the so-called "TOF (Time Of Flight) delay". If this TOF delay is D samples, the radius R can be calculated by the following equation.

図６（Ｆ）は、ｄＧａｔｅ＝０の場合のＡＤゲート信号ＧＴを例示するものである。前述したとおり、ゲート抽出部１４は、ＡＤＣ出力信号から、ＡＤゲート信号ＧＴのアサート区間のみを抽出する。後述するＤＳＰ１６は、この抽出区間のみに基づいて、物体に関するパラメータ推定を行う。したがって、ＴＯＦ遅延時間が大きい場合には、物体からの戻りパルス成分がゲートからはみ出してしまい正当なパラメータ推定が行えない。正当なパラメータ推定が行われるためにはＴＯＦ遅延時間Ｄが次式を満たしていることが必要となる。

FIG. 6F exemplifies the AD gate signal GT when dGate=0. As described above, the gate extraction unit 14 extracts only the asserted period of the AD gate signal GT from the ADC output signal. The DSP 16, which will be described later, performs parameter estimation regarding the object based only on this extracted section. Therefore, when the TOF delay time is long, the return pulse component from the object protrudes from the gate, making proper parameter estimation impossible. In order to perform valid parameter estimation, the TOF delay time D must satisfy the following equation.

ここでＬ_ＩＲはシステムの総合インパルス応答の長さであり、Ｄ_ＭＡＸは正当なパラメータ推定が可能な最大ＴＯＦ遅延時間として定義される。図６（Ｅ）は、ＴＯＦ遅延時間がこの最大ＴＯＦ遅延時間に等しい場合の受信セグメント信号ＲＳを例示している。

where _LIR is the length of the total impulse response of the system and _DMAX is defined as the maximum TOF delay time for which valid parameter estimation is possible. FIG. 6(E) illustrates the received segment signal RS when the TOF delay time is equal to this maximum TOF delay time.

Note that instead of the example of FIG. 6, the gate delay dGate may be set equal to the system delay time. By setting in this way, correct parameter estimation is possible even for objects at a longer distance.

The scanning control unit 21 monitors the output of a rotary encoder 67 installed outside the ASIC 10 and controls the rotation of the motor 54 based on this. Specifically, the scanning control unit 21 supplies the torque control signal TC to the motor 54 based on the scanning direction information SDI output from the rotary encoder 67 (scanning direction detection unit 53) of the scanning optical unit 50. FIG. The rotary encoder 67 in this embodiment outputs two pulse trains of A phase and Z phase (hereinafter referred to as "encoder pulses"). The temporal relationship between both pulse trains is shown in FIG. 7(A). As shown, for the A phase, one pulse is generated and output for every 1° rotation of the motor 54 . Therefore, 360 A-phase encoder pulses are generated and output for each rotation of the motor 54 . On the other hand, for the Z phase, one pulse is generated and output for one rotation of the motor 54 corresponding to a predetermined rotation angle.

The scanning control unit 21 measures the rise time of the encoder pulse as the counter value of the system clock SCK, and controls the torque of the motor 54 so that this counter value becomes a desired value. That is, the scanning control unit 21 performs PLL control of the motor 54 so that the encoder pulse and the segment slot have a desired time relationship.

The time relationship between encoder pulses and segment slots can be set by the W register shown in FIG. 7(B). The number of A-phase encoder pulses per motor rotation is set in "nPpr". This value is determined by the specifications of the rotary encoder 67, and is set to 360 as described above in this embodiment. "nRpf" gives the number of rotations per frame, and "nSpf" gives the number of segments per frame. "dSmpA" and "dSmpZ" are prepared to adjust the time relationship between the rise of the encoder pulse and the segment slot in units of sample clocks, and can define the delay of the encoder pulse with respect to the start point of the segment slot. . On the other hand, "dSegZ" is prepared for adjusting the temporal relationship between the rise of the Z-phase pulse and the frame in units of segments.

FIG. 8 shows the time relationship between encoder pulses and segment slots in a steady state. As shown, in the default setting, one frame consists of 1800 segments, and the motor 54 makes one revolution in one frame.

(6) DSPs
The DSP 16 sequentially reads the received segments _{y_frm, seg} from the received segment memory 15 and processes them. Here, "frm" is a frame index and "seg" is a segment index. In the following, the notation of these indices is omitted to the extent that there is no fear of misunderstanding. The receiving segment y is a real vector of vector length wGate and is expressed by the following equation.

ＤＳＰ１６の詳細な構成については、第１～第３実施例で説明する。ＤＳＰ１６は、本発明における「情報処理装置」の一例である。

A detailed configuration of the DSP 16 will be described in the first to third embodiments. The DSP 16 is an example of an "information processing device" in the present invention.

Here, the reception segment y (that is, the reception segment signal RS) is superimposed with interference synchronized with the segment period (also simply called "synchronization interference") due to the effects of electromagnetic jumping, current flowing to the ground, and the like. be done. FIG. 9 shows the relationship between the pulse trigger signal PT and the received segment signal RS. The example of FIG. 9 shows the received segment signal RS when the intensity of the return light Lr received by the APD 41 is assumed to be 0 with respect to the emitted light Lo emitted by asserting the pulse trigger signal PT.

In the example of FIG. 9, high-frequency components included in the rising and falling edges of the pulse trigger signal PT are superimposed on the received segment signal RS. Here, the TTL level of the pulse trigger signal PT is normally 3.3V or 5V, and the current level output by the APD 41 is on the order of nA or pA. Therefore, in this case, the influence of the synchronization disturbance on the output of the APD 41 is relatively large, and the synchronization disturbance may cause deterioration of the object detection performance and range finding performance of the rider 1 . In consideration of the above, in the first and third embodiments described below, the DSP 16 preferably reduces the influence of the synchronization disturbance by estimating the synchronization disturbance. The pulse trigger signal PT is an example of the "first signal" in the present invention, and the synchronization disturbance is an example of the "noise signal" in the present invention.

<First embodiment>
First, the first embodiment will be explained. Schematically, the lidar 1 has an absorber that absorbs the emitted light Lo in a particular scan direction, and the DSP 16 averages the received segment y corresponding to the scan direction incident on the absorber to achieve synchronization Estimate interference. DSP 16 then subtracts the estimated synchronization disturbance from received segment y. This preferably reduces the influence of synchronization disturbance.

FIG. 10(A) is a diagram schematically showing the arrangement of the absorber 7. FIG. In FIG. 10A, the absorber 7 is arranged near the housing 25 of the substantially cylindrical lidar 1 that accommodates the scanning unit 55 and the like. Here, the absorber 7 is a non-detection direction (arrow A1). In the example of FIG. 10A, the absorber 7 exists on the wall surface of the housing 25 behind the rider 1 irradiated with the emitted light Lo for the angle "θa" (for example, 60 degrees). In this case, the absorber 7 is provided, for example, inside the transparent cover of the housing 25 that transmits the emitted light Lo and the return light Lr. In another example, the absorber 7 may be a portion of the transparent cover of the housing 25 that has been processed (for example, painted black) so as to absorb the emitted light Lo. Hereinafter, a period during which the APD 41 receives the emitted light Lo irradiated to the absorber 7 within a period in which one scan is performed by the scanning unit 55 (that is, one frame period) is also referred to as a “reflection suppression period Ttag1”. . The antireflection period Ttag1 includes a plurality of segment periods corresponding to each scanning angle at which the absorber 7 is irradiated with the emitted light Lo. Information about the scanning angle at which the light beam Lo is irradiated onto the absorber 7 (eg, segment index, etc.) is pre-stored in a W register or the like so that the DSP 16 can refer to it.

FIG. 10(B) shows a state in which the emitted light Lo is emitted in the direction in which the absorber 7 is arranged in the example of FIG. 10(A). When the emitted light Lo is incident, the absorber 7 absorbs at least part of the emitted light Lo. A reflectance (for example, 0.1%) that produces a small level of the output signal of the APD 41 is achieved. In this case, the return light Lr, which is the emitted light Lo reflected by the absorber 7 , reaches the scanning unit 55 with sufficiently low intensity relative to the output level of the synchronization disturbance detected by the APD 41 . If the absorber 7 is made of a material that completely absorbs the emitted light Lo, no return light Lr is generated.

Here, the absorber 7 has a reflecting surface for the exiting light Lo, which is made of, for example, a material with a very low reflectance. In another example, the absorber 7 may be a beam damper or the like that has a multiple reflection structure and the inner surface (reflection surface) of each reflection structure has a low reflectance.

FIG. 11 shows a block diagram of signal processing performed by the DSP 16 in the first embodiment. As shown in FIG. 11, the DSP 16 according to the first embodiment includes a synchronization disturbance estimator 71, a subtractor 72, a receive filter 73, a peak detector 74, a determiner 75, and a formatter . The DSP 16 sequentially reads out the received segments y from the received segment memory 15 and processes them.

The synchronization disturbance estimation unit 71 averages the received segments y generated within the reflection suppression period Ttag1, and subtracts the averaged reception segment y as an estimated synchronization disturbance (also referred to as "estimated synchronization disturbance w"). 72. Details of the synchronization disturbance estimator 71 will be described later. The subtractor 72 subtracts the estimated synchronization disturbance w supplied from the synchronization disturbance estimation unit 71 from the received segment y, and obtains the received segment y from which the estimated synchronization disturbance w has been subtracted (also referred to as “corrected received segment y _dash ”). is supplied to the receive filter 73 . The synchronization disturbance estimator 71 is an example of the "estimator" in the present invention, and the subtractor 72 is an example of the "subtracter" in the present invention.

The reception filter 73 convolves the corrected reception segment y _dash with a predetermined impulse response "h" (cyclic convolution) to calculate the filtered segment "z". The peak detector 74 detects the point at which the amplitude is maximum within the filtered segment z, that is, the peak point, and outputs the delay "D" and the amplitude "A" of the peak point. The determination unit 75 selectively sends to the formatter 76 only the points where the amplitude A is greater than the predetermined threshold “tDet”. The formatter 76 converts the delay D, the amplitude A, the frame index frm of the segment, and the segment index seg into an appropriate format, and outputs it to the system CPU 5 as target information TI.

Each block will be described in detail below.

Synchronization disturbance estimator 71 includes a switch 77 and an averaging processor 78 . The switch 77 is a switch that is controlled to be ON only during the antireflection period Ttag1, and supplies the received segment y generated during the antireflection period Ttag1 to the averaging processor 78. FIG. Note that the switch 77 does not need to be turned on during the entire antireflection period Ttag1, and may be set to be turned on during a part of the antireflection period Ttag1.

The averaging processor 78 averages the received segments y supplied while the switch 77 is on, and supplies the averaged received segments y to the subtractor 72 as the estimated synchronization disturbance w. The estimated synchronization disturbance w is a real vector of vector length wGate. In this case, for example, the averaging processing unit 78 integrates received segments y sequentially supplied from the switch 77 within one frame period, and divides the integrated received segments y by the integrated number of received segments y. Compute the averaged received segment y as the estimated synchronization disturbance w. In another example, the averaging unit 78 averages the average of the received segments y calculated within one frame period in the frame direction (i.e., between different frame indexes) using an IIR filter or the like, and calculates the estimated synchronization. Calculate as disturbance w.

The receive filter 73 convolves the impulse response h with the corrected receive segment _{y_dash} to calculate the filtered segment z. The impulse response h of the reception filter section 73 can be set by the W register, and is set in advance by the system CPU 5, for example, so that the SNR at the filter output is increased. For example, impulse response h is set to satisfy the following equation. This setting achieves optimal performance (high SNR) when the noise is white and when the overall system impulse response is significantly shorter than wGate.

上式（「関係式Ａ」とも呼ぶ。）において、基準受信パルス「ｇ」は走査原点（Ｒ＝０ｍ）に物体を置いた場合に観測される受信セグメント波形であり、トランスミッタ３０とレシーバ４０を含むシステム全体の総合インパルス応答を代表している。実際に走査原点に物体を置くことが困難な場合には、例えば「Ｒ＝１ｍ」での受信セグメント波形を観測し、これを数学的に時間シフトすることで、等価的に基準受信パルスを測定すれば良い。

In the above equation (also referred to as “relational equation A”), the reference received pulse “g” is a received segment waveform observed when an object is placed at the scanning origin (R=0 m), and transmitter 30 and receiver 40 are It represents the overall impulse response of the entire system including If it is difficult to actually place the object at the scanning origin, for example, observe the received segment waveform at "R = 1 m" and shift it mathematically in time to equivalently measure the reference received pulse. do it.

FIG. 12A shows the waveforms of the reference received pulse g and the impulse response h within one segment period. As shown in FIG. 12(A), the reference received pulse g and the impulse response h have a time-inverted relationship within the segment period, as defined by the above equation. FIG. 12(B) shows waveforms of corrected received segment y _dash and filtered segment z observed when an object is placed at the scanning origin (R=0 m), and FIG. , and the waveforms of the corrected received segment _{y_dash} and the filtered segment z observed when . The reception filter 73 convolves the impulse response h with the corrected reception segment y _dash to perform noise removal filtering, and as shown in FIGS. 12B and 12C, the system delay D _SYS (FIG. (C)), the phase is adjusted accordingly.

Note that the cyclic convolution operation of the receive filter 71 may be implemented in the frequency domain using DFT. By doing so, the amount of calculation can be greatly reduced. In this case, instead of allowing the impulse response h to be set by the W register, the frequency response H may be set by previously obtaining the frequency response H by performing a DFT operation on the impulse response h.

The peak detection unit 74 detects the point where the amplitude becomes maximum in the filtered segment, that is, the peak point with sub-sample accuracy, and outputs the delay D and the amplitude A of the peak point. FIG. 13 illustrates a filtered segment for "R=10m". Curves in the figure represent continuous-time waveforms before sampling, and round dots indicate sampling points. A peak detector 74 calculates a peak position in a continuous time system based on the sampling series. In the example of FIG. 13, the peak position in sample units on the filtered segment {z _k : k=0, 1, . . . , wGate−1} is “k=34”. On the other hand, the peak position in a continuous-time waveform, that is, with sub-sample accuracy, is
D=R·Fsmp/(c/2)=34.157
is. The peak detector 74 estimates and calculates the delay D and the amplitude A of the peak point with sub-sample accuracy.

Various algorithms can be applied to peak point detection processing with sub-sampling precision. An example is shown below.
(Procedure 1) A sample point (point P in FIG. 13) having the maximum amplitude is obtained.
(Procedure 2) For the point (P point) obtained in Procedure 1 and the points before and after it (A point and B point), a quadratic curve passing through these three points is obtained.
(Procedure 3) Obtain the delay D and the amplitude A as the maximum point of the quadratic curve obtained in the procedure 2.

Based on the peak point information D, A (delay D, amplitude A) output from the peak detection unit 74, the determination unit 75 determines whether an object exists at the detection point. This determination is made by comparing the amplitude A of the peak point and the determination threshold "tDec". Specifically, when A>tDec, the determination unit 75 determines that “an object exists” and outputs the peak point information. On the other hand, when A≦tDec, the determination unit 75 determines that “the object does not exist” and does not output the peak point information.

The formatter 76 converts the peak point information D, A output from the determination unit 75 and the scanning information (frame index frm, segment index seg) corresponding to the peak points into a format that is easy for the user (upper system) to use. The formatter 76 in this embodiment performs the following format conversions.
(1) The frame index frm is output as it is.
(2) The segment index seg is converted into a horizontal scanning angle θ and output.
(3) Delay D is converted into radius (distance) R and output (R=D(c/2)/Fsmp).
(4) Amplitude A is output as it is.
(5) Calculate reflectance U from amplitude A and radius R and output (U=A/Ψ(R)).

Here, the function "Ψ(R)" for calculating the reflectance U is a reflectance conversion table, which can be set from the external CPU. By setting the same table to the expected value of the peak amplitude obtained from a Lambertian diffuser with a reflectance of 100% set at the radius R, it is possible to estimate the reflectance with little error.

As described above, the lidar 1 according to the first embodiment includes the scanning unit 55 that irradiates the emitted light Lo according to the pulse trigger signal PT while changing the irradiation direction, and the emitted light Lo that is arranged in a predetermined irradiation direction. It includes an absorber 7 that absorbs Lo, an APD 41 that receives return light Lr of the emitted light Lo, and a DSP 16 . The DSP 16 calculates an estimated synchronous disturbance w, which is a noise signal caused by the pulse trigger signal PT or the like, based on the output signal of the APD 41 when the irradiation direction of the emitted light Lo is the direction in which the absorber 7 is irradiated. A disturbance estimator 71 is provided. As a result, the rider 1 can suitably suppress deterioration in object detection performance and ranging performance due to synchronization disturbance.

<Second embodiment>
Next, a second embodiment will be described. The lidar 1 according to the second embodiment has a reflector instead of the absorber 7 . Then, the DSP 16 estimates the reference received pulse g by averaging the received segment y for the return light Lr reflected by the reflector. In the following description, the same elements as in the first embodiment will be appropriately given the same elements, and the description thereof will be omitted.

First, the effect of estimating the reference received pulse g will be described. When using the reference received pulse g that is measured in advance as a specific known reference pulse g and stored in the W register, errors due to individual differences, aging, etc. occur, and as a result, object detection performance and distance measurement performance deteriorate. There is a problem of lowering Specifically, when measuring the reference received pulse g in the development process of the lidar 1, there is a problem that individual differences in the transmission pulse shape and time delay of the LD 35 and individual differences in the receiver 40 cannot be accommodated. On the other hand, even when the reference received pulse g is measured in the manufacturing process of the lidar 1, there is a problem that it is not possible to cope with the above-described changes in the shape of the transmitted pulse over time and changes due to the ambient environment such as temperature. In consideration of the above, the DSP 16 according to the second embodiment estimates the reference received pulse g suitable for the current state of the rider 1, the surrounding environment, etc., thereby suitably suppressing deterioration in object detection performance and ranging performance. do.

FIG. 14A is a diagram schematically showing the arrangement of reflectors 8. FIG. In FIG. 14A, the reflector 8 is arranged near the housing 25 of the substantially cylindrical lidar 1 that accommodates the scanning unit 55 and the like. Here, the reflector 8 is directed in a non-detection direction (arrow A1). In the example of FIG. 14A, the reflector 8 exists on the wall surface of the housing 25 behind the rider 1 irradiated with the emitted light Lo of the angle "θb" (for example, 60 degrees). In this case, the reflector 8 is provided, for example, inside the transparent cover of the housing 25 that transmits the emitted light Lo and the returned light Lr. In another example, the reflector 8 may be a portion of the transparent cover of the housing 25 described above that is processed to reflect the emitted light Lo. Hereinafter, the period during which the APD 41 receives the emitted light Lo irradiated to the reflector 8 within the period in which one scan is performed by the scanning unit 55 (that is, one frame period) is also referred to as the "reference reflection period Ttag2". . The reference reflection period Ttag2 includes a plurality of segment periods corresponding to each scanning angle at which the reflector 8 is irradiated with the emitted light Lo. Information (for example, segment index, etc.) related to the scanning angle at which the reflector 8 is irradiated with the emitted light Lo is pre-stored in a W register or the like so that the DSP 16 can refer to it.

FIG. 14B shows a state in which the emitted light Lo is emitted in the direction in which the reflector 8 is arranged in the example of FIG. 14A. is reflected as return light Lr by the reflectance of , and the return light Lr reaches the scanning unit 55 . Here, the reflectance of the reflector 8 is such that when the return light Lr reflected by the reflector 8 is incident on the APD 41, the level of the analog voltage signal input to the ADC 20 falls within the input dynamic range of the ADC 20 (that is, , not too small for noise and no saturation occurs). In other words, the reflectance of the reflector 8 is such that the signal input to the ADC 20 when the APD 41 receives the return light Lr reflected by the reflector 8 can be distinguished from noise, and The reflectivity is such that the ADC 20 does not saturate. Since the reflector 8 has such a reflectance, the DSP 16 can accurately estimate the reference received pulse g.

FIG. 15 shows a block diagram of signal processing performed by the DSP 16 in the second embodiment. As shown in FIG. 15, the DSP 16 according to the second embodiment includes a reference received pulse estimator 80, a matched filter 81, a peak detector 74, a determiner 75, and a formatter . The DSP 16 sequentially reads out the received segments y from the received segment memory 15 and processes them.

The reference received pulse estimator 80 averages the received segments y generated within the reference reflection period Ttag2, and supplies the averaged received segments y to the matched filter 81 as the reference received pulse g. The reference received pulse estimator 80 has a switch 77A and an averaging processor 78A.

The switch 77A is a switch that is controlled to be ON only within the reference reflection period Ttag2, and supplies the reception segment y generated during the reference reflection period Ttag2 to the averaging processor 78A. Note that the switch 77A need not be turned on during the entire reference reflection period Ttag2, and may be set to be turned on during a part of the reference reflection period Ttag2.

The averaging processor 78A averages the received segments y supplied while the switch 77A is on, and supplies the averaged received segments y to the matched filter 81 as an estimated reference received pulse g. In this case, for example, the averaging processing unit 78A integrates received segments y sequentially supplied from the switch 77A within one frame period, and divides the integrated received segments y by the number of integrated received segments y. Compute the averaged received segment y as the estimated reference received pulse g. In another example, the averaging processing unit 78A averages the average of the received segments y calculated within one frame period in the frame direction (that is, between different frame indexes) using an IIR filter or the like to obtain an estimated value. calculated as a reference received pulse g.

Matched filter 81 includes a receive filter 73A and a time reversing section 79 . The time reversing section 79 generates an impulse response h from the reference received pulse g based on the relational expression A described above. In this case, the reference received pulse g and the impulse response h have a time-reversed relationship within the segment period, as shown in FIG. 12(A). The reception filter 73A convolves the impulse response h output by the time reversing unit 79 with the reception segment y to calculate the filtered segment z. In this case, the impulse response h supplied to the reception filter 73A is generated from the reference reception pulse g suitable for the current state of the rider 1, the surrounding environment, and the like. Therefore, the reception filter 73A outputs to the peak detection unit 74 the filtered segment z in which noise, time delay, etc. are accurately corrected, regardless of changes in the shape of the transmission pulse over time or changes due to the ambient environment such as temperature. be able to.

A peak detector 74 detects a peak point with the maximum amplitude within the filtered segment z, and outputs the delay D and the amplitude A of the peak point. The determination unit 75 selectively sends to the formatter 76 only points where the amplitude A is greater than the threshold value tDet. The formatter 76 converts the delay D, the amplitude A, the frame index frm of the segment, and the segment index seg into an appropriate format, and outputs it to the system CPU 5 as target information TI.

As described above, the lidar 1 according to the second embodiment includes the scanning unit 55 that irradiates the emitted light Lo while changing the irradiation direction, and the reflector 8 that is arranged in a predetermined irradiation direction and reflects the emitted light Lo. , an APD 41 for receiving return light Lr of the emitted light Lo reflected by the object, and a DSP 16 . The DSP 16 includes a reference received pulse estimator 80 for estimating the reference received pulse g used by the matched filter 81 based on the output signal of the APD 41 when the APD 41 receives the return light Lr reflected by the reflector 8. . As a result, the lidar 1 can appropriately estimate the reference received pulse g corresponding to changes in the shape of the transmitted pulse over time and changes in the surrounding environment such as temperature, thereby realizing a high SNR.

<Third embodiment>
Next, a third embodiment will be described. In the third embodiment, the DSP 16 performs both the calculation processing of the estimated synchronization disturbance w described in the first embodiment and the estimation processing of the reference received pulse g described in the second embodiment. In this case, the DSP 16 performs processing to eliminate the influence of the return light Lr reflected by the absorber 7, assuming that the reflectance of the absorber 7 is not the ideal value (that is, substantially 0%). As a result, the DSP 16 accurately estimates the estimated synchronization disturbance w and the reference received pulse g even when the return light Lr from the absorber 7 affects the received segment y.

First, an arrangement example of the absorber 7 and the reflector 8 will be described. FIG. 16A is a diagram schematically showing the arrangement of absorbers 7 and reflectors 8. FIG.

In FIG. 16A, the absorber 7 and the reflector 8 are arranged near the housing 25 of the substantially cylindrical lidar 1 . In this case, the distance of the absorber 7 with respect to the scanning unit 55 and the distance of the reflector 8 with respect to the scanning unit 55 are approximately equal. Further, the absorber 7 and the reflector 8 are the target for the rider 1 to detect the object in the 360-degree irradiation direction of the emitted light Lo scanned by the scanning unit 55, as in the first and second embodiments. are provided in the non-detection direction (see arrow A1) other than the direction in which they are detected. Although the absorber 7 and the reflector 8 are provided adjacent to each other in FIG. 16A, they may be provided apart from each other by a predetermined distance.

Next, the influence of the return light Lr reflected by the absorber 7 will be described. FIG. 16(B) shows a state in which the emitted light Lo is emitted in the direction in which the absorber 7 is arranged in the example of FIG. 16(A).

In the case of FIG. 16B, ideally, it is desirable that the reflectance of the absorber 7 is substantially zero and the intensity of the return light Lr received by the APD 41 is substantially zero. However, in practice, it is difficult to prepare an absorber 7 having such a reflectance, and even if an absorber 7 having such a reflectance is provided, it will not change over time. As a result, it is conceivable that the reflectance gradually deviates from the ideal value. In this case, the return light Lr from the absorber 7, which has a non-negligible amount of light, reaches the scanning unit 55 and affects the receiving segment y.

17A to 17C respectively show the reflection component from the absorber 7, the synchronous interference component, and the reflection component from the reflector 8, which are superimposed on the receiving segment y. Here, in the reflection suppression period Ttag1, the waveform of the reception segment y is the sum of the reflection component from the absorber 7 shown in FIG. noise is also added). Therefore, in the method of calculating the estimated synchronization disturbance w by the synchronization disturbance estimation unit 71 described in the first embodiment, in addition to the synchronization disturbance component shown in FIG. 17(B) in the estimated synchronization disturbance w, Since the reflection component from the absorber 7 is included, it becomes impossible to accurately estimate the synchronous interference component.

On the other hand, since the absorber 7 and the reflector 8 are arranged at approximately the same distance from the scanning unit 55, the reflected component from the absorber 7 is and the peak position of the reflected component from the reflector 8 are substantially equal. In consideration of the above, as will be described later, the DSP 16 responds to the reception segment y (that is, the estimated synchronization interference w on which the reflection component from the absorber 7 is superimposed) obtained in the reflection suppression period Ttag1 as shown in FIG. 17(C). By calculating the correlation with the reflected component from the reflector 8 shown in FIG. 17A, a replica of the reflected component from the absorber 7 shown in FIG. Subtracting the replicas of the components preferably estimates only the synchronous disturbance.

FIG. 18 shows a block diagram of signal processing performed by the DSP 16 in the third embodiment. As shown in FIG. 18, the DSP 16 according to the third embodiment includes a synchronization disturbance first estimator 71X, a synchronization disturbance second estimator 71Y, a subtractor 72, a reference received pulse estimator 80A, and a matched filter 81A. , a peak detection unit 74 , a determination unit 75 , and a formatter 76 . The DSP 16 sequentially reads out the received segments y from the received segment memory 15 and processes them.

The synchronization interference first estimating unit 71X performs the same processing as the synchronization interference estimating unit 71 of the first embodiment, so that the received segment y within the reflection suppression period Ttag1 corresponds to the estimated synchronization interference w of the first embodiment. to generate an uncorrected estimated synchronous disturbance 'v'. Specifically, the synchronization disturbance first estimator 71X has a switch that is turned on only during the reflection suppression period Ttag1, and extracts the reception segment y during the reflection suppression period Ttag1. Then, the synchronization disturbance first estimating unit 71X performs the same averaging processing as the averaging processing unit 78 of the first embodiment on the extracted reception segment y in the reflection suppression period Ttag1, so that the real number of the vector length wGate Generate the uncorrected estimated synchronous disturbance v, which is a vector.

At the end of the reflection suppression period Ttag1, the synchronization disturbance second estimator 71Y calculates the reference received pulse g estimated by the reference received pulse estimator 80A described later and the pre-correction estimated synchronization disturbance v calculated by the first synchronization disturbance estimator 71X. , the estimated synchronous interference w is calculated by removing the reflection component from the absorber 7 shown in FIG. 17A from the pre-correction estimated synchronous interference v. A block configuration example of the synchronization disturbance second estimation unit 71Y will be described later. In the period until the second synchronous disturbance estimator 71Y calculates the estimated synchronous disturbance w, the second synchronous disturbance estimator 71Y supplies the pre-correction estimated synchronous disturbance v to the subtractor 72 as the estimated synchronous disturbance w.

As in the first embodiment, the subtractor 72 subtracts the estimated synchronization disturbance w from the received segment y and supplies the corrected received segment _{y_dash} to the reference received pulse estimator 80A and the matched filter 81A, respectively. In this case, the corrected received segment y _dash has the synchronization disturbing component shown in FIG. Note that during the period in which the pre-correction estimated synchronous disturbance v including the reflection component from the absorber 7 shown in FIG. 17A is input to the subtractor 72 as the estimated synchronous disturbance w, the subtractor 72 , the reflection component of the absorber 7 shown in FIG. 17(A) is subtracted in addition to the synchronous interference component shown in FIG. 17(B). As a result, the corrected receive segment _{y_dash} will be over-subtracted by the reflection component of the absorber 7 shown in FIG. 17(A).

The reference received pulse estimator 80A estimates the reference received pulse g by performing the same processing as the reference received pulse estimator 80 of the second embodiment on the corrected received segment _{y_dash} . Specifically, the reference received pulse estimator 80A has elements corresponding to the switch 77A and the averaging processor 78A of the second embodiment, and extracts the corrected received segment _{y_dash} generated within the reference reflection period Ttag2. and estimate the averaged corrected receive segment _{y_dash} as the reference receive pulse g. The reference received pulse estimator 80A supplies the estimated reference received pulse g to the synchronization disturbance second estimator 71Y and the matched filter 81A.

Here, the corrected received segment _{y_dash} supplied to the reference received pulse estimator 80A is absorbed as shown in FIG. Only the reflection component of the body 7 is excessively subtracted. Therefore, during this period, the reference received pulse estimator 80A supplies the reference received pulse g obtained by excessively subtracting the reflection component of the absorber 7 shown in FIG. 17(A) to the synchronization disturbance second estimator 71Y.

The matched filter 81 A performs the same processing as the matched filter 81 of the second embodiment on the corrected received segment y _dash to generate a filtered segment z and supplies it to the peak detector 74 . Specifically, the matched filter 81A has elements corresponding to the reception filter 73A and the time reversing section 79 of the second embodiment, and generates an impulse response h based on the relational expression A from the reference reception pulse g, and then corrects it. Compute the filtered segment z by convolving the impulse response h with the received segment y _dash .

A peak detector 74 detects a peak point with the maximum amplitude within the filtered segment z, and outputs the delay D and the amplitude A of the peak point. The determination unit 75 selectively sends to the formatter 76 only points where the amplitude A is greater than the threshold value tDet. The formatter 76 converts the delay D, the amplitude A, the frame index frm of the segment, and the segment index seg into an appropriate format, and outputs it to the system CPU 5 as target information TI.

FIG. 19 shows a signal processing block executed by the synchronization disturbance second estimator 71Y. As shown in FIG. 19, the synchronization disturbance second estimator 71Y has a correlation calculator 86, a replica generator 87, and a subtracter 88. Also, in FIG. 19, the waveforms of each input and output during the period in which the pre-correction estimated synchronization disturbance v is input to the subtractor 72 as the estimated synchronization disturbance w are shown within dashed-dotted lines 90 to 93 .

The correlation calculator 86 calculates the correlation between the pre-correction estimated synchronous disturbance v supplied from the first synchronous disturbance estimator 71X and the reference received pulse g supplied from the reference received pulse estimator 80A. 7 (see FIG. 17 ₍ A)). Specifically, as shown in the following equation, the correlation calculator 86 calculates the pre-correction estimated synchronization disturbance v, which is a real vector whose vector length is equal to the gate width wGate (here, 1024), and the normalized reference received pulse g The amplitude estimate value A _dark is calculated by calculating the scalar product (inner product) with .

ここで、上式に基づく振幅推定値Ａ_ｄａｒｋの算出の妥当性について説明する。基準受信パルスｇは、図１７（Ａ）に示す吸収体７の反射成分だけ過剰に減算された状態で基準受信パルス推定部８０Ａから相関算出部８６に供給される。従って、この場合、基準受信パルスｇは、一点鎖線枠９１内に示されるように、図１７（Ｃ）に示す反射体８からの反射成分と吸収体７からの反射成分に起因した過剰減算分との和に相当する。また、相関算出部８６に供給される補正前推定同期妨害ｖは、一点鎖線枠９０内に示されるように、図１７（Ａ）に示した吸収体７からの反射成分と、図１７（Ｂ）に示した同期妨害成分との和に相当する。この場合、反射体８からの反射成分と吸収体７からの反射成分に起因した過剰減算分とは、ピーク位置が略一致し、これらの和に相当する基準受信パルスｇのピーク位置は、吸収体７からの反射成分のピーク位置と略一致することになる。従って、相関算出部８６は、上式に基づき、基準受信パルスｇと補正前推定同期妨害ｖとの相関を算出することで、図１７（Ａ）に示した吸収体７からの反射成分のピーク位置の振幅を的確に推定することができる。なお、上述の振幅推定値Ａ_ｄａｒｋの算出式では、ルート演算を避け、かつ、後述するレプリカｕの算出の際の基準受信パルスｇの大きさによる除算を省くため、補正前推定同期妨害ｖと基準受信パルスｇとのスカラー積が基準受信パルスｇの大きさの２乗により除算されている。

Here, the validity of the calculation of the amplitude estimated value A _dark based on the above equation will be explained. The reference received pulse g is supplied from the reference received pulse estimator 80A to the correlation calculator 86 in a state in which the reflection component of the absorber 7 shown in FIG. 17A is excessively subtracted. Therefore, in this case, the reference received pulse g is, as shown in a dashed-dotted line frame 91, an excessive subtraction resulting from the reflected component from the reflector 8 and the reflected component from the absorber 7 shown in FIG. 17(C). corresponds to the sum of Further, the pre-correction estimated synchronization disturbance v supplied to the correlation calculator 86 is, as shown in a dashed-dotted line frame 90, the reflected component from the absorber 7 shown in FIG. ) corresponds to the sum with the synchronous disturbance component shown in ). In this case, the peak positions of the reflection component from the reflector 8 and the excess subtraction due to the reflection component from the absorber 7 substantially coincide, and the peak position of the reference received pulse g corresponding to the sum of these peak positions is the absorption The peak position of the reflection component from the body 7 is approximately matched. Therefore, the correlation calculator 86 calculates the correlation between the reference received pulse g and the pre-correction estimated synchronization disturbance v based on the above equation, thereby obtaining the peak of the reflection component from the absorber 7 shown in FIG. 17(A). The amplitude of the position can be accurately estimated. In the above equation for calculating the amplitude estimated value A _dark , in order to avoid the root calculation and to omit the division by the size of the reference received pulse g when calculating the replica u, which will be described later, the pre-correction estimated synchronization disturbance v and The scalar product with the reference received pulse g is divided by the square of the magnitude of the reference received pulse g.

The replica generator 87 generates a replica “u” of the reflected component from the absorber 7 shown in FIG. 17A based on the amplitude estimated value A _dark output from the correlation calculator 86 and the reference received pulse g. Replica u is a real vector whose vector length is equal to gate width wGate. Considering that the peak position of the reference received pulse g substantially coincides with the peak position of the reflected component from the absorber 7 shown in FIG. Then, a replica u is generated so that the peak position coincides with the reference received pulse g and the amplitude at the peak position corresponds to the amplitude estimated value A _dark .

このようにすることで、レプリカ生成部８７は、一点鎖線枠９２内に示すように、図１７（Ａ）に示した吸収体７からの反射成分のレプリカを好適に生成することができる。

By doing so, the replica generator 87 can suitably generate a replica of the component reflected from the absorber 7 shown in FIG.

A subtractor 88 subtracts the replica u generated by the replica generator 87 from the pre-correction estimated synchronization disturbance v. FIG. In this case, as shown in a dashed-dotted line frame 93, the estimated synchronous disturbance w has the reflection component from the absorber 7 shown in FIG. 17(A) preferably removed from the estimated synchronous disturbance v before correction. After that, the estimated synchronization disturbance w is supplied to the subtractor 72, which subtracts the estimated synchronization disturbance w from the received segment y to generate the corrected received segment _{y_dash} , which is applied to the reference received pulse estimator 80A and the matched filter 81A. supply. In this case, the reference received pulse estimator 80A generates a reference received pulse g (see FIG. 17(C)) from which the influence of the reflected component from the absorber 7 shown in FIG. 17(A) is removed. The matched filter 81A filters the correction reception segment _{y_dash} from which the synchronization interference component has been removed, based on the reference reception pulse g from which the influence of the reflected component from the absorber 7 has been removed. As a result, the matched filter 81A outputs to the peak detector 74 the filtered segment z in which noise, time delay, etc. have been accurately corrected.

Here, in the block configuration of the DSP 16 shown in FIG. 18, the effect of providing the synchronization disturbance second estimator 71Y will be additionally explained. Instead of the configuration shown in FIG. 18, if there is a configuration without the synchronization interference second estimator 71Y (that is, the synchronization interference estimator 71 of the first embodiment and the reference received pulse estimator 80 of the second embodiment are 17(A) from the absorber 7 shown in FIG. 17(A) in addition to the synchronous interference component shown in FIG. A reflection component is included. In this case, even during the scanning period of the detection target direction of the lidar 1, the corrected reception segment y _dash is generated based on the estimated synchronization disturbance w including the reflection component from the absorber 7, so the reflection based on the corrected reception segment y _dash Errors will occur in rate and range estimates. Further, the reference reception pulse g calculated in the reference reflection period Ttag2 includes an excessive subtraction due to the reflection component from the absorber 7, as shown in the dashed-dotted line frame 91 in FIG. The estimation accuracy of the pulse g is degraded. In consideration of the above, the DSP 16 according to the third embodiment includes the synchronous disturbance second estimator 71Y for removing the reflected component from the absorber 7 from the estimated synchronous disturbance w. It is possible to suitably suppress the occurrence of errors and the like caused by

As described above, the lidar 1 according to the third embodiment includes the scanning unit 55 that irradiates the emitted light Lo while changing the irradiation direction, and the reflector 8 that is arranged in the first irradiation direction and reflects the emitted light Lo. , arranged in the second irradiation direction, and includes an absorber 7 that absorbs the emitted light Lo, an APD 41 that receives the return light Lr, and a DSP 16 . The DSP 16 generates a replica u of the reflection component of the absorber 7 based on the output signals of the APD 41 when the irradiation direction of the emitted light Lo is the first irradiation direction and the second irradiation direction. As a result, the rider 1 can estimate the estimated synchronous disturbance w by removing the replica u from the pre-correction estimated synchronous disturbance v, and suitably suppresses the occurrence of errors due to the reflection component from the absorber 7. be able to.

1 Rider 5 System CPU
7 absorber 8 reflector 10 ASIC
30 transmitter 40 receiver 50 scanning optics

Claims (1)

an irradiation unit that irradiates laser light according to the first signal while changing the direction of irradiation;
an absorber arranged in a predetermined irradiation direction and absorbing the laser light;
An information processing device for processing an output signal of the light receiving unit of an optical device, comprising:
An information processing apparatus comprising an estimation unit that estimates a noise signal caused by the first signal based on the output signal when the irradiation direction of the laser light is the predetermined irradiation direction.

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