JP2022051152A - Open winding motor drive device and refrigeration cycle device - Google Patents

Abstract

To provide an open winding motor drive device capable of suppressing the generation of a zero-axis current and further increasing an output voltage.SOLUTION: An open winding motor drive device includes primary and secondary inverters connected to the terminals of an open winding structure motor, and a control unit that controls a current and rotation speed that energize the motor, and the control unit includes a zero-axis current suppression unit that controls the current and rotation speed of the motor and suppress a zero-axis current that flows between the primary and secondary inverters, and the zero-axis current suppression unit divides a space voltage vector consisting of 64 voltage vectors into 12 sectors, and only one first switching pattern that generates a voltage applied to the motor without generating a zero-axis voltage to be used according to each sector is selected during a control cycle.SELECTED DRAWING: Figure 1

Description

An embodiment of the present invention relates to a device for driving a motor having an open winding structure, and a refrigeration cycle device including the device.

For example, when driving an AC motor such as a permanent magnet synchronous motor, it is necessary to convert a DC power source into three-phase AC power by using an inverter. However, as the capacity of the motor increases, the current flowing through the inverter also increases, which causes problems such as heat generation in the power device constituting the inverter.

To solve this problem, for example, Patent Document 1 proposes a system in which the windings of a three-phase motor are opened in an open state without being connected in a star shape, and inverters are connected to both ends of the three-phase windings to drive the motors. There is. According to this system, by using two inverters, the voltage that can be applied to both ends of the three-phase winding can be doubled, so that the motor can be driven at a higher speed. Alternatively, by increasing the number of turns of the winding, it is possible to drive a motor that outputs a high torque with a small current.

Further, Patent Document 1 also proposes a technique for suppressing a zero-axis current commonly flowing in three phases of a motor, which is generated by adopting a configuration in which a DC link voltage is shared between two inverters.

Japanese Unexamined Patent Publication No. 2020-31458

However, although the details will be described later, in the configuration of Patent Document 1, when the output voltage of a normal inverter is set to "1", it can output only up to √3 times that.
Therefore, we provide an open winding motor drive device capable of suppressing the generation of zero-axis current and further increasing the output voltage, and a refrigeration cycle device including the device.

In the open winding motor drive device of the embodiment, the three-phase windings are independent of each other, and are connected to three output terminals of the six output terminals of the motor having an open winding structure having six output terminals. With the next inverter
A secondary inverter connected to the remaining three output terminals of the motor,
A control unit that controls the current and rotation speed that energize the motor based on the line-to-line duty ratio of each of the primary and secondary inverters in PWM control.
It is equipped with a current detector that detects the current energized in the motor.
The control unit controls the current and rotation speed of the motor based on the duty between the lines of the primary side and secondary side inverters, and has three phases in the same direction between the primary side and secondary side inverters. It has a zero-axis current suppression unit that suppresses the zero-axis current flowing through the inverter.
The zero-axis current suppression unit has a space voltage vector composed of 64 voltage vectors, which is a combination of on / off patterns of the primary side and secondary side inverters.
Centering on the point where two second switching patterns that do not generate a zero-axis voltage and do not generate a voltage that acts between the phases of the motor are located, and do not generate a zero-axis voltage that acts equally on the three phases of the motor. The first switching pattern that generates the voltage applied to the motor is divided into 12 sectors, which are further divided into 12 sectors, with the 6 regions divided by the points where two of them are located as the vertices, and used according to each sector. Only one switching pattern is selected during the control cycle.
Further, in the refrigeration cycle apparatus of the embodiment, a motor having an open winding structure in which three-phase windings are independent and has six winding terminals, and a motor having an open winding structure,
The open winding motor drive device of the embodiment is provided.

The figure which is 1st Embodiment and shows the structure of the motor drive system. Functional block diagram showing the configuration of the current control unit Functional block diagram showing the configuration of an air conditioner The figure which shows the waveform of a motor U-phase current and a zero-axis current. A magnified view of a part of FIG. The figure which shows the change of the zero-axis voltage which occurs with the switching of the primary side and secondary side inverters. The figure which shows the space voltage vector corresponding to the configuration which drives a general three-phase motor. The figure which shows the space voltage vector corresponding to the configuration which drives an open winding motor. The figure which divides the space voltage vector into 6 sectors and shows the 1st and 2nd vector patterns used in each sector. The figure which divides the space voltage vector into 12 sectors and shows the 1st vector pattern used in each sector. Figure showing the first vector pattern used in each sector in chronological order (No. 1) Figure showing the first vector pattern used in each sector in chronological order (Part 2) The figure which is 2nd Embodiment and shows the structure of the motor drive system. Functional block diagram showing the configuration of the zero-axis voltage generator The figure which shows the vector pattern used when the zero axis current is reduced in a sector 6. The figure which shows the vector pattern used when increasing the zero axis current in sector 6. The figure which shows the vector pattern used when the zero axis current is reduced in a sector 7. The figure which shows the vector pattern used when increasing the zero axis current in a sector 7. Functional block diagram showing the configuration of the space voltage vector modulator The figure which shows the waveform of each phase current and zero axis current of a motor in the control of 1st Embodiment. The figure which shows the waveform of each phase current and zero axis current of a motor in the control of 2nd Embodiment FIG. 3 is a diagram showing a configuration of a motor drive system according to a third embodiment. Functional block diagram showing the configuration of the dq current control unit Flow chart showing switching process from normal pulse control to synchronous pulse control Flow chart showing switching process from synchronous pulse control to normal pulse control The figure which conceptually shows the process of steps S5 to S7. A diagram conceptually showing the process of increasing the voltage amplitude from the initial value to the target value. A diagram conceptually showing the process of switching from the space voltage vector used for normal pulse control to the space voltage vector used for synchronous pulse control. The figure which shows the simulation result in the case of transitioning like normal pulse control-> synchronous pulse control-> normal pulse control.

(First Embodiment)
Hereinafter, the first embodiment will be described with reference to FIGS. 1 to 12. The present embodiment is based on the technique disclosed in Patent Document 1, and is an improvement thereof. FIG. 1 is a diagram showing a circuit configuration of the motor drive system of the present embodiment. The motor M is assumed to be a three-phase permanent magnet synchronous motor, an inducer, or the like, but in the present embodiment, it is a permanent magnet synchronous motor. The three-phase windings of the motor M are not connected to each other and both terminals are in an open state. That is, the motor M includes six winding terminals Ua, Va, Wa, Ub, Vb, and Wb.

The primary side inverter 1 and the secondary side inverter 2 are configured by connecting N-channel MOSFETs 3 which are switching elements in a three-phase bridge, respectively, and these are connected in parallel to the DC power supply 4. The DC power supply 4 may be a DC power source converted into a direct current. Each phase output terminal of the inverter 1 is connected to the winding terminals Ua, Va, Wa of the motor M, respectively, and each phase output terminal of the inverter 2 is connected to the winding terminals Ub, Vb, Wb, respectively.

The position sensor 6 is a sensor that detects the rotor rotation position and rotation speed of the motor M, and the current sensor 7 (U, V, W) is a sensor that detects each phase current Iu, Iv, Iw of the motor M. , Corresponds to a current detector. The voltage sensor 8 detects the voltage V _DC of the DC power supply 4.

The control device 11 is given a speed command value ω _Ref from a higher-level control device in the system for driving the motor, and controls so that the detected motor speed ω matches the speed command value ω _Ref . The control device 11 switches the gates of the FETs 3 constituting the inverters 1 and 2 based on the phase currents Iu, Iv, Iw detected by the current sensor 7 and the DC voltage _VDC detected by the voltage sensor 8. Generate a signal. The control device 11 corresponds to a control unit.

The current detection / coordinate conversion unit 12 converts the detected phase currents Iu, Iv, and Iw into the currents Id and Iq of the d and q-axis coordinates used for vector control by the equation (1). The conversion of the zero-axis current I0 shown in the equation (1) is performed by the current detection / coordinate conversion unit 43 in the second embodiment.

The speed / position detection unit 13 detects the motor speed ω and the rotor rotation position θ from the signal detected by the position sensor 6. The rotation position θ is input to the current detection / coordinate conversion unit 12 and the dq / αβ conversion unit 17. Further, the speed / position detection unit 13 may be configured to estimate the speed and position from the voltage / current of the motor M. The speed control unit 14 generates and outputs the q-axis current command I _qRef from the input speed command ω _Ref and the speed ω, for example, by performing PI calculation on the difference between the two.

The current control unit 16 generates and outputs d, q-axis voltage commands Vq, Vd from the input q-axis current commands I _qRef , DC voltage V _DC , detected currents Id, Iq, and motor speed ω. .. The dq / αβ conversion unit 17 converts the dq-axis voltage commands Vq and Vd into αβ-axis voltages Vα and Vβ by the equation (2).

The space vector modulation unit 18 performs a space vector operation from the αβ axis voltages Vα and Vβ, and performs space vector operations on the phase duty _Du1 , D _v1 , D _w1 of the inverter 1 and the phase duty D _u2 , D _v2 , D _w2 of the inverter 2. Is generated and input to the PWM signal generation unit 19. The PWM signal generation unit 19 generates switching signals, PWM signals U1 ±, V1 ±, W1 ±, U2 ±, V2 ±, W2 ±, which are given to the gates of each FET 3 constituting the inverters 1 and 2 from the input phase duty. Generate and output.

In this embodiment, the modulation factor of the inverters 1 and 2 is 1.0. FIG. 2 shows the detailed configuration of the current control unit 16. The subtractor 21 takes a difference between the q-axis current command I _qRef obtained as a result of the speed control and the q-axis current Iq, and the PI control unit 22 performs a PI control operation on the difference. The calculation result is output as the voltage phase θ _{V_PI} . The non-interference term calculation unit 23 sets the voltage phase non-interference control term θ _{V_FF} from the motor constant, the d-axis current command I _dRef , the q-axis current command I _qRef , and the speed ω, which are set equal to the d-axis current Id. Calculate.

Further, the adder 24 adds the non-interference control term θ _{V_FF} to the voltage phase θ _{V_PI} to generate the final voltage phase θ _V. The d and q-axis voltage commands Vd and Vq are generated by the dq-axis voltage calculation unit 25 by the output voltage amplitude V _amp corresponding to the DC voltage V _DC and the voltage phase θ _V. As a result, the DC voltage can be directly applied to the motor M, and when the load changes, the motor M can be controlled at high speed by adjusting the voltage phase.

FIG. 3 shows the configuration of the air conditioner 30 to which the motor drive system of the present embodiment is applied. The compressor 32 constituting the heat pump system 31 is configured by accommodating the compression unit 33 and the motor M in the same iron closed container 35, and the rotor shaft of the motor M is connected to the compression unit 33. The compressor 32, the four-way valve 36, the indoor heat exchanger 37, the decompression device 38, and the outdoor heat exchanger 39 are connected so as to form a closed loop by a pipe serving as a heat transfer medium flow path. The compressor 32 is, for example, a rotary type compressor. The air conditioner 30 includes the heat pump system 31 described above.

At the time of heating, the four-way valve 36 is in the state shown by the solid line, and the high-temperature refrigerant compressed by the compression unit 33 of the compressor 32 is supplied from the four-way valve 36 to the indoor heat exchanger 37 to condense, and then the decompression device. The pressure is reduced by 38, the temperature becomes low, and the heat flows to the outdoor heat exchanger 39, where it evaporates and returns to the compressor 32. On the other hand, during cooling, the four-way valve 36 is switched to the state shown by the broken line. Therefore, the high-temperature refrigerant compressed by the compression unit 33 of the compressor 32 is supplied from the four-way valve 6 to the outdoor heat exchanger 39 to be condensed, and then decompressed by the decompression device 38 to become low temperature and indoor side. It flows through the heat exchanger 37, where it evaporates and returns to the compressor 32. The fans 40 and 41 blow air to the indoor and outdoor heat exchangers 37 and 39, respectively, and the heat exchange between the heat exchangers 37 and 39 and the indoor air and outdoor air is efficient. It is configured to be done well.

Next, the operation of this embodiment will be described with reference to FIGS. 4 to 12. In order to operate the open winding motor M, a voltage is applied to each terminal Ua, Va, Wa, Ub, Vb, Wb by two inverters 1 and 2. The voltage obtained as a result of the speed control and the current control is divided into voltage commands to the inverters 1 and 2 by the dq / αβ conversion unit 17, the space voltage vector modulation unit 18, and the PWM signal generation unit 19. The phase duty _Du1 , D _v1 , D _w1 of the inverter 1 and the phase duty D _u2 , D _v2 , D _w2 of the inverter 2 are energized as PWM signals having a phase difference of 180 ° from each other. In this way, by applying a voltage of opposite phase to the motor M by the two inverters 1 and 2, the voltage amplitude per phase can be increased, and the motor M can be rotated at a higher speed.

As described above, in the configuration in which the inverters 1 and 2 share the DC link portion, the zero-axis current flowing in the same direction in the three phases becomes a problem. The zero-axis current is a low-frequency current that flows at a frequency component that is three times the fundamental frequency of the phase current that is energized in the motor M, and a carrier frequency component that flows in synchronization with the switching of the inverters 1 and 2. Divided into. FIG. 4 shows the U-phase current Iu of the open winding motor that flows when the zero-axis current is not suppressed, and FIG. 5 shows the zero-axis current I0. FIG. 5 is an enlargement of the time axis of FIG. It is a current waveform shown by. Ripple that changes at the same timing can be confirmed in the U-phase current Iu and the zero-axis current I0, which is the zero-axis current of the carrier frequency component. Further, since the zero-axis current I0 pulsates at a component three times the fundamental wave frequency of the phase current, the distortion of the U-phase current Iu is large.

V0_ripple can be obtained by subtracting the average of the three-phase voltage of the inverter 2 from the average of the three-phase voltage of the inverter 1 as in the equation (3). The phase voltages Vu1, Vv1, Vw1, Vu2, Vv2, and Vw2 are _VDC when the FET 3 is on, and 0 when the FET 3 is off.

As shown in FIG. 6, the waveform of V0_ripple fluctuates positively and negatively according to the switching state, the zero-axis current I0 increases during the period occurring on the positive side, and zero during the period occurring on the negative side. The shaft current I0 is decreasing. Therefore, when V0_ripple becomes zero, the ripple of the zero-axis current I0, that is, the fluctuation of the carrier frequency component disappears. Further, the generation state of V0_ripple depends on the number of phases in which the FET 3 of the inverters 1 and 2 is on, and when the number of on-phases of the inverters 1 and 2 is different, it is generated positively or negatively according to the difference. That is, if the number of on-phases of the inverters 1 and 2 is the same, V0_ripple will not occur.

Here, in order to examine the switching pattern of the inverters 1 and 2 for achieving the above-mentioned object, the space voltage vector is examined. FIG. 7 shows a space vector when a general motor is energized by a three-phase inverter. For example, V1 (100) indicates that the U-phase upper arm is on and the V and W-phase upper arms are off, and there are eight vectors V0 to V7.

On the other hand, FIG. 8 shows the space voltage vector of the open winding motor, and since there are two inverters, the switching pattern is 8 × 8 = 64 patterns. For example, the combination in which the inverter 1 is V1 and the inverter 2 is V4 is described as "V14". In the space vector of an open winding motor, there are innumerable patterns of voltage vectors for outputting a certain command voltage. For example, in order to output the vector indicated by the arrow in FIG. 8, the energization time of any of the voltage vectors V06, V21, V30, V37, V45, V76 and the energization time of the voltage vector V01, V32, V40, V47, V56, V71 It can be output by adjusting the energization time of any of the voltage vectors.

Here, considering the relationship between the zero-axis voltage and the space vector, among the 64 patterns, the vector pattern that generates the voltage applied to the motor M and does not generate the zero-axis voltage that acts equally on the three phases, that is, on. There are 12 patterns of V15, V24, V26, V35, V31, V46, V42, V51, V53, V62, V64, and V13 in which at least two of the phases that have the same number of phases and are turned on do not match. FIG. 9 shows these patterns represented by spatial vectors.

In FIG. 9, the PWM waveform corresponding to each voltage vector is also shown. The above 12 patterns are paired with each other and placed at the vertices to draw a regular hexagon and divided into 6 sectors. For example, in order to output the vector belonging to the sector 4 indicated by the arrow in FIG. 9, the energization time of each of the voltage vectors V42 and V31 is adjusted. The PWM waveform of each voltage vector is
V42: Inverter 1 (U, V, W) = (off, on, on)
Inverter 2 (U, V, W) = (on, on, off)
V31: Inverter 1 (U, V, W) = (off, on, off)
Inverter 2 (U, V, W) = (on, off, off)
Is. To these, V77 for turning on all phases of the inverters 1 and 2 and V00 for turning off all phases are added. As can be seen from the PWM waveforms of each vector, since the on-phase numbers of the inverters 1 and 2 match, the zero-axis voltage V0 does not occur. That is, if energization is performed using this PWM switching pattern, the ripple of the carrier component of the zero-axis current shown in FIGS. 4 and 5 can be suppressed.

In FIG. 9, the following first and second vector patterns are used. The first and second vector patterns correspond to the first and second switching patterns.
<First vector pattern>
A pattern that generates a voltage applied to the motor M and does not generate a zero-axis voltage that acts equally on the three phases. 12 patterns of the above V15, V24, V26, V35, V31, V46, V42, V51, V53, V62, V64, V13 <second vector pattern>
A pattern in which a voltage acting between the phases of the motor M is not generated, and a zero-axis voltage acting equally on the three phases is not generated. V77 and V00 are second vector patterns in all sectors.

FIG. 9 corresponds to FIG. 8 of Patent Document 1, and the output order of the vector patterns in one control cycle is “second, first, first, second”. However, in the PWM switching pattern shown in FIG. 9, when the output of the normal inverter is 1, the inside of the inscribed circle of a regular hexagon connecting the points where two first switching patterns in FIG. 9 are located is driven. It becomes a range. With this, the voltage can only be output up to √3 times that of a normal inverter.

Therefore, in the present embodiment, the space voltage vector modulation unit 18 generates only one 12th first vector pattern during the control cycle. FIG. 10 shows these patterns represented by spatial vectors. In FIG. 10, the six sectors shown in FIG. 9 are further divided into 12 sectors, and the voltage vector is switched every 30 degrees of the electric angle according to each sector. The control cycles shown in FIGS. 9 and 10 have the same length.

For example, in order to output the vector belonging to the sector 7 indicated by the arrow in FIG. 10, only the voltage vector V42 is selected. The PWM waveform of the voltage vector is
V42: Inverter 1 (U, V, W) = (off, on, on)
Inverter 2 (U, V, W) = (on, on, off)
Is. As a result, the six points on the circumscribed circle of the regular hexagon shown in FIG. 10 become the driving points, and √3 × 2 / √3 = 2, so that the output voltage can be doubled that of a normal inverter. Since the on-phase numbers of the inverters 1 and 2 match, the zero-axis voltage V0 does not occur, and the ripple of the carrier component of the zero-axis current can be suppressed. The space voltage vector modulation unit 18 corresponds to a zero-axis current suppression unit.

FIG. 11 shows a motor three-phase voltage waveform corresponding to each voltage vector for one round of the electric angle. The motor phase voltage waveform obtained by the switching pattern of the inverters 1 and 2 becomes a square wave voltage of 120 degrees, and a DC voltage can be directly applied to the motor to increase the output as compared with the conventional pseudo sine wave voltage. In addition, since the number of switchings is small, the switching loss of the FET 3 can be reduced.

Further, a part of the switching pattern can be changed for simplification. In FIG. 12, for example, the voltage vector of the odd-numbered sector is replaced with the voltage vector of the next even-numbered sector.

As described above, according to the present embodiment, the motor M having an open winding structure in which the three-phase windings are independent and has six output terminals Ua to Wb is the primary side inverter 1 and the secondary side inverter 2. In the configuration driven by the above, the control device 11 controls the current and the rotation speed of the motor M based on the duty between the lines of the inverters 1 and 2, and includes the space voltage vector modulation unit 18.

The space voltage vector modulation unit 18 has a space voltage vector consisting of 64 voltage vectors, which is a combination of on / off patterns of the inverters 1 and 2, centered on a point where two second switching patterns are located, and the first switching pattern is each. The 6 regions divided by the points located two by two as vertices are further divided into 12 sectors, and only one pattern is selected during the control cycle as the first switching pattern to be used according to each sector.

As a result, the zero-axis current of the carrier component flowing in synchronization with the switching of the inverters 1 and 2 is suppressed, and the output voltage to the motor M is compared with the configuration in which a normal three-phase motor is driven by one inverter. Can be doubled. Further, since the space voltage vector modulation unit 18 selects the same first switching pattern for two adjacent sectors, control can be performed more easily. In addition, by applying the open winding motor drive device of the present embodiment to the air conditioner 30, air conditioning operation can be performed with high output and high efficiency.

(Second Embodiment)
Hereinafter, the same parts as those in the first embodiment are designated by the same reference numerals, the description thereof will be omitted, and different parts will be described. First, suppression of the zero-axis current flowing at a component three times the fundamental frequency will be described. Equation (4) is a relational expression between the dq0 axis voltage and the current of the open winding motor.

Here, it can be seen that when the dq-axis currents Id and Iq flow, the zero-axis voltage V0 is generated due to the influence of the diagonal elements shown in Eq. (4). This is the interference from the dq axis to the 0 axis, and the zero axis current I0 flows as a result of the generation of the zero axis voltage V0.

The control device 42 of the second embodiment shown in FIG. 13 has a current detection / coordinate conversion unit 43 that replaces the current detection / coordinate conversion unit 12, a zero-axis voltage generation unit 44, and a space voltage that replaces the space voltage vector modulation unit 18. It is provided with a vector modulation unit 45. As described above, the current detection / coordinate conversion unit 43 converts the detected phase currents Iu, Iv, and Iw into the currents Id, Iq, and I0 of the axis coordinates of d, q, and 0 used for vector control (1). Convert by expression.

FIG. 14 shows the detailed configuration of the zero-axis voltage generation unit 44. The zero-axis voltage generation unit 44 includes a P control unit 44A and a resonance control unit 44B. In the P control unit 44A, the proportional gain K _p0 is multiplied by the difference value between the zero-axis current command I 0 _Ref and the detection current I 0. The resonance control unit 44B is configured to improve the followability to the square value of 3ω, which is three times the fundamental wave frequency ω of the phase current. The difference value is multiplied by the resonance gain Kr, the difference from the integration result of the integrator 46 is taken by the subtractor 50, and the difference is input to the integrator 47.

The integration result of the integrator 47 is input to the adder 48 and the multiplier 49. In the multiplier 49, the product with the frequency (3ω) ² is taken, and the result is input to the integrator 46. In the adder 48, the result of multiplying the difference value by the proportional gain K _p0 is added, and a (1 / √2) multiple value of the zero-phase voltage V0 is output.

However, if only the space voltage vector pattern that does not generate the zero-axis voltage V0 shown in the first embodiment is used, the zero required for the control for suppressing the zero-axis current I0 flowing due to the interference described in the equation (4). The shaft voltage V0 cannot be generated. Therefore, in the second embodiment, the following third vector pattern is additionally used. The third vector pattern corresponds to the third switching pattern.
<Third vector pattern>
A pattern in which a voltage acting between the phases of the motor M is generated, and a zero-axis voltage acting equally on the three phases is generated.

Then, the third vector pattern is inserted before or after outputting only one pattern of the first vector pattern used according to each sector during the control cycle. As a result, the zero-axis voltage V0 is controlled on average.

Here, a method of controlling the zero-axis voltage V0 in the case of even-numbered sectors will be described with reference to FIGS. 15 and 16. For example, in sector 6, V31 is the first vector pattern. Then, as shown in FIG. 15, when the zero-axis current I0 is reduced in the sector 6, as the third vector pattern, a vector pattern that turns off the FET 3 constituting the primary side inverter 1 from the first vector pattern V31. V01 is selected. When the third vector pattern is inserted after the first vector pattern, the switching pattern changes as follows.
Primary side inverter (UVW) (010) → (000)
Secondary side inverter (UVW) (100) → (100)
That is, in the third vector pattern V01, the number of phases that the FET 3 of the inverter 2 is turned on is one more than that of the inverter 1. This reduces the zero-axis current I0.

When the zero-axis voltage V0 is negative, a value 6 times the duty Vv calculated by the equation (5) is added to the V-phase duty of the inverter 1. The reason why the value is set to 6 times here is that, as shown in the third embodiment described later, the control amount is made equal when the drive method is switched from the time of asynchronous PWM drive. During asynchronous PWM drive, ± 1/3 _VDC zero-axis voltage is applied three times. During synchronous PWM drive, a zero-axis voltage of ± 1/3 _VDC is applied once in even-numbered sectors. Since the double value of the duty is further tripled, it becomes 6 times.
Here, the "double value of duty" is related to the method of generating the PWM signal pulse. In the asynchronous PWM drive, the pulse is generated so as to expand and contract both sides with respect to the intermediate phase of the PWM cycle. On the other hand, in the synchronous PWM drive, the pulse is generated so as to expand and contract to one side with respect to one end of the PWM cycle. Therefore, the pulse width value to be added or subtracted is twice that of the former.

Further, as shown in FIG. 16, when the zero-axis current I0 is increased in the sector 6, as the third vector pattern, a vector pattern that turns off the FET 3 constituting the secondary side inverter 2 from the second vector pattern V31. Select V30. When the third vector pattern is inserted after the first vector pattern, the switching pattern changes as follows.
Primary side inverter (UVW) (010) → (010)
Secondary side inverter (UVW) (100) → (000)
That is, in the third vector pattern V30, the number of phases that the FET 3 of the inverter 1 is turned on is one more than that of the inverter 2. This increases the zero-axis current I0. When the zero-axis voltage V0 is positive, the value 6 times the duty Vu calculated by the equation (5) is subtracted from the U-phase duty of the inverter 2.

Next, a method of controlling the zero-axis voltage V0 in the case of odd-numbered sectors will be described with reference to FIGS. 17 and 18. For example, in sector 7, V42 is the first vector pattern. As shown in FIG. 17, when the zero-axis current I0 is reduced in the sector 7, as the third vector pattern, a vector pattern V02 that turns off the FET 3 constituting the primary side inverter 1 is used from the first vector pattern V42. select. When the third vector pattern is inserted after the first vector pattern, the switching pattern changes as follows.
Primary side inverter (UVW) (011) → (000)
Secondary side inverter (UVW) (110) → (110)
That is, in the third vector pattern V02, the number of phases that the FET 3 of the inverter 2 is turned on is two more than that of the inverter 1. This reduces the zero-axis current I0. When the zero-axis voltage V0 is negative, 6 times the duty Vv and Vw calculated by the equation (5) are added to the V and W phase duty of the inverter 1. During asynchronous PWM drive, ± 1/3 _VDC zero-axis voltage is applied three times. During synchronous PWM drive, a zero-axis voltage of ± 2/3 _VDC is applied once in odd-numbered sectors. By V0 / 2, which will be described later, the size will be the same as that of even-numbered sectors.

Further, as shown in FIG. 18, when the zero-axis current I0 is increased in the sector 7, as the third vector pattern, a vector pattern that turns off the FET 3 constituting the secondary side inverter 2 from the second vector pattern V42. Select V40. When the third vector pattern is inserted after the first vector pattern, the switching pattern changes as follows.
Primary side inverter (UVW) (011) → (011)
Secondary side inverter (UVW) (110) → (000)

In the third vector pattern V40 in this case, the number of phases that the FET 3 of the inverter 1 is turned on is two more than that of the inverter 2. This increases the zero-axis current I0. When the zero-axis voltage V0 is positive, the U and V phase duty of the inverter 2 is subtracted from the 6 times value of the duty Vu and Vv calculated by the equation (5).

By controlling in this way, the zero-axis voltage V0_ripple is continuously generated only on the positive side and the negative side. Therefore, as shown in FIG. 6, the ripple of the zero-axis current I0 due to the positive and negative fluctuations of V0_ripple does not occur, and the triple frequency component can be suppressed. In the control cycle, the third vector pattern may be inserted before the first vector pattern.

FIG. 19 shows the internal configuration of the space voltage vector modulation unit 45 based on the above-mentioned control principle, and includes the space vector calculation unit 45A and the zero-axis voltage synthesis unit 45B. The space vector calculation unit 45A determines which sector of the twelve belongs to according to the magnitude of the input voltage commands Vα and Vβ, and selects the first vector pattern according to the sector. For example, in the case of sector 7, it is V42.

The voltage value of one selected vector pattern and the zero-axis voltage V0 are input to the zero-axis voltage synthesizer 45B together with the DC voltage _VDC . As shown in FIGS. 15 to 18, the zero-axis voltage synthesizer 45B selects and inserts a third vector pattern according to the increase / decrease of the sector and the zero-axis current I0. In the case of sector 7, the first vector pattern is V42, the third vector pattern to be inserted when the zero axis current is decreased is V02, and the third vector pattern to be inserted when the zero axis current is increased is V40.

In FIGS. 15 to 18, the voltage vector of the third vector pattern is inserted at only one place, but the magnitude of the zero-axis voltage generated in the even-numbered sector and the odd-numbered sector is twice as different. The zero-axis voltage generated by the third vector pattern needs to match the zero-axis voltage command V0 output from the zero-axis voltage generation unit 44. Therefore, in order to align the control amount for the zero-axis voltage command V0 for each sector, for example, in the odd sector, the size of the voltage vector of the third vector pattern is set to V0 / 2.

Since the magnitude of the three-phase voltage of each of the inverters 1 and 2 can be obtained by the above calculation, the duty Du1, Dv1, Dw1, Du2, Dv2, Dw2 of each phase is determined and output by dividing by the _DC voltage VDC. To.

20 and 21 show waveforms simulating the results of suppressing the zero-axis current by controlling the first and second embodiments, respectively. In FIG. 21, it can be seen that the component three times the fundamental wave frequency of the phase current can be suppressed as compared with FIG.

As described above, according to the second embodiment, the space voltage vector modulation unit 45 inserts the third vector pattern before or after outputting only one pattern in the control cycle for the one vector pattern used according to each sector. .. When generating a negative zero-axis voltage, select a vector pattern that turns off all FETs 3 constituting the inverter 1 based on the first vector pattern as the third vector pattern, and generate a positive zero-axis voltage. As the third vector pattern, a vector pattern that turns off all the FETs 3 constituting the inverter 2 is selected based on the first vector pattern.

As a result, both the low-frequency zero-axis current that flows at a component three times the fundamental wave frequency of the phase current and the zero-axis current of the carrier component that flows in synchronization with the switching of the inverters 1 and 2 can be suppressed, and the inverter 1 and 2. It is possible to realize low current and low loss of the motor M.

(Third Embodiment)
The third embodiment relates to the switching control of the drive system. The first and second embodiments are effective in setting the modulation factor to be larger than 0.866 and increasing the output voltage, particularly in the region where the motor M is operated at high speed. Therefore, in a region where the motor M is operated at a low speed, a different drive method, for example, a drive method similar to that in Patent Document 1, is adopted. Hereinafter, the drive system of the first embodiment adopted in the high-speed operation region is referred to as "synchronous pulse control", and the drive system adopted in the low-speed operation region is referred to as "normal pulse control".

FIG. 22 shows the configuration of the control device 51 when executing normal pulse control. Based on the configuration of the second embodiment, it includes a d-axis current command generation unit 52, a dq-axis current control unit 53 that replaces the current control unit 16, and a space voltage vector modulation unit 54 that replaces the space voltage vector modulation unit 45. The d-axis current command generation unit 52 generates and outputs a d-axis current command value for field weakening control from the DC voltage _VDC and the voltage amplitude V _dq of the dq axis by performing PI calculation on the difference between the two. ..

FIG. 23 shows a detailed configuration of the dq-axis current control unit 53. The dq-axis current control unit 53 includes PI control units 55d and 55q and a non-interference term calculation unit 56. The PI control unit 55d calculates the d-axis voltage V _{d_PI} by the PI control calculation from the difference between the d-axis current command value I _dRef and the d-axis current Id. Similarly, the PI control unit 55q calculates the q-axis voltage V _{q_PI} by the PI control calculation from the difference between the q-axis current command value I _qRef and the q-axis current Iq.

The non-interference term calculation unit 56 obtains the non-interference terms V _{d_FF} and V _{q_FF} by the formula shown in the figure in order to prevent the interference of the dq axis, and the d-axis voltage V _{d_PI} and the q-axis voltage are obtained by the adders 58d and 58q. Add to V _{q_PI} respectively. Then, the addition result of the adders 58d and 58q becomes the final d-axis voltage command value Vd and the q-axis voltage command value Vq.

In this way, the reason for separating the configuration of the current control unit used in the normal pulse control and the synchronous pulse control will be described. The normal pulse control is a general vector control in which the d-axis voltage Vd and the q-axis voltage Vq are directly generated by the current control, and the current is made to follow the current command value obtained from the results of the field weakening control and the speed control. This realizes motor control with good responsiveness between current phase lead control and motor output torque control. That is, current control that emphasizes controllability is applied.

On the other hand, since the synchronous pulse control is used in the high output region of the motor, it is necessary to further advance the current phase by setting the output voltage to the maximum of the DC voltage as compared with the normal pulse control. In other words, it is a control that emphasizes output. In this case, it is easier to control the motor by calculating the d, q-axis voltage Vd, Vq from the voltage amplitude and the voltage phase and instructing the output voltage. If the modulation factor is 1.0, the maximum value of the voltage amplitude is input, and only the voltage phase is controlled to control the q-axis current, and as a result, the speed is controlled. As a result, field weakening control is performed.

Next, the operation of the third embodiment will be described with reference to FIGS. 24 to 29. As shown in FIG. 24, for example, when the signal instructing the switching from the normal pulse control to the synchronous pulse control, which is input from the upper control device, is turned ON (S1; YES), the current control unit 16 in the synchronous pulse control is turned on. After confirming that the system is stopped (S2; YES), the voltage amplitude V _dq and the voltage phase θ in the current normal pulse control are calculated (S3, S4). The voltage amplitude V _dq is the square root of the sum of squares of the d-axis voltage Vd and the q-axis voltage Vq, and the voltage phase θ is Atan (Vd / Vq).

Here, in order to switch from normal pulse control to synchronous pulse control, for example, it is necessary to set a modulation factor of 0.866 as a threshold value and set the modulation factor to a value larger than 0.866 according to the operating state of the motor M. At that time, the above switching instruction signal is turned ON.

Subsequently, the initial values _Vamp and θv of the voltage amplitude and the voltage phase at the time of starting the synchronous pulse control are set (S5). The initial value _Vamp of the voltage amplitude is set to the voltage amplitude V _dq , and the initial value θv of the voltage phase does not set the integral term of θ _{V_PI} output by the PI control unit 22 of the current control unit 16 from the above voltage phase θ. It is the value obtained by subtracting the interference term θ _{V_FF} . The non-interference term θ _{V_FF} starts the calculation at the time of switching. As a result, the initial value of the voltage phase θv becomes the voltage phase θ calculated from the voltages Vd and Vq before switching. As a result, the magnitudes of the voltages Vd and Vq can be matched before and after the switching of the current control unit.

Next, the operation of the current control unit 16 used for the synchronous pulse control is started (S6), and the operation of the dq-axis current control unit 53 is stopped only by the PI control unit 55 as shown in FIG. 26 (S7). FIG. 26 conceptually shows the processing of steps S5 to S7. The non-interference term calculation unit 56 is continuously operated because the calculation result is used when switching in the reverse direction, which will be described later. Then, it is determined whether or not the voltage amplitude is equal to or greater than the initial value V _dq ( _VDC × 2 / √3), that is, the phase voltage conversion value which is twice the DC voltage _VDC (S8, FIG. 27). reference).

If V _amp does not reach ( _VDC × 2 / √3) (S8; NO), the voltage amplitude is increased (S11). When V _amp reaches ( _VDC × 2 / √3) (S8; YES → S9), synchronous pulse control is performed using a space voltage vector as in the first embodiment (see S10 and FIG. 28). That is, the space voltage vector modulation unit 54 is switched to the space voltage vector modulation unit 18.

FIG. 25 shows the switching process from the synchronous pulse control to the normal pulse control, contrary to FIG. 24. When the signal instructing the switching from the synchronous pulse control to the normal pulse control is turned ON (S21; YES), the space voltage vector modulation unit 18 is switched to the space voltage vector modulation unit 54 (S22; YES → S23). Then, the target value of the voltage amplitude is calculated (S24). The target value V _dq is the square root of the sum of squares of V _{d_FF} and V _{q_FF} calculated by the non-interference term calculation unit 56 in normal pulse control.

Then, the output voltage _Vamp is gradually reduced from the maximum value ( _VDC × 2 / √3) to the target value V _dq (S25, S30). When the output voltage V _amp reaches the target value V _dq (S25; YES → S26), the operation of the dq axis current control unit 53 is started (S28) and the operation of the current control unit 16 is stopped (S29), but prior to that. The initial values of the d-axis voltage Vd, the q-axis voltage _Vq , and the d-axis current command I default are set (S27). As described above, since the non-interference term is input as the target value and the initial value, the initial value of the integral term of the PI control unit 55 is set to 0. Further, since the d-axis current at the time of synchronous pulse control is overwritten as the d-axis current command I _default , it is used as the initial value of the integral term of the field weakening control unit when switching to the normal pulse control.

FIG. 29 shows a simulation result in the case of transitioning from normal pulse control → synchronous pulse control → normal pulse control. With a 6-pole motor, the PWM control carrier frequency is 5 kHz, the rotation speed is 80 rps, the output torque is 3 N · m, and the rate of change of the output voltage _Vamp is 0.7 V / ms. When switching the control, the rotation speed of the motor is less than 1 rps, and the switching can be performed smoothly with almost no fluctuation.

In the switching of the drive method of the third embodiment, the functional blocks used in the control devices 11, 42, and 51 are exchanged, but when actually configuring these control devices, a microcomputer or a DSP (Digital) is used. The software of Signal Processor) can be used, or FPGA (Field Programmable Gate Array) or the like can be used. Therefore, the replacement of functional blocks can be flexibly performed even with real-time control.

As described above, according to the third embodiment, until the modulation factor of the signals driving the inverters 1 and 2 reaches the threshold value, the first switching pattern is repeated twice following the output of the second switching pattern during the control cycle. Normal pulse control is performed to output and then output the second switching pattern again. Then, when the modulation factor reaches the threshold value, the control is switched to the synchronous pulse control. This makes it possible to selectively use a drive method suitable for the operating state of the motor M. Further, by appropriately setting the initial values of the parameters used for control when switching the drive method, it is possible to suppress fluctuations in the rotation speed of the motor M and perform the switching smoothly.

(Other embodiments)
The current sensor 7 may be a shunt resistor or CT.
The AC power supply may be single-phase.
The switching element is not limited to the MOSFET, and other wide bandgap semiconductors such as IGBTs, power transistors, SiC, and GaN may be used.
Not limited to air conditioners, it may be applied to other products and the like.

Although some embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other embodiments, and various omissions, replacements, and changes can be made without departing from the gist of the invention. These embodiments and variations thereof are included in the scope and gist of the invention, and are also included in the scope of the invention described in the claims and the equivalent scope thereof.

In the drawings, M is an open structure winding motor, 1 is a primary side inverter 1, 2 is a secondary side inverter, 11 is a control device, 18 is a space voltage vector modulator, and 30 is an air conditioner.

In the open winding motor drive device of the embodiment, the three-phase windings are independent of each other, and are connected to three of the six terminals of the motor having an open winding structure having six terminals . Primary side inverter to be
A secondary inverter connected to the remaining three terminals of the motor,
A control unit that controls the current and rotation speed that energize the motor based on the line-to-line duty ratio of each of the primary and secondary inverters in PWM control.
It is equipped with a current detector that detects the current energized in the motor.
The control unit controls the current and rotation speed of the motor based on the duty between the lines of the primary side and secondary side inverters, and has three phases in the same direction between the primary side and secondary side inverters. It has a zero-axis current suppression unit that suppresses the zero-axis current flowing through the inverter.
The zero-axis current suppression unit has a space voltage vector composed of 64 voltage vectors, which is a combination of on / off patterns of the primary side and secondary side inverters.
Centering on the point where two second switching patterns that do not generate a zero-axis voltage and do not generate a voltage that acts between the phases of the motor are located, and do not generate a zero-axis voltage that acts equally on the three phases of the motor. The first switching pattern that generates the voltage applied to the motor is divided into 12 sectors, which are further divided into 12 sectors, with the 6 regions divided by the points where two of them are located as the vertices, and used according to each sector. Only one switching pattern is selected during the control cycle.
Further, in the refrigeration cycle apparatus of the embodiment, a motor having an open winding structure in which three-phase windings are independent and has six winding terminals, and a motor having an open winding structure,
The open winding motor drive device of the embodiment is provided.

Claims (5)

A primary inverter connected to three of the six output terminals of a motor with an open winding structure, each of which has three independent windings and has six output terminals.
A secondary inverter connected to the remaining three output terminals of the motor,
A control unit that controls the current and rotation speed that energize the motor based on the line-to-line duty ratio of each of the primary and secondary inverters in PWM control.
It is equipped with a current detector that detects the current energized in the motor.
The control unit controls the current and rotation speed of the motor based on the duty between the lines of the primary side and secondary side inverters, and has three phases in the same direction between the primary side and secondary side inverters. It has a zero-axis current suppression unit that suppresses the zero-axis current flowing through the inverter.
The zero-axis current suppression unit has a space voltage vector composed of 64 voltage vectors, which is a combination of on / off patterns of the primary side and secondary side inverters.
Centering on the point where two second switching patterns that do not generate a zero-axis voltage and do not generate a voltage that acts between the phases of the motor are located, and do not generate a zero-axis voltage that acts equally on the three phases of the motor. The first switching pattern that generates the voltage applied to the motor is divided into 12 sectors, which are further divided into 12 sectors, with the 6 regions divided by the points where two points are located as the vertices, and the first is used according to each sector. An open winding motor drive that selects only one switching pattern during the control cycle. The open winding motor drive device according to claim 1, wherein the zero-axis current suppression unit selects the same first switching pattern for two adjacent sectors. The zero-axis current suppression unit generates a zero-axis voltage before or after outputting only one pattern during the control cycle of the first switching pattern used according to each sector, and generates a voltage acting between the phases of the motor. Insert the third switching pattern to generate,
When generating a negative axis zero-axis voltage, as the third switching pattern, only a switching pattern that turns off all the switching elements constituting the primary side inverter is selected based on the first switching pattern.
A claim that when generating a positive zero-axis voltage, as the third switching pattern, only a switching pattern that turns off all the switching elements constituting the secondary inverter based on the first switching pattern is selected. The open winding motor drive device according to 1 or 2. Assuming that the control for selecting only one pattern during the control cycle as the first switching pattern used according to each sector is the synchronous pulse control.
The zero-axis current suppression unit outputs the first switching pattern twice following the output of the second switching pattern during the control cycle until the modulation factor of the signal driving the inverter reaches the threshold value. After that, normal pulse control for outputting the second switching pattern again is performed.
The open winding motor drive device according to claim 3, wherein when the modulation factor reaches a threshold value, the control is switched to the synchronous pulse control. A motor with an open winding structure in which each of the three-phase windings is independent and has six winding terminals.
A refrigeration cycle device including the open winding motor drive device according to any one of claims 1 to 4.

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