JP2015119317A - Receiving device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a receiving device capable of achieving communication with high receiving quality when a propagation path is largely fluctuated during a time length of inverse diffusion or symbol repetition.SOLUTION: A receiving device includes: a channel estimation part 4 which estimates a state of a wireless propagation path from an already-known pilot signal and outputs a channel estimation result for every designated time length; a variable delay line 5 which, based on the channel estimation result, compensates the delay time and phase of a receiving signal for every time length; a multiplication part 6; an addition part 7 which performs cyclic addition of the compensated signal by a unit of the time length; an integration part 8; a demodulation part 9 which performs demodulation processing using the integration result from the integration part 8; and a control part 10 which controls timing for outputting the integration result of the demodulation part 9.

Description

  The present invention relates to a receiving apparatus.

  Conventionally, a receiver in a spread spectrum communication system calculates a partial correlation by dividing a spread sequence into a plurality of partial sequences as a parallel method for improving the despreading processing speed of a received signal, and calculates individual partial correlations. By detecting the correlation peak timing, determining the phase of the partial correlation value at the correlation peak timing, and adding all the partial correlation values after compensation with the phase of the correlation peak timing that calculated the phase of each partial correlation value, The despread value of the spreading sequence length is calculated. Such a technique is disclosed in Patent Document 1 below.

JP-A-8-8780

  However, according to the above conventional technique, it is necessary to detect the correlation peak timing of a partial sequence shorter than the original spread sequence. For this reason, when the received signal-to-noise power ratio (SNR) is low, the correlation peak detection error increases, and it is difficult to correctly obtain the despread value of the original spread sequence.

  Further, according to the conventional technique, only the phase value of the partial correlation value is compensated, and the process of adjusting the timing of all the partial correlation values is not performed. Therefore, when the time variation of the transmission / reception propagation distance is fast due to the fact that the receiving device is moving at high speed, the timing between the partial correlations is shifted, the despread output may be reduced, and the reception quality may be deteriorated. There was a problem.

  The present invention has been made in view of the above, and in a spread spectrum communication system having a large spreading factor or a communication system that repeatedly transmits transmission symbols, the propagation path varies greatly between the despread length or the symbol repetition length. An object of the present invention is to obtain a receiving apparatus capable of realizing communication with high reception quality.

  In order to solve the above-described problems and achieve the object, the present invention estimates a state of a radio propagation path from a known pilot signal and outputs a channel estimation result for each specified time length; Compensation means for compensating the delay time and phase of the received signal for each time length based on the channel estimation result, and cyclic addition means for cyclically adding the signal compensated by the compensation means in units of the time length; Demodulating means for performing demodulation processing using the integration result from the cyclic adding means, timing for designating the time length, and timing for the cyclic adding means to output the integration result to the demodulating means based on the time length And a control means for controlling.

  According to the present invention, in a spread spectrum communication system having a large spreading factor or a communication system in which transmission symbols are repeatedly transmitted, communication with high reception quality can be performed when a propagation path greatly varies between despread lengths or symbol repetition lengths. There is an effect that it can be realized.

FIG. 1 is a diagram illustrating a configuration example of a receiving apparatus according to the first embodiment. FIG. 2 is a diagram illustrating an operation example in the case where four symbols are subjected to in-phase addition. FIG. 3 is a diagram illustrating a configuration example of a receiving apparatus according to the second embodiment. FIG. 4 is a diagram showing a relationship between a transmission bit sequence, a spreading code phase, and a correlation peak when transmitting 3 bits of information per symbol using a spreading code having a spreading code length of 8 chips. FIG. 5 is a diagram illustrating an operation example in the case where the correlation waveforms for four symbols are added in phase with compensation of timing and phase. FIG. 6 is a diagram illustrating a configuration example of a receiving apparatus according to the third embodiment.

  Embodiments of a receiving apparatus according to the present invention will be described below in detail with reference to the drawings. Note that the present invention is not limited to the embodiments.

Embodiment 1 FIG.
FIG. 1 is a diagram illustrating a configuration example of a receiving apparatus according to the present embodiment. In the present embodiment, in a wireless communication system in which a certain number of transmission symbols are continuously transmitted repeatedly, and symbols are added on the receiving side to improve communication quality on a low S / N (Signal / Noise) line. A receiving apparatus will be described.

  The receiving apparatus includes a receiving antenna 1, an RF (Radio Frequency) unit 2, an A / D (Analog / Digital) conversion unit 3, a channel estimation unit 4, a variable delay line 5, a multiplication unit 6, and an addition unit. 7, an integration unit 8, a demodulation unit 9, and a control unit 10. In the receiving apparatus, the variable delay line 5 and the multiplication unit 6 constitute a compensation unit, and the addition unit 7 and the integration unit 8 constitute a cyclic addition unit.

  In the receiving apparatus, the RF unit 2 converts the received signal received by the receiving antenna 1 into an intermediate frequency (IF) signal or a complex baseband signal by amplification processing, filter processing, and frequency conversion processing. The A / D conversion unit 3 converts the signal after the conversion by the RF unit 2 into a quantized digital signal.

  In the following description, the operation will be described based on the configuration of the receiving apparatus when the input signal to the A / D converter 3 is a complex baseband signal. When the output of the RF unit 2 is an IF signal, an orthogonal frequency conversion unit by digital signal processing is provided after the A / D conversion unit 3, and the orthogonal frequency conversion unit converts the digital IF signal into a complex baseband signal. The signal may be regarded as an output signal from the A / D conversion unit 3.

  The A / D conversion unit 3 is a quantized digital signal (when the input of the A / D conversion unit 3 is an IF signal, the orthogonal frequency conversion unit digitally outputs the output of the A / D conversion unit 3 as described above. To the channel estimation unit 4 and the variable delay line 5.

  The channel estimation unit 4 estimates the amplitude and phase of the propagation path and the reception timing, and outputs symbol timing information 11 and a phase / amplitude compensation value 12. In general, since the propagation path fluctuates with time, the channel estimation unit 4 performs the symbol timing information 11 and the phase / amplitude compensation value at a period (time length) specified by the channel estimation unit control information 13 output from the control unit 10. 12 is updated and output. Note that in wireless communication schemes that repeatedly transmit symbols, it is common to match the update period of the estimated value to the symbol length of the transmission signal, but when the fluctuation of the propagation path is sufficiently slow compared to the symbol length, The update cycle does not necessarily have to match the symbol cycle, and may be an integer multiple of the symbol cycle, for example.

  The control unit 10 outputs the channel estimation unit control information 13 based on the communication method performed by the receiving device. However, when the receiving device moves, the moving speed information of the receiving device is input and the moving speed information is used. Then, the channel estimation unit control information 13 may be generated and output.

  Further, in the channel estimation unit 4, channel estimation is generally performed using a known pattern (generally called a pilot signal, preamble, etc.) included in the received signal. The known patterns are time-multiplexed and code-multiplexed with respect to a signal for transmitting data, and the channel estimation unit 4 extracts these known patterns and calculates a channel estimation value.

  The channel estimation value updating method in the channel estimation unit 4 may be a method of calculating the channel estimation value individually for each update period using only known patterns included in the update period. However, the accuracy of the channel estimation value is improved. Therefore, the channel estimation values of several symbols before and after the update period may be weighted or cyclic addition with a forgetting factor may be performed.

  The variable delay line 5 adjusts the delay time of the output from the A / D conversion unit 3 in accordance with the symbol timing information 11 output from the channel estimation unit 4. With this processing, the receiving apparatus compensates for the propagation delay time for each symbol with respect to fluctuations in reception timing due to fluctuations in propagation delay time, and can add a plurality of consecutive symbols at the same stage at the same timing.

  The multiplication unit 6 complex-multiplies the output from the variable delay line 5 by the phase / amplitude compensation value 12 output from the channel estimation unit 4. With this processing, in the receiving apparatus, for the signal output from the variable delay line 5, the phase for each symbol is aligned with the accuracy of channel estimation with respect to the phase fluctuation caused by the fluctuation of the propagation path. Is possible. The amplitude compensation value may be a value weighted according to a signal-to-noise power value for each symbol estimated by the channel estimation unit 4 or a value indicating communication quality such as a signal-to-noise / interference power value. When the noise and the interference power are constant within a plurality of symbol sections to be added and these can be regarded as additive white Gaussian noise, the multiplication unit 6 performs complex multiplication on the complex conjugate of the complex channel estimation value. Good.

  In addition, the multiplication unit 6 may perform phase-only compensation when the amplitude of the propagation path can be regarded as a constant value in a plurality of symbol sections to be added, or when it is desired to simplify the compensation process.

  In addition, in the multiplication unit 6, when performing symbol addition in the subsequent stage, in addition to coherent addition, power addition, absolute value addition, etc., instead of multiplying the phase / amplitude compensation value 12, the variable delay line 5 The output power value and absolute value may be obtained.

  The multiplier 6 outputs the signal after complex multiplication to the adder 7 that constitutes the cyclic adder together with the integrator 8. The case where four consecutive symbols {S1, S2, S3, S4} are added in units of the above-described time length will be described as an example. When the first symbol S1 is input, the adding unit 7 includes two inputs. The input signal on the side connected to the integrating unit 8 becomes zero. As a result, the symbol S 1 is output as it is from the adder 7 and is stored in the integrator 8. Next, when the second symbol S2 is input, in the addition unit 7, the symbol S2 is connected to the side connected to the multiplication unit 6 and the side connected to the integration unit 8 in the two inputs. Is the value of the integration unit 8, that is, the symbol S1 is input. As a result, the symbol addition value S1 + S2 is output from the adding unit 7 and input to the integrating unit 8.

  When this operation is repeated up to the fourth symbol S4, the symbol addition value S1 + S2 + S3 + S4 for four symbols is finally stored in the integrating unit 8. The integrator 8 outputs the stored value to the demodulator 9 and clears it to zero according to the integrator control information 14 based on the time length unit output from the controller 10 when the symbol addition values S1 + S2 + S3 + S4 for four symbols are accumulated. To do.

  Similarly to the channel estimation unit control information 13, the control unit 10 outputs the integration unit control information 14 based on the communication method performed by the receiving device. However, when the receiving device moves, information on the moving speed of the receiving device is input. Then, the integration unit control information 14 may be generated and output using the information on the moving speed.

  The demodulator 9 performs a demodulation process using the symbol addition values S1 + S2 + S3 + S4 for four symbols output from the integrator 8.

  Operations of timing compensation and phase / amplitude compensation in the receiving apparatus according to the present embodiment will be described in detail with reference to FIG. FIG. 2 is a diagram illustrating an operation example in the case where four symbols are subjected to in-phase addition. 2, channel estimation values 15-1 to 15-4, data symbols 16-1 to 16-4, and compensated symbols 17-1 to 17-4 are shown, respectively. Branch numbers -1 to -4 distinguish four consecutive symbols and are symbol numbers in time order. The receiving apparatus according to the present embodiment is not limited to 4-symbol repeated transmission, and any number of symbols can be applied. Here, in order to simplify the explanation, explanation using an example of four symbols will be given.

  In FIG. 2, in the receiving apparatus, channel estimation unit 4 updates channel estimation values 15-1 to 15-4 in symbol periods. In FIG. 2, the amplitude and phase are illustrated on the IQ plane, and the timing for each symbol is represented on the time axis. In the example of FIG. 2, in the first symbol, the channel estimation value 15-1 is on the I axis, and in the subsequent symbols, the phase of the channel estimation value turns to the Q axis side, and the symbol timing is gradually delayed. .

  In the receiving apparatus, the variable delay line 5 performs delay compensation on the data symbols 16-1 to 16-4 in symbol units, and the multiplication unit 6 performs phase / amplitude compensation to obtain compensated symbols 17-1 to 17-17. -4.

  For the purpose of facilitating the explanation of the operation for compensating the timing, the data symbols 16-1 to 16-4 are shown here when the symbol timing does not vary. As shown in FIG. 2, the second to fourth compensated symbols 17-2 to 17-4 include the signal components of the subsequent symbols in the symbol interval as a result of the symbol timing compensation. . The adder 7 and the integrator 8 add these four symbols according to the above-described operation, and output the result to the demodulator 9.

  The description of the operation of the receiving apparatus so far relates to a case where the transmission symbol is not spread spectrum, but the receiving apparatus according to the present embodiment is also applicable when the symbol is spread spectrum. In this case, the variable delay line 5 performs despreading in symbol units at the timing designated by the channel estimation unit 4 and outputs the result to the multiplication unit 6. In the variable delay line 5, rake synthesis may be performed by internal despreading processing. In addition, it may be considered that the four symbols shown in FIG. 2 represent one symbol of spread spectrum. In this case, the receiving apparatus divides one symbol into four to calculate partial correlation, and the amplitude / phase In addition, the correlation value of the entire symbol is calculated after the timing is compensated.

  In the case of spread spectrum, the same sequence may be used between repeated symbols, or a different sequence may be used for each symbol. In either case, demodulation processing can be performed by using an appropriate spreading code on the receiving device side.

  In addition, when it is necessary to simultaneously demodulate a plurality of code-multiplexed signals such as CDMA (Code Division Multiple Access), the receiving apparatus has a configuration from the variable delay line 5 having a despreading function to the demodulator 9. The number of multiplexed signals may be provided in parallel, or even if the number of components is smaller than the number of multiplexed signals, the time division processing may be performed while changing the spreading code corresponding to the signal.

  In the receiving apparatus according to the present embodiment, the configuration example in the case where there is one receiving antenna 1 in FIG. 1 is shown, but the present invention is not limited to this, and the configuration using a plurality of receiving antennas is also possible. Applicable.

  As described above, according to the present embodiment, the receiving apparatus, in the spread spectrum communication system having a large spreading factor or the communication system that repeatedly transmits transmission symbols, in addition to the phase and amplitude, the timing fluctuation accompanying the propagation path length fluctuation. Is also compensated for, and partial correlation values are added or symbols are added. Thereby, even when the propagation path varies greatly between the despreading length or the symbol repetition length, high quality reception can be realized.

Embodiment 2. FIG.
FIG. 3 is a diagram illustrating a configuration example of the receiving device according to the present embodiment. In this embodiment, a receiving apparatus will be described in which a CSK (Code Shift Keying) signal is continuously transmitted repeatedly for a plurality of symbols and symbol addition is performed on the receiving side.

  The receiving apparatus includes a receiving antenna 21, an RF unit 22, an A / D conversion unit 23, despreading units 24 and 25, a channel estimation unit 26, a cyclic shift unit 27, a multiplication unit 28, and an addition unit 29. And an integration unit 30, a demodulation unit 31, and a control unit 32. In the receiving apparatus, the cyclic shift unit 27 and the multiplication unit 28 constitute compensation means, and the addition unit 29 and the integration unit 30 constitute cyclic addition means.

  Before explaining the operation of the receiving apparatus shown in FIG. 3, the CSK that is the object of this embodiment will be explained with reference to FIG. CSK is a type of spread spectrum communication method, and is a method of putting information on the code phase of a spread code. FIG. 4 is a diagram showing a relationship between a transmission bit sequence, a spreading code phase, and a correlation peak when transmitting 3 bits of information per symbol using a spreading code having a spreading code length of 8 chips. Taking the spreading code for transmission bit sequence 000 (binary representation) as a reference, the spreading code phase varies depending on the value of the transmission bit sequence. For example, when transmitting a transmission bit sequence 001 (binary representation), spread modulation is performed using a spreading code sequence obtained by cyclically shifting the original spreading code to the right by one chip. Similarly, the cyclic shift amount is changed in accordance with the value of the transmission bit sequence, such as a 2-chip cyclic shift in 010 and a 3-chip cyclic shift in 011. Although FIG. 4 shows an example in which a transmission bit sequence is assigned to all code phase states of a spread code having a code length of 8 chips, the code phase to which information is assigned may be less than or equal to the possible code phase state. , You do not have to use everything.

  In FIG. 4, when the sliding correlation is performed on the spreading code sequence that is cyclically shifted with reference to the timing of the spreading code sequence serving as a reference on the receiving side, the correlation peak of the received signal is transmitted as shown on the right side of FIG. The peak timing is shifted according to the value of the bit series. If the reference timing is known in advance on the receiving side (receiving device), the original transmission bit sequence can be estimated from the position of the correlation peak. Thus, in CSK, a transmission bit sequence can be estimated from the position of the correlation peak of the received signal.

  When the spreading factor is large and the side lobe of the correlation peak after despreading is lower than the noise level in the operation S / N, CSK can be regarded as equivalent to M-ary orthogonal code transmission. The M-ary orthogonal code transmission is an effective method when communication is desired at a low S / N because a line is established with a lower required Eb / N0 as the multi-value number M is larger. On the other hand, in M-ary orthogonal code transmission, if the bit rate is the same, the symbol length becomes longer as the multi-value number M is increased, so that it is vulnerable to transmission path fluctuations. Even when the symbol length is shortened and the bit rate is equivalently lowered by repeated symbol transmission, if the transmission path varies between symbols, the S / N improvement effect by symbol addition is reduced. Therefore, a method of adding symbols while compensating for the phase, amplitude, and timing between symbols is effective.

  Returning to FIG. 3, the operation of the receiving apparatus of this embodiment will be described. In the receiving apparatus, the RF unit 22 converts the received signal received by the receiving antenna 21 into an intermediate frequency (IF) signal or a complex baseband signal by amplification processing, filter processing, and frequency conversion processing. The A / D conversion unit 23 converts the signal converted by the RF unit 22 into a quantized digital signal.

  In the following description, the operation will be described based on the configuration of the receiving apparatus when the input signal to the A / D converter 23 is a complex baseband signal. When the output of the RF unit 22 is an IF signal, an orthogonal frequency conversion unit by digital signal processing is provided after the A / D conversion unit 23, and the orthogonal frequency conversion unit converts the digital IF signal into a complex baseband signal. The signal may be regarded as an output signal from the A / D conversion unit 23.

  The A / D conversion unit 23 is a quantized digital signal (when the input of the A / D conversion unit 23 is an IF signal, the orthogonal frequency conversion unit digitally outputs the output of the A / D conversion unit 23 as described above. The signal is converted to a complex baseband by orthogonal frequency conversion to the despreading unit 24 and the despreading unit 25.

  The despreading unit 24 performs despreading with a spreading code corresponding to a known pilot signal in order to estimate the propagation path, and outputs a correlation waveform to the channel estimation unit 26.

  The despreading unit 25 performs despreading with a spreading code corresponding to the data signal carrying the transmission data sequence, and outputs a correlation waveform to the cyclic shift unit 27. In the despreading unit 25, a channel estimation unit 26 described later performs despreading at the same cycle (time length) as the cycle (time length) designated by the control unit 32. Although not clearly shown in FIG. 3, the despreading unit 25 may acquire information on the specified period (time length) from the control unit 32 or may acquire via the channel estimation unit 26. The method is not limited.

  Channel estimation unit 26 estimates the amplitude and phase of the propagation path and the reception timing from the despread output of the pilot signal in despreading unit 24, and outputs symbol timing information 33 and phase / amplitude compensation value 34. In general, since the propagation path fluctuates with time, the channel estimator 26 uses the symbol timing information 33 and the phase / amplitude compensation value at a period (time length) specified by the channel estimator control information 35 output from the controller 32. 34 is updated and output. As in the first embodiment, in the wireless communication system that repeatedly transmits symbols, the update period of the estimated value is generally matched with the symbol length of the transmission signal, but the propagation path varies compared to the symbol length. If it is sufficiently slow, etc., the update period does not necessarily match the symbol period, and may be an integral multiple of the symbol period, for example.

  The control unit 32 outputs the channel estimation unit control information 35 based on the communication method performed by the receiving device. However, when the receiving device moves, the moving speed information of the receiving device is input and the moving speed information is used. Then, the channel estimation unit control information 35 may be generated and output.

  In addition, channel estimation in channel estimation unit 26 is generally performed using a known pattern (generally referred to as a pilot signal, preamble, etc.) included in the received signal. The known patterns are time-multiplexed and code-multiplexed with respect to a signal for transmitting data, and the channel estimation unit 26 extracts these known patterns and calculates a channel estimation value.

  The channel estimation value updating method in the channel estimation unit 26 may be a method in which only the known pattern included in the update period is used and the channel estimation value is calculated individually for each update period, but the accuracy of the channel estimation value is improved. Therefore, the channel estimation values of several symbols before and after the update period may be weighted or cyclic addition with a forgetting factor may be performed.

  The cyclic shift unit 27 cyclically shifts the correlation output output from the despreading unit 25 according to the value indicated by the symbol timing information 33. With this processing, the reception apparatus compensates for the propagation delay time for each symbol with respect to fluctuations in reception timing due to fluctuations in the propagation delay time, and the despread outputs (correlation waveforms) of a plurality of consecutive symbols in the subsequent stage are the same. It can be added at the timing.

  Such a cyclic shift can be realized by, for example, a method in which, in the cyclic shift unit 27, the correlation waveform for one symbol output from the despreading unit 25 is once written in the memory and the read address is controlled when reading again.

  The multiplication unit 28 complex-multiplies the output from the cyclic shift unit 27 by the phase / amplitude compensation value 34 output from the channel estimation unit 26. With this processing, in the receiving apparatus, the phase of each correlation symbol output from the cyclic shift unit 27 with respect to the phase fluctuation due to the fluctuation of the propagation path is aligned with the accuracy of the channel estimation. Addition is possible. The amplitude compensation value may be a value weighted according to a signal-to-noise power value for each symbol estimated by the channel estimation unit 26 or a value indicating communication quality such as a signal-to-noise / interference power value. When the noise and the interference power are constant within the plurality of symbol sections to be added and these can be regarded as additive white Gaussian noise, the multiplier 28 performs complex multiplication on the complex conjugate of the complex channel estimation value. Good.

  In addition, the multiplication unit 28 may perform phase-only compensation when the amplitude of the propagation path can be regarded as a constant value in a plurality of symbol sections to be added, or when it is desired to simplify the compensation process.

  In addition, in the multiplication unit 28, when adding symbols in the subsequent stage instead of coherent addition, power addition, absolute value addition, etc., instead of multiplying the phase / amplitude compensation value 34, the cyclic shift unit 27 The output power value and absolute value may be obtained.

  The multiplier 28 outputs the signal after complex multiplication to an adder 29 that constitutes a cyclic addition means together with the integrator 30. The case where the correlation waveforms {S1, S2, S3, S4} for four consecutive symbols are added in units of the above-described time length will be described as an example. When the correlation waveform S1 of the first symbol is input, the addition unit 29, the input signal on the side connected to the integrating unit 30 out of the two inputs is zero. As a result, the correlation waveform S1 is output as it is from the adder 29 and stored in the integrator 30. Next, when the correlation waveform S2 of the second symbol is input, in the addition unit 30, the correlation waveform S2 is connected to the integration unit 30 on the side connected to the multiplication unit 28 of the two inputs. The value of the integration unit 30, that is, the correlation waveform S1 is input to the closed side. As a result, the correlation waveform addition value S1 + S2 is output from the adding unit 30 and input to the integrating unit 30. Note that the correlation waveform of each symbol is different from the symbol in the first embodiment, but in order to facilitate the explanation, a symbol such as “S1” is used here as in the first embodiment.

  When this operation is repeated up to the correlation waveform S4 of the fourth symbol, the integration waveform 30 finally stores the correlation waveform addition values S1 + S2 + S3 + S4 for four symbols. The integration unit 30 outputs the stored value to the demodulation unit 31 according to the integration unit control information 36 based on the time length unit output from the control unit 32 when the correlation waveform addition values S1 + S2 + S3 + S4 for four symbols are accumulated. Clear to zero.

  Similarly to the channel estimation unit control information 35, the control unit 32 outputs the integration unit control information 36 based on the communication method performed by the receiving device. However, when the receiving device moves, information on the moving speed of the receiving device is input. Then, the integration unit control information 36 may be generated and output using the information on the moving speed.

  The demodulator 31 performs demodulation using the correlation waveform addition values S1 + S2 + S3 + S4 for four symbols output from the integrator 30.

  The operation of timing compensation and phase / amplitude compensation in the receiving apparatus according to the present embodiment will be described in detail with reference to FIG. FIG. 5 is a diagram illustrating an operation example in the case where the correlation waveforms for four symbols are added in phase with compensation of timing and phase. In FIG. 5, data symbols 37-1 to 37-4, despread outputs 38-1 to 38-4, cyclic shift outputs 39-1 to 39-4, phase compensation outputs 40-1 to 40-4, integration, respectively. Output 40 is shown. Branch numbers -1 to -4 distinguish four consecutive symbols and are symbol numbers in time order. The receiving apparatus according to the present embodiment is not limited to 4-symbol repeated transmission, and any number of symbols can be applied. Here, in order to simplify the explanation, explanation using an example of four symbols will be given.

  In FIG. 5, in the receiving apparatus, the despread outputs 38-1 to 38-4 of the data symbols 37-1 to 37-4 by the despreading unit 25 are correlation waveforms obtained by sliding correlation or the like, and the correlation peak timing. Send information is on. Here, the same transmission bit sequence is transmitted continuously for 4 symbols, and the code phase cyclic shift rule in CSK shows the same case over 4 symbols. If there is no fluctuation in the propagation path length between symbols, the position of the correlation peak timing should be the same as the symbol boundary, but FIG. 5 shows a situation in which the propagation path length changes with time, so the peak for each symbol. The timing is slightly off. Note that here, in order to make the explanation of the figure easy to understand, the correlation peak can be clearly discriminated in the despread outputs 38-1 to 38-4, but in reality, the S / N before symbol addition is Since it is low, the correlation peak cannot always be determined.

  In the receiving apparatus, in order to align the timings of the despread outputs 38-1 to 38-4 from the despreading unit 25, the cyclic shift is performed according to the propagation timing separately obtained from the pilot signal in the despreading unit 24 and the channel estimation unit 26. The unit 27 performs a cyclic shift. Thereafter, the phase is compensated by the multiplication unit 28, the correlation peak timings of 4 symbols are aligned, and the addition unit 29 and the integration unit 30 perform symbol integration by in-phase addition. The demodulator 31 detects the peak position of the symbol-integrated correlation waveform, compares it with the reference timing obtained from the pilot, obtains the code phase shift amount, and estimates the transmission bit sequence.

  Further, when it is necessary to simultaneously demodulate a plurality of code-multiplexed signals as in CDMA, the receiving apparatus has a configuration from the despreading units 24 and 25 having a despreading function to the demodulating unit 31 for the number of multiplexed signals. Even if the number of components is less than the number of multiplexed signals, time division processing may be performed while changing the spreading code corresponding to the signal. When the pilot signals are common, it is not necessary to prepare a plurality of despreading unit 24 systems.

  Further, in the receiving apparatus according to the present embodiment, the configuration example in the case where there is one receiving antenna 21 in FIG. 3 is shown, but the present invention is not limited to this, and the configuration using a plurality of receiving antennas is also possible. Applicable.

  Although not explicitly described in the above description, when symbols are repeatedly transmitted, the same spread sequence may be used for the repeated symbols, or different symbols may be used for each symbol. In any case, demodulation processing can be performed by using an appropriate spreading code on the receiving side.

  As described above, according to the present embodiment, in the CSK communication system to which symbol repetition is applied, the receiving apparatus compensates for timing fluctuations associated with propagation path length fluctuations in addition to phase and amplitude, and then performs partial correlation values. Or symbol addition. Thereby, even when the propagation path varies greatly between the despreading length or the symbol repetition length, high quality reception can be realized.

Embodiment 3 FIG.
FIG. 6 is a diagram illustrating a configuration example of the receiving device according to the present embodiment. In this embodiment, a description will be given of a receiving apparatus that performs despreading processing in the frequency domain when CSK signals are continuously transmitted repeatedly for a plurality of symbols and symbol addition is performed on the receiving side. In the case of a CSK signal having a large spreading factor, the computation scale is reduced by signal processing in the frequency domain, so that it is possible to reduce the size of the receiver and reduce power consumption.

  The reception apparatus includes a reception antenna 41, an RF unit 42, an A / D conversion unit 43, an FFT (Fast Fourier Transform) unit 44, an interference measurement unit 45, an interference removal unit 46, and an FFT unit 48. Unit 49, FFT unit 51, multiplication unit 52, IFFT (Inverse Fast Fourier Transform) unit 53, channel estimation unit 54, FFT unit 55, weight calculation unit 56, multiplication unit 57, and addition unit 58 An integration unit 59, an IFFT unit 60, a demodulation unit 61, and a control unit 62. In the receiving apparatus, the multiplication unit 57 constitutes compensation means, the addition unit 58 and integration unit 59 constitute cyclic addition means, and the interference measurement unit 45 and interference removal unit 46 constitute interference processing means. .

  In the receiving apparatus, the RF unit 42 converts the received signal received by the receiving antenna 41 into an intermediate frequency (IF) signal or a complex baseband signal through amplification processing, filter processing, and frequency conversion processing. The A / D conversion unit 43 converts the signal after the conversion by the RF unit 42 into a quantized digital signal.

  In the following description, the operation will be described based on the configuration of the receiving apparatus when the input signal to the A / D converter 43 is a complex baseband signal. When the output of the RF unit 42 is an IF signal, an orthogonal frequency conversion unit by digital signal processing is provided after the A / D conversion unit 43, and the orthogonal frequency conversion unit converts the digital IF signal into a complex baseband signal. The signal may be regarded as an output signal from the A / D converter 43.

  The A / D conversion unit 43 is a quantized digital signal (when the input of the A / D conversion unit 43 is an IF signal, the orthogonal frequency conversion unit digitally outputs the output of the A / D conversion unit 43 as described above. The signal obtained by orthogonal frequency conversion to a complex baseband is output to the FFT unit 44.

  The FFT unit 44 converts the quantized digital signal input from the A / D conversion unit 43 into a frequency domain signal by discrete Fourier transform.

  When the received signal is a spread spectrum signal having a large spreading factor, a narrow-band interference that is strong enough to affect the reception quality can be easily detected by performing a discrete Fourier transform on the received signal. The interference measurement unit 45 detects such narrowband interference, and the interference removal unit 46 removes or suppresses the interference component.

  In the interference measurement unit 45, as a measurement of interference, for example, based on an instruction from the control unit 62, using a threshold coefficient (not shown in FIG. 6), the calculated average power of the received signal is multiplied by the threshold coefficient. Alternatively, a method such as detecting a frequency component having a large amplitude may be used.

  In the interference removal unit 46, as a method of removing / reducing the interference, a method of replacing the frequency component in which the interference is detected with “0” or another fixed value, and interference detecting a coefficient designated by the control unit 62. A method may be used in which the frequency component is multiplied.

  Note that, when it is not necessary to deal with such interference, the interference measuring unit 45 and the interference removing unit 46 may be omitted from the configuration of the receiving device.

  The interference removing unit 46 outputs the frequency domain signal from which the interference is removed to the multiplying units 49 and 52.

  Multiplier 49 multiplies the received signal from interference canceler 46 by the signal obtained by converting the spread code 47 of the pilot signal into the frequency domain representation by FFT unit 48, and outputs the result to IFFT unit 53. This series of processing is the same as performing sliding correlation on the received signal with a pilot spreading code.

  Similarly, the multiplication unit 52 multiplies the reception signal from the interference removal unit 46 and the signal obtained by converting the spread code 50 of the data signal into the frequency domain representation by the FFT unit 51, and outputs the result to the multiplication unit 57. The output from the multiplier 52 becomes one input in the multiplier 57.

  Note that the spread code multiplied in the frequency domain by the received signal may be preliminarily subjected to discrete Fourier transform, or may be subjected to discrete Fourier transform one by one as shown in FIG. An optimal method may be used in consideration of the above.

  The pilot-side IFFT unit 53 converts the output from the multiplication unit 49 into a time domain representation by inverse discrete Fourier transform, and outputs it to the channel estimation unit 54.

  The channel estimation unit 54 periodically calculates a channel estimation value for the time-domain representation signal converted by the IFFT unit 53 using a weighted moving average process for improving estimation accuracy. The control unit 62 controls the calculation period control of the estimated value in the channel estimation unit 54 and the control of the averaging time for improving the estimation accuracy through the channel estimation unit control information 63. The channel estimation unit 54 updates and outputs the propagation path estimated value at a period (time length) specified by the channel estimation unit control information 63 output from the control unit 62. As in the first embodiment, in the wireless communication system that repeatedly transmits symbols, the update period of the estimated value is generally matched with the symbol length of the transmission signal, but the propagation path varies compared to the symbol length. If it is sufficiently slow, etc., the update period does not necessarily match the symbol period, and may be an integral multiple of the symbol period, for example.

  The control unit 62 outputs the channel estimation unit control information 63 based on the communication method performed by the receiving device. However, when the receiving device moves, the moving speed information of the receiving device is input and the moving speed information is used. Then, the channel estimation unit control information 63 may be generated and output.

  The FFT unit 55 converts the propagation path estimation value in the channel estimation unit 54 into a frequency domain representation again and outputs it to the weight calculation unit 56.

  The weight calculation unit 56 calculates timing and phase / amplitude compensation values for the signal in the frequency domain representation converted by the FFT unit 55. Control of the output update cycle, calculation method, and the like in the weight calculation unit 56 is controlled by the control unit 62 via the weight calculation unit control information 64 based on the time length unit.

  The control unit 62 outputs the weight calculation unit control information 64 based on the communication method performed by the receiving device. However, when the receiving device moves, the moving speed information of the receiving device is input and the moving speed information is used. The weight calculation unit control information 64 may be generated and output.

  The multiplier 57 multiplies the compensation value output from the weight calculator 56 and the output from the multiplier 52. By this processing, the timing, phase and amplitude of the received signal are collectively compensated in the frequency domain. That is, as in the first and second embodiments, in the receiving apparatus, the propagation delay time for each symbol is compensated for the variation in the reception timing due to the variation in the propagation delay time, and the phase variation due to the variation in the propagation path is compensated. On the other hand, the phase of each symbol is aligned with the accuracy of channel estimation, so that in-phase addition can be performed at a later stage.

  The multiplier 57 outputs the multiplied signal to the adder 58 that constitutes a cyclic adder together with the integrator 59. The case where the signals {S1, S2, S3, S4} for four consecutive symbols are added in units of the above-described time length will be described as an example. When the signal S1 of the first symbol is input, the adder 58 Of the two inputs, the input signal on the side connected to the integrator 59 becomes zero. As a result, the signal S1 is output as it is from the adder 58 and stored in the integrator 59. Next, when the signal S2 of the second symbol is input, in the adder 58, the signal S2 of the symbol is connected to the integrator 59 on the side connected to the multiplier 57 of the two inputs. The value of the integration unit 59, that is, the signal S1 is input to the closed side. As a result, a signal addition value S1 + S2 is output from the adding unit 58 and input to the integrating unit 59. In addition, although the signal of each symbol is different from the symbol in the first embodiment, in order to facilitate the description, here, a symbol such as “S1” is used as in the first embodiment.

  When this operation is repeated up to the signal S4 of the fourth symbol, the signal adding value S1 + S2 + S3 + S4 for four symbols is finally stored in the integrating unit 59. The integration unit 59 outputs the stored value to the IFFT unit 60 and clears it to zero according to the integration unit control information 65 based on the time length unit output from the control unit 62 when the signal addition values S1 + S2 + S3 + S4 for four symbols are accumulated. To do.

  The control unit 62 outputs the integration unit control information 65 based on the communication method performed by the receiving device. When the receiving device moves, the information on the moving speed of the receiving device is input and the moving speed information is used. The integration unit control information 65 may be generated and output.

  IFFT unit 60 converts the stored value (signal addition value S1 + S2 + S3 + S4) output from integration unit 59 into a time domain signal and outputs the time domain signal to demodulation unit 61.

  The demodulator 61 performs demodulation processing using the time domain correlation waveform added for four symbols.

  The receiving apparatus according to the present embodiment is not limited to 4-symbol repeated transmission, and any number of symbols can be applied.

  Further, in the receiving apparatus according to the present embodiment, the configuration example in the case where there is one receiving antenna 41 in FIG. 6 is shown, but the present invention is not limited to this, and the configuration using a plurality of receiving antennas is also possible. Applicable.

  In the receiving apparatus, there are a plurality of FFT units and IFFT units in FIG. 6, but in implementation, the number may be reduced by sharing by time division processing.

  Further, when it is necessary to simultaneously demodulate a plurality of code-multiplexed signals as in CDMA, the receiving apparatus parallelizes the configurations from the multipliers 49 and 52 that perform despread processing to the demodulator 61 by the number of multiplexed signals. Even if the number of components is less than the number of multiplexed signals, the time division processing may be performed while changing the spreading code corresponding to the signal. When the pilot signals are common, it is not necessary to prepare a plurality of multiplication units 49.

  Further, in the receiving apparatus, in FIG. 6, the spread codes 47 and 50 are subjected to discrete Fourier transform by the FFT units 48 and 51 in the despreading process in the frequency domain, but the present invention is not limited to this. The outputs of the FFT units 48 and 51 may be calculated in advance and stored in a memory inside the receiving apparatus.

  As described above, in this embodiment, in the CSK communication system to which symbol repetition is applied, the receiving apparatus compensates for timing fluctuations associated with propagation path length fluctuations in the frequency domain in addition to phase and amplitude. Integration of partial correlation values or symbol addition was performed. As a result, even when the propagation path greatly fluctuates between the despreading length or the symbol repetition length, high-quality reception can be realized, and further, the receiving apparatus can be reduced in size and power consumption. Have

  Furthermore, since the receiving apparatus according to the present embodiment can remove or suppress the interference signal, in addition to the effect of the receiving apparatus according to the second embodiment, there is an effect that can realize high-quality reception resistant to interference.

  As described above, the receiving apparatus according to the present invention is useful for wireless communication, and is particularly suitable when the propagation path between the transmitting side and the transmitting side varies.

  1, 21, 41 Receiving antenna, 2, 22, 42 RF unit, 3, 23, 43 A / D conversion unit, 4, 26, 54 Channel estimation unit, 5 Variable delay line, 6, 28, 49, 52, 57 Multiplier, 7, 29, 58 Adder, 8, 30, 59 Integrator, 9, 31, 61 Demodulator, 10, 32, 62 Controller, 24, 25 Despreader, 27 Cyclic shifter, 44, 48 , 51, 55 FFT unit, 45 interference measurement unit, 46 interference removal unit, 53, 60 IFFT unit, 56 weight calculation unit.

Claims (11)

Channel estimation means for estimating the state of a radio propagation path from a known pilot signal and outputting a channel estimation result for each specified time length;
Compensation means for compensating the delay time and phase of the received signal for each time length based on the channel estimation result;
Cyclic addition means for cyclically adding the signal compensated by the compensation means in units of the time length;
Demodulation means for performing demodulation processing using the integration result from the cyclic addition means;
Control means for designating the time length, and for controlling the timing at which the cyclic addition means outputs the integration result to the demodulation means based on the time length;
A receiving apparatus comprising: A receiving device for receiving a spread spectrum signal,
Channel estimation means for estimating the state of a radio propagation path from a known pilot signal and outputting a channel estimation result for each specified time length;
Despreading means for despreading the received signal for each time length;
Compensation means for compensating the delay time and phase of the received signal after despreading for each time length based on the channel estimation result;
Cyclic addition means for cyclically adding the signal compensated by the compensation means in units of the time length;
Demodulation means for performing demodulation processing using the integration result from the cyclic addition means;
Control means for designating the time length, and for controlling the timing at which the cyclic addition means outputs the integration result to the demodulation means based on the time length;
A receiving apparatus comprising: The despreading means, the compensation means, and the cyclic addition means each perform processing in the frequency domain.
The receiving device according to claim 2. The time length is the length of the symbol period of the transmission signal,
The receiving apparatus according to claim 1, 2, or 3. The time length is set to a length that is an integral multiple of the symbol period of the transmission signal.
The receiving apparatus according to claim 1, 2, or 3. The compensation means compensates reception timing variation of a reception signal due to variation in propagation delay time for each time length.
The receiving apparatus according to claim 1, wherein the receiving apparatus includes: The compensation means compensates for a phase variation of a received signal due to a propagation path variation for each time length.
The receiving apparatus according to claim 1, wherein the receiving apparatus includes: The compensation means multiplies the amplitude value of the received signal by a weighting coefficient according to the channel estimation result.
The receiving apparatus according to claim 1, wherein the receiving apparatus includes: The compensation means performs compensation of the reception timing fluctuation using the received signal or a signal obtained by cyclically shifting the signal obtained by converting the received signal within the time length.
The receiving apparatus according to claim 6. further,
Interference processing means for detecting interference from a signal obtained by converting the received signal into a frequency domain, and multiplying a frequency component in which interference is detected by a coefficient for reducing interference;
The receiving apparatus according to claim 3, further comprising: further,
Interference processing means for detecting interference from a signal obtained by converting the received signal into a frequency domain, and replacing a frequency component in which the interference is detected with a fixed value;
The receiving apparatus according to claim 3, further comprising:

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