JP2010068585A - Power conversion device - Google Patents

Power conversion device Download PDF

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Publication number
JP2010068585A
JP2010068585A JP2008230887A JP2008230887A JP2010068585A JP 2010068585 A JP2010068585 A JP 2010068585A JP 2008230887 A JP2008230887 A JP 2008230887A JP 2008230887 A JP2008230887 A JP 2008230887A JP 2010068585 A JP2010068585 A JP 2010068585A
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Japan
Prior art keywords
power conversion
smoothing capacitor
connected
terminals
semiconductor switching
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JP2008230887A
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Inventor
Matahiko Ikeda
Akihiko Iwata
Masaru Kobayashi
Hirotoshi Maekawa
Tatsuya Okuda
Giichi Tsunoda
Takahiro Urakabe
博敏 前川
達也 奥田
勝 小林
明彦 岩田
又彦 池田
隆浩 浦壁
義一 角田
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Mitsubishi Electric Corp
三菱電機株式会社
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Priority to JP2008230887A priority Critical patent/JP2010068585A/en
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Abstract

<P>PROBLEM TO BE SOLVED: To obtain a power conversion device which can reduce a current flowing to a smoothing capacitor even if a capacity as the power conversion device is the same, can reduce a power loss of a current detection means for detecting the current, can suppress the heat generation of the current detection means, and can make the current detection means small in size and low in price. <P>SOLUTION: The power conversion device includes: a power conversion circuit which has a serial body of an IGBT, is connected between terminals of the smoothing capacitor CH, and performs power conversion by on/off-driving the IGBT at a prescribed cycle; a shunt resistor 140 which detects a current flowing to the smoothing capacitor CH; and a transistor which turns off the IGBT when an output of the shunt resistor 140 exceeds a prescribed set value. The power conversion circuit is constituted of two (unit) power conversion circuits 110, 120 which are connected in parallel with each other with respect to the smoothing capacitor CH in the power conversion circuit, and phases for on/off-driving the IGBTs of both the power conversion circuits 110, 120 are displaced from each other at π (180°). <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a power conversion device using semiconductor switching elements such as a metal-oxide-semiconductor field-effect transistor (MOSFET) and an insulated gate bipolar transistor (IGBT), and an excessive current flows due to a failure of the semiconductor switching element. It is related with the power converter device provided with the overcurrent protection circuit which can be stopped safely, without damaging the components in a power converter device.

  A conventional power converter having an overcurrent protection circuit includes a smoothing capacitor that smoothes a DC voltage between DC terminals and a semiconductor switching element that is bridge-connected between the DC terminals, and detects a current that flows through the capacitor. A detection unit, a comparator for comparing the output of the current detection unit and a predetermined overcurrent detection level, and detecting an overcurrent when the output of the current detection unit is greater than the overcurrent detection level; and When the comparator detects an overcurrent, the gate potential of the semiconductor switching element is lowered below the normal ON gate potential and the gate potential of the semiconductor switching element is lowered to the normal OFF gate potential. A circuit is provided (see, for example, Patent Document 1).

Japanese Patent Laid-Open No. 7-15948 (see claim 1, FIG. 1)

  In the power converter provided with the overcurrent protection circuit disclosed in Patent Document 1, since the current value of the current detection means for detecting the current flowing through the smoothing capacitor is large, a shunt resistor with relatively low cost and good detection accuracy is used. In such a case, there is a problem that heat generation becomes large and the current detecting means becomes large and expensive. In addition, a non-contact type current sensor using a Hall element is used, which is relatively expensive and difficult to detect, but has the advantage of not requiring an interface circuit with the control circuit because the detection signal is isolated and can be extracted. Even in this case, since the size of the magnetic circuit unit used for the detecting means increases or decreases in accordance with the current value, there is a similar problem.

  The present invention has been made to solve the conventional problems as described above. Even if the capacity of the power converter is the same, the current flowing through the smoothing capacitor can be reduced, and the current for detecting this current can be reduced. An object of the present invention is to obtain a power conversion device that can make the detection means small and inexpensive with low power loss and reduced heat generation.

A power conversion device according to a first aspect of the present invention has a series body of a first smoothing capacitor and a semiconductor switching element, is connected between terminals of the first smoothing capacitor, and drives the semiconductor switching element on and off at a predetermined cycle. Power provided with a power conversion circuit for performing power conversion, current detection means for detecting a current flowing through the first smoothing capacitor, and a cutoff means for turning off the semiconductor switching element when the output of the current detection means exceeds a predetermined set value In the conversion device,
The power conversion circuit is composed of a plurality of unit power conversion circuits connected in parallel to each other to the first smoothing capacitor, and the timing in the cycle of turning on and off the semiconductor switching element is different in each unit power conversion circuit. It is a thing.

A power conversion device according to a second invention has a series body of first, second and third smoothing capacitors and a series body of semiconductor switching elements, and is connected between terminals of the first smoothing capacitor at a predetermined cycle. A first power conversion circuit that performs power conversion by driving the semiconductor switching element on and off, and has a series body of semiconductor switching elements and is connected between terminals of a second smoothing capacitor and is driven on and off at a predetermined cycle. A second power conversion circuit that performs power conversion by connecting, a series body of semiconductor rectifier elements that are connected between terminals of the third smoothing capacitor and that can conduct in one direction, and an intermediate between the series bodies of semiconductor switching elements A semiconductor device comprising a series body of a resonance inductor and a resonance capacitor connected between a connection point and an intermediate connection point of a series body of each semiconductor rectifier Using the charging / discharging of the resonance capacitor based on the on / off drive of the switching element, the DC voltage is applied between the serial connection body terminals of the first, second and third smoothing capacitors and between the terminals of the third smoothing capacitor. In a power conversion device that performs conversion,
A current detecting means for detecting a current flowing through the first smoothing capacitor and a current flowing through the second smoothing capacitor; and a blocking means for turning off the semiconductor switching element when the output of the current detecting means exceeds a predetermined set value. ,
The first power conversion circuit is composed of a plurality of first unit power conversion circuits connected in parallel to the first smoothing capacitor, and the second power conversion circuit is connected in parallel to the second smoothing capacitor. A plurality of second unit power conversion circuits, and a series body of semiconductor rectifier elements connected to a third smoothing capacitor in parallel with each other, the number of first unit power conversion circuits and the second unit power conversion Consists of a unit series body of the sum of the number of circuits and
A cycle in which the semiconductor switching elements in the first power conversion circuit are driven to be turned on and off at different timings in each first unit power conversion circuit, and the semiconductor switching elements in the second power conversion circuit are driven to be turned on and off. Are different from each other in each of the second unit power conversion circuits.

  In the power conversion device according to the first aspect of the present invention, the power conversion circuit is composed of a plurality of unit power conversion circuits, and the timing in the cycle for turning on and off the semiconductor switching elements is made different between the unit power conversion circuits. , The timing of the current flowing from the first smoothing capacitor to the semiconductor switching element of each unit power conversion circuit is shifted, and as a result, the effective current value flowing through the first smoothing capacitor is reduced, and the current detecting means reduces the power loss. There is an effect that heat generation is suppressed and the device is small and inexpensive.

  In the power conversion device according to the second aspect of the invention, the power conversion device converts DC voltage between the series connection body terminals of the first, second and third smoothing capacitors and between the terminals of the third smoothing capacitor. According to the same principle as in the first invention, the effective current value flowing through the first and second smoothing capacitors is reduced, and the current detecting means has an effect that the power loss is small and heat generation is suppressed, and the size is low. .

Embodiment 1 FIG.
Hereinafter, a power converter according to Embodiment 1 of the present invention will be described.
FIG. 1 shows a configuration of a power conversion device according to Embodiment 1 of the present invention. This power converter can convert voltage from a high DC voltage VH to a low DC voltage VL, and can convert voltage from a low DC voltage VL to a high DC voltage VH. In the power conversion device 100 shown in FIG. 1, a power conversion circuit that has a serial body of semiconductor switching elements to be described later and performs power conversion by ON / OFF driving thereof includes two unit power conversion circuits 110 connected in parallel to each other, 120. In the following description, for example, the unit power conversion circuit 110 is appropriately abbreviated as the power conversion circuit 110 within a range where no misunderstanding occurs.

  The power conversion apparatus 100 further includes a control circuit 130 that outputs a gate signal to the power conversion circuits 110 and 120, a high-voltage side smoothing capacitor CH that is a first smoothing capacitor, and a low voltage that is a second smoothing capacitor. Side smoothing capacitor CL, shunt resistor 140 as current detecting means connected in series to the smoothing capacitor CH, an inverting amplifier circuit 150 for inverting and amplifying the current detection voltage signal from the shunt resistor 140, and the inverting and amplified voltage The signal is compared with a reference voltage Vref which is a predetermined set value, and when the voltage signal is large (overcurrent), the comparator 160 outputs a high voltage, and the high voltage output from the comparator 160 is latched to power the conversion circuit And a latch circuit 170 for outputting to 110 and 120.

  The power conversion circuits 110 and 120 include ground terminals 110TG and 120TG, low voltage side input / output terminals 110TVL and 120TVL, high voltage side input / output terminals 110TVH and 120TVH, and a gate signal terminal for driving a high voltage side semiconductor switching element. 110TSH, 120TSH, gate signal terminals 110TSL, 120TSL for driving the low voltage side semiconductor switching elements, cutoff signal terminals 110TCH, 120TCH for the high voltage side semiconductor switching elements, and cutoff signal terminals 110TCL for the low voltage side semiconductor switching elements, 120 TCL.

Next, details of the connection configuration will be described.
One terminal of the smoothing capacitor CH is connected to one terminal of the shunt resistor 140 and the input terminal of the inverting amplifier circuit 150, and the other terminal is connected to the high-voltage side input / output terminals 110 TVH and 120 TVH of the power conversion circuits 110 and 120. Connected. The other terminal of the shunt resistor 140 is connected to the ground terminals 110TG and 120TG of the power conversion circuits 110 and 120 and one terminal of the smoothing capacitor CL and is grounded. The other terminal of the smoothing capacitor CL is connected to the low voltage side input / output terminals 110TVL and 120TVL of the power conversion circuits 110 and 120. The voltage VH is input or output between the input / output voltage terminals (110TVH, 110TG) and (120TVH, 120TG), and the voltage VL is output between the input / output voltage terminals (110TVL, 110TG), (120TVL, 120TG). Or it has been entered.

The output terminal of the inverting amplifier circuit 150 is connected to the + side input terminal of the comparator 160. The reference voltage Vref is input to the − side input terminal of the comparator 160, and the output terminal is connected to the input terminal of the latch circuit 170. The output terminal of the latch circuit 170 is connected to the cutoff signal terminals 110TCL, 110TCH, 120TCL, and 120TCH.
The control circuit 130 outputs gate signals Gate1H, Gate1L, Gate2H, and Gate2L according to the required voltage conversion specifications of the voltages VH and VL, and the output gate signals are the gate signal terminals of the power conversion circuits 110 and 120. 110 TSH, 110 TSL, 120 TSH, 120 TSL.

Next, details of internal connections of the power conversion circuits 110 and 120 will be described. However, since the configurations of the power conversion circuits 110 and 120 are the same, only the connections in the power conversion circuit 110 will be described here.
The power conversion circuit 110 includes IGBTs 110SH and 110SL, which are semiconductor switching elements, diodes 110DH and 110DL, an inductor 110L, and gate drive circuits 110DrH and 110DrL, which will be described in detail in the subsequent overcurrent protection operation. , And 110SLH and 110TrL for fixing the input voltage of the gate drive circuits 110DrH and 110DrL to the emitter terminal voltage, and the IGBT 110SH and the transistor 110TrH arranged on the high voltage side for the purpose of turning off the gate voltage of the 110SL. And photocouplers 110PH and 110PTr for electrically isolating signals between the control circuit 130 and the latch circuit 170, respectively.

  In this application, the interruption means of the present application responsible for the overcurrent protection operation corresponds to the inverting amplifier circuit 150, the comparator 160, the latch circuit 170, and the transistors 110TrH and 110TrL in FIG.

  The collector terminal of the IGBT 110SH is connected to the cathode terminal of the diode 110DH and the high voltage side input / output terminal 110TVH, and the emitter terminal is the anode terminal of the diode 110DH, one terminal of the inductor 110L, and the collector terminal of the IGBT 110SL. And the cathode terminal of the diode 110DL and the emitter terminal of the transistor 110TrH. The emitter terminal of the IGBT 110SL is connected to the anode terminal of the diode 110DL, the ground terminal 110TG, and the emitter terminal of the transistor 110TrL. The other terminal of the inductor 110L is connected to the low voltage side input / output terminal 110TVL.

  The gate terminal of the IGBT 110SH is connected to the output terminal of the gate drive circuit 110DrH, and the input terminal of the gate drive circuit 110DrH is connected to the output terminal of the photocoupler 110PH and the collector terminal of the transistor 110TrH. The input terminal of the photocoupler 110PH is connected to the gate signal terminal 110TSH. The base terminal of the transistor 110TrH is connected to the output terminal of the photocoupler 110PTr, and the input terminal of the photocoupler 110PTr is connected to the cutoff signal terminal 110TCH.

  The gate terminal of the IGBT 110SL is connected to the output terminal of the gate drive circuit 110DrL, and the input terminal of the gate drive circuit 110DrL is connected to the collector terminal of the transistor 110TrL and the gate signal terminal 110TSL. The base terminal of the transistor 110TrL is connected to the cutoff signal terminal 110TCL.

Next, an operation related to DC voltage conversion will be described.
The power conversion circuit 110 adjusts the voltage VH input between the input / output voltage terminals (110TVH, 110TG) to a voltage VL between the input / output voltage terminals (110TVL, 110TG) by driving the IGBT 110SH on and off. Output. Further, by turning on / off the IGBT 110SL, the voltage VL input between the input / output voltage terminals (110TVL, 110TG) is boosted and adjusted to the voltage VH between the input / output voltage terminals (110TVH, 110TG). The power conversion circuit 110 is a non-insulated step-up / step-down DC / DC converter, and details of the operation are described in, for example, Non-Patent Document 1 (Electrical Society / Semiconductor Power Conversion System Research Special Committee: “Power Electronics Circuit”). , Ohm Co., pp. 245-265, 2000), etc., only the outline of the operation will be described below.

The step-down adjustment operation will be described as an example.
Gate signals Gate1H and Gate1L are output from the control circuit 130, the gate signal Gate1H is transmitted to the gate terminal of the photocoupler 110PH → gate drive circuit 110DrH → IGBT110SH, and the gate signal Gate1L is transmitted to the gate terminal of the gate drive circuit 110DrL → IGBT110SL. 110SL are turned on and off. The gate voltage is on at a high voltage and off at a low voltage. Here, the gate signals Gate1H and Gate1L are also at a high voltage, and the IGBT is turned on, and at a low voltage, the IGBT is turned off. At this time, the IGBT 110SL may always be in an off state by setting the gate signal Gate1L to a low voltage, but here, the gate signal Gate1L is an inverted voltage signal of the gate signal Gate1H and is in an on / off operation. By adjusting the on-duty ratio of the on / off operation of the IGBT, the step-down adjustment of VH → VL is performed.

  Similarly, in the power conversion circuit 120, an IGBT 120SH (not shown) corresponding to the IGBT 110SH of the power conversion circuit 110 and an IGBT 120SL (not shown) corresponding to the IGBT 110SL by the gate signals Gate2H and Gate2L output from the control circuit 130. Performs an on / off operation and adjusts the on-duty ratio to perform step-down adjustment from VH to VL.

As an example, the voltage VH is 600 V, the voltage VL is 200 V, the output power is 13 kW, the inductors 110 L and 120 L (components of the power conversion circuit 120, not shown) have an inductance value of 500 μH and a driving frequency of 10 kHz. The operation of the power conversion circuits 110 and 120 will be described.
FIG. 2 shows voltage waveforms of the gate signals Gate1H and Gate2H, input current waveforms of the power conversion circuits 110 and 120, and a current waveform of the smoothing capacitor CH. As shown in the figure, the power conversion circuits 110 and 120 operate with phases shifted from each other by π (180 degrees), which is the timing in the cycle of on-off driving. When such an operation is performed, the input current is dispersed, and as a result, the effective current value of the smoothing capacitor CH can be kept small. In this case, the effective current value of the smoothing capacitor is 18 Arms.

As a comparative example, a case where the same power is voltage-converted by one power conversion circuit will be described. The conditions are a voltage VH of 600 V, a voltage VL of 200 V, an output power of 13 kW, an inductor inductance value of 250 μH (equivalent to two 500 μH inductors in parallel), and a drive frequency of 10 kHz.
FIG. 3 shows the gate signal, the waveform of the input current of the conversion circuit, and the waveform of the current of the smoothing capacitor. As can be seen from the figure, the input current is concentrated without being dispersed. As a result, the effective current value of the smoothing capacitor is increased. In this case, the effective current value of the smoothing capacitor is 33.4 Arms.

  From the above, the current flowing through the shunt resistor 140 of the power conversion device of the present invention can be 0.54 times that of the conventional one. This has the effect of reducing the heat generation of the shunt resistor to 30% of that of the conventional one. The shunt resistor can be made small and inexpensive, can be made safe and have a long life, and at the same time, the efficiency of the power converter is improved. Can do.

Next, the overcurrent protection operation will be described.
The overcurrent protection operation will be described by taking as an example a case where the IGBT 110SL has a short circuit failure during the period when the IGBT 110SH of the power conversion circuit 110 is on.
FIG. 4 shows the voltage waveform of the gate signal Gate1H, the voltage waveform corresponding to the voltage Vref at the negative input terminal of the comparator 160 and the detected current at the positive input terminal, and the current waveform of the IGBT 110SH.

  The current flowing through the smoothing capacitor CH is converted into a voltage signal by the shunt resistor 140, and inverted and amplified by the inverting amplifier circuit 150 (in order to output the voltage signal when the overcurrent is generated as a positive voltage with reference to the ground potential, the inverting amplification is performed. Circuit is used) and input to the + input terminal of the comparator 160. When a short circuit failure occurs in the IGBT 110SL, the voltage at the + input terminal of the comparator 160 increases rapidly as shown in the figure (time t1⇒t2 in the figure). When the voltage at the + input terminal exceeds the voltage Vref at the −input terminal (time t2 in the figure), the comparator 160 detects an overcurrent and outputs a high voltage. This high voltage is maintained by the latch circuit 170, and when the output voltage of the latch circuit 170 changes from low to high, the transistors are connected via the cutoff signal terminals 110TCH, 110TCL, 120TCH, 120TCL of the power conversion circuits 110, 120. A high voltage signal is transmitted to the bases of 110TrH, 110TrL, 120TrH (not shown), and 120TrL (not shown). When each transistor is turned on, the inputs of the gate driving circuits 110DrH, 110DrL, 120DrH (not shown), 120DrL (not shown) become a low voltage, and the IGBTs 110SH, 110SL, 120SH (not shown), 120SL (not shown). ) Decreases to the emitter potential, and each IGBT is turned off (time t3 → t4 in the figure).

  As described above, by detecting the current of the smoothing capacitor, the overcurrent state is detected, and the IGBT in the power conversion circuit is turned off, thereby preventing the power conversion device from being damaged. The current flowing into the power conversion circuit is not only from the smoothing capacitor but also from the connected power supply. However, the high-frequency impedance between the power supply and the power conversion circuit is large compared to that between the smoothing capacitor and the power conversion circuit. In order to detect such an overcurrent phenomenon, a method of detecting the current of the smoothing capacitor may be used. Another advantage is that only one current detection point is required.

In the above description, the phase difference of the on / off drive operation between the power conversion circuits 110 and 120 is π. However, the effect of reducing the current of the smoothing capacitor CH is reduced, but the phase difference is set to other than π. However, it goes without saying that the current is reduced as compared with the prior art.
In the above description, the configuration in which the power conversion circuit is divided into the two (unit) power conversion circuits 110 and 120 has been described. However, this is divided into three or more, and each (unit) power conversion circuit is divided. By making the phase difference of operation different, the current of the smoothing capacitor is further reduced, and further, the heat generation of the shunt resistor is suppressed, and the size can be reduced.

In the above description, it has been described that the current detection is performed by the shunt resistor 140. However, a non-contact type using a magnetic circuit that forms a magnetic field based on a current flowing through the smoothing capacitor and a Hall element provided in the magnetic field. The same effect can be obtained by using this current detector. In this case, the magnetic circuit (core material) used for the current detector can be reduced in size, and the current detector can be reduced in size and cost.
Further, in the above description, the case where the power conversion device performs the step-down operation from the voltage VH to VL has been described. However, the power conversion circuit also performs the step-up operation from the voltage VL to VH as in the above description. Is divided and operated with different phases, the current of the smoothing capacitor can be reduced, and the current detection portion can be made smaller and less expensive.

Embodiment 2. FIG.
Next, a power converter according to Embodiment 2 of the present invention will be described.
This power conversion device 200 is a voltage doubler DC / DC converter that boosts an input voltage VL twice and outputs it as a voltage VH. FIG. 5 shows a configuration of power conversion device 200 according to Embodiment 2 of the present invention. The power conversion apparatus 200 includes (unit) power conversion circuits 210 and 220, a control circuit 230, a smoothing capacitor Cs1 that is a first smoothing capacitor, a smoothing capacitor Cs2 that is a second smoothing capacitor, a shunt resistor 140, and an inverting amplifier circuit 150. , A comparator 160, a reference voltage Vref, and a latch circuit 170.
The shunt resistor 140, the inverting amplifier circuit 150, the comparator 160, the reference voltage Vref, and the latch circuit 170 are the same as those in the first embodiment.

  The power conversion circuits 210 and 220 include ground terminals 210TG and 220TG, low-voltage side input voltage terminals 210TVL and 220TVL, high-voltage side output voltage terminals 210TVH and 220TVH, and a high-voltage side semiconductor switching element driving gate signal terminal. 210TSH and 220TSH, gate signal terminals 210TSL and 220TSL for driving low-voltage side semiconductor switching elements, and cutoff signal terminals 210TC and 220TC for semiconductor switching elements.

Next, details of the connection will be described.
One terminal of the smoothing capacitor Cs1 is connected to one terminal of the shunt resistor 140 and the input terminal of the inverting amplifier circuit 150, and the other terminal is input voltage terminals 210TVL and 220TVL on the low voltage side of the power conversion circuits 210 and 220. And connected to one terminal of the smoothing capacitor Cs2. The other terminal of the shunt resistor 140 is connected to the ground terminals 210TG and 220TG of the power conversion circuits 210 and 220 and grounded. The other terminal of the smoothing capacitor Cs2 is connected to the output voltage terminals 210TVH and 220TVH on the high voltage side of the power conversion circuits 210 and 220. The voltage VL is input between the input voltage terminals (210TVL, 210TG) and (220TVL, 220TG), and the voltage VH is output between the output voltage terminals (210TVH, 210TG) and (220TVH, 220TG).

Similarly to the first embodiment, the output terminal of the inverting amplifier circuit 150 is connected to the + side input terminal of the comparator 160. The reference voltage Vref is input to the − side input terminal of the comparator 160, and the output terminal is connected to the input terminal of the latch circuit 170. The output terminal of the latch circuit 170 is connected to the cutoff signal terminals 210TC and 220TC.
The control circuit 230 outputs gate signals Gate1H, Gate1L, Gate2H, and Gate2L, and the output gate signals are input to the gate signal terminals 210TSH, 210TSL, 220TSH, and 220TSL of the power conversion circuits 210 and 220, respectively.

Next, details of internal connections in the power conversion circuits 210 and 220 will be described. However, since the configurations of the power conversion circuits 210 and 220 are the same, only the connections in the power conversion circuit 210 will be described here.
The power conversion circuit 210 includes MOSFETs 210SH and 210SL which are semiconductor switching elements (in the figure, a parasitic diode of the MOSFET is shown between the drain terminal and the source terminal), a resonance inductor 210Lr, a resonance capacitor 210Cr, A gate drive circuit 210Dr that drives the MOSFETs 210SH and 210SL, and a transistor 210TrH that fixes the two input voltages of the gate drive circuit 210Dr to the source terminal voltage of the MOSFET 210SL for the purpose of turning off the gate voltage of the MOSFET as the source voltage. 210TrL and diodes 210DH and 210DL which are semiconductor rectifier elements.

  In this application, the interruption means of the present application responsible for the overcurrent protection operation corresponds to the inverting amplifier circuit 150, the comparator 160, the latch circuit 170, and the transistors 210TrH and 210TrL in FIG.

  The drain terminal of the MOSFET 210SH is connected to the anode terminal of the diode 210DL and the input voltage terminal 210TVL on the low voltage side, and the source terminal is connected to the drain terminal of the MOSFET 210SL and one terminal of the resonance capacitor 210Cr. . The source terminal of the MOSFET 210SL is connected to the emitter terminals of the transistors 210TrH and 210TrL and the ground terminal 210TG. The other terminal of the resonance capacitor 210Cr is connected to one terminal of the resonance inductor 210Lr, and the other terminal of the resonance inductor 210Lr is connected to the cathode terminal of the diode 210DL and the anode terminal of the diode 210DH. . The cathode terminal of the diode 210DH is connected to the output voltage terminal 210TVH on the high voltage side.

  In other words, the connection configuration described above includes a resonance inductor 210Lr and a resonance capacitor 210Cr between an intermediate connection point of the series body of the MOSFET 210SH and the MOSFET 210SL and an intermediate connection point of the series body of the diode 210DH and the diode 210DL. Will be connected.

  The gate terminals of the MOSFETs 210SH and 210SL are respectively connected to the output terminal of the gate drive circuit 210Dr, and the input terminals of the gate drive circuit 210Dr are connected to the gate signal terminals 210TSH and 210TSL, respectively. The gate signal Gate1H for driving the MOSFET 210SH and the gate signal Gate1L for driving 210SL are input to the input terminal of the gate driving circuit 210Dr. The voltage level of the gate signal of the high-voltage side MOSFET 210SH is converted from the ground potential to the source potential of the MOSFET 210SH in the gate drive circuit 210Dr.

  The gate signal terminal 210TSH is connected to the collector terminal of the transistor 210TrH, and the gate signal terminal 210TSL is connected to the collector terminal of the transistor 210TrL. The emitter terminals of the transistors 210TrH and 210TrL are connected to the ground terminal 210TG, and the base terminal is connected to the cutoff signal terminal 210TC.

Next, an operation related to DC voltage conversion will be described. However, since the operations of the power conversion circuits 210 and 220 are the same, only the operation of the power conversion circuit 210 will be described here.
The voltage doubler DC / DC converter of the present invention outputs the input voltage VL as a voltage VH of the voltage doubled. In the steady state, the smoothing capacitor Cs1 is charged with the voltage VL, and the smoothing capacitor Cs2 is charged with the voltage (VH−VL). The voltage VH has a relationship of VH <VL × 2 when power is supplied from the output terminal. The resonance capacitor 210Cr is charged with the voltage VL on average.

  When the MOSFET 210SL is turned on (210SH is off), the energy is transferred from the smoothing capacitor Cs1 to the resonance capacitor 210Cr in the order of Cs1⇒210DL⇒210Lr⇒210Cr⇒210SL⇒shunt resistor 140. At this time, the on-time is about ½ of the resonance period determined by the resonance capacitor 210Cr and the inductor 210Lr. By setting the on-time in this way, the MOSFET can be turned on / off in a current zero state, and energy can be transferred with high efficiency.

Next, when the MOSFET 210SH is turned on (210SL is off), the energy is transferred from the resonance capacitor 210Cr to the smoothing capacitor Cs2 in the order of 210Cr → 210Lr → 210DH → Cs2 → 210SH. Similarly, in this case, the on-time is about ½ of the resonance period determined by 210Cr and 210Lr, and the energy is transferred with high efficiency. In this way, voltage double boosting is performed by sending energy from Cs1 to Cs2.
The power conversion circuits 210 and 220 have the same resonance period determined by their Lr and Cr.

Next, the overall operation including the power conversion circuits 210 and 220 will be described.
FIG. 6 shows voltage waveforms of the gate signals Gate1H and Gate2H. The gate signal is a high voltage, and the corresponding MOSFET is turned on. As in the first embodiment, the power conversion circuits 210 and 220 operate with a phase difference of operation of π (180 degrees). By performing such an operation, the current of the smoothing capacitor Cs1 can be reduced as in the first embodiment as compared with the case of one conventional power conversion circuit.

  When the voltage VL is 300 V and the output power is 13 kW, the current of the smoothing capacitor Cs1 having a configuration in which the power conversion circuit of the present invention is divided into two and a configuration in which it is not divided is compared. FIG. 7 shows the current of the smoothing capacitor Cs1 in both. As can be seen from the figure, the current of the smoothing capacitor of the present invention is small. Compared with the current effective value, the present invention is 10.5 Arms, whereas the conventional method has 26.2 Arms.

  Since the overcurrent protection operation is the same as that in the first embodiment, the description thereof is omitted.

  From the above, the current flowing through the shunt resistor 140 of the power conversion device according to the second embodiment of the present invention can be 0.40 times that of the conventional one. This has the effect of reducing the heat generation of the shunt resistor to 16% of the conventional one. The shunt resistor can be made small and inexpensive, can be safe and have a long life, and at the same time, the efficiency of the power converter is improved. Can do.

In the above description, the phase difference of the on / off drive operation between the power conversion circuits 210 and 220 is π. However, although the effect of reducing the current of the smoothing capacitor is reduced, the phase difference may be set to other than π. Needless to say, the current is reduced as compared with the prior art.
Further, in the above description, the configuration in which the power conversion circuit is divided into two (unit) power conversion circuits 210 and 220 has been described. However, this is divided into three or more and between each (unit) power conversion circuit. By making the phase difference of operation different, the current of the smoothing capacitor is further reduced, and further, the heat generation of the shunt resistor is suppressed, and the size can be reduced.

In the second embodiment of the present invention, the diodes 210DH and 210DL are used as the semiconductor rectifier elements. However, they may be replaced with MOSFETs so as to be turned on in accordance with the conduction timing. Replacing with MOSFET makes the device expensive, but there is an advantage that various operation modes are possible and the usefulness as the device is improved. In that case, it arrange | positions so that the conduction direction of the parasitic diode of MOSFET and the said diode may become the same.
Further, in the above description, it has been described that the current detection is performed by the shunt resistor, but the same effect can be obtained by using a non-contact type current detector using a Hall element. In this case, the magnetic circuit (core material) used for the current detector can be reduced in size, and the current detector can be reduced in size and cost.

Embodiment 3 FIG.
Next, a power converter according to Embodiment 3 of the present invention will be described.
Unlike the previous second embodiment in which the boosting operation is performed, the power conversion device 300 in the third embodiment steps down the input voltage VH by a factor of 1/2 and outputs it as a voltage VL. It is a DC converter. FIG. 8 shows the configuration of power conversion device 300 according to Embodiment 3 of the present invention. The power conversion device 300 includes (unit) power conversion circuits 310 and 320, a control circuit 330, a smoothing capacitor Cs1 that is a second smoothing capacitor, Cs2 that is a first smoothing capacitor, a shunt resistor 140, an inverting amplification circuit 150, and a comparator. 160, a reference voltage Vref, and a latch circuit 170.
The shunt resistor 140, the inverting amplifier circuit 150, the comparator 160, the reference voltage Vref, and the latch circuit 170 are the same as those described above except that the operation reference voltage of the circuit is changed from the ground terminal to the output voltage terminals 310TVL and 320TVL. This is the same as in the first and second embodiments.

  The power conversion circuits 310 and 320 include ground terminals 310TG and 320TG, low-voltage side output voltage terminals 310TVL and 320TVL, high-voltage side input voltage terminals 310TVH and 320TVH, and a high-voltage side semiconductor switching element driving gate signal terminal. 310TSH, 320TSH, gate signal terminals 310TSL, 320TSL for driving low-voltage side semiconductor switching elements, and cutoff signal terminals 310TC, 320TC for semiconductor switching elements.

Next, details of the connection will be described.
One terminal of the smoothing capacitor Cs1 is connected to the ground terminals 310TG and 320TG and grounded, and the other terminal is one of the output voltage terminals 310TVL and 320TVL on the low voltage side of the power conversion circuits 310 and 320 and one of the shunt resistors 140. Connected to the terminal. The other terminal of the shunt resistor 140 is connected to the input terminal of the inverting amplifier circuit 150 and one terminal of the smoothing capacitor Cs2. The other terminal of the smoothing capacitor Cs2 is connected to the input voltage terminals 310TVH and 320TVH on the high voltage side of the power conversion circuits 310 and 320. The voltage VH is input between the input voltage terminals (310TVH, 310TG) and (320TVH, 320TG), and the voltage VL is output between the output voltage terminals (310TVL, 310TG) and (320TVL, 320TG).

Further, as in the first and second embodiments, the output terminal of the inverting amplifier circuit 150 is connected to the + side input terminal of the comparator 160. The reference voltage Vref is input to the − side input terminal of the comparator 160, and the output terminal is connected to the input terminal of the latch circuit 170. The output terminal of the latch circuit 170 is connected to the cutoff signal terminals 310TC and 320TC.
The control circuit 330 outputs gate signals Gate1H, Gate1L, Gate2H, and Gate2L, and the output gate signals are input to the gate signal terminals 310TSH, 310TSL, 320TSH, and 320TSL of the power conversion circuits 310 and 320, respectively.

Next, details of internal connections of the power conversion circuits 310 and 320 will be described. However, since the configurations of the power conversion circuits 310 and 320 are the same, only the connections in the power conversion circuit 310 will be described here.
The power conversion circuit 310 includes MOSFETs 310SH and 310SL which are semiconductor switching elements (in the drawing, a parasitic diode of the MOSFET is shown between the drain terminal and the source terminal), a resonance inductor 310Lr, a resonance capacitor 310Cr, A gate drive circuit 310Dr for driving the MOSFETs 310SH and 310SL, and a transistor 310TrH for fixing two input voltages of the gate drive circuit 310Dr to the source terminal voltage of the MOSFET 310SL for the purpose of turning off the gate voltage of the MOSFET as a source voltage. 310TrL, diodes 310DH and 310DL which are semiconductor rectifiers, MOSFETs 310SH and 310SL and transistors 310TrH and 310T arranged on the high voltage side L and each control circuit 330 and the photo coupler for electrically insulating the signal between the latch circuit 170 310PH, is composed of a 310PL and 310PTr.

  In the present application, the cutoff means of the present application responsible for the overcurrent protection operation corresponds to the inverting amplifier circuit 150, the comparator 160, the latch circuit 170, and the transistors 310TrH and 310TrL in FIG.

  The drain terminal of the MOSFET 310SH is connected to the input voltage terminal 310TVH on the high voltage side, and the source terminal is connected to the drain terminal of the MOSFET 310SL and one terminal of the resonance inductor 310Lr. The source terminal of the MOSFET 310SL is connected to the emitter terminals of the transistors 310TrH and 310TrL, the output voltage terminal 310TVL on the low voltage side, and the cathode terminal of the diode 310DH. The other terminal of the resonance inductor 310Lr is connected to one terminal of the resonance capacitor 310Cr, and the other terminal of the resonance capacitor 310Cr is connected to the cathode terminal of the diode 310DL and the anode terminal of the diode 310DH. . The anode terminal of the diode 310DL is connected to the ground terminal 310TG.

  The gate terminals of the MOSFETs 310SH and 310SL are respectively connected to the output terminal of the gate drive circuit 310Dr, and the input terminals of the gate drive circuit 310Dr are connected to the gate signal terminals 310TSH and 310TSL via the photocouplers 310PH and 310PL, respectively. The gate signal Gate1H for driving the MOSFET 310SH and the gate signal Gate1L for driving 310SL are input to the input terminal of the gate driving circuit 310Dr. In the gate drive circuit 310Dr, the voltage level of the gate signal of the high voltage side MOSFET 310SH is converted from the potential of the voltage terminal 310TVL to the source potential of the MOSFET 310SH.

  The gate signal terminal 310TSH is connected to the collector terminal of the transistor 310TrH via the photocoupler 310PH, and the gate signal terminal 310TSL is connected to the collector terminal of the transistor 310TrL via the photocoupler 310PL. The emitter terminals of the transistors 310TrH and 310TrL are connected to the voltage terminal 310TVL, and the base terminal is connected to the cutoff signal terminal 310TC via the photocoupler 310PTr.

Next, an operation related to DC voltage conversion will be described. However, since the operations of the power conversion circuits 310 and 320 are the same, only the operation of the power conversion circuit 310 will be described here.
The 1/2 voltage DC / DC converter of the present invention outputs the input voltage VH as the voltage VL of the 1/2 voltage. In the steady state, the smoothing capacitors Cs1 and Cs2 are combined and charged with the voltage VH. The voltage VL has a relationship of VL <VH × (1/2) when power is supplied from the output terminal. The resonance capacitor 310Cr is charged with a voltage (VH−VL) on average.

  When the MOSFET 310SH is turned on (310SL is off), energy is transferred from the smoothing capacitor Cs2 to the resonance capacitor 310Cr in the order of Cs2 → 310SH → 310Lr → 310Cr → 310DH → shunt resistor 140. At this time, as in the second embodiment, the on-time is about ½ of the resonance period determined by the resonance capacitor 310Cr and the inductor 310Lr. By setting the on-time in this way, the MOSFET can be turned on / off in a current zero state, and energy can be transferred with high efficiency.

Next, when the MOSFET 310SL is turned on (310SH is turned off), energy is transferred from the resonance capacitor 310Cr to the smoothing capacitor Cs1 in the order of 310Cr → 310Lr → 310SL → Cs1 → 310DL. In this case as well, the on-time is about ½ of the resonance period determined from 310Cr and 310Lr, and the energy is transferred with high efficiency. Thus, the voltage is stepped down by a factor of 1/2 by sending energy from Cs2 to Cs1.
The power conversion circuits 310 and 320 have the same resonance period determined by their Lr and Cr.

Next, the overall operation including the power conversion circuits 310 and 320 will be described.
FIG. 9 shows voltage waveforms of the gate signals Gate1H and Gate2H. The gate signal is a high voltage, and the corresponding MOSFET is turned on. As in the first and second embodiments, the power conversion circuits 310 and 320 operate with a phase difference of operation of π (180 degrees). By performing such an operation, the current of the smoothing capacitor Cs2 can be reduced as in the first and second embodiments, as compared with the case of one conventional power conversion circuit.

  When the voltage VH is 600 V and the output power is 13 kW, the current of the smoothing capacitor Cs2 having a configuration in which the power conversion circuit of the present invention is divided into two and a configuration in which it is not divided is compared. FIG. 10 shows the current of the smoothing capacitor Cs2 in both. As can be seen from the figure, the current of the smoothing capacitor of the present invention is small. Compared with the current effective value, the present invention is 10.5 Arms, whereas the conventional method has 26.2 Arms.

  Since the overcurrent protection operation is the same as that in the first embodiment, the description thereof is omitted.

  From the above, the current flowing through the shunt resistor 140 of the power conversion device according to the third embodiment of the present invention can be 0.40 times that of the conventional one. This has the effect of reducing the heat generation of the shunt resistor to 16% of the conventional one. The shunt resistor can be made small and inexpensive, can be safe and have a long life, and at the same time, the efficiency of the power converter is improved. Can do.

In the above description, the phase difference of the on / off drive operation between the power conversion circuits 310 and 320 is π. However, although the effect of reducing the current of the smoothing capacitor is reduced, the phase difference may be set to other than π. Needless to say, the current is reduced as compared with the prior art.
Further, in the above description, the configuration in which the power conversion circuit is divided into two (unit) power conversion circuits 310 and 320 has been described. However, this is divided into three or more and each (unit) power conversion circuit is divided. By making the phase difference of operation different, the current of the smoothing capacitor is further reduced, and further, the heat generation of the shunt resistor is suppressed, and the size can be reduced.

In the third embodiment of the present invention, the diodes 310DH and 310DL are used as the semiconductor rectifier elements. However, they may be replaced with MOSFETs so as to be turned on in accordance with the conduction timing. In that case, it arrange | positions so that the conduction direction of the parasitic diode of MOSFET and the said diode may become the same.
Further, in the above description, it has been described that the current detection is performed by the shunt resistor, but the same effect can be obtained by using a non-contact type current detector using a Hall element. In this case, the magnetic circuit (core material) used for the current detector can be reduced in size, and the current detector can be reduced in size and cost.

Embodiment 4 FIG.
Next, a power converter according to Embodiment 4 of the present invention will be described.
The power conversion device 400 is a triple voltage DC / DC converter that boosts an input voltage VL three times and outputs the boosted voltage as a voltage VH. FIG. 11 shows the configuration of power conversion device 400 according to Embodiment 4 of the present invention. The power conversion device 400 includes (unit) power conversion circuits 410 and 420, a control circuit 430, a smoothing capacitor Cs1 that is a first smoothing capacitor, Cs2 and Cs3 that are second and third smoothing capacitors, a shunt resistor 140, an inversion The circuit includes an amplifier circuit 150, a comparator 160, a reference voltage Vref, and a latch circuit 170.
The shunt resistor 140, the inverting amplifier circuit 150, the comparator 160, the reference voltage Vref, and the latch circuit 170 are the same as those in the first embodiment.

  The power conversion circuits 410 and 420 include ground terminals 410TG and 420TG, low-voltage side input voltage terminals 410TVL and 420TVL, intermediate voltage-side voltage terminals 410TVM and 420TVM, high-voltage side voltage terminals 410TVH and 420TVH, Voltage-side semiconductor switching element driving gate signal terminals 410TSH and 420TSH, low-voltage side semiconductor switching element driving gate signal terminals 410TSL and 420TSL, and semiconductor switching element blocking signal terminals 410TC and 420TC are provided.

Next, details of the connection will be described.
One terminal of the smoothing capacitor Cs1 is connected to one terminal of the shunt resistor 140 and the input terminal of the inverting amplifier circuit 150, and the other terminal is an input voltage terminal 410TVL on the low voltage side of the power conversion circuits 410 and 420, 420TVL, a voltage terminal 410TVM on the intermediate voltage side, and one terminal of the smoothing capacitor Cs2. The other terminal of the shunt resistor 140 is connected to the ground terminals 410TG and 420TG of the power conversion circuits 410 and 420 and grounded. The other terminal of the smoothing capacitor Cs2 is connected to the high voltage side voltage terminal 410TVH of the power conversion circuit 410, the intermediate voltage side voltage terminal 420TVM of the power conversion circuit 420, and one terminal of the smoothing capacitor Cs3. . The voltage VL is input between the input voltage terminals (410TVL, 410TG) and (420TVL, 420TG), and the voltage VH is output between the output voltage terminals (420TVH, 420TG).

Similarly to the first embodiment, the output terminal of the inverting amplifier circuit 150 is connected to the + side input terminal of the comparator 160. The reference voltage Vref is input to the − side input terminal of the comparator 160, and the output terminal is connected to the input terminal of the latch circuit 170. The output terminal of the latch circuit 170 is connected to the cutoff signal terminals 410TC and 420TC.
The control circuit 430 outputs gate signals Gate1H, Gate1L, Gate2H, and Gate2L, and the output gate signals are input to the gate signal terminals 410TSH, 410TSL, 420TSH, and 420TSL of the power conversion circuits 410 and 420, respectively.

Next, details of internal connections of the power conversion circuits 410 and 420 will be described. However, since the configurations of the power conversion circuits 410 and 420 are the same, only the connections in the power conversion circuit 410 will be described here.
The power conversion circuit 410 includes MOSFETs 410SH and 410SL which are semiconductor switching elements (in the figure, a parasitic diode of the MOSFET is shown between the drain terminal and the source terminal), a resonance inductor 410Lr, a resonance capacitor 410Cr, A gate drive circuit 410Dr for driving the MOSFETs 410SH and 410SL; and a transistor 410TrH for fixing two input voltages of the gate drive circuit to the source terminal voltage of the MOSFET 410SL for the purpose of turning off the gate voltage of the MOSFET as a source voltage. 410TrL and diodes 410DH and 410DL which are semiconductor rectifiers.

  The drain terminal of the MOSFET 410SH is connected to the input voltage terminal 410TVL on the low voltage side, and the source terminal is connected to the drain terminal of the MOSFET 410SL and one terminal of the resonance capacitor 410Cr. The source terminal of the MOSFET 410SL is connected to the emitter terminals of the transistors 410TrH and 410TrL and the ground terminal 410TG. The other terminal of the resonance capacitor 410Cr is connected to one terminal of the resonance inductor 410Lr, and the other terminal of the resonance inductor 410Lr is connected to the cathode terminal of the diode 410DL and the anode terminal of the diode 410DH. . The cathode terminal of the diode 410DH is connected to the voltage terminal 410TVH on the high voltage side. The anode terminal of the diode 410DL is connected to the voltage terminal 410TVM on the intermediate voltage side.

  The gate terminals of the MOSFETs 410SH and 410SL are respectively connected to the output terminal of the gate drive circuit 410Dr, and the input terminals of the gate drive circuit 410Dr are connected to the gate signal terminals 410TSH and 410TSL, respectively. The gate signal Gate1H for driving the MOSFET 410SH and the gate signal Gate1L for driving 410SL are input to the input terminal of the gate driving circuit 410Dr. The voltage level of the gate signal of the high voltage side MOSFET 410SH is converted from the ground potential to the source potential of the MOSFET 410SH in the gate drive circuit 410Dr.

  The gate signal terminal 410TSH is connected to the collector terminal of the transistor 410TrH, and the gate signal terminal 410TSL is connected to the collector terminal of the transistor 410TrL. The emitter terminals of the transistors 410TrH and 410TrL are connected to the ground terminal 410TG, and the base terminal is connected to the cutoff signal terminal 410TC.

Each component constituting the power conversion circuit 420 corresponds to the component constituting the power conversion circuit 410 as follows.
MOSFET 410SH⇒420SH, 410SL⇒420SL
Diode 410DH⇒420DH, 410DL⇒420DL
Resonant inductor 410Lr⇒420Lr
Resonance capacitor 410Cr⇒420Cr
Gate drive circuit 410Dr⇒420Dr
Transistor 410TrH⇒420TrH, 410TrL⇒420TrL

Next, an operation related to DC voltage conversion will be described.
The triple voltage DC / DC converter of the present invention outputs the input voltage VL as a voltage VH of the triple voltage. In the steady state, the smoothing capacitor Cs1 is charged with the voltage VL, and the smoothing capacitors Cs2 and Cs3 are charged with the voltage (VH−VL) / 2. The voltage VH has a relationship of VH <VL × 3 when power is supplied from the output terminal. The resonance capacitor 410Cr is charged with the voltage VL on average, and the voltage VL + (VH−VL) / 2 is charged on the average with 420Cr.

First, the operation of the power conversion circuit 410 will be described.
When the MOSFET 410SL is turned on (410SH is turned off), energy is transferred from the smoothing capacitor Cs1 to the resonance capacitor 410Cr in the order of Cs1 → 410DL → 410Lr → 410Cr → 410SL → shunt resistor 140. At this time, as in the second and third embodiments, the on-time is about ½ of the resonance period determined by the resonance capacitor 410Cr and the inductor 410Lr. By setting the on-time in this way, the MOSFET can be turned on / off in a current zero state, and energy can be transferred with high efficiency.

  Next, when the MOSFET 410SH is turned on (410SL is turned off), energy is transferred from the resonance capacitor 410Cr to the smoothing capacitor Cs2 in the order of 410Cr → 410Lr → 410DH → Cs2 → 410SH. In this case as well, the on-time is about ½ of the resonance period determined by 410Cr and 410Lr, and the energy is transferred with high efficiency. In this way, voltage double boosting is performed by sending energy from Cs1 to Cs2.

Next, the operation of the power conversion circuit 420 will be described.
When the MOSFET 420SL is turned on (420SH is turned off), the energy is transferred from the smoothing capacitors Cs1 and Cs2 to the resonance capacitor 420Cr in the order of Cs1 → Cs2 → 420DL → 420Lr → 420Cr → 420SL → shunt resistor 140. In this case as well, the on-time is approximately ½ of the resonance period determined by the resonance capacitor 420Cr and the resonance inductor 420Lr. By setting the on-time in this way, the MOSFET can be turned on / off in a current zero state, and energy can be transferred with high efficiency.

Next, when the MOSFET 420SH is turned on (420SL is off), energy is transferred from the resonance capacitor 420Cr to the smoothing capacitors Cs2 and Cs3 in the order of 420Cr → 420Lr → 420DH → Cs3 → Cs2 → 420SH. In this case as well, the on-time is about ½ of the resonance period determined by 420Cr and 420Lr, and the energy is transferred with high efficiency. Thus, the voltage is boosted by a factor of 3 by sending energy from Cs1 to Cs2 and Cs3.
The power conversion circuits 410 and 420 have the same resonance period determined by their Lr and Cr.

Next, the overall operation including the power conversion circuits 410 and 420 will be described.
FIG. 12 shows voltage waveforms of the gate signals Gate1H and Gate2H. The gate signal is a high voltage, and the corresponding MOSFET is turned on. As in the previous embodiments, the power conversion circuits 410 and 420 are operating with a phase difference of operation of π (180 degrees). By performing such an operation, the current of the smoothing capacitor Cs1 can be reduced as in the previous embodiments as compared with the case of one conventional power conversion circuit.

  When the voltage VL is 200 V and the output power is 13 kW, the current of the smoothing capacitor Cs1 having a configuration in which the power conversion circuit of the present invention is divided into two and a configuration in which it is not divided is compared. FIG. 13 shows the current of the smoothing capacitor Cs1 in both. As can be seen from the figure, the current of the smoothing capacitor of the present invention is small. Compared with the current effective value, the present invention is 10.5 Arms, whereas the conventional method has 26.2 Arms.

  Since the overcurrent protection operation is the same as that in the first embodiment, the description thereof is omitted.

  As mentioned above, the electric current which flows into the shunt resistance 140 of the power converter device of Embodiment 4 of this invention can be 0.40 time compared with the past. This has the effect of reducing the heat generation of the shunt resistor to 16% of the conventional one. The shunt resistor can be made small and inexpensive, can be safe and have a long life, and at the same time, the efficiency of the power converter is improved. Can do.

In the above description, the phase difference of the on / off drive operation between the power conversion circuits 410 and 420 is π. However, although the effect of reducing the current of the smoothing capacitor is reduced, the phase difference may be set to other than π. Needless to say, the current is reduced as compared with the prior art.
In the above description, the configuration in which the power conversion circuit is divided into two (unit) power conversion circuits 410 and 420 has been described. However, this is divided into three or more (increasing the step-up ratio), and each power By making the phase difference of the operation between the conversion circuits different, the current of the smoothing capacitor is further reduced, the heat generation of the shunt resistor is further suppressed, and the size can be reduced.

In the fourth embodiment of the present invention, the diodes 410DH and 410DL are used as the semiconductor rectifier elements. However, they may be replaced with MOSFETs so as to be turned on in accordance with the conduction timing. In that case, it arrange | positions so that the conduction direction of the parasitic diode of MOSFET and the said diode may become the same.
Further, in the above description, it has been described that the current detection is performed by the shunt resistor. However, the same effect can be obtained by using a non-contact type current detector using a Hall element. In this case, the magnetic circuit (core material) used for the current detector can be reduced in size, and the current detector can be reduced in size and cost.

By the way, in the fourth embodiment of the present invention, the DC / DC converter that performs the triple boosting from the voltage VL to the voltage VH has been described. With reference to the third embodiment, the MOSFETs 410SH, 410SL, DC / DC that performs step-down of 1/3 times from voltage VH to voltage VL by replacing 420SH and 420SL with diodes, and replacing diodes 410DH, 410DL, 420DH, and 420DL with MOSFETs and turning these MOSFETs on and off It can also be operated as a converter.
At that time, similarly, by operating each power conversion circuit with the phase difference of the operation of the MOSFET being π, the same effect can be obtained.

In the above-described power converter, the series part of the MOSFET corresponds to the first and second power converter circuits connected to the first and second smoothing capacitors, respectively, according to claim 6 of the present application. To do. The shunt resistor is inserted into the first and second smoothing capacitors. The series body of diodes corresponds to the series body connected to the third smoothing capacitor of claim 6.
Further, each of the first and second power conversion circuits is composed of two unit power conversion circuits connected in parallel with each other, and the series body of the diodes includes four units that are the total number of the unit power conversion circuits. It is necessary to configure the unit series body connected in parallel.

It is a figure which shows the structure of the power converter device in Embodiment 1 of this invention. It is the figure which showed the gate signal waveform for driving IGBT which comprises the power converter circuit in Embodiment 1 of this invention, the current waveform input into a power converter circuit, and the current waveform of a smoothing capacitor. The gate signal waveform for driving the IGBT when configured with one power conversion circuit for comparison in the first embodiment of the present invention, the current waveform input to the power conversion circuit, the current waveform of the smoothing capacitor, FIG. It is the figure which showed the gate signal waveform for driving IGBT which comprises the power converter circuit in Embodiment 1 of this invention, the input voltage waveform of the comparator which determines overcurrent, and the current waveform which flows into IGBT. . It is a figure which shows the structure of the power converter device in Embodiment 2 of this invention. It is the figure which showed the gate signal waveform for driving MOSFET which comprises the power converter circuit in Embodiment 2 of this invention. It is the figure which showed the electric current waveform of the smoothing capacitor of Embodiment 2 of this invention, and the case where it comprises with one power converter circuit for a comparison. It is a figure which shows the structure of the power converter device in Embodiment 3 of this invention. It is the figure which showed the gate signal waveform for driving MOSFET which comprises the power converter circuit in Embodiment 3 of this invention. It is the figure which showed the electric current waveform of the smoothing capacitor of Embodiment 3 of this invention and the case where it comprises with one power converter circuit for a comparison. It is a figure which shows the structure of the power converter device in Embodiment 4 of this invention. It is the figure which showed the gate signal waveform for driving MOSFET which comprises the power converter circuit in Embodiment 4 of this invention. It is the figure which showed the current waveform of the smoothing capacitor of Embodiment 4 of this invention, and the case where it comprises with one power converter circuit for a comparison.

Explanation of symbols

100, 200, 300, 400 power converter,
110, 120, 210, 220, 310, 320, 410, 420 (unit) power conversion circuit,
CH, CL, Cs1, Cs2, Cs3 smoothing capacitor,
130, 230, 330, 430 control circuit, 140 shunt resistor,
150 inverting amplifier circuit, 160 comparator, 170 latch circuit,
Vref reference voltage,
110SH, 110SL, IGBT, 210SH, 210SL, 310SH, 310SL, 410SH, 410SL MOSFET,
210DH, 210DL, 310DH, 310DL, 410DH, 410DL diode,
110TrH, 110TrL, 210TrH, 210TrL, 310TrH, 310TrL, 410TrH, 410TrL transistors,
110Dr, 210Dr, 310Dr, 410Dr gate drive circuit,
210Cr, 310Cr, 410Cr resonant capacitors,
210Lr, 310Lr, 410Lr Resonant inductor.

Claims (10)

  1. A power conversion circuit having a serial body of a first smoothing capacitor and a semiconductor switching element, connected between terminals of the first smoothing capacitor, and performing power conversion by driving the semiconductor switching element on and off at a predetermined period; In a power conversion device comprising: current detection means for detecting a current flowing through the first smoothing capacitor; and a cutoff means for turning off the semiconductor switching element when the output of the current detection means exceeds a predetermined set value.
    The power conversion circuit includes a plurality of unit power conversion circuits connected to the first smoothing capacitor in parallel with each other, and the timing in the period for driving the semiconductor switching element on and off is the unit power conversion circuit. A power conversion device characterized in that they are different from each other.
  2. An inductor provided for each unit power conversion circuit, one end of which is connected to an intermediate connection point of the series body of the semiconductor switching elements, and the other end of the inductor connected to each other and one end of the first smoothing capacitor And a second smoothing capacitor connected between the terminals of the first smoothing capacitor using the charge / discharge of the second smoothing capacitor based on on / off driving of the semiconductor switching element and the second smoothing capacitor. The power converter according to claim 1, wherein a DC voltage is converted between terminals of the smoothing capacitor.
  3. A second smoothing capacitor connected in series with the first smoothing capacitor; a semiconductor rectifying element provided for each unit power conversion circuit and connected between terminals of the second smoothing capacitor and capable of conducting in one direction; A series body of a resonance inductor and a resonance capacitor connected between an intermediate connection point of the series body of the semiconductor switching elements and an intermediate connection point of the series body of the semiconductor rectifier elements. , Between the terminals of the first smoothing capacitor and the serially connected body terminals of the first and second smoothing capacitors by using charge / discharge of the resonance capacitor based on the on / off drive of the semiconductor switching element. The power converter according to claim 1, wherein DC power conversion is performed.
  4. A second smoothing capacitor connected in series with the first smoothing capacitor; a semiconductor rectifying element provided for each unit power conversion circuit and connected between terminals of the second smoothing capacitor and capable of conducting in one direction; A series body of a resonance inductor and a resonance capacitor connected between an intermediate connection point of the series body of the semiconductor switching elements and an intermediate connection point of the series body of the semiconductor rectifier elements. , Between the terminals of the second smoothing capacitor and the series connection body terminals of the first and second smoothing capacitors by using charging / discharging of the resonance capacitor based on the on / off drive of the semiconductor switching element. The power converter according to claim 1, wherein DC power conversion is performed.
  5. A second body of the second and third smoothing capacitors connected in series with the first smoothing capacitor, and the second smoothing provided for each of the unit power conversion circuits of the plurality of unit power conversion circuits. A series body of semiconductor rectifier elements that are connected between capacitors and are conductive in one direction, and a terminal of the third smoothing capacitor provided for each of the remaining unit power conversion circuits of the plurality of unit power conversion circuits A serial body of semiconductor rectifier elements that are electrically connected in one direction and connected between an intermediate connection point of the serial bodies of the semiconductor switching elements and an intermediate connection point of the serial bodies of the semiconductor rectifier elements. The first smoothing capacitor using a charge / discharge of the resonance capacitor based on the on / off drive of the semiconductor switching element. Wherein the inter-terminal first, power converter according to claim 1, characterized in that for converting a DC voltage between the inter-series circuit of terminals of the second and third smoothing capacitor.
  6. By having a series body of first, second and third smoothing capacitors and a series body of semiconductor switching elements connected between the terminals of the first smoothing capacitor and driving the semiconductor switching elements on and off at a predetermined cycle A first power conversion circuit that performs power conversion and a serial body of semiconductor switching elements are connected between the terminals of the second smoothing capacitor, and power conversion is performed by driving the semiconductor switching elements on and off at a predetermined cycle. A second power conversion circuit, a series body of semiconductor rectifier elements connected between terminals of the third smoothing capacitor and capable of conducting in one direction, and an intermediate connection point between the series bodies of the semiconductor switching elements A semiconductor device comprising a series body of a resonance inductor and a resonance capacitor connected between an intermediate connection point of a series body of semiconductor rectifier elements, Utilizing charging / discharging of the resonance capacitor based on on / off driving of the switching element, between the serial connection body terminals of the first, second and third smoothing capacitors and between the terminals of the third smoothing capacitor In a power converter that converts DC voltage,
    Current detecting means for detecting a current flowing through the first smoothing capacitor and a current flowing through the second smoothing capacitor; and a cutoff for turning off the semiconductor switching element when an output of the current detecting means exceeds a predetermined set value. With means,
    The first power conversion circuit is composed of a plurality of first unit power conversion circuits connected in parallel to the first smoothing capacitor, and the second power conversion circuit is used as the second smoothing capacitor. A plurality of second unit power conversion circuits connected in parallel to each other, and the number of the first unit power conversion circuits connected in parallel to the third smoothing capacitor in the series body of the semiconductor rectifier elements. And the number of unit serial bodies of the sum of the number of the second unit power conversion circuits and
    The semiconductor switching elements in the second power conversion circuit are configured such that the timings in the cycle of driving the semiconductor switching elements in the first power conversion circuit in the first power conversion circuit are different from each other in the first power conversion circuit. The power conversion device according to claim 1, wherein the second unit power conversion circuits have different timings in the period for driving the elements on and off.
  7. The power conversion device according to claim 3, wherein the semiconductor rectifying element is configured by a diode.
  8. The power conversion device according to claim 3, wherein the semiconductor rectifier element is configured by a MOSFET.
  9. 2. The current detecting means is constituted by a shunt resistor connected in series to the smoothing capacitor, and detects a current flowing through the smoothing capacitor from an output between terminals of the shunt resistor. The power converter device of any one of thru | or 8.
  10. The current detection means includes a magnetic circuit that forms a magnetic field based on a current flowing through the smoothing capacitor and a Hall element provided in the magnetic field, and a current flowing through the smoothing capacitor from an output between terminals of the Hall element. The power converter according to any one of claims 1 to 8, wherein the power converter is detected.
JP2008230887A 2008-09-09 2008-09-09 Power conversion device Pending JP2010068585A (en)

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JP2008230887A JP2010068585A (en) 2008-09-09 2008-09-09 Power conversion device

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JP2013110855A (en) * 2011-11-21 2013-06-06 Sanden Corp Fault detection device for inverter circuit

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JP2005051927A (en) * 2003-07-29 2005-02-24 Matsushita Electric Ind Co Ltd Switching power unit
JP2006262619A (en) * 2005-03-17 2006-09-28 Mitsubishi Electric Corp Switched-capacitor type dc/dc converter device
JP2007060805A (en) * 2005-08-24 2007-03-08 Toshiba Tec Corp Power supply
WO2008032424A1 (en) * 2006-09-15 2008-03-20 Mitsubishi Electric Corporation Dc/dc power converter

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JPH0715948A (en) * 1993-06-24 1995-01-17 Toshiba Corp Overcurrent protective circuit based on voltage-driven self arc-extinguishing element
JP2005051927A (en) * 2003-07-29 2005-02-24 Matsushita Electric Ind Co Ltd Switching power unit
JP2006262619A (en) * 2005-03-17 2006-09-28 Mitsubishi Electric Corp Switched-capacitor type dc/dc converter device
JP2007060805A (en) * 2005-08-24 2007-03-08 Toshiba Tec Corp Power supply
WO2008032424A1 (en) * 2006-09-15 2008-03-20 Mitsubishi Electric Corporation Dc/dc power converter

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JP2013110855A (en) * 2011-11-21 2013-06-06 Sanden Corp Fault detection device for inverter circuit
US9360515B2 (en) 2011-11-21 2016-06-07 Sanden Corporation Fault detection device for inverter system

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