JP2009017662A - Power converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To allow use of a small and inexpensive capacitor by suppressing heat generation, since a carrier frequency used for PWM control is mixed into a DC power supply as a ripple component and this turns into loss component in a capacitor of a power supply part and causes heat generation in the capacitor, in a power converter that drives a load while controlling it with PWM. <P>SOLUTION: A power converter is provided with an observer 111 that acquires load control conditions in an inverter 104 so as to generate an AC command value, which becomes a waveform that cancels ripple component contained in phase voltage or phase current by prescribed procedures, on the basis of the load control conditions; and the power converter has a converter 102, provided with a controller that receives an AC command value being output of the observer so as to execute processings for making the AC command value superimpose on a step-up command value. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention has one or more converters and one or more inverters as a power supply unit, and the inverters drive a load connected to the output side, and these converters and inverters share a capacitor on the power supply side. It concerns on the connected power converter.

  A configuration in which a predetermined DC voltage is generated by a DC-DC converter (hereinafter abbreviated as a converter), a capacitor is connected between the DC voltage output unit and the ground potential, and power is supplied to the inverter from both ends of the capacitor. In the power conversion device that performs PWM (Pulse Width Modulation) control of a load such as an electric motor with this inverter, the ripple current component due to the carrier used for PWM or the signal used for driving the load is the above capacitor. The ripple component is consumed in the capacitor as a loss. The capacitor generates heat due to this loss. Therefore, it is necessary to use a high-performance and large-sized capacitor having a large ripple resistance so as to provide a sufficient margin for this heat generation in the circuit configuration. For this reason, the example of the power converter device which reduced the following ripple current is disclosed.

  That is, in the following “Patent Document 1”, the power conversion device is composed of two sets of power conversion devices for driving two loads, and each power conversion device drives a motor as a connected load. Each of the inverters is a three-phase bridge circuit. Each inverter is provided with two sets of triangular wave generators for PWM control of each load connected to these inverters. For the two sets of triangular wave generating means as a carrier generating device for operating each phase of these three-phase inverters, the phase difference of these triangular waves is switched so as to have a predetermined phase difference from each other. The ripple current component is canceled out. According to this configuration, there is an effect in suppressing the carrier frequency component of the ripple current, but there is a problem that there is no effect on the frequency of the phase voltage component of the motor drive other than the carrier frequency component.

Further, in the following “Patent Document 2”, a bridge-connected inverter circuit is connected in parallel to a circuit in which two diodes are connected in series or two anti-parallel circuits of a diode and a switching element are connected in series. In the frequency conversion circuit using the AC switch, the energy of the snubber circuit attached to the AC switch is regenerated to the load side or the power source side. In this configuration, the energy regeneration of the snubber circuit, that is, the ripple component of each phase current in the power converter is detected, and feedback control is performed to match the command value specifying the target output voltage of the power converter. This configuration is affected by the frequency characteristics of the feedback system, so that the control gain decreases as the frequency of the ripple current component increases, and as a result, there is a problem that the effect of reducing the capacitor current cannot be obtained sufficiently.
JP 2002-300800 A Japanese Patent Laid-Open No. 10-80147

The power converter includes a converter that performs DC-DC conversion and an inverter that generates a power waveform for driving a load. The ripple current component generated from the carrier frequency used in this converter and inverter is superimposed on the DC voltage of the power source, and this superimposed ripple current component flows into the capacitor interposed between the converter and the inverter as a capacitor current. Absorbed. This capacitor current becomes a loss in the capacitor and causes a rise in the temperature of the capacitor. In order to solve this problem, an expensive capacitor that can withstand this heat generation and that has low high-frequency loss is required. In addition, as a countermeasure, a method of setting a carrier wave for generating a PWM signal used in the power converter to a predetermined frequency and phase difference, or detecting a ripple component of the phase voltage or phase current of the power converter, A method of reducing a ripple current component by a method of feedback-controlling a detected component to a command value setting unit of a power converter has been disclosed. However, it cannot be said that a sufficient reduction effect is obtained by any of the methods.
Therefore, an object of the present invention is to provide a power conversion device that uses a small and inexpensive capacitor and is not limited by the frequency characteristics of the control loop in the power conversion device, and can provide a sufficiently large reduction effect. Is.

  In order to achieve the above object, in the present invention, parameters of a capacitor voltage, a phase voltage command value, a phase current command value, and a carrier frequency used in a controller of an inverter that drives a load are obtained and obtained. An AC command value that is a component that cancels the ripple component is generated using the parameters, and an observer that is sent to the converter is connected to the control unit of the inverter. The AC command value sent from the observer is sent to the controller in the converter, and this AC command value is superimposed on the boost command value in the converter to cancel the ripple component.

  In the power conversion device according to the present invention, the AC command value for suppressing the AC component, that is, the ripple component, is first superposed on the boost command value of the converter based on the control variable of the inverter. Furthermore, since this AC command value superimposition uses a feedforward by an observer, there is no limit on gain reduction due to the frequency characteristics of the feedback loop as in the past, and the control gain is made infinite for the entire frequency band. Therefore, the ripple current component including the harmonic component and the sideband component can be sufficiently reduced.

  FIG. 1 shows a configuration of a power converter according to the present invention. 1 includes a converter 102 that is a DC-DC converter as a first power converter, and a voltage-type three-phase PWM inverter 104 (hereinafter, referred to as a three-phase bridge as a second power converter). Abbreviated as an inverter), a DC power supply unit 101, an observer 111 for generating a waveform used for ripple current component suppression, and a smoothing capacitor connected in common to the DC power supply 101 of the converter 102 and the inverter 104 103 is connected.

  In FIG. 1, a DC power source 101 is, for example, a battery (storage battery) or a fuel cell, and this DC voltage is boosted by a converter 102 that is a DC-DC converter, and then drives a motor 105 by an inverter 104 via a capacitor 103.

  The converter 102 includes a reactor 106, a semiconductor device 107 including two IGBTs (Insulated Gate Bipolar Transistors), and a controller 108 on the converter side. The controller 108 turns on and off the power supply 101. Alternatively, conduction control of the semiconductor device 107 is performed based on a voltage state such as a power supply voltage, a voltage state of the capacitor 103, and an AC command value from an observer 111 described later.

  The inverter 104 includes a semiconductor device 109 using six IGBTs (two for each phase) and a controller 110 for the inverter 104. The inverter controller 110 performs conduction control of the semiconductor device 109 based on the voltage state of the capacitor 103 and the rotation speed, rotation angle, and torque of the electric motor 105, and sets control variables related to conduction control of the semiconductor device 109 to the observer 111. send. The observer 111 refers to a MAP as a preset reference table based on the control variable acquired from the controller 110 of the inverter 104, or performs arithmetic processing using a preset mathematical formula described later, A command for controlling the conduction of the semiconductor device 107 thus obtained is sent to the controller 108 on the 102 side. That is, an inverter control variable for suppressing a ripple current component caused by the inverter is acquired, and an AC command value for ripple current component suppression is generated in the observer 111 based on this control variable. This AC command value is sent from the observer 111 to the controller 108 of the converter 102, superimposed on the boost command value used by the converter 102, and the boost command value (final value of the converter 102 is finally set to a waveform in which the ripple current component is suppressed. Is generated). By comparing this boost command value with the carrier and the voltage comparator, the load driving power in which the ripple current component is suppressed is obtained.

  In the following, the operation of the power conversion device having the configuration shown in FIG. 1 will be described first with respect to the ripple current due to the converter 102 alone, then with respect to the ripple current due to the inverter, and then with the integrated ripple current components. The case will be described.

  First, general control of the converter will be described with reference to FIG. Here, the controller 108 on the converter 102 side in FIG. 1 has a boost command value generation unit 201, a carrier generation unit 202, a voltage comparator 208, a buffer circuit 209, and an inverting circuit 210 as shown in FIG. The outputs of the buffer circuit 209 and the inverting circuit 210 are connected to the control electrodes of the semiconductor elements (upper arm 204 and lower arm 205) constituting the semiconductor device 203 of the converter 102, respectively.

Converter 102 determines boost command value D in boost command value generation unit 201 based on power supply voltage V _in 206 and capacitor voltage V _c 207, and determines the boost command value D and the triangular wave generated in carrier generation unit 202. Comparison with the carrier is performed by the voltage comparator 208, and conduction control of the semiconductor device 203 of the converter is performed based on the comparison result. In this conduction control, when the upper arm 204 in the semiconductor device 203 is in the conduction state, the lower arm 205 is in the non-conduction state, and when the upper arm 204 is in the non-conduction state, the lower arm 205 is in the conduction state. It is controlled so as not to be in a conductive state.

FIG. 3 shows a state in which the boost command value D301 and the carrier 302 are compared by the voltage comparator 208 in FIG. 2 in the boost converter 102 shown in FIG. In FIG. 3, the horizontal axis is a time axis, and the vertical axis is a voltage axis indicating amplitude. The carrier 302 is a triangular wave having a predetermined frequency and an amplitude having a value from 0 to 1 on an arbitrary scale. Further, the boost command value D301 takes a value between 0 and 1 at an arbitrary scale according to the boost ratio, and the boost command value D301 and the value of the carrier 302 are compared by the voltage comparator 208 in FIG. Is greater than the carrier 302, the output of the voltage comparator 208 is logic "0", which causes the upper arm 204 to be non-conductive and the lower arm to be conductive. Therefore, when the boost command value D301 is 0, the output of the voltage comparator 208 is always logic “1”. Therefore, the power supply voltage V _in 206 and the capacitor voltage V _c 207 are the same voltage, and the boost command value D301 is The closer to 1, the closer the duty ratio α (α = T _ON / T) to 1, and the higher the capacitor voltage V _c 207. Here, T is the period of the carrier _{302, T ON} indicates time the upper arm 204 is in a conductive state.

Next, the operation of a general inverter will be described for any one of the three phases with reference to FIG. The inverter 104 determines the phase voltage command value V _u ^* by the phase voltage command value generation unit 401 based on the capacitor voltage V _c 406 and the phase current I _u 407 of the target phase, and this phase voltage command value The voltage comparator 408 compares V _u ^* and the triangular waveform carrier 402 generated by the carrier generation unit 402, and the conduction control of each switching element of the semiconductor device 403 is performed based on the comparison result. In the semiconductor device 403 as well, in the case of the converter of FIG. 2, when the upper arm 404 is conductive, the lower arm 405 is non-conductive, and when the upper arm 404 is non-conductive, the lower arm is conductive and both arms Are controlled so as not to become conductive at the same time.

FIG. 5 illustrates the relationship between the phase voltage command value V _u ^* 501 on the input side of the voltage comparator 408 and the triangular waveform carrier 502 in the inverter shown in FIG. 4 (104 in FIG. 1). Time is plotted on the vertical axis. The carrier 502 is a triangular wave that takes a value from 0 to the capacitor voltage V _c , and the phase voltage command value V _u ^* 501 changes according to the AC voltage on the output side connected to the motor 105, and its amplitude is the maximum. a waveform the voltage _{V c.} Then, the phase voltage command value V _u ^* 501 and the carrier 502 are compared. For example, when the phase voltage command value V _u ^* 501 is large, the upper arm 404 is in a conductive state and the lower arm is in a non-conductive state. Become.

  FIG. 6 shows the frequency on the horizontal axis and the effective frequency component of the capacitor current, that is, the ripple current on the vertical axis. The capacitor caused by the boost converter 102 when the converter 102 operates at a carrier frequency of 20 kHz. The frequency analysis result of the ripple current which flows into 103 is shown. The ripple current flowing through the capacitor 103 is composed of a component 601 obtained by multiplying the carrier frequency of the converter 102.

  FIG. 7 shows the ripple caused by the inverter when the frequency is plotted on the horizontal axis and the effective value of the frequency component of the ripple current which is the capacitor current is plotted on the vertical axis, and the inverter 104 is operating at a carrier frequency of 10 kHz. The frequency analysis result of an electric current is shown. The ripple current flowing through the capacitor 103 is composed of a carrier frequency × 2 multiplication component 701 of the inverter 104 and upper and lower sidebands 702 and 703 having a phase voltage frequency × 3 multiplication frequency centered on the carrier frequency multiplication frequency. ing.

  By using the same method as in Patent Document 1, the ripple current can be reduced by canceling out the carrier frequency component 601 of the converter 102 and the carrier frequency × 2 component 701 of the inverter 104. Since there is no effect of canceling frequency components other than the frequency, the ripple current reduction effect is insufficient.

  Therefore, in the present invention, the AC voltage component corresponding to the ripple current is superimposed on the boost command value of the converter 102, so that the sideband wave of the phase voltage frequency × 3 centered on the carrier frequency of the inverter 104 and the multiplied frequency thereof is used. A method of canceling out components (for example, 702 and 703 in FIG. 7) is disclosed.

Here, the frequency, amplitude, and phase of the AC component are determined using control variables of a power converter other than the converter. For this purpose, the observer 111 acquires the following parameters as control variables from the inverter.
・ Capacitor voltage V _c
-Phase voltage command value ^{_{V u * Vcos (2πf o t}} + θ v)
-Phase current command value I _u ^* I cos (2πf _o t + θ _i )
Carrier frequency f _c
However, the phase current command value I _u ^* indicates a case where the phase voltage command value V _u ^* is expressed by the above formula. Each variable in the above equation is as follows.

f _o : phase voltage, phase current frequency (Hz)
f _c : carrier frequency (Hz)
V: Phase voltage amplitude (V)
I: Phase current amplitude (A)
θ _v : phase voltage phase angle (rad)
θ _i : Phase current phase angle (rad)
t: Time (sec)
Subsequently, when canceling the lower sideband 702, the observer 111 in FIG. 1 performs the calculation shown in the following (Equation 3) based on the inverter control variable, and the frequency Freq, amplitude Amp, and phase of the lower sideband 702 are calculated. Estimate Phase. Here, the constant 0.992 in the amplitude estimation formula is a value obtained experimentally, and may be any value that is substantially close to this value.

その後、下記（数４）式の演算を行うことによりコンバータの昇圧指令値に重畳する下側帯波成分を打ち消すための交流指令値Ｄ₋を算出する。

Thereafter, an AC command value D ₋ for canceling the lower sideband component superimposed on the boost command value of the converter is calculated by performing the calculation of the following (Equation 4).

ここで、Ｉ_ｄｃは昇圧コンバータと三相インバータの間を流れる直流電流であり、Ｉ_ｄｃは上記インバータ制御変数を元に下記（数５）式の演算により算出される。次に

Here, I _dc is a direct current flowing between the boost converter and the three-phase inverter, and I _dc is calculated by the following equation (5) based on the inverter control variable. next

上記（数５）式を用いて（数４）式を書き直して下記（数６）式を得る。つまり、下側帯波７０２を打ち消したい場合には、オブザーバ１１１にて上記インバータ制御変数を元に下記の演算を行い、昇圧コンバータの昇圧指令値に（数６）式で示す交流指令値Ｄ₋を重畳することになる。

Using the above equation (5), the equation (4) is rewritten to obtain the following equation (6). That is, when canceling the lower sideband 702, the observer 111 performs the following calculation based on the inverter control variable, and sets the boost command value of the boost converter to the AC command value D ₋ expressed by the equation (6). It will be superimposed.

これにより、昇圧コンバータ起因のコンデンサ電流の周波数成分が図９に示すようになり、この側帯波成分９０１により相電圧周波数×３の側帯波成分の図７における７０２で示される周波数成分を打ち消すことができる。図８は横軸に時間をとり，昇圧指令値Ｄに交流指令値Ｄ₋を重畳した昇圧コンバータの指令値（Ｄ＋Ｄ₋）とキャリアの比較を示したもので、図９は横軸に周波数、縦軸に電流値（実効値）をとり、昇圧コンバータに起因するコンデンサ電流の周波数分析結果を示したものである。コンデンサの電流は本来の昇圧コンバータのキャリアの周波数の逓倍の他に、０倍を含むキャリア周波数の逓倍を中心としたＤ₋の周波数の逓倍の側帯波の成分により構成される。なお、上記（数６）式は請求項２における（数１）式と同じである。

As a result, the frequency component of the capacitor current caused by the boost converter becomes as shown in FIG. 9, and the sideband component 901 can cancel the frequency component indicated by 702 in FIG. 7 of the sideband component of the phase voltage frequency × 3. it can. 8 the horizontal axis represents time, the AC command value D to boost command value D _- instruction value of the boost converter by superimposing the (D + D _-) and shows a comparison of the carrier, Figure 9 is a frequency on the horizontal axis, The current value (effective value) is taken on the vertical axis, and the frequency analysis result of the capacitor current caused by the boost converter is shown. Current of the capacitor to other multiple of the frequency of the carrier of the original boost converter, multiplying D around the carrier frequency including 0 times _- composed of component sidebands of the multiplication of the frequency. In addition, the said (Formula 6) Formula is the same as the (Formula 1) Formula in Claim 2.

Further, when canceling the other upper sideband component 703, the AC command value D ₊ expressed by the following (Equation 7) is superimposed on the following AC command value in the boost command value of the boost converter in the same manner as described above.

このようにして、コンバータ起因のコンデンサ電流、すなわちリプル電流の周波数成分は図１１のようになり、周波数（１１０１）と（１１０２）とにより図７における側帯波周波数成分（７０２）と（７０３）とを打ち消すことができる。当然であるが、上側帯波（７０３）のみを打ち消したい場合には交流指令値Ｄ_＋のみを昇圧指令値に重畳すればよい。上記（数７）式は請求項２における（数２）式と同じである。

Thus, the frequency component of the capacitor current caused by the converter, that is, the ripple current is as shown in FIG. 11, and the sideband frequency components (702) and (703) in FIG. 7 are obtained from the frequencies (1101) and (1102). Can be countered. Of course, when only the upper side band (703) is to be canceled, only the AC command value D ₊ may be superimposed on the boost command value. The above equation (7) is the same as the equation (2) in claim 2.

Figure 10 is a horizontal axis is a time axis, the AC command value D corresponding to the boost command value D vertical side bands _- and D ₊ and total command value obtained by adding the (D + D - _{+ D +)} and shows the relationship between the carrier wave ing. FIG. 11 shows the frequency analysis result of the capacitor current in the converter 102. From this analysis result, the AC command value corresponding to the ripple current component flowing in the capacitor is the lower side band centering on the multiplication of the carrier frequency including 0 times (DC component) in addition to the multiplication of the original carrier frequency of the converter 102. AC command value D corresponding to the waves _- of the sideband wave component of the multiplied frequency, 0-fold multiplication of the AC command value D ₊ having a frequency corresponding to the side bands around the multiplication of the carrier frequency that contains the (DC component) It can be seen that this is a configuration including the sideband component.

As described above, as shown in FIG. 1, the present invention has been described by taking as an example a case where a power conversion device using an IGBT and an observer is configured. For example, a configuration using a power MOSFET instead of an IGBT, or a method for calculating by an arithmetic process using the observer 111 in this embodiment has been disclosed, but the relationship between an inverter control variable acquired in advance and an AC command value including a DC component is disclosed. It is also possible to use a method in which the observer 111 uses the MAP to obtain the AC command values D ₋ and D ₊ and cancels the ripple current component using the MAP.

The circuit block diagram which shows the structure of the power converter device by this invention. The circuit diagram of the converter for demonstrating the general operation | movement of converter control. The voltage level comparison figure in the voltage comparator input terminal in a converter. The circuit diagram regarding the inverter part one phase for general operation | movement description of an inverter part. These waveform diagrams for comparing the inverter part carrier and the phase voltage waveform per phase. Frequency analysis diagram of capacitor current caused by converter. The frequency analysis figure of the capacitor current resulting from an inverter part. The voltage level comparison figure which shows the command value of the converter which considered the lower sideband component, and the voltage relationship with a carrier. FIG. 9 is a frequency analysis diagram of the voltage level comparison waveform shown in FIG. 8. The voltage level comparison figure which shows the command value of the converter which considered the upper and lower sideband component, and the voltage relationship with a carrier. FIG. 11 is a frequency analysis diagram of the voltage level comparison waveform shown in FIG. 10.

Explanation of symbols

DESCRIPTION OF SYMBOLS 101: Power supply 102: Converter 103: Capacitor 104: Inverter 105: Electric motor 106: Reactor 107: Converter side semiconductor device 108: Converter controller 109: Inverter side semiconductor device 110: Inverter controller 111: Observer 201: Boost command value Generation unit 202: Carrier generation unit 203: Semiconductor device 204: Upper arm 205: Lower arm 206: Power supply 207: Capacitor voltage
208: Voltage comparator 401: Phase voltage command value generation unit 402: Carrier generation unit 403: Semiconductor device 404: Upper arm 405: Lower arm 406: Capacitor voltage 407: Phase current 408: Voltage comparator

Claims (3)

The power converter includes a plurality of power converters including at least one DC-DC converter and an inverter that drives a load, and the plurality of power converters are connected with a power supply smoothing capacitor in common. In the power conversion device that controls the drive by PWM,
A waveform that suppresses an AC component flowing into the smoothing capacitor that is generated when the PWM control is performed is generated as an AC command value by an observer, and the AC command value is superimposed on a boost command value of the converter. Power conversion device. The power conversion device according to claim 1,
The frequency, amplitude, and phase of the AC component are determined by the following procedure using control variables of a power converter other than the converter,
The control variable is the capacitor voltage V _c
Phase voltage command value V _u ^* = V cos (2πf _o t + θ _v )
Phase current command value I _u ^* = I cos (2πf _o t + θ _i )
The carrier frequency _{f c}
And, also, D _- and D ₊ AC command value for canceling the sideband component and the upper sideband component under each, theta _v, theta _i is the phase component of the phase voltage command value and the phase current command value, the f _o As the frequency of the phase voltage and phase current, the AC command value D _{− in the} case of canceling the lower sideband component is expressed by the equation (1).

In addition, the AC command value D ₊ when canceling the upper side band is expressed by the following equation (2):

A power conversion device, wherein the AC command value to be superimposed on the boost command value is set by performing arithmetic processing in each observer. The power conversion device according to claim 1,
A power conversion device, wherein a relationship between the AC command value to be superimposed and the control variable is determined with reference to a MAP set in an observer in advance as a reference table.

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