JP2006345682A - Annular winding motor - Google Patents

Annular winding motor Download PDF

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JP2006345682A
JP2006345682A JP2005171650A JP2005171650A JP2006345682A JP 2006345682 A JP2006345682 A JP 2006345682A JP 2005171650 A JP2005171650 A JP 2005171650A JP 2005171650 A JP2005171650 A JP 2005171650A JP 2006345682 A JP2006345682 A JP 2006345682A
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winding
winding arrangement
arrangement
slots
annular
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JP4657820B2 (en
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Toshiyuki Baba
Takanori Komatsu
Toshinori Tanaka
孝教 小松
敏則 田中
俊行 馬場
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Mitsubishi Electric Corp
三菱電機株式会社
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Abstract

PROBLEM TO BE SOLVED: To obtain an annular winding motor having a small torque ripple by improving a winding coefficient.
SOLUTION: A winding 14 wound in a toroidal shape on a yoke portion 9 of a stator core 8 is arranged in a predetermined order in accordance with a phase order of a multiphase alternating current applied to the winding 14, and a slot 11 of the stator core 8 is arranged. , 13 are inserted in turn from one of the remaining slots 11 and 13 in the same order as the winding group 14 of the winding arrangement A inserted every other in turn. The winding arrangement A is composed of the winding group 14 of the winding arrangement B having a phase difference, and the multi-phase alternating current is applied to the winding group 14 of the winding arrangement A and the winding arrangement B. The synthetic magnetomotive force distribution was made to approximate a sine wave.
[Selection] Figure 10

Description

  According to the present invention, a rotor with a magnetic pole attached is opposed to the teeth of the stator core provided with teeth projecting on the outer side and the inner side of the annular yoke, respectively, and the winding wound around the yoke part in a toroidal shape is provided. The present invention relates to an annular winding motor that is energized by a multiphase alternating current to rotate the rotor.

There are concentrated windings and distributed windings, full-pitch windings and short-pitch windings in the winding system of the motor. When the windings are distributed in the peripheral direction of the armature, an electrical phase difference is generated. In general, a winding that distributes windings so as to function similarly to a winding method called distributed winding is defined as distributed winding here, and the winding coefficient at this time is called distributed winding coefficient.
A case where the number of slots per phase per pole q is 1 is defined as concentrated winding, and a case where q is 1 or more is defined as distributed winding. Accordingly, the distributed winding is realized by a two-layer winding, and the distributed winding coefficient is a value determined by the number of magnetic poles and the number of slots.
On the other hand, the short-pitch winding coefficient is a value determined by the winding pitch and the magnetic pole pitch. A value obtained by multiplying the distributed winding coefficient and the short-pitch winding coefficient is called a winding coefficient.

In recent years, there is a winding method in which winding is concentrated around one tooth and called concentrated winding. Actually, the number of slots per phase per pole q is 1/4 to 1/2. In order to distinguish the winding method called concentrated winding in this way from general concentrated winding, it is hereinafter referred to as magnetic pole concentrated winding. For example, in the case of a three-phase motor with 14 poles and 12 slots, the number of slots per pole per phase q is q = 12 / (14 × 3) = 2/7. Even in concentrated magnetic pole winding, the number of slots per phase q is a fractional slot smaller than 1, and distributed winding coefficients are obtained by distributing windings, and short-pitch winding coefficients are obtained by phase differences of winding sides. It is done. Therefore, the winding coefficient of the magnetic pole concentrated winding is winding coefficient = (distributed winding coefficient × short-pitch winding coefficient).
Even in the case of the annular winding, the windings can be distributed by setting the number of slots per phase q to a fractional slot as in the case of concentrated magnetic winding, and a distributed winding coefficient can be obtained.

  A conventional annular winding motor is shown in FIGS. FIG. 38 shows an annular winding motor similar to the motor described in Patent Document 1, and exemplifies a 4-pole 12-slot. FIG. 39 is a cross-sectional view taken along the line AA in FIG. In both figures, a stator core 8 is fixed to the bracket 1. The stator core 8 includes an annular yoke 9, an outer tooth 10 projecting from the outer periphery of the yoke 9, and an inner tooth 12 projecting from the inner periphery of the yoke 9 corresponding to the outer tooth 10. ing. Outer slots 11 are formed between the outer teeth 10, and inner slots 13 are formed between the inner teeth 12 corresponding to the outer slots 11. A winding 14 inserted in the outer slot 11 and the inner slot 13 is wound around the yoke 9 in a toroidal shape, and a three-phase alternating current is supplied.

A rotating shaft 2 concentric with the yoke 9 is rotatably supported on the bracket 1 via a bearing 3. An outer rotor 4 including a stator core 8 is fixed to the rotating shaft 2. An outer magnetic pole 5 facing the outer teeth 10 is attached to the inner surface of the outer rotor 4. Further, the inner rotor 6 included in the stator core 8 is also fixed to the rotating shaft 2. An inner magnetic pole 7 facing the inner teeth 12 is attached to the outer surface of the inner rotor 6.
By supplying three-phase alternating current to the winding 14, torque is generated in the outer rotor 4 by the current flowing in the outer winding side 14, and torque is generated in the inner rotor 6 by the current flowing in the inner winding side 14. . For this reason, a large torque can be obtained with the same current.

JP 2001-37133 A (paragraph number 27, FIG. 10)

The conventional annular winding motor is configured as described above, and it is difficult to perform short-pitch winding with the annular winding. Therefore, a winding arrangement of integer slots of concentrated winding or distributed winding is provided. Since short-pitch winding cannot be applied to integer slots of concentrated winding or distributed winding, there is a problem that the winding coefficient of higher-order harmonics is increased and torque ripple due to induced voltage harmonics is increased.
That is, in the above-described conventional example, there are 4 poles and 12 slots, and the distributed winding coefficient Kd is calculated by the following equation (1). The results are shown in FIG. The winding coefficients Kd of the fundamental wave (n = 1) and higher harmonics (n = 5, 7, 11, 13) are all Kd = 1, and a large torque ripple is generated.

  The present invention has been made to solve the above problems, and an object of the present invention is to obtain an annular winding motor having a small torque ripple by improving the winding coefficient.

  An annular winding motor according to the present invention includes a stator core slot in which windings wound in a toroidal shape on a yoke portion of a stator core are arranged in a predetermined order according to the phase order of the multiphase alternating current applied to the winding. The winding group of winding arrangement A inserted every other in turn, and arranged in the same order as the order of this winding arrangement A and inserted in order from any of the remaining slots, winding arrangement A and Is composed of the winding group of the winding arrangement B having a phase difference, and the resultant magnetomotive force distribution is approximated to a sine wave when a multiphase alternating current is applied to the winding group of the winding arrangement A and the winding arrangement B. It is what I did.

  In the annular winding motor according to the present invention, the winding group of the winding arrangement A and the winding group of the winding arrangement B having a phase difference with respect to the winding arrangement A are arranged in a toroidal shape on the yoke portion of the stator core. The composite magnetomotive force distribution can be brought close to a sine wave because the coil is wound around and energized with multiphase alternating current. For this reason, there exists an effect that the torque ripple of an annular winding motor can be reduced.

  Embodiments of the present invention will be described below with reference to the drawings. In addition, in each figure, the same code | symbol is attached | subjected to the part which is the same or it corresponds, and duplication of description was omitted.

Embodiment 1 FIG.
First, the winding coefficient of the electric motor will be described.
A general distributed winding coefficient Kd will be described with reference to FIGS. In the figure, in the case of distributed winding, since the winding 14 is inserted into a plurality of different slots, the induced voltages e1, e2, e3 of the winding sides 14-1, 14-2, 14-3 correspond to the slot pitch. The phase difference α is generated. For this reason, the combined induced voltage er ′ for each pole and phase of the distributed winding is a vector sum of the induced voltages e1, e2, and e3 of each winding side. The ratio of the combined induction voltage er for each pole and each phase in the case of concentrated winding and the combined induction voltage er ′ of the distributed winding is the distributed winding coefficient Kd.

That is, if the number of phases is m, the number of magnetic poles is P, the number of armature slots is Q, the number of slots is q for each pole, and the harmonic order is n.
Number of slots per phase per pole q = Q / (mP)
Phase difference α = πP / Q = π / (mq)
Distributed winding coefficient Kd = sin {nπ / (2m)} / [q · sin {nπ / (2mq)}] for n-order harmonics ----------- (1)
However, in the case of fractional slots, the numerator value of an improper fraction of the number of slots per phase per phase is used as the number of slots q per phase per pole.
That is, when the windings 14 are distributed, a phase difference α is generated between the induced voltages e1, e2, and e3. Therefore, the combined induced voltage er ′ has a distributed winding coefficient Kd expressed by the equation (1). Applies.

Next, the distributed winding coefficient Kd of the annular winding will be described with reference to FIGS. The distributed winding coefficient Kd of the winding 14 shown in the equation (1) was calculated assuming that the winding 14 is distributed in the circumferential direction of the armature. In the annular winding, as shown in FIG. 4, the winding sides 14 are spaced apart in the radial direction, which is different from the arrangement of the windings 14 shown in FIG. 1. However, for example, if only the inner winding side 14 is considered, the same idea as in FIG. 1 can be applied.
That is, FIG. 4 shows the arrangement of the annular winding 14. When there is a phase difference αt in the in-phase winding side 14 among the inner side winding sides 14, the combined induced voltage er ′ is as shown in FIG. 5. Accordingly, the distributed winding coefficient Kd of the annular winding is also obtained from the relationship between the number of magnetic poles P and the number of slots Q, and is similarly expressed by the above equation (1).
Incidentally, since the conventional annular winding motor shown in FIG. 38 has the number of magnetic poles P = 4, the number of armature slots Q = 12, and the number of phases m = 3, the number of slots per phase per pole q = 1. Therefore, as shown in FIG. 40, all the harmonic orders n are full-pitch winding with distributed winding coefficient Kd = 1.

Next, the winding coefficient Kw of the annular winding motor will be described. FIG. 6A is a development view of an annular winding motor constituted by magnetic poles 5 and 7 made of permanent magnets having the number of magnetic poles P and an armature 8 having a number Q of slots. As shown in FIG. 39, the armature 8 is fixed to the frame, and hence the armature 8 is also referred to as a stator core 8 hereinafter.
FIG. 6B shows a winding in which the stator core 8A in which the slots 11 and 13 corresponding to the odd numbers in FIG. FIG. 6 is a development view of the annular winding motor of arrangement A. Therefore, if the number of slots is Q ′, Q ′ = Q / 2. Assuming that the number of phases is m (m = 3 in the figure), the number of slots per phase per pole q ′ = Q ′ / (mP). The distributed winding coefficient Kd of the annular winding motor of FIG. 6B is expressed by equation (1).
Further, FIG. 6C shows an annular shape of the winding arrangement A in which the stator core 8B in which slots corresponding to the even numbers in FIG. It is an expanded view of a winding motor. Also in this case, the number of slots Q ′ = Q / 2. Assuming that the number of phases is m (m = 3 in the figure), the number of slots per phase per pole q ′ = Q ′ / (mP). The distributed winding coefficient Kd of the annular winding motor shown in FIG. 6C can also be expressed by equation (1).

Incidentally, the winding 14 accommodated in the stator core 8B of FIG. 6C has a phase difference of an electrical angle (πP / Q) with respect to the winding 14 of the stator core 8A of FIG. 6B. This phase difference corresponds to one slot. Therefore, the annular winding motor of FIG. 6A is obtained by superimposing the winding arrangement A of FIG. 6B and the winding arrangement B of FIG. 6C.
Therefore, the induced voltages generated in the windings 14 for the # 1, # 2, # 3, and # 4 slots are e1, e2, e3, and e4. A phase difference α = (mP / Q ′) is generated between the induced voltages e1 and e3 of the windings 14 in the # 1 slot and the # 3 slot of the stator core 8A. A phase difference α = (mP / Q ′) also occurs between the induced voltages e2 and e4 of the windings 14 in the # 2 slot and the # 4 slot of the stator core 8B. Further, a phase difference α ′ = (mP / Q) = α / 2 occurs between the induced voltages e1 and e2 of the windings 14 in the # 1 slot and the # 2 slot.

FIG. 7 is a vector diagram showing the induced voltages e1, e2, e3, e4.
Here, a coefficient multiplied by the distributed winding coefficient Kd by the phase difference α ′ is defined as a coupling coefficient Kc. The coupling coefficient Kc of the nth order component where the fundamental wave is n = 1 is
Kc = cos {nπP / (2Q)} -------------- (2)
Therefore, the winding coefficient Kw is expressed by the following expression (3) obtained by multiplying the distributed winding coefficient Kd of expression (1) by the coupling coefficient Kc of expression (2).
Kw = Kd · Kc = [sin {nπ / (2m)} / [q ′ · sin {nπ / (2mq ′)}]] cos {nπP / (2Q)} ---------- -------- (3)
However, (nπP / Q) is in the range of 0 to 2π.
By selecting the number of magnetic poles P and the number of slots Q so that the higher-order component winding coefficient Kw is reduced from the above equation (3), torque ripple can be suppressed.

FIG. 8 shows a winding in which a winding 14 having a phase difference of an electrical angle {(πP / Q) · Qn} with respect to the winding 14 of the stator core 8A is housed in a stator core 8B in which even-numbered slots are formed. FIG. 6 is a development view of the annular winding motor in arrangement B.
That is, the number of slots Qn that can be shifted in the winding 14 of the stator core 8B with respect to the winding 14 of the stator core 8A is an odd number of 1, 3, 5 to (2Q'-1). The phase difference regarding the coupling coefficient Kc at this time is (πP / Q) × Qn, that is, (πP / Q) × 1, (πP / Q) × 3, (πP / Q) × 5 to (πP / Q). X (2Q'-1).

Therefore, the coupling coefficient Kc when the winding 14 of the stator core 8A is shifted by the number of slots Qn is expressed by the following equation (4). Further, the winding coefficient Kw is expressed by equation (5). However, {(πP / Q) × Qn} is 0 to 2π.
Kc = cos [{nπP / (2Q)} × Qn] -------------- (4)
Kw = Kd · Kc = [sin {nπ / (2m)} / [q ′ · sin {nπ / (2mq ′)}]] cos [{nπP / (2Q)} × Qn] ------ ------------ (5)
That is, by selecting a winding arrangement in which the winding 14 is appropriately shifted with respect to the combination of the number of magnetic poles P and the number of slots Q, the winding coefficient Kw can be improved, and the magnetomotive force distribution can be further increased. The sine wave can be approximated and concentration of the reaction magnetic flux can be avoided. Therefore, it is effective for improving the torque / current ratio, and torque ripple can be reduced.

FIG. 9 is a development view of the annular winding motor of the winding arrangement B in which the winding 14 whose phase is reversed with respect to the winding 14 of FIG. 8 is housed in the stator core 8B.
That is, the winding 14 of the stator core 8B has an odd number of slots Qn that can be shifted with respect to the winding 14 of the stator core 8A, and is 1, 3, 5 to (2Q'-1). . Further, in order to invert the phase of the winding 14, the phase is shifted by π with respect to the phase between the slots. The phase difference regarding the coupling coefficient Kc at this time is {(πP / Q) × Qn−π} in consideration of inverting the phase of the winding 14. That is, {(πP / Q) × 1-π}, {(πP / Q) × 3-π}, {(πP / Q) × 5-π} to {(πP / Q) × (2Q′−1) ) −π}.

Accordingly, the coupling coefficient Kc when the winding 14 of the stator core 8A is shifted by the number of slots Qn and the phase of the winding 14 is further inverted is expressed by the following equation (6). Further, the winding coefficient Kw is expressed by equation (7). However, {(πP / Q) × Qn−π} is 0 to 2π.
Kc = cos [n {πP / (2Q) × Qn−π}] ---------------- (6)
Kw = Kd · Kc = [sin {nπ / (2m)} / [q ′ · sin {nπ / (2mq ′)}]] · cos n [{πP / (2Q)} × Qn−π] --- --------------- (7)
That is, for the combination of the number of magnetic poles P and the number of slots Q, the winding coefficient Kw can be similarly improved by selecting an appropriate winding arrangement in consideration of the shift of the winding 14 and the phase inversion. Yes, the magnetomotive force distribution can be made closer to a sine wave, and the concentration of the reaction magnetic flux can be avoided. Therefore, it is effective for improving the torque / current ratio, and torque ripple can be reduced.

In the above description, the odd numbered slot and the even numbered slot are divided into two. However, the present invention is not limited to this.
For a motor having an annular winding, with respect to the configuration of the number of magnetic poles P and the number of slots Q, the winding arrangement by the fractional slots of the distributed winding in the number of magnetic poles P and the number of slots Q ′ = (Q / 4), respectively (πP / Q) Three winding arrangements with fractional slots of distributed winding in the number of magnetic poles P having a phase difference and the number of slots Q ′ = (Q / 4) are arranged for an annular winding motor having the number of magnetic poles P and the number of slots Q. Following the above, it is possible to improve the winding coefficient Kw by applying four windings 14. In addition, each mathematical expression relating to the winding coefficient Kw can also be applied to the following embodiments. The winding coefficient Kw in the annular winding motor has been described above.

Next, the annular winding motor according to the first embodiment of the present invention will be described with reference to FIGS.
FIG. 10 is a cross-sectional view of an annular winding motor provided on the outer side and the inner side with magnetic cores 5 and 7 having a magnetic pole number P = 8 and a stator core 8 constituting an armature having a slot number Q = 18. The stator core 8 is fixed to the bracket. The stator core 8 includes an annular yoke 9, an outer tooth 10 projecting from the outer periphery of the yoke 9, and an inner tooth 12 projecting from the inner periphery of the yoke 9 corresponding to the outer tooth 10. ing. Between the outer teeth 10, outer slots 11 having a slot number Q = 18 from # 1 to # 18 are formed. Inner slots 13 are also formed between the inner teeth 12 so as to correspond to the outer slots 11. A winding 14 inserted in the outer slot 11 and the inner slot 13 is wound around the yoke 9 in a toroidal shape, and a three-phase alternating current is supplied.

Further, an outer rotor 4 including a stator core 8 is fixed to the rotary shaft 2 concentric with the yoke 9. An outer magnetic pole 5 facing the outer teeth 10 is attached to the inner surface of the outer rotor 4. Further, the inner rotor 6 included in the stator core 8 is also fixed to the rotary shaft 2. An inner magnetic pole 7 facing the inner teeth 12 is attached to the outer surface of the inner rotor 6.
By supplying three-phase alternating current to the winding 14, torque is generated in the outer rotor 4 by the current flowing in the outer winding side 14, and torque is generated in the inner rotor 6 by the current flowing in the inner winding side 14. .

FIG. 11 is a cross-sectional view of an annular winding motor including the magnetic poles 5 and 7 having the number of magnetic poles P = 8 and the stator core 18 having the number of slots Q ′ = 9. Outer slots 11 and inner slots 13 corresponding to odd numbers # 1, # 3, # 5 to # 17 are formed in the stator core 18 with the same reference position as in FIG. A winding 14 is inserted into each of the slots 11 and 13 and is wound around the yoke 9 in a toroidal shape, which corresponds to the winding arrangement A. The winding 14 is supplied with a three-phase alternating current composed of a U phase, a V phase, and a W phase.
The distributed winding coefficient Kd of the annular winding motor of FIG. 11 can be obtained by the equation (1). The calculation result is shown in FIG.

  FIG. 12 is a cross-sectional view of an annular winding motor including the magnetic poles 5 and 7 having the number of magnetic poles P = 8 and the stator core 28 having the number of slots Q ′ = 9. Outer teeth 30 and inner teeth 32 protrude from the stator core 28, and outer slots 11 and inner slots 13 corresponding to even numbers # 2, # 4, and # 6 to # 18 are formed. Therefore, the slots 11 and 13 of # 2, # 4, and # 6 to # 18 are (πP / Q) = π × 8 with respect to the slots 11 and 13 of # 1, # 3, and # 5 to # 17. The phase is delayed by / 18 = 4π / 9. In addition, the windings 14 are accommodated in the slots 11 and 13 of # 2, # 4, and # 6 to # 18 in accordance with the winding arrangement shown in FIG. This distributed winding coefficient Kd is also the same as in the case of FIG. 11 and has the value shown in FIG.

The annular winding motor of FIG. 10 has the number of magnetic poles P = 8 and the number of slots Q = 18, and is obtained by superimposing the winding arrangement A of FIG. 11 and the winding arrangement B of FIG. When the winding arrangement A and the winding arrangement B are overlapped, the reference position of the winding arrangement B in FIG. 12 is arranged at # 12, so that the phase difference of the induced voltage is {180 × (P / Q) × Qn −180} = (180 × 8 × 11/18) −180 = −20 °. The coupling coefficient Kc due to this phase difference can be obtained by equation (6). The calculation result is shown in FIG.
FIG. 14 shows the coupling coefficient Kc when the winding arrangement A of FIG. 11 and the winding arrangement B of FIG. 12 are overlapped, which corresponds to the short-winding coefficient of general distributed winding or magnetic pole concentrated winding. It is.
Therefore, the winding coefficient Kw of the annular winding motor of FIG. 10 is obtained by the distributed winding coefficient Kd obtained by the winding arrangement A of FIG. 11 and the winding arrangement B of FIG. 12 having a phase difference with respect to the reference position. Multiplication with the coupling coefficient Kc. For example, the fifth-order winding coefficient Kw is Kw = 0.218 × 0.643 = 0.140. The winding coefficients Kw of other orders are shown in FIG.

  According to the first embodiment, the winding 14 of the winding arrangement A and the winding 14 of the winding arrangement B having a phase difference with respect to the winding arrangement A are arranged in a toroidal shape on the yoke 9 of the stator core 8. The winding coefficient Kw of the harmonic component is reduced because the coil is wound around and energized with multiphase alternating current. That is, the resultant magnetomotive force distribution by the windings 14 of the winding arrangement A and the winding arrangement B can be made close to a sine wave. For this reason, the torque ripple of an annular winding motor can be reduced.

Embodiment 2. FIG.
The second embodiment describes an annular winding motor having the number of magnetic poles P = 10 and the number of slots Q of the stator core 8 = 36 (# 1 to # 36). Based on the same concept as in the first embodiment, the winding arrangement A (# 1, # 3 to # 35) and the winding arrangement B (# 2, # 4 to ##) with the number of magnetic poles P = 10 and the number of slots Q ′ = 18. 36). Winding arrangement B is an arrangement in which the reference position is arranged at # 4 and the phase of each winding is inverted by 180 °. The winding arrangement A and the winding arrangement B are overlapped to obtain an annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 36.
FIG. 16 shows the distributed winding coefficient Kd, the coupling coefficient Kc, and the winding coefficient Kw of the annular winding motor. Here, in FIG. 16, the fifth- and seventh-order coupling coefficients Kc are 0.259 and −0.259, respectively, and the winding coefficient Kw can be improved as compared with the conventional one.

Next, the magnetomotive force distribution of the winding will be described. As a reference for comparison, the winding arrangement used for evaluation is a distributed winding and an integer number of magnetic poles P = 10 and the number of slots Q = 30. Therefore, the number of slots per phase per pole q = 1.
First, how to draw the magnetomotive force distribution will be described. The number of magnetic poles P = 10 and the number of slots Q = 30 are considered to be a continuation of the number of magnetic poles P = 2 and the number of slots Q = 6. The winding configuration is shown in FIG.
FIG. 18 shows the current of each phase of the three-phase alternating current. The U phase has a peak value (in this case, 1 A) when the electrical angle phase is 90 °, and the magnetomotive force distribution at that time is considered. Both the V-phase current and the W-phase current having an electrical angle phase of 90 ° are −0.5 A. When the winding 14 entering one slot is defined as one turn, the magnetomotive force of one winding 14 is represented by a current value of 1 because the number of turns is 1.
FIG. 19 shows a magnetomotive force distribution generated by each phase current of FIG. Since there are two gaps in one closed magnetic circuit including the magnetomotive force in one winding 14, the amplitude of the magnetomotive force is ½ each. By synthesizing the magnetomotive force distribution of each phase shown in FIG. 19, a three-phase magnetomotive force distribution is obtained, and the resultant magnetomotive force distribution is shown in FIG. As is clear from FIG. 20, the peak value of the synthetic magnetomotive force distribution = 1. Further, since the number of slots per pole per phase q = 1, the winding coefficient Kw becomes the same value as shown in FIG. 40, and the amplitude value of each order is also 1.

Next, the magnetomotive force distribution of the annular winding having the number of magnetic poles P = 10 and the number of slots Q = 36 will be described.
Since the currents in the U, V, and W phases have the same values and the same torque is generated, the number of magnetic poles P in the comparative reference model is the same as described above. On the other hand, since the slot number Q is different from that of the comparison reference model, the current value is multiplied by (30/36) with respect to the comparison reference in order to generate the same torque. The current peak value of the U phase is 1 × (30/36) = 0.833. Similarly, when the electrical angle phase is 90 °, the V-phase current is −0.417 and the W-phase current is −0.417. The relationship of the current phase at this time is equivalent to FIG.

FIG. 21 is a diagram modeling a magnetic circuit for calculating the magnetomotive force distribution. The magnetoresistance of the gap between the magnetic poles 5 and 7 and the teeth 10 and 12 is R, the magnetomotive force generated by the current flowing through the windings 14 of the slots 11 and 13 is Fk, and the teeth 10 and 12 generated by the magnetomotive force Fk The magnetic flux is φk, and the magnetic potential at each contact is Uk. Each magnetic flux φk is expressed by the following equation.
φ 1 = U 1 / R
φ 2 = U 2 / R = (F 1 −U 1 ) / R
φ 3 = U 3 / R = (F 2 + F 1 −U 1 ) / R
…………
φ n = U n / R = (F n-1 + ...... + F 2 + F 1 -U 1) / R
Since the total amount of magnetic flux flowing into the point y is 0, the condition for determining the magnetic potential Uk is given by the following formula from the above formula. That is,
U 1 + U 2 + U 3 + …… U n = 0 ---------- (8)

FIG. 22 shows the winding arrangement. When the direction of the current of each phase is the back direction of the drawing, it is U, V, W, and when the direction is the front side of the drawing, it is U ′, V ′, W ′.
FIG. 23 is a diagram showing the magnetomotive force Fk, magnetic potential Uk, and offset magnetomotive force Fo of each of the slots 11 and 13. First, the magnetomotive force Fk of the winding arrangement A and the winding arrangement B is calculated.
First, the magnetomotive force Fk generated by each winding 14 is expressed by each phase current × current direction. Here, when the current direction of each phase is represented by symbols U, V, and W, it is plus, and when it is represented by symbols U ′, V ′, and W ′, it is minus. For example, the magnetomotive force F1 at # 1 of the winding arrangement A is 0.83 × 1 = 0.83, and the magnetomotive force F3 at # 3 is −0.42 × (−1) = 0.42. Including the winding arrangement B, # 2, # 4 to # 36 can be similarly obtained.

Next, the magnetic potential Uk of the winding arrangement A and the winding arrangement B is calculated.
The magnetic potential Uk of the winding arrangement A is obtained by adding # 3, # 5 and magnetomotive force Fk sequentially from # 1. When the magnetomotive force Fk up to # 35 is finally obtained and the total value is calculated, 0.21 is obtained. Similarly, when the magnetic potential Uk of the winding arrangement B is added to # 2, # 3... # 36, it is 0.21. In the winding arrangement A and the winding arrangement B, each has a magnetic potential Uk of 0.21. However, since the condition of the equation (8) needs to be satisfied, these values correspond to the magnetomotive force Fk for each winding 14. It becomes an offset. The magnetic potential Uk from which the offset is removed is the offset magnetomotive force Fo, which is the magnetomotive force distribution of this model. Hereinafter, the magnetomotive force distribution will be described as the offset magnetomotive force Fo.

24 shows the magnetomotive force distribution of the winding arrangement A, and FIG. 25 shows the magnetomotive force distribution of the winding arrangement B. The winding arrangement A has a magnetomotive force distribution biased in the positive direction, and the winding arrangement B has a magnetomotive force distribution biased in the negative direction.
FIG. 26 shows a combined magnetomotive force distribution obtained by synthesizing the magnetomotive force distribution of the winding arrangement A and the magnetomotive force distribution of the winding arrangement B, which is the magnetomotive force distribution of the annular winding motor according to the second embodiment. As is clear from the figure, it is confirmed that the resultant magnetomotive force distribution is balanced in the vertical direction with respect to 0 as a boundary, and is not biased toward the specific slots 11 and 13.

  Also according to the second embodiment, since the windings are arranged like the winding arrangements A and B, the winding coefficient Kw can be improved and the torque ripple can be reduced.

Embodiment 3 FIG.
In the second embodiment, the annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 36 has been described. In this example, the reference position of the winding arrangement B is arranged in the # 2 slot with respect to the winding arrangement B as in the case of the winding arrangement A with the number of magnetic poles P = 10 and the number of slots Q ′ = 18. Without being reversed, the winding arrangement A and the winding arrangement B were overlapped to form an annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 36. FIG. 27 shows the distributed winding coefficient Kd, the coupling coefficient Kc, and the winding coefficient Kw of the annular winding motor.
FIG. 28 shows the arrangement of the windings 14 housed in the slots 11 and 13 in the winding arrangement A and the winding arrangement B. FIG. 29 shows the magnetomotive force distribution of the winding arrangement A, and FIG. 30 shows the magnetomotive force distribution of the winding arrangement B. FIG. 31 shows a combined magnetomotive force distribution obtained by combining the magnetomotive force distributions of the winding arrangement A and the winding arrangement B.

  As can be confirmed from the fact that the magnetomotive force distribution from # 1 of the winding arrangement A is the same as the magnetomotive force distribution from # 2 of the winding arrangement B, the distribution changes by one slot. For this reason, the resultant magnetomotive force distribution is concentrated in certain slots 11 and 13 as shown in FIG. For example, in the # 10 slot and the # 18 slot, the gap magnetomotive force is 1.67, which is larger than that of the second embodiment shown in FIG. For this reason, it causes local magnetic saturation, so that the torque / current ratio is deteriorated compared to the second embodiment and the torque ripple is larger than that of the second embodiment, but the winding coefficient Kw is improved. .

  Also according to the third embodiment, the winding coefficient Kw can be improved and the torque ripple can be reduced. That is, the third embodiment is an annular winding motor having the same number of magnetic poles P = 10 and the number of slots Q = 36 as in the second embodiment, but by selecting the winding arrangements A and B, the gap magnetomotive force is set. The torque ripple is reduced by approaching different sine waves.

Embodiment 4 FIG.
Here, an annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 48 (# 1 to # 48) will be described. Based on the same concept as in the first embodiment, the winding arrangement A and the winding arrangement B with the number of magnetic poles P = 10 and the number of slots Q ′ = 24 are arranged as shown in FIG.
That is, the reference position of the winding arrangement B is arranged at # 6, and each winding 14 inverts the phase by 180 °. By winding the winding arrangement A and the winding arrangement B, an annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 48 is obtained. The distributed winding coefficient Kd, coupling coefficient Kc, and winding coefficient Kw at this time are collectively shown in FIG. The coupling coefficients of the fifth and seventh order components are 0.947 and 0.897, respectively. Therefore, although the winding coefficient Kw is larger than that in the second embodiment shown in FIG. 16, it is improved as compared with the conventional one.

  FIG. 34 shows a combined magnetomotive force distribution synthesized by superimposing the magnetomotive force distributions of the winding arrangement A and the winding arrangement B. FIG. As in the second embodiment, the magnetomotive force distribution of the winding arrangement A is obtained as a magnetomotive force biased in the positive direction, and the magnetomotive force distribution of the winding arrangement B is obtained as a magnetomotive force biased in the negative direction. Therefore, the magnetomotive force distribution is balanced in the vertical direction with respect to 0, and the magnetomotive force is not biased to the specific slots 11 and 13.

  Also in the fourth embodiment, since the winding coefficient Kw of the harmonic component is reduced by superimposing the windings 14 arranged according to the winding arrangement A and the winding arrangement B, torque ripple can be reduced. .

Embodiment 5. FIG.
An annular winding motor having the number of magnetic poles P = 10 and the number of slots Q = 48 (# 1 to # 48) will be described. Here, the winding arrangement A and the winding arrangement B with the number of magnetic poles P = 10 and the number of slots Q ′ = 24 are set as shown in FIG.
That is, the reference position of the winding arrangement B is arranged at # 2, and each winding 14 does not invert the phase, and the winding arrangement A and the winding arrangement B are overlapped so that the number of magnetic poles P = 10 and the number of slots Q = 48 annular winding motors are obtained. FIG. 35 collectively shows the distributed winding coefficient Kd, the coupling coefficient Kc, and the winding coefficient Kw at this time. The coupling coefficients of the fifth-order component and the seventh-order component are −0.065 and −0.659, respectively. Therefore, winding coefficient Kw is improved as compared with the fourth embodiment shown in FIG.

  FIG. 37 shows the gap magnetomotive force distribution. As in the third embodiment, the magnetomotive force distribution from # 1 of the winding arrangement A and the magnetomotive force distribution from # 2 of the winding arrangement B are the same, so the transition is caused by a difference of one slot. Yes. Therefore, the magnetomotive force distribution is concentrated in a specific slot 11 or 13. For example, in the slots 11 and 13 of # 24 or # 32, the magnetomotive force is 1.88, which is larger than that of the fourth embodiment. This causes local magnetic saturation of the stator core 8, the torque / current ratio is deteriorated as compared with the fourth embodiment, and the torque ripple is increased as compared with the fourth embodiment.

  Also in the fifth embodiment, since the winding coefficient Kw of the harmonic component is reduced by superimposing the winding arrangement A and the winding arrangement B, the torque ripple can be reduced as compared with the conventional example.

Explanatory drawing which shows the winding condition of a general electric motor. The vector diagram which shows the induced voltage e1, e2, e3 of a common electric motor. The vector diagram which shows the distributed winding coefficient Kd of a general electric motor. An explanatory view showing a winding situation of an annular winding motor. The vector diagram which shows the induced voltages e1 and e2 of an annular winding motor. The expanded view of the annular winding motor in Embodiment 1 of this invention. The vector diagram which shows the coupling coefficient Kc and winding coefficient Kw of an annular winding motor. The expanded view which shows the coil | winding arrangement | positioning B of an annular winding motor. The expanded view which shows the coil | winding arrangement | positioning B of an annular winding motor. BRIEF DESCRIPTION OF THE DRAWINGS FIG. FIG. 3 is a cross-sectional view perpendicular to the axis of an annular winding motor according to winding arrangement A in Embodiment 1 of the present invention. Sectional drawing orthogonal to the axis | shaft of the annular winding motor by the coil | winding arrangement | positioning B in Embodiment 1 of this invention. The figure which shows the calculation result of the distributed winding coefficient Kd of the annular winding motor in Embodiment 1 of this invention. The figure which shows the calculation result of the coupling coefficient Kc of the ring winding motor in Embodiment 1 of this invention. The figure which shows the calculation result of the winding coefficient Kw of the annular winding motor in Embodiment 1 of this invention. The figure which shows the distributed winding coefficient Kd of the annular winding motor in Embodiment 2 of this invention, the coupling coefficient Kc, and the winding coefficient Kw. The figure which shows the coil | winding structure for the comparison reference | standard of the magnetomotive force distribution of the coil | winding in Embodiment 2-5 of this invention. Explanatory drawing which shows the electric current of each phase of a three-phase alternating current. Fig. 3 is a distribution diagram of magnetomotive force generated by each phase current. The figure which shows synthetic | combination magnetomotive force distribution. The figure which modeled the magnetic circuit for calculating a magnetomotive force distribution. The figure which shows the winding arrangement | positioning of the stator core 8. FIG. The figure which shows a magnetomotive force, a magnetic potential, and an offset magnetomotive force. The figure which shows the magnetomotive force distribution of the coil | winding arrangement | positioning A. FIG. The figure which shows the magnetomotive force distribution of the coil | winding arrangement | positioning B. FIG. The figure which shows synthetic | combination magnetomotive force distribution. The figure which shows the distributed winding coefficient Kd, the coupling coefficient Kc, and the winding coefficient Kw of the annular winding motor in Embodiment 3 of this invention. The figure which shows the winding arrangement | positioning of the stator core 8. FIG. The figure which shows the magnetomotive force distribution of the coil | winding arrangement | positioning A. FIG. The figure which shows the magnetomotive force distribution of the coil | winding arrangement | positioning B. FIG. The figure which shows synthetic | combination magnetomotive force distribution. The figure which shows the distributed winding coefficient Kd of the annular winding motor in Embodiment 4 of this invention, the coupling coefficient Kc, and the winding coefficient Kw. The figure which shows the winding arrangement | positioning of the stator core 8. FIG. The figure which shows the synthetic | combination magnetomotive force distribution which synthesize | combined the magnetomotive force distribution of the coil | winding arrangement | positioning A and the coil | winding arrangement | positioning B. FIG. The figure which shows the distributed winding coefficient Kd of the annular winding motor in Embodiment 5 of this invention, the coupling coefficient Kc, and the winding coefficient Kw. The figure which shows the winding arrangement | positioning of the stator core 8. FIG. The figure which shows the synthetic | combination magnetomotive force distribution which synthesize | combined the magnetomotive force distribution of the coil | winding arrangement | positioning A and the coil | winding arrangement | positioning B. FIG. Sectional drawing orthogonal to the axis | shaft of the conventional annular winding motor. Sectional drawing which looked at the AA line cross section of FIG. The figure which shows the distributed winding coefficient Kd of the conventional annular winding motor.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Bracket, 2 Rotating shaft, 3 Bearing, 4 Outer rotor, 5 Outer magnetic pole, 6 Inner rotor, 7 Inner magnetic pole, 8 Stator core, 9 Yoke, 10 Outer teeth, 11 Outer slot, 12 Inner teeth, 13 Inner slot, 14 Coil , 18 stator core, 20 outer teeth, 22 inner teeth, 28 stator core, 30 outer teeth, 32 inner teeth.

Claims (3)

  1.   A stator core having an outer tooth projecting on the outer periphery of an annular yoke, an inner tooth projecting to correspond to the outer tooth inside the yoke, and a stator core that encloses the stator core and faces the outer tooth. An outer rotor having a magnetic pole disposed on the inner surface, an inner rotor having a magnetic pole enclosed in the stator core and opposed to the inner teeth, and the outer teeth and the inner teeth. Each of the outer rotor and the outer rotor, and a winding wound in a toroidal shape around the yoke portion, and the winding is energized by a multiphase alternating current to generate a rotating magnetic field. In the annular winding motor for rotating the inner rotor, the windings are arranged in a predetermined order according to the phase order of the polyphase alternating current, and the slot Are arranged in such a manner that the winding groups of the winding arrangement A are inserted in turn in the same order, and in the same order as the order of the winding arrangement A, and the current direction is the same. The winding arrangement A is composed of a winding group of a winding arrangement B inserted in order from one of the slots and has a phase difference from the winding arrangement A, and the winding arrangement A and the winding arrangement B An annular winding motor characterized in that the resultant magnetomotive force distribution by the wire group is made to approximate a sine wave.
  2.   The winding arrangement B is a winding arrangement in which the winding arrangement B is inserted into the remaining slot so that windings having the same phase as the winding arrangement A and having the same current direction are adjacent to each other. Ring winding motor.
  3.   The annular winding motor according to claim 1, wherein the winding arrangement B is changed to a winding arrangement in which the current direction is the same, and the current direction is reversed in the same order as the winding arrangement A. An annular winding motor with a winding arrangement.
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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2009033952A (en) * 2007-07-05 2009-02-12 Panasonic Corp Electric motor
JP2009284626A (en) * 2008-05-21 2009-12-03 Daihatsu Motor Co Ltd Stator of rotating machine and motor
US8552609B2 (en) 2009-08-06 2013-10-08 Panasonic Corporation Synchronous motor and system for driving synchronous motor
WO2014167877A1 (en) * 2013-04-10 2014-10-16 株式会社日立産機システム Permanent magnet synchronous machine and compressor using same
WO2015018083A1 (en) * 2013-08-09 2015-02-12 深圳市配天电机技术有限公司 Birotor motor as well as fan and compressor using same
CN106887908A (en) * 2016-11-16 2017-06-23 南方电机科技有限公司 A kind of stator, motor, motor and control device, equipment

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JP2000050550A (en) * 1998-07-28 2000-02-18 Sankyo Seiki Mfg Co Ltd Toroidal shading coil motor
JP2001037133A (en) * 1999-07-23 2001-02-09 Matsushita Electric Ind Co Ltd Stator and motor

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JPS63167379U (en) * 1987-04-20 1988-10-31
JPH10271782A (en) * 1997-03-28 1998-10-09 Matsushita Electric Ind Co Ltd Motor
JP2000050550A (en) * 1998-07-28 2000-02-18 Sankyo Seiki Mfg Co Ltd Toroidal shading coil motor
JP2001037133A (en) * 1999-07-23 2001-02-09 Matsushita Electric Ind Co Ltd Stator and motor

Cited By (11)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2009033952A (en) * 2007-07-05 2009-02-12 Panasonic Corp Electric motor
US8970080B2 (en) 2007-07-05 2015-03-03 Panasonic Intellectual Property Management Co., Ltd. Electric motor having reduced cogging torque
JP2009284626A (en) * 2008-05-21 2009-12-03 Daihatsu Motor Co Ltd Stator of rotating machine and motor
US8552609B2 (en) 2009-08-06 2013-10-08 Panasonic Corporation Synchronous motor and system for driving synchronous motor
WO2014167877A1 (en) * 2013-04-10 2014-10-16 株式会社日立産機システム Permanent magnet synchronous machine and compressor using same
JP2014204646A (en) * 2013-04-10 2014-10-27 株式会社日立産機システム Permanent magnet synchronous machine and compressor employing the same
CN105075071A (en) * 2013-04-10 2015-11-18 株式会社日立产机系统 Permanent magnet synchronous machine and compressor using same
CN105075071B (en) * 2013-04-10 2017-09-22 株式会社日立产机系统 Permagnetic synchronous motor and use its compressor
WO2015018083A1 (en) * 2013-08-09 2015-02-12 深圳市配天电机技术有限公司 Birotor motor as well as fan and compressor using same
CN106887908A (en) * 2016-11-16 2017-06-23 南方电机科技有限公司 A kind of stator, motor, motor and control device, equipment
CN106887908B (en) * 2016-11-16 2019-03-29 南方电机科技有限公司 A kind of stator, motor, motor and control equipment, equipment

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