JP2006148521A - Cofdm modulation system receiver and adjacent channel disturbance excluding method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a technology of enhancing a resistance to an adjacent signal wave disturbance characteristic in the case of receiving a COFDM modulation broadcast wave and improving the oscillation purity of a local oscillator. <P>SOLUTION: An unnecessary power calculation section 502 detects unnecessary power of an adjacent channel depending on a result of the detection of a scalar quantity by a scalar quantity detection section 501 of a demodulation section 50. A system controller 80 carries out shift control of the pass band by changing a phase comparison frequency of PLLs 10, 4 of frequency synthesizers 61, 62 (double synthesizer) in response to the detected power so as to change frequencies of first and second IF signals thereby enhancing the resistance to the adjacent signal wave disturbance characteristic at the reception of the COFDM modulation broadcast wave and also improving the oscillation purity of the local oscillator through the adoption of the double synthesizer system, resulting in improving the C/N of a base band signal transmitted to the demodulation section. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a COFDM modulation system (broadcast) receiver, and more particularly to an adjacent channel interference elimination method for eliminating interference from adjacent channels.

  FIG. 6 is a diagram showing a block configuration example of a tuner unit of an existing DAB mobile receiver as an example of a conventional mobile receiver (see, for example, Patent Document 1). This example corresponds to a digitally modulated radio, but the tuner is not significantly different from a radio, analog TV, FM, or AM, and has an amplitude level that is easy to electrically process while maintaining C / N. The purpose is to extract only the desired signal by amplification and to send it faithfully (without distortion) to the demodulator.

  Next, the basic operation of the conventional mobile receiver will be described with reference to FIG. First, power induced in an antenna element (not shown) is taken in from an RF input terminal 45, and two bands of L-Band (1452-1492MHz) and Band3 (175-250MHz) currently used in DAB are extracted by the diplexer 1. Then, they are sent to the L-Band gain variable RF amplifier 25 and the combiner 2, respectively.

  The L-BandRF signal sent to the L-Band gain variable RF amplifier 25 is amplified there, and then in the L-Band mixer 26, the reference local oscillator 34, the first bandpass filter 33, and the first PLL block. 30, the second low-pass filter 28, the L-Band local oscillator 27, and the first buffer 8 are mixed with a signal having a fixed oscillation frequency, and are transmitted to the combiner 2 as a Band 3 band RF signal.

  Also, the output signal from the L-Band mixer 31 is smoothed after envelope detection by the L-Band AGC block 31, and then converted to a DC voltage, and the signal controls the gain of the L-Band gain variable RF amplifier 25. The AGC loop of the L-Band downconverter block is formed.

  The Band 3 RF signal sent to the combiner 2 is sent to the Band 3 first tracking double-tuned filter 3, band-limited, amplified by the Band 3 gain variable RF amplifier 4, and then again by the second tracking double-tuned filter 5. The band 3 is sent to the first mixer 6 after being limited.

  The Band3 RF signal band-limited by the Band3 second tracking double-tuned filter 5 includes a PLL block 10, a first low-pass filter 15, a second buffer 9, an RF stage local oscillator 7, a first DATA 36, A transmission line composed of one CLOCK 37 receives an N value, which is one of the control signals from the system controller, and generates a signal having a variable oscillation frequency generated by performing synthesized tuning, and the first mixer 6. Is mixed. In addition, here, the tuning voltage which is the control signal of the oscillation frequency of the RF stage local oscillator 7 sent from the first low-pass filter 15 is the first tracking double-tuned filter 3 and the second tracking double-tuned filter 5. Is also used as a control signal for adjusting each center frequency to a desired reception frequency.

  The signal down-converted to the first IF frequency by the first mixer 6 is amplified by the first IF amplifier 12 and then subjected to narrowband limitation by the first IF bandpass filter 13. After being amplified by one IF amplifier 14, it passes through an attenuator 16 and is sent to the second mixer 17.

  On the other hand, the first IF signal is envelope-detected by the RF stage AGC block 11 and then smoothed and converted into a DC voltage, which is given to the gain variable RF amplifier 4 as a gain control signal. An AGC loop is formed. The first IF signal sent from the attenuator 16 is further mixed in the second mixer 17 with a fixed oscillation frequency signal generated by the crystal 20 and the IF local oscillator 32, thereby down-converting to the second IF frequency. Then, it is sent to the second IF amplifier 22, the IF stage AGC block 18, and the envelope detector (RSSI block) 19.

  In the IF stage AGC block 18, the output signal from the second mixer 17 is smoothed after envelope detection, and a direct current serving as a control signal for the attenuation amount of the attenuator 16 is sent out. The second IF signal sent to the second IF amplifier 22 is amplified there, and then subjected to band limitation by the second IF bandpass filter 23, and the second IF amplifier 24 matches the subsequent demodulator. After being amplified to the signal level, it is sent to the IF output terminal 40.

  Further, the second IF signal sent to the envelope detector (RSSI block) 19 is subjected to envelope detection there, and the fourth low-pass filter 41 at the next stage limits the band above the frequency of the second IF. . Thereafter, it is sent to the RSSI output terminal 42 via the third buffer 21 as an RSSI signal which is a synchronization signal of a demodulator (not shown).

An AFC control signal having frequency offset information is sent from the demodulator to the AFC control terminal 39, and fine control of the oscillation frequency of the reference local oscillator 34 is performed. Then, the output signal of the reference local oscillator 34 is subjected to removal of harmonics by the first band pass filter 33, and then supplied to the second PLL block 10 and the first PLL block 30 as a reference frequency source. Therefore, the tuning to the desired signal is performed with high accuracy. The PLL lock information is transmitted from the LOCK 38 to the outside (for example, a microcomputer).
JP 11-46154 A (page 5-6, FIG. 1)

  However, in the conventional mobile receiver, (1) like the existing DAB broadcast, when the gap (gap) on the frequency axis with the undesired adjacent wave is narrower than the self-occupied bandwidth of the broadcast wave, Cases in which selectivity characteristics are insufficient in the out-of-band exclusion capability of the bandpass filter provided in the IF stage are likely to occur. (2) When performing synthesized tuning, the reference side for performing PLL phase comparison on the standard of the reception channel step of 16 kHz minimum with respect to the reception frequency of 1.5 GHz of L-Band, In addition to the increased frequency division value on the control side, the use of FFT as the demodulation method makes it difficult to achieve the required oscillation purity of the local oscillator. (3) Trying to improve the ability to eliminate adjacent interfering waves and making the filter multi-stage prevents miniaturization of the receiver. (4) Further attempts to improve the ability to eliminate adjacent interfering waves and making the filter multi-stage will increase costs. (5) Furthermore, trying to improve the ability to eliminate adjacent interference waves and making the filter multi-stage increases the number of components and leads to an increase in the failure rate of the receiver.

  The present invention has been devised in view of the above circumstances, and an object of the present invention is to improve the resistance to adjacent signal waves without impairing the downsizing of the receiver, the failure rate, and the suppression of the cost increase. It is another object of the present invention to provide an adjacent channel interference elimination method capable of improving the oscillation purity of a local oscillator and a COFDM modulation system receiver employing the adjacent channel interference elimination method.

  In order to achieve the above object, the present invention converts a first frequency conversion unit that converts a COFDM modulated received signal into a first intermediate frequency signal, and converts the first intermediate frequency signal into the second intermediate frequency signal. A COFDM modulation type receiver having a second frequency converting unit and a demodulating unit for demodulating the second intermediate frequency signal, and a channel adjacent to a tuning target channel when demodulating the second intermediate frequency signal Power detecting means for detecting the power of the first frequency changing means, first frequency changing means for changing the frequency of the first intermediate frequency signal in accordance with the detected power, and in response to a change in the first intermediate frequency signal And second frequency changing means for changing the frequency of the second intermediate frequency signal.

  The present invention is also a method for eliminating adjacent channel interference in a COFDM modulation type receiver, wherein a part of power of a channel adjacent to a tuning target channel is detected and converted from a received COFDM modulated signal according to the detected power. It is characterized by changing the frequency of the intermediate frequency signal.

  As described above, the present invention detects the power (unnecessary power) of the channel adjacent to the channel to be tuned, and changes the frequency of the intermediate frequency signal of the received COFDM modulated signal in the direction in which the detected power decreases. In addition, it is possible to improve the anti-adjacent signal wave interference characteristics without impairing the downsizing of the receiver, the failure rate, and the suppression of the cost increase. For this purpose, the received COFDM modulated signal is converted into a first intermediate frequency signal by a double synthesizer, and then the power is detected and detected when demodulating it by converting it into a second intermediate frequency signal. The frequency of the first intermediate frequency signal is changed according to the power, and the frequency of the second intermediate frequency signal is changed according to the frequency change of the first intermediate frequency signal. In addition to the above effects, the oscillation purity of the local oscillator can be improved by changing the center frequency to be equivalent to the case of the center frequency.

According to the present invention, the power (unnecessary power) of a channel adjacent to the channel to be tuned is detected, and the frequency of the first intermediate frequency signal of the received COFDM modulated signal is changed in a direction in which the detected power decreases. As a result, it is possible to improve the anti-adjacent signal wave interference characteristics without impairing the downsizing of the receiver, the failure rate, and the suppression of the cost increase.
Further, the received COFDM modulated signal is converted into first and second intermediate frequency signals by a double synthesizer method, and the first intermediate frequency signal frequency is changed according to the detected unnecessary power, and a local oscillator The oscillation purity can be improved.

  Detects the power (unnecessary power) of the channel adjacent to the channel to be tuned for the purpose of improving anti-adjacent signal wave interference characteristics when receiving COFDM modulated broadcast waves without impairing downsizing, failure rate, and cost increase suppression. And it implement | achieved by changing the frequency of the 1st intermediate frequency signal of the received COFDM modulation signal in the direction where the detected electric power decreases.

  FIG. 1 is a block diagram showing a configuration of a COFDM modulation system receiver according to an embodiment of the present invention. However, the same parts as those in the conventional example will be described with the same reference numerals. The COFDM modulation type receiver includes a diplexer (distributor) 1, a combiner 2, a first tracking double-tuned filter 3, a variable gain RF amplifier 4, a second tracking double-tuned filter 5, a first mixer 6, and an RF stage local part. Oscillator 7, first buffer 8, second buffer 9, second PLL block 10, RF stage AGC block 11, first IF amplifier 12, first IF bandpass filter 13, first IF amplifier 14, first low-pass filter 15, attenuator 16, second mixer 17, IF stage AGC block 18, envelope detector (RSSI block) 19, second low-pass filter 20, third buffer 21, second IF amplifier 22, second IF bandpass filter 23, second IF amplifier 24, L-Band gain variable RF amplifier 25 L-Band mixer 26, L-Band local oscillator 27, third low-pass filter 28, fourth buffer 29, first PLL block 30, L-Band AGC block 31, IF local oscillator 32, first band-pass filter 33, reference local oscillator 34, antenna power supply terminal 35, DATA 36, CLOCK 37, LOCK 38, AFC control terminal 39, IF output terminal 40, fourth low-pass filter 41, RSSI output terminal 42, AND circuit 43, and third PLL block 44 , An RF input terminal 45, a demodulator 50, a system controller 80, a scalar quantity calculator 501 and an unnecessary power detector 502, and the reception frequency band and the reception frequency global width are exemplified by DAB. However, numerical values are different between DVB and one-segment terrestrial digital television.

  Next, the operation of the present embodiment will be described. First, the power induced in the antenna element is taken in from the RF input terminal 45, and two bands of L-Band (1452-1492MHz) and Band3 (175-250MHz) are extracted by the diplexer 1, and each L-Band gain variable RF is extracted. It is sent to the amplifier 25 and the combiner 2.

  After the L-BandRF signal sent to the L-Band gain variable RF amplifier 25 is amplified, the L-Band mixer 26 uses a fourth buffer 29, a first PLL block 30, a third low-pass filter 28, L-Band local oscillator 27, reference local oscillator 34, mixed with a signal generated by AFC composed of first bandpass filter 33 and subsequent demodulator 60, down-converted to Band 3 band RF signal, It is sent to the combiner 2. The output signal from the L-Band mixer 26 is smoothed after envelope detection by the L-Band AGC block 31 and converted to a DC voltage to control the gain of the L-Band gain variable RF amplifier 25. -AGC loop of Band downconverter block is formed.

  The Band 3 RF signal sent from the diplexer 1 or from the L-Band mixer 26 to the combiner 2 is band-limited by the first tracking double-tuned filter 3 and amplified by the gain variable RF amplifier 4. The amplified Band 3 RF signal is further band-limited by the second tracking double-tuned filter 5 and sent to the first mixer 6.

  Synthesizing by a DATA 36, CLOCK 37 bus and second PLL block 10, first low-pass filter 15, first buffer 8 and RF stage local oscillator 7 for receiving an N value which is a control signal from the system controller 80. A signal having a variable oscillation frequency generated by the size tuning is sent to the first mixer 6. The band-limited Band 3 RF signal from the second tracking double-tuned filter 5 is mixed by the first mixer 6.

  The signal down-converted to the first IF frequency by the first mixer 6 is amplified by the first IF amplifier 12 and then, for example, a first IF band suitable for a surface acoustic wave filter, a ceramic filter, or the like. The narrow band limitation corresponding to the occupied bandwidth of a certain broadcast signal (for example, 1.536 MHz in the case of DAB) is received by the pass filter 13, amplified again by the first IF amplifier 14, passed through the attenuator 16, It is sent to the second mixer 17.

  On the other hand, the first IF signal is envelope-detected by the RF stage AGC block 11, smoothed and converted to a DC voltage, which is supplied to the gain variable RF amplifier 4 as a gain control signal. This closed loop becomes the AGC of the front end of Band3.

The first IF signal sent from the attenuator 16 is further supplied to the third PLL from the system controller 80 in the second mixer 17 in the same manner as the first D / C (down converter) including the first mixer 6. DATA 36 for receiving another N value which is a control signal of the block 44, a bus composed of the CLOCK 37, and the IF local oscillator 32 and the second buffer 9, the third PLL block 44, the second low-pass filter 20, And a second synthesized tuning portion (same as the synthesizer 61 in FIG. 3) composed of the reference local oscillator 34 and the first bandpass filter 33, and a demodulator 60 that follows the second synthesized tuning portion. Is mixed with the signal. Then, it is sent to the IF stage AGC block 18, the second IF amplifier 22, and the envelope detector 19.

  The second IF signal sent to IF stage AGC block 18 is then smoothed after envelope detection, converted to a direct current, and sent to attenuator 16 as a signal for controlling the attenuation. This closed loop is an IF-stage AGC centered on the second mixer 17. The second IF signal sent to the second IF amplifier 22 is amplified there, and then subjected to a narrowband band limitation close to the characteristics of the first IF bandpass filter 13 by the second IF bandpass filter 23, and finally Specifically, the signal is amplified by the second IF amplifier 24 to a signal level suitable for the demodulator 50 (for example, an AD converter configured therein) and sent to the IF output terminal 40.

  The second IF signal sent to the envelope detector 19 is envelope-detected there, and is limited by the fourth low-pass filter 41 in the next stage to a band equal to or higher than the frequency of the second IF. Thereafter, an RSSI signal (signal strength detection signal), which is also a signal for the demodulator 50 to obtain time synchronization, is sent from the RSSI output terminal 42 to the demodulator 50 via the third buffer 21.

  An AFC control signal having a frequency offset amount is sent from the demodulator 50 to the AFC (automatic frequency control) control terminal 39 (DC voltage when the reference local oscillator is controlled by voltage such as TCVCXO). With respect to the Doppler shift of the center frequency of the desired wave, fine control of the oscillation frequency of the reference local oscillator 34 is performed every moment.

  The output signal of the reference local oscillator 34 is given to the second PLL block 10 as a reference frequency source after unnecessary components other than the fundamental wave are removed by the first band pass filter 33. As a result, high-precision tuning to the desired signal is performed. The signal output to the terminal of the LOCK 38 is a logical synthesis sum that notifies that the PLL of each of the second PLL block 10 and the third PLL block 44 has reached the locked state.

  In the demodulator 50 at the subsequent stage, the multi-carrier second IF signal transmitted from the IF output terminal 40 is converted into a digital signal by an AD converter, and IQ demodulation (demodulation of an orthogonal component), FFT (time axis-frequency expansion) ), A series of OFDM waves are demodulated. As one of the demodulation results, the level of each carrier constituting the multicarrier is also calculated, but the number of FFT points is 2N (N is an integer) by performing DFT (discrete Fourier transform processing), and demodulation is performed. The unit 50 performs the FFT processing with the number of points equal to or greater than the number of multi-carriers of the OFDM modulated wave and 2N. In the DAB standard, when the transmission mode is MODE1, the maximum number of multi-carriers is 1,536, the occupied bandwidth is 1.536 MHz, and more than 2048 FFT points are required. As can be seen, there are at least 256 kHz bandwidths above and below the desired received wave, in other words, 256 free FFT points.

  The scalar quantity detector 501 of the demodulator 50 detects the scalar quantity by the FFT processing in the above-described band located above and below. The unnecessary power calculation unit 502 calculates unnecessary power such as other DAB signals and TV signals of channels adjacent to the desired wave reception channel based on the detected scalar quantity by detecting FFT points other than the desired signal, and this is calculated by the system. Output to the controller 80. The calculation is performed by changing the control of two kinds of N values sent from the system controller 80 to the second PLL block 10 and the third PLL block 44 in accordance with the degree of the level. An operation of shifting the frequency in the direction in which the unnecessary power is reduced is performed.

  FIG. 3 is a diagram showing the relationship between the unnecessary power detection value in the demodulator 50 and the shift amount of the N value of the second PLL block 10. Control is performed such that the amount of shift increases stepwise as the unnecessary power increases.

  FIG. 4 is a block diagram showing an embodiment (DAB MODE 1) of the two frequency synthesizers (double synthesizers) 61 and 62 described above. Basically, the phase comparison frequency of the frequency synthesizer 61 that down-converts the first-stage RF signal to the first IF signal is set to the IF shift step value, and is shifted by an integer multiple of the step value. Enable operation. In this embodiment, the frequency step is 64 kHz, which is a setting in which the divided value of the five values N1 can be varied before and after. Then, since the phase comparison frequency is 64 kHz with respect to 16 kHz (DAB transmission frequency grid standard) of the conventional example (PLL block 10 in FIG. 6), the divided value becomes 1/4. Further, in another frequency synthesizer 62 that downconverts the first IF signal at the second stage to the second IF signal, the second IF signal is shifted in accordance with the shift amount of the first IF signal. The setting can return to the original second IF frequency.

  FIG. 5 shows an example of setting in FIG. 3 (DAB MODE1) performed on the two PLLs 10 and 44 when performing frequency shift, that is, the relationship between the two N values of N1 and N2 and the shift amount of the first IF frequency. FIG. In FIG. 5, dN1 is a variable relative value of N1 in the PLL block 2 that shifts the frequency of the RF local oscillator, and dN2 is a variable relative value of N2 in the PLL block 3 that shifts the frequency of the RF local oscillator. Is shown. In FIG. 6, the range of the shift value (dN1) of N1 is ± 5, and the range of the shift value (dN2) of N2 is ± 22.

  Thus, for example, when the first IF frequency is shifted by +132 kHz, the N value of the second PLL block 10 (referred to as the frequency division value N1 of the RF stage local oscillator in the PLL block 2) is set to N1 + 3, and the PLL block 3 If N2 + 3 × 4, the second IF frequency is the original center frequency. The shift of 132 kHz, which is this example, is actually only one control example in the case where the adjacent wave exists above the occupied band of the desired signal as shown in FIG.

  According to the present embodiment, the center frequency of the DAB desired received signal that is passed through the filter having a fixed frequency, specifically, the frequencies of the first and second IF signals of the frequency synthesizers 61 and 62 are shifted. Thus, the anti-adjacent signal wave interference characteristics are improved, and the same effect as changing the frequency characteristics of the filter is obtained. At this time, of course, the adjacent wave is not only filtered and attenuated, but it is not an ideal filter, so the signal of the own wave is also filtered. However, COFDM, which is a digital modulation method such as DAB, is large. Interleaving is performed on both the frequency axis and the time axis, and large redundancy and strong error code correction (Viterbi code, etc.) are performed, so the quality of information deteriorates even if part of the desired signal reception band is lost. The margin is large up to the threshold value. Such a feature of the COFDM system is also a major factor that can realize the present scheme.

  Actually, the minimum sensitivity degradation (several dB) due to the total power loss of the multi-carrier group (Ensembles) of the self-wave over the bandwidth of about 300 kHz in this setting is slight, and due to the adjacent wave. To improve the disturbance (deterioration of bit error rate) characteristic, specifically, the ratio of the power of the undesired signal and the unwanted signal to the power of the desired signal (DU ratio) (Amplitude dynamic range) can be greatly improved, and at this time, the above-mentioned effect can be obtained while using an existing fixed IF filter as it is.

  Further, since the filtering is performed by changing the N value of the two PLLs 10 and 44 and shifting the first IF frequency, it is structurally adjacent to the N value control from the system controller 80. The use of the FFT point calculation result that is not used for the demodulation of the COFDM signal for detecting the unnecessary power of the channel is expensive, and the required value is high (such as a SAW filter with a steep skirt characteristic) or large in size The use of a filter can be avoided, and the circuit scale of the tuner can be prevented from becoming large.

  Further, since two frequency synthesizers 61.62 for tuning the frequency of the received signal are owned, the local oscillator of the local oscillator that depends on the increase of the frequency division value when generating the reference signal for phase comparison caused by the fineness of the channel step of the broadcast signal is used. By dividing the frequency division value by the two PLLs 10 and 44 against the deterioration of the oscillation purity, the effect of improving the total signal purity (SSB phase noise) of the local transmitter can be obtained.

  Furthermore, in comparison with the conventional system configuration, only one PLL block and one AND circuit are added to the analog signal processing unit (digital) without using a multi-stage filter to try to improve the ability to eliminate adjacent interference waves. (There is no signal processing unit and system control unit) Since the above effect is obtained, it is possible to prevent the downsizing of the apparatus, the failure rate from being increased, or the apparatus from being expensive.

  In addition, this invention is not limited to the said embodiment, In the range which does not deviate from the summary, it can implement also with another various form in a concrete structure, a function, an effect | action, and an effect.

It is the block diagram which showed the structure of the COFDM modulation system receiver which concerns on one embodiment of this invention. It is the figure which showed the band limitation characteristic example of the band pass filter (SAW filter) 13 shown in FIG. FIG. 3 is a diagram showing a relationship between an unnecessary power detection value in the demodulator shown in FIG. 1 and an N value shift amount of a second PLL block 10. It is the block diagram which showed the Example of the two frequency synthesizers shown in FIG. It is the figure which showed the relationship between two N value supplied to PLL10 and PLL44 shown in FIG. 1, and 1st IF frequency shift amount. It is the figure which showed the block structural example of the tuner part of the existing DAB mobile receiver as an example of the conventional mobile receiver.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Diplexer (distributor), 2 ... Combiner, 3 ... 1st tracking double-tuned filter, 4 ... Variable gain RF amplifier, 5 ... 2nd tracking double-tuned filter, 6 ... 1st Mixer, 7 ... RF stage local oscillator, 8 ... first buffer, 9 ... second buffer, 10 ... second PLL block, 11 ... RF stage AGC block, 12, 14 ... first IF amplifier, 13 ... 1st IF band pass filter, 15 ... 1st low pass filter, 16 ... Attenuator, 17 ... 2nd mixer, 18 ... IF stage AGC block, 19 ... Envelope Line detector (RSSI block), 20 ... second low-pass filter, 21 ... third buffer, 22, 24 ... second IF amplifier, 23 ... second IF bandpass filter, 25 ... L-Band gain variable RF amplifier, 26 ... L-Band mixer, 27 ... L-Band local oscillator, 28 ... Third low-pass filter, 29 ... Fourth buffer, 30 ... First PLL block , 31... L-BandAGC block, 32... IF local oscillator, 33... First bandpass filter, 34... Reference local oscillator, 35. 38... LOCK, 39... AFC control terminal, 40... IF output terminal, 41... 4th low-pass filter, 42... RSSI output terminal, 43. 45 …… RF input terminal, 50 …… Demodulator, 61, 62 …… Frequency synthesizer, 80 …… System controller, 501 …… Scalar amount detection , 502 ...... unnecessary power calculation unit.

Claims (8)

A first frequency converter that converts a COFDM modulated received signal into a first intermediate frequency signal; a second frequency converter that converts the first intermediate frequency signal into the second intermediate frequency signal; A COFDM modulation type receiver having a demodulator that demodulates two intermediate frequency signals,
Power detection means for detecting the power of a channel adjacent to the tuning target channel when demodulating the second intermediate frequency signal;
First frequency changing means for changing the frequency of the first intermediate frequency signal according to the detected power;
Second frequency changing means for changing the frequency of the second intermediate frequency signal in response to the change of the first intermediate frequency signal;
A COFDM modulation system receiver comprising:   The second frequency change means has a frequency of the second intermediate frequency signal according to a frequency change of the first intermediate frequency signal, and the first intermediate frequency signal is a center frequency of the channel to be tuned. 2. The COFDM modulation system receiver according to claim 1, wherein the receiver is changed to be equivalent to the case.   The power detection means detects unnecessary power of an adjacent channel using a calculated value of an unused frequency point for demodulation among calculated values in time axis frequency expansion (FFT) at the time of demodulation. The COFDM modulation system receiver according to claim 1.   The first frequency changing means changes the frequency of the first intermediate frequency signal stepwise according to the detected power, and the second frequency changing means changes the frequency of the second intermediate frequency signal. 2. The COFDM modulation type receiver according to claim 1, wherein the COF modulation type receiver is changed in steps according to the detected power.   The frequency division value for generating a reference signal for phase comparison used in the first frequency conversion unit and the second frequency conversion unit is respectively assigned to two PLL circuits, and both are made variable. The COFDM modulation system receiver according to Item 1. A method for eliminating adjacent channel interference in a COFDM modulation receiver, comprising:
A method of eliminating adjacent channel interference, comprising detecting a part of power of a channel adjacent to a tuning target channel and changing a frequency of an intermediate frequency signal converted from a received COFDM modulated signal according to the detected power.   Detecting the power of a channel adjacent to the channel to be tuned when converting the received COFDM modulated signal into a first intermediate frequency signal, and further converting the first intermediate frequency signal into a second intermediate frequency signal; The adjacent channel interference elimination method according to claim 6, wherein the frequencies of the first and second intermediate frequency signals are changed in accordance with the detected power.   8. The adjacent channel interference elimination method according to claim 7, wherein there is a predetermined relationship between a frequency change of the first intermediate frequency signal and a frequency change of the second intermediate frequency signal.

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