JP2003052190A - Resonant inverter controller, and vehicle device using inverter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an inverter controller, which controls a resonant soft- switching inverter circuit having reduced inductance for resonance, according to its principle of operation. SOLUTION: When a rotational angle θ of a motor exists, for example, in the area 1 of a space vector, a control CPU 5 outputs an output voltage vector V1 at the time t00 for an interval (T1); and next at a time t01, it transits to an output voltage vector V4 and outputs the vector for a time (T3/4), and at the time t02, it transits to an output voltage vector V3 and outputs the vector for a time (T2+T3/2). After outputting the output voltage vector V3, it retransits to an output voltage vector V4 at a time t03 and outputs the vector for an interval (T3/4), and then it repeats the output, in order of output voltage vectors V1, V4, V3, and V4. Since the zero vector is a value obtained by subtracting the output time of the output voltage vector from the output time of the output voltage vector V3, it follows that T2+T3/2-(T3/4+T3/4)=T2.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter controller for controlling an inverter circuit for driving a motor, and more particularly, a resonance type inverter controller for controlling an inverter circuit having a buffer capacitor for soft switching. Regarding

[0002]

2. Description of the Related Art Conventionally, an inverter circuit for driving a load such as a motor is disclosed in US Pat. No. 5,710,698.
There are techniques described in Japanese Patent No. 5,642,273, US Pat. No. 5,047,913, and the like. According to these, for example, as shown in FIG. 15, the soft switching inverter of the conventional example has, for example, an IGBT (Insulated Gate Bipolar Transistor) to which a motor 1 including a three-phase induction motor or a DC brushless motor is connected as a load.
istor: insulated gate bipolar transistor) Q1
To Q6 are used as switching elements. The inverter section is provided at both ends of the DC power supply 3,
The IGBTs Q1 to Q6 are connected in a three-phase bridge structure composed of U-phase, V-phase, and W-phase, and a regenerative motor 1 or other inductive load is generated between the collector and emitter terminals of each IGBT. Commutation diodes (FWD: Free Wheeling Diodes) D1 to D6 are connected for the purpose of circulating energy or current energy stored in an inductive load. A buffer capacitor C1 is provided between the collector terminal and the emitter terminal of each IGBT to absorb a surge voltage applied between the collector terminal and the emitter terminal of the IGBT when the IGBT is turned on or turned off.
~ C6 are also connected.

Further, the connection point between the U-phase buffer capacitors C1 and C2 of the inverter section, and the V-phase buffer capacitor C3.
And C4, and between the W-phase buffer capacitors C5 and C6, respectively, between the buffer capacitors C1 and C2.
Inductance L4 and buffer capacitor C that resonate with
3, an inductance L5 that resonates with C4, an inductance L6 that resonates with the buffer capacitors C5 and C6, and bidirectional switch units SU4 to SU6 for flowing a resonance current through the inductances are connected. A smoothing capacitor C9 is connected to the DC power supply 3.

The above-described structure is generally called an auxiliary resonance AC link type snubber inverter.
In the soft switching inverter of 5, for example, the IGBT Q1 of the U phase is turned off, and the IGBTQ1
The charge current of the buffer capacitor C1 and the discharge current of the buffer capacitor C2 when it is desired to turn on 2 are made to flow to the V phase or the W phase through the inductance L4 or L6, and at the same time, the IGBT Q4 and Q6 are turned off.
However, when it is desired to turn on the IGBTs Q3 and Q5 with a slight delay, the charging current of the buffer capacitors C4 and C6 and the discharging current of the buffer capacitors C3 and C5 are the inductance L
Supply from U phase through L4 and L6.

Therefore, when the IGBT is turned off and the buffer capacitor is charged by charging and discharging the buffer capacitor by the resonance current of the buffer capacitor and the inductance as described above, the time constant given by the buffer capacitor is given. From the delay in the rise in the saturation voltage of the IGBT due to
When the ZVS (Zero Voltage Switching) of the IGBT is realized and, conversely, the buffer capacitor is discharged before the IGBT is turned ON, the voltage applied to the IGBT by the conduction of the commutation diode,
Since the current becomes "zero", the IGBT ZVS (Ze
Since ro voltage switching (Zero current switching) and ZCS (Zero current Switching) are realized, the turn-off of the switching element is achieved.
It is possible to reduce the loss generated at F or turn-on.

Further, Japanese Patent Publication No. 6-69305, Japanese Patent No. 2689575, Japanese Patent No. 3115795, etc. disclose techniques relating to a space vector PWM (Pulse Width Modulation) control method for controlling an inverter circuit as described above. It is disclosed. In the space vector PWM control, in order to establish the soft switching of the switching elements of each phase forming the inverter circuit, a time-division PWM control method using a space vector is used to generate timings for turning on and off the switching elements of each phase. Is used as an easy-to-use control method, and can also be used for a soft switching inverter as shown in FIG.

Therefore, a general time-division PWM control method using a space vector will be described with reference to the drawings. First, Fig. 1
IGBT forming a three-phase bridge of inverter circuit 5
When Q1, Q3, and Q5 are in a conductive state, the state of that phase is represented by a logical value "1", and IGBTs Q2, Q4, and
When Q6 is in the conductive state, the state of that phase is changed to the logical value "
Assuming that it is represented by "0", eight states can be combined as shown in Table 1 by combining the states of the three phases, and each state is represented by vectors V0 to V7.

[Table 1] Furthermore, it is known that the output voltage vector expression of the three-phase inverter with respect to the rotation angle θ (rotation position) of the motor is expressed by a space vector as shown in FIG.

Under such a premise, the time-division PWM control by the space vector is, for example, the rotation angle θ of the motor.
When the (rotation position) is within the area 1 in FIG. 16, the vectors V1 and V at both ends sandwiching the area 1 within a certain fixed time.
This is a PWM control method in which 3 and the zero vector V0 or V7 are divided and output. At this time, this fixed time is generally set to a value sufficiently shorter than one cycle of the rotation speed. Here, FIG. 17 shows an example of time-division PWM control by a space vector. In this example, the area elapsed time is set to be four times the sampling time Ts. However, in practice, the sampling time Ts is shortened and the number of switching operations within the area elapsed time is controlled to increase. In this way, time-division PWM using space vectors
In the control, for example, when the rotation angle θ of the motor is within the area 1, this sampling time Ts is time-divided and the output voltage vector in the area 1 is V1->V3-> V0.
PWM that sequentially outputs (V7) and then sequentially outputs the output voltage vector in the area 2 as V3->V2-> V0 (V7) when the rotation angle θ of the motor is within the area 2.
This is the output method.

[0009]

In the conventional soft switching inverter as described above, forming the resonance circuit by the buffer capacitor and each inductance is performed by the current flowing through the IGBT (switching element) and the applied voltage. Is effective for reducing the loss at the time of turn-on or turn-off that occurs in the switching element. However, since the core volume required for the inductance is determined by the conduction peak current, the weight and volume of the inductance increase as the load current to be controlled increases, and a current equal to or greater than the load current flows. In the conventional soft switching inverter that requires three inductances, there is a problem in that the weight and volume cannot be reduced and the size cannot be reduced due to the inductance. In addition, it is clear that reducing the inductance is the most effective for reducing the weight and size, but in order to perform the soft switching after reducing the inductance, it is necessary to control the inverter different from the conventional one. Therefore, there is a problem that the control means must be clarified.

The present invention has been made in view of the above problems, and provides an inverter control device for realizing soft switching by concretely controlling a resonance type inverter circuit having reduced resonance inductance according to its operating principle. The purpose is to provide.

[0011]

In order to solve the above-mentioned problems, the invention according to claim 1 proposes six main switching elements (for example, the IGBT of the embodiment) connected in a three-phase bridge.
TQ1 to Q6) and six commutation diodes (for example, the commutation diodes D1 to D6 of the embodiment) connected in parallel between the two terminals of the main switching element that are turned on or off by switching control, and 6 Individual buffer capacitors (for example, the buffer capacitors C1 to C1 of the embodiment)
C6) and 2 connected in series at each end of the power supply, 3
Three connection points between the three sets of main switching elements forming each phase of the phase bridge are used as three-phase output terminals for connecting a three-phase motor (for example, the motor 1 of the embodiment), and three-phase bridge connection is performed. A bridge circuit composed of six auxiliary switching elements (for example, IGBTs Q7 to Q12 of the embodiment) that connect the respective connection points of the pair of auxiliary switching elements to the three-phase output terminals and pass current in a single direction. An inverter control device for controlling an inverter circuit having a resonance inductance (for example, the resonance inductance Lr of the embodiment) connected to the opposite side of the connection point between the three sets of auxiliary switching elements of the bridge circuit. The pattern of switching control is such that the angle differs by π / 3 [rad] depending on the switching control pattern of the inverter circuit. The three-phase control means (for example, the control CPU 5 of the embodiment) that switches and outputs six output voltage vectors represented by the three-phase control signal according to the output voltage vector. At the time of switching, a reverse voltage vector whose angle is π [rad] different from the output voltage vector before switching is output for a predetermined time, and then the output voltage vector after switching is output. With such a configuration, the number of resonance inductances for generating the charging / discharging current of the buffer capacitor can be reduced, and moreover, simultaneous switching of two or more phases always occurs in the soft switching operation. P
In order to solve the control condition of the three-phase control signal (output voltage vector) that is restricted when performing WM control (zero vector cannot be output, consecutive adjacent vectors cannot be output), non-zero vector (arbitrary When a neighboring vector is output by using a combination of the output voltage vector and its reverse voltage vector), the reverse voltage vector of either one of the neighboring vectors is output and then the transition to the adjacent vector is performed. It is possible to realize the soft switching operation using the time-division PWM control by the space vector, which is equivalent to the control of outputting the zero vector of 1 or outputting the adjacent vector continuously.

According to a second aspect of the present invention, the three-phase control means controls the six regions represented by the output voltage vector,
Define a lead angle region starting from a point advanced by a predetermined angle in the motor rotation direction from the start point of the region, assigning an optimum single output voltage vector to each lead angle region, and setting the motor rotation angle to the lead angle. While existing in any of the regions, the output voltage vector assigned to the advance region is continuously output, and the rotation angle moves to another advance region in accordance with the rotation of the motor,
Switching to the output voltage vector assigned to the advanced angle region that has moved, stopping the output of the output voltage vector for a predetermined time when switching the advanced angle region, and outputting before or after switching during the stop period. Voltage vector and π [ra
d] It is characterized by outputting reverse voltage vectors having different angles.

According to a third aspect of the present invention, in the inverter control device according to the second aspect, the three-phase control means stops the output of the output voltage vector a predetermined time before the switching of the advance angle region, The reverse voltage vector is output from the stop time to the advance angle switching time.

According to a fourth aspect of the present invention, in the inverter control device according to the second aspect, the three-phase control means stops the output of the output voltage vector at the time of switching the advance region, and from the switching time. The reverse voltage vector is output for a predetermined time, and the output voltage vector of the next advance angle region is output after the predetermined time. In the inventions of claims 2 to 4, the soft switching operation using the time-division PWM control by the space vector is realized in the same manner as the conventional control of outputting the zero vector or continuously outputting the adjacent vector. Further, it is possible to reduce the switching operation in the inverter circuit and increase the maximum output of the inverter circuit.

According to a fifth aspect of the present invention, in the inverter control device according to the third or fourth aspect, the predetermined time is equal to or longer than the time from the start to the end of the switching operation per auxiliary switching element. It is characterized by With such a configuration, it is possible to sufficiently obtain the soft switching effect by the resonance inductance in the control of outputting the reverse voltage vector.

According to a sixth aspect of the present invention, in the inverter control device according to any of the second to fifth aspects, the three-phase control means has a motor rotation speed of a predetermined rotation speed or more and an advance angle. It is characterized in that the output voltage vector is controlled by the region. With such a configuration, by applying the control of the inverter control device according to the above-mentioned claim 2 to claim 5 after the rotation operation of the motor is stabilized, the output voltage vector in the advance region is stabilized after the operation is stabilized. It is possible to rotate the motor more stably and efficiently by performing the control of.

According to a seventh aspect of the present invention, there is provided an electric storage device for supplying electric energy to an electric motor driving a drive shaft of a vehicle, and an inverter for adjusting an electric energy supply amount supplied from the electric storage device to the electric motor. In the equipped vehicle device, the inverter includes six main switching elements connected in three-phase bridge and six commutations connected in parallel between the two terminals of the main switching elements that are turned on or off by switching control. Connect a diode, six buffer capacitors, and two connection points at each end of the power supply in series to each of the three sets of main switching elements forming each phase of the three-phase bridge. As a three-phase output terminal for connection, each connection point of three sets of auxiliary switching elements connected in a three-phase bridge is connected to the three-phase output terminal, respectively. An inverter circuit including a bridge circuit composed of six auxiliary switching elements for passing a current in a direction, and a resonance inductance connected to a side opposite to a connection point between the three sets of auxiliary switching elements of the bridge circuit, Three-phase control for switching and outputting six output voltage vectors represented by three-phase control signals according to the switching control pattern, the angles of which differ by π / 3 [rad] according to the switching control pattern of the inverter circuit. The three-phase control means has means, and with respect to the six regions represented by the output voltage vector, an advance angle region whose start point is a point advanced from the start point of the region by a predetermined angle in the rotation direction of the motor. Defined and assigned a single optimum output voltage vector to each advance region, and the output assigned to that advance region while the rotation angle of the motor exists in any of the advance regions. If the rotation angle moves to another advance angle region in accordance with the rotation of the motor, the force voltage vector is continuously output, and the output voltage vector is switched to the output voltage vector assigned to the advance angle region after the movement. At the time of switching, a reverse voltage vector whose output angle vector is different from the output voltage vector before switching by π [rad] or which is different from the output voltage vector after switching by π [rad] Time output,
Then, the inverter control device outputs the switched output voltage vector, and the inverter charges the power storage device with regenerative energy generated from the electric motor when the reverse voltage vector is output. With such a configuration, when the inverter circuit and the inverter control device of the present invention are used as a drive device for an electric motor of a vehicle device, a soft switching operation using time-division PWM control by a space vector is realized by using a combination of non-zero vectors. Since the regenerative energy is charged to the power storage device when the reverse voltage vector is output in the switching operation, the electric motor of the vehicle device can be efficiently driven.

According to an eighth aspect of the invention, in the vehicle apparatus according to the seventh aspect, the inverter uses a reverse voltage vector when the regenerative energy generated from the electric motor is charged into the power storage device as a reverse voltage vector when the electric motor is driven. It is characterized by outputting for a longer time than the vector. With such a configuration, it is possible to efficiently acquire regenerative energy and charge the power storage device.

[0019]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a circuit diagram showing an embodiment of the present invention. In FIG. 1, an inverter circuit controlled by the inverter control device according to the present embodiment has, for example, an IGBTQ to which a three-phase motor 1 is connected as a load.
A main switching circuit 2A that constitutes an inverter unit using 1 to Q6 as main switching elements, and an IGB, for example.
It is composed of an auxiliary switching element that uses TQ7 to Q12 as a unidirectional switching element and an auxiliary switching circuit 2B that constitutes a resonance section composed of a resonance inductance Lr. In addition, as a switching element, Q1 to Q1
The element used for 2 is not limited to an IGBT, but a reverse blocking thyristor, a GTO (Gate Turn Off Thyristor), a bipolar transistor, a MOSFET, or the like may be used, and an induction motor or a DC brushless motor is used for the motor 1.

The main switching circuit 2A has IGBTs Q1 to Q6 connected to both ends of a DC power source 3 in a three-phase bridge structure composed of U-phase, V-phase and W-phase, and has collector terminals of each IGBT. Between the emitter terminals, the commutation diodes D1 to D6 are connected to the collector terminals of the IGBT in order to circulate the regenerative energy generated by the inductive load such as the motor 1 and the current energy stored in the inductive load. The anode terminal of the current diode, the emitter terminal of the IGBT and the cathode terminal of the commutation diode are connected to each other. Further, between the collector terminal and the emitter terminal of each IGBT, there are also buffer capacitors C1 to C6 for absorbing a surge voltage applied between the collector terminal and the emitter terminal of the IGBT when the IGBT is turned on or turned off. Connected.

I in the main switching circuit 2A
The emitter terminal of the GBQQ1 and the collector terminal of the IGBTQ2, the emitter terminal of the IGBTQ3 and the collector terminal of the IGBTQ4, the emitter terminal of the IGBTQ5 and the IGBTQ6.
Each of the connection points of the collector terminals of the motors is a three-phase output terminal of U phase, V phase, W phase of the inverter circuit of the present embodiment, and each terminal of U phase, V phase, W phase of the motor 1 is The main switching circuit 2A is connected to each other, and the auxiliary switching circuit 2B is connected to the three-phase output terminal of the main switching circuit 2A. The auxiliary switching circuit 2B has an IGBTQ at both ends of a resonance inductance Lr for forming a resonance circuit together with the buffer capacitors C1 to C6 used in the main switching circuit.
A circuit in which a bridge circuit 2B1 in which 7 to Q12 are connected to a three-phase bridge structure composed of a U ′ phase, a V ′ phase, and a W ′ phase is connected (in other words, a connection point between three sets of auxiliary switching elements of the bridge circuit 2B1 (Where the resonance inductance Lr is connected to the opposite side), the IGBTs Q7, Q9,
On the collector terminal side of Q11, protection diodes D7 and D
9, D11 are connected in series so that the collector terminal of the IGBT and the anode terminal of the protection diode are connected. Similarly, the protection diodes D8, D10, D1 are connected to the emitter terminals of the IGBTs Q8, Q10, Q12.
2 are connected in series so that the emitter terminal of the IGBT and the cathode terminal of the protection diode are connected.

The connection between the three-phase output terminal of the main switching circuit 2A and the auxiliary switching circuit 2B is such that the U-phase terminal of the three-phase output terminal and U'of the auxiliary switching circuit 2B.
The IGBT Q7 and the IGBT Q8, which are the connection points of the phases, are connected to each other, and similarly, the V phase terminal of the three-phase output terminal and the V'phase of the auxiliary switching circuit 2B are the connection points of the IGBTs.
The connection point of TQ9 and IGBTQ10, the W-phase terminal of the three-phase output terminal, and the connection point of IGBTQ11 and IGBTQ12, which is the connection point of W'phase of the auxiliary switching circuit 2B, are connected to each other. A smoothing capacitor C9 is connected to the DC power supply 3.

The above is the configuration of the inverter circuit in which the resonance inductance for soft switching is omitted from the conventional three for three phases to one, and the inverter control device of the present embodiment has the above-mentioned inverter circuit. In order to control the rotation speed, a rotation sensor 4 for measuring the rotation angle (rotational position) and the rotation speed of the motor 1 connected to the inverter circuit, and the rotation angle and the rotation speed Ps of the motor 1 measured by the rotation sensor 4.
And a three-phase control signal (PWM signal) by a time-division PWM control method using a space vector according to the output control signal Os.
And a U output from the control CPU 5
Each switching element IGBTQ1 to Q12 of the inverter circuit is based on the three-phase control signal composed of s, Vs, and Ws.
Drive signal generating means 6 for generating the switching control signals S1 to S12, and a resonance current for supplying the drive signal generating means 6 with the turn-off timing of each switching element IGBTQ1 to Q6 of the main switching circuit 2A of the inverter circuit. Similarly to the arrival determination means 7, a zero voltage detection means 8 for supplying the drive signal generation means 6 with the turn-on timing of each of the switching elements IGBTQ1 to Q6 of the main switching circuit 2A of the inverter circuit,
Further, it is composed of a drive circuit 9 which converts the switching control signals S1 to S12 of the switching elements IGBTQ1 to Q12 of the inverter circuit output from the drive signal generating means 6 into drive signals for driving the IGBT.

Further, since the current flowing through the load of the inverter circuit is measured and notified to the resonance current arrival determination means 7, the current sensor Is1 is provided between the U-phase terminal of the motor 1, which is the load, and the U-phase output terminal of the inverter circuit. The measured value I1 of the current sensor Is1 is input to the resonance current arrival determination means 7. Similarly, the current sensors Is2 and Is3 are respectively connected between the V phase terminal and the V phase output terminal of the motor 1 which is a load,
The measured value I2 of the current sensor Is2 and the measured value I3 of the current sensor Is3 are inserted between the W-phase terminal and the W-phase output terminal, respectively, and are input to the resonance current arrival determination means 7. Further, since the current flowing through the resonance inductance Lr of the auxiliary switching circuit 2B of the inverter circuit is measured and notified, the current sensor Is4 causes the resonance inductance Lr to be measured.
The measured value I4 of the current sensor Is4 is input to the resonance current arrival determination means 7. Further, in order to notify the zero voltage detecting means 8 of the operating state of the main switching element of the inverter circuit, a collector terminal is provided between the collector terminals and the emitter terminals of the IGBTs Q1 to Q6 forming the main switching circuit 2A of the inverter circuit. Voltage sensor V for measuring and notifying the voltage between emitter terminals
s1 to Vs6 are connected, and the measured values V1 to V6 of the voltage sensors Vs1 to Vs6 are input to the zero voltage detecting means 8.

The resonance current arrival judging means 7 confirms that the inductance current I4 flowing through the resonance inductance Lr becomes larger than the current showing the maximum value among the input load currents I1, I2 and I3. The drive signal generating means is notified as the instruction value I. Further, the currents I1, I2, and I3 flowing through the three-phase load change the direction of current flow, and the signs of the indicated values of the current sensors Is1, Is2, and Is3 output both positive and negative values. The resonance current arrival determination means 7 takes the absolute values of the input I1, I2, I3 and compares them with the inductance current I4. Further, the zero voltage detecting means 8 indicates that the voltage becomes zero at each voltage value of the input voltage sensors V1 to V6 by Z
The drive signal generation means 6 is notified by each instruction value of 1 to Z6. The detailed operation of each unit will be described later.

Further, the driving circuit 9 outputs the control signal for controlling each switching element of the inverter circuit to the switching control output from the driving signal generating means 6 in order to actually drive (drive) the control terminal of the switching element. The switching elements that make up the inverter circuit
In the case of current drive such as reverse blocking thyristor, GTO, and bipolar transistor, the control terminal of the switching element is converted into a current that can be sufficiently driven, and similarly, the switching control signal output from the drive signal generating means 6 is converted into an inverter. The switching element that constitutes the circuit is an IGBT
In the case of voltage drive such as a MOSFET or MOSFET, the voltage is converted into a voltage that can sufficiently drive the control terminal of the switching element.

Next, the operation of the inverter circuit of this embodiment will be described with reference to the drawings. In explaining the operation of the circuit, the voltage and current of each part in the circuit diagram of FIG.
The notation of ON / OFF of each switching element is defined first. First, regarding the voltage and current of each part, (1) the voltage with the collector side of Q1 applied to both ends of the parallel circuit of the IGBT Q1, the commutation diode D1, and the buffer capacitor C1 in the positive direction is the same as V1, and (2) Q2 , Q2 added to both ends of the parallel circuit of D2, C2
A voltage with the collector side of V3 in the positive direction is applied to both ends of the parallel circuit of V2 (3) Q3, D3, and C3.
A voltage with the collector side of V4 as the positive direction is applied to both ends of the parallel circuit of V3 (4) Q4, D4, and C4.
A voltage with the collector side of V5 in the positive direction is applied to both ends of the parallel circuit of V4 (5) Q5, D5, and C5.
A voltage with the collector side of V6 in the positive direction is applied to both ends of the parallel circuit of V5 (6) Q6, D6, and C6.
A voltage whose positive side is the collector side of is defined as V6. Furthermore, three-phase currents I1 and I flowing through the load
2 and I3 are defined as the direction in which the load flows.

Further, ON / OFF of the IGBTs Q1 to Q6
For the definition of, the state in which the IGBT Q1 on the upper side (plus side) of the U phase of the main switching circuit 2A is ON and the IGBTQ2 on the lower side (minus side) is OFF is "1", and the IGBT Q1 on the upper side of the U phase is OFF. And the lower IGBTQ
When 2 is ON, “0” is indicated. Similarly, when the IGBT Q3 on the upper side of the V phase is ON and the IGBTQ4 on the lower side is OFF, it is “1”, and the IGBTQ3 on the upper side of the V phase is OFF and the lower side IGBTQ3 is The state in which the IGBT Q4 is ON is set to "0". W
The upper phase IGBTQ5 is ON and the lower phase IGBTQ
6 is OFF, "1", WQ phase upper side IGBTQ
When 5 is OFF and the lower IGBT Q6 is ON "
0 ". Also, I on the upper side of the U'phase of the auxiliary circuit 2B
The state where the IGBTQ7 is ON and the lower IGBTQ8 is OFF is "1", and the lower IGBTQ8 of the U'phase is ON and the upper IGBTQ7 is OFF is "0". Similarly, the V'phase is the upper stage. Side IGBT Q9 is ON and the lower side I
When the IGBTQ10 is OFF, the IGBTQ10 on the lower stage side of the V'phase is ON and the IGBTQ11 on the upper stage side is O.
The state of FF is set to "0". W'phase is also the upper IGBT
When Q11 is ON and the lower IGBT Q12 is OFF, the state is "1", and when the W'phase lower IGBT Q12 is ON and the upper IGBT Q11 is OFF, the state is "0".
Therefore, for example, when the three-phase control signal (Us, Vs, Ws) output by the control CPU 5 is expressed as (1, 0, 0): output voltage vector V1, the IGBTQ1 is ON, and the IGBT is ON.
Q2 is OFF, IGBTQ3 is OFF, IGBTQ4 is ON, IGBTQ5 is OFF, and IGBTQ6 is ON.

Further, the switching control signal for each switching element of the inverter circuit output from the drive signal generating means 6 turns off the logical value "0" and turns on "1". Further, the operation shown in FIG. 2 is performed by setting the three-phase control signals (Us, Vs, Ws) to (1, 0, 0): output voltage vector V1 as an example for explaining the control mode of the inverter circuit of the present embodiment. -> (0, 0, 1): Output voltage vector V
4-> (1,1,0): The case of controlling with the output voltage vector V3 will be described. The operation of the circuit is the same for controls other than the above.

Next, based on the definitions of voltage and current of each portion and ON / OFF of each switching element defined above, the inverter control device of the present embodiment will be described with reference to FIGS. 2 and 3 to 6. The operation will be described. First, at time t1,
Since the steady state of (U, V, W) = (1, 0, 0), as shown in (a) Mode 1, the DC power supply 3 to the IG
The current that has flowed toward the U-phase terminal of the motor 1 via the BTQ1 is IG from the V-phase terminal and the W-phase terminal of the motor 1, respectively.
It flows through BTQ4 and Q6 and returns to the DC power supply 3. In the steady state of mode 1, the upper side (plus side) switching elements IGBTQ7, Q9, Q12 of the auxiliary circuit 2B are in the ON state, and the lower side (minus side) switching elements IGBTQ8, Q10, Q11 are in the OFF state. But
Since no energy is stored in the resonance inductance Lr, no current flows in the resonance inductance Lr.

This (Us, Vs, Ws) = (1, 0,
0): From the steady state of the output voltage vector V1, the control CP
When U5 changes the three-phase control signal to the state of (Us, Vs, Ws) = (0, 0, 1): output voltage vector V4, as shown in (b) mode 2, the drive signal generation means 6
Are the IGBTs Q8 and Q11 of the auxiliary switching circuit 2B.
The switching control signals S8 and S11 for the above are switched from "0" to "1" to turn on the IGBTs Q8 and Q11. At this time, from the IGBT Q1 to the motor 1
A part of the current flowing to the U-phase terminal flows through the resonance inductance Lr and returns to the DC power supply 3 via the IGBTs Q4 and Q6, and energy having the current ILr as an initial current is accumulated in the resonance inductance Lr.

When the IGBTs Q8 and Q11 are turned on and the inductance current I4 begins to flow, and the inductance current I4 becomes larger than the maximum absolute value of the load current (I1 in the example of FIG. 2) at the time t2, the resonance current arrival determination is made. The output I of the means 7 changes from the logical value "0" to "1", and in response to this, the drive signal generating means 6
(C) As shown in mode 3, the main switching circuit 2A
Switching control signal S for the IGBTs Q1 and Q6 of
1 and S6 are switched from the logical value "1" to "0", and the IGB
Turn off TQ1 and Q6. At this time, IGBTQ
In 1 and Q6, the voltages V1 and V6 between the collector and emitter terminals of the IGBTs Q1 and Q6 cannot be rapidly increased due to the time constants of the buffer capacitors C1 and C6, so that ZVS in the IGBTs Q1 and Q6 is realized. .

The IGBTs Q1 and Q6 are turned off.
Then, along with charging the buffer capacitors C1 and C6,
The voltages V2 and V5 across the buffer capacitors C2 and C5, which have been supplied with a voltage close to the power supply voltage VB, start discharging the buffer capacitors C2 and C5 by connecting the buffer capacitors C1 and C6, and To descend. The charging currents of the buffer capacitors C1 and C6 and the discharging currents of C2 and C5 flow as resonance currents through the resonance inductance Lr and become a resonance mode in which they circulate in the circuit. Further, when this resonance mode is continued, the resonance current further flows due to the energy accumulated in the resonance inductance Lr, and the voltages V2 and V5 across the buffer capacitors C2 and C5 are applied.
Energy becomes almost "zero", the energy stored in the resonance inductance Lr flows through the commutation diodes D2 and D5.

Next, at time t3, voltage sensors Vs2 and Vs5 for measuring the voltage between the collector and emitter terminals of the IGBTs Q2 and Q5 of the main switching circuit 2A.
Detects that the voltage between the collector terminal and the emitter terminal of the IGBTs Q2 and Q5 has become "zero", and the outputs Z2 and Z5 of the zero voltage detection means 8 are output from the logical values "0" to "1", respectively. Switch. In response to this, the drive signal generating means 6 switches the switching control signals S2 and S5 for the IGBTs Q2 and Q5 of the main switching circuit 2A from the logical value "0" to "1" to turn ON the IGBTs Q2 and Q5, Us, Vs, Ws) = (0, 0,
1): The output voltage vector V4 shifts to the regeneration mode shown in (d) mode 4. At this time, in the IGBTs Q2 and Q5, the voltages V2 and V5 between the collector terminals and the emitter terminals of the IGBTs Q2 and Q5 are "zero", and the currents are flowing through the commutation diodes D2 and D5, respectively. ZVS and ZCS in Q5 are realized.

In the regenerative mode of the mode 4, the regenerative energy of the motor 1 and the energy stored in the resonance inductance Lr are applied from the W-phase terminal of the motor 1 to the IGBT.
The regenerative current flowing to the positive side of the DC power supply 3 via the TQ5, the regenerative current flowing from the V-phase terminal of the motor 1 to the negative side of the DC power supply 3 via the IGBTQ4, the IGBTQ2 flowing to the U-phase terminal of the motor 1. Regenerative current, and further I
GBTQ8, resonance inductance Lr, IGBTQ1
A current flowing with 1 is generated. However, since the power supply voltage of the DC power supply 3 is applied as a reverse voltage to the resonance inductance Lr in the direction of decreasing the current ILr, the current ILr is reduced.
Gradually decreases to zero. When the current ILr becomes zero, the protection diodes D8 and D11 block the currents flowing to the emitter side to the IGBTs Q8 and Q11 due to the power supply voltage of the DC power supply 3, and shift to the steady mode (e) mode 5.

Next, as in the above-described operation, at time t4, (Us, Vs, Ws) = (0, 0, 1): From the steady state of the output voltage vector V4, the control CPU 5 (U
s, Vs, Ws) = (1,1,0): When the three-phase control signal is changed to the state of the output voltage vector V3, (f) the drive signal generation means 6 is an auxiliary device as shown in mode 6. The logic values of the switching control signals S7, S9 and S12 for the IGBTs Q7, Q9 and Q12 of the switching circuit 2B
Switching from 0 "to" 1 ", IGBTs Q7, Q9 and Q1
Turn 2 on. The IGBTs Q7, Q9 and Q12 are turned on, and the inductance current I4 flows out,
At time t5, when the inductance current I4 becomes larger than the maximum value of the absolute value of the load current (I1 in the example of FIG. 2) and the state of (g) mode 7 is reached, the output I of the resonance current arrival determination means 7 is a logical value " The output is switched from 0 "to" 1 ", and in response to this, the drive signal generation means 6 sets the switching control signals S2, S4 and S5 for the IGBTs Q2, Q4 and Q5 of the main switching circuit 2A from" 1 "to" logical value ". Switching to 0 ", the IGBTs Q2, Q4 and Q5 are turned off (h) and the mode 8 is entered. At this time, IGB
For TQ2, Q4 and Q5, buffer capacitors C2 and C4
IGBTs Q2 and Q4 depend on the time constants of C5 and C5.
And V5 between collector and emitter terminals V2, V
4 and V5 cannot rise rapidly, so IGBTQ
ZVS in 2 and Q4 and Q5 is realized.

When the IGBTs Q2, Q4, and Q5 are turned off, the buffer capacitors C2, C4, and C5 are charged, and the buffer capacitors C1, C3, and C6, to which a voltage close to the power supply voltage VB has been applied, have been applied. Both end voltage V1
And V3 and V6 are connected to the buffer capacitors C1, C3 and C5 by connecting the buffer capacitors C2, C4 and C5.
The discharge of 6 begins and therefore drops. The charging current of these buffer capacitors C2, C4, C5 and C1, C3 and C
The discharge current of No. 6 flows in the resonance inductance Lr as a resonance current and becomes a resonance mode in which it circulates in the circuit. Further, when this resonance mode is continued, the resonance current further flows due to the energy accumulated in the resonance inductance Lr, and the voltage V1 across the buffer capacitors C1, C3, and C6.
When V3 and V6 become "zero", the energy stored in the resonance inductance Lr flows through the commutation diodes D1, D3 and D6 as shown in (i) Mode 9. .

Next, at time t6, voltage sensors Vs1 and Vs for measuring the voltage between the collector and emitter terminals of the IGBTs Q1, Q3 and Q6 of the main switching circuit 2A.
s3 and Vs6 detect that the voltage between the collector terminal and the emitter terminal of the IGBTQ1, Q3 and Q6 has become "zero", and the outputs Z1, Z3 and Z6 of the zero voltage detecting means 8 are detected.
, Respectively, switch the output from the logical value "0" to "1". In response to this, the drive signal generating means 6 sets the switching control signals S1, S3 and S6 for the IGBTs Q1, Q3 and Q6 of the main switching circuit 2A to the logical value "1" to turn on the IGBTs Q1, Q3 and Q6, (U
s, Vs, Ws) = (1,1,0): The output voltage vector V3 shifts to the steady state shown in the (j) mode 10. At this time, in the IGBTs Q1, Q3 and Q6, the voltage V between the collector and emitter terminals of the IGBTs Q1, Q3 and Q6 is
1, V3, and V6 are "zero", and the energy stored in the inductance is commutation diodes D1 and D3.
No current flows in the IGBTs Q1 and Q3 and Q6 because they are flowing as currents in D6 and D6, respectively.
S and ZCS are realized.

As described above, in the inverter control device of the present embodiment, (Us, Vs, Ws) from the time t1 to the time t6 shown in FIG. 2 is (1, 0,
0): output voltage vector V1-> (0, 0, 1): output voltage vector V4-> (1, 1, 0): the operation of the inverter circuit when controlling with the output voltage vector V3 has been described. In performing the space vector PWM control on the circuit, the operation of the inverter circuit at the transition between other control vectors is also the above (Us, Vs, Ws) (1,
This is the same as the case of controlling 0,0)->(0,0,1)-> (1,1,0).

It should be noted that the main switching element is electrically connected only in the direction in which a current is passed to each connection point between the main switching elements, and the first, second and third upper stages corresponding to the respective phases of the three-phase bridge are connected. Switching element (IGBTQ1, Q
3, Q5) and the fourth, fifth, and sixth switching elements (IGBTQ2) on the lower side corresponding to each phase of the three-phase bridge, which conducts only in the direction in which current flows from each connection point between the main switching elements. , Q4, Q6), and the auxiliary switching elements are connected to the three-phase output terminals that conduct only in the direction in which the current flows to the connection points of the auxiliary switching elements, respectively 9 switching elements (IGBT Q7, Q9, Q11) and the 10th, 11th, and 11th lower-stage sides respectively connected to the three-phase output terminals that conduct only in the direction in which current flows from each connection point between the auxiliary switching elements. 12 switching elements (I
When classified into GBQQ8, Q10, Q12), the drive signal generation means 6 is a three-phase instructing output of a switching control signal for conducting the first, second and third switching elements (IGBTQ1, Q3, Q5). In synchronization with the control signal,
The seventh, eighth, and ninth switching elements (IGBTQ
7, Q9, Q11) to turn on the switching control signal, and the fourth, fifth, and sixth switching elements (IG
BTQ2, Q4, Q6) in synchronization with the three-phase control signal for instructing the output of a switching control signal to turn on the tenth, eleventh, and twelfth switching elements (IGBs).
A switching control signal for making TQ8, Q10, Q12) conductive is output.

Next, a description will be given of a control method for realizing soft switching of an inverter circuit in all combinations of outputs of output voltage vectors when performing time-division PWM control by a space vector in an inverter circuit, with reference to the drawings. To do. First, the IGBTs Q1, Q3, Q5
Is all conductive, or the IGBTs Q2, Q4, Q6 are all conductive, no current can flow through the resonance inductance Lr, and soft switching cannot be performed. Therefore, as shown in FIG. 7A, the three-phase control signals (Us, V
s, Ws) can be output based on whether the control signals Us, Vs, W
If each s is represented by a logical value "0" or "1", U
It is output only when the exclusive OR of s, Vs, and Ws has a logical value of "1".

In order to perform soft switching, two phases, that is, a phase for supplying a current to the resonance inductance Lr and a phase for discharging the current are required. Therefore, a state transition of two or more phases must occur. Of the three phases, state transitions between vectors whose states change only in one phase are excluded. Therefore,
The transition of the three-phase control signals (Us, Vs, Ws) is shown in FIG.
As shown in (b), when the control signals Us, Vs, and Ws are represented by logical values “0” and “1”, respectively, Us and V
The transition can be made only when the logical sum of s and Ws is the same. For example, three-phase control signal (1) (0, 0, 1): output voltage vector V4 (2) (1, 0, 0): output voltage vector V1 (3) (0, 1, 0): output voltage vector V2 can transit between each other, and similarly, three-phase control signal (4) (0, 1, 1): output voltage vector V6 (5) (1, 1, 0): output voltage vector V3 ( 6) (1, 0, 1): The output voltage vector V5 can transit between each other.

Three-phase control signals (Us, Vs, Ws)
The transition of the control signal Us, as shown in FIG.
When Vs and Ws are represented by logical values “0” and “1”, it is possible to transit to signals represented by logical inversion of Us, Vs, and Ws. For example, three-phase control signal (a) (0, 0, 1): output voltage vector V4 and (1, 1, 0): output voltage vector V3 (b) (0, 1, 0): output voltage vector V2 (1, 0, 1): output voltage vector V5 (c) (1, 0, 0): output voltage vector V1 and (0, 1, 1): output voltage vector V6 can transit between each other. is there.

Therefore, the control CPU 5 of the inverter circuit shown in FIG. 1 outputs the zero vector of the three-phase control signal (output voltage vector) which is restricted when performing the time division PWM control by the space vector as described above. It is not possible, or solves the control condition that consecutive adjacent vectors cannot be output, and when expressing a zero vector with a combination of non-zero vectors or when outputting adjacent vectors, either one of the adjacent vectors By outputting the reverse voltage vector and then transiting to the adjacent vector, the same control as outputting the zero vector or continuously outputting the adjacent vector is performed.

Next, an example of the control of each output voltage vector by the control CPU 5 will be specifically described with reference to FIG. Note that, in FIG. 8 as well, as in FIG. 17, the area elapsed time of the time division PWM control by the space vector is described as being four times the sampling time Ts. However, in reality, the sampling time Ts is shortened and the area elapsed time is reduced. The number of times of switching within is controlled to be large. In FIG. 8, when the rotation angle θ of the motor exists in the area 1 of the space vector shown in FIG. 16, the control CPU 5 first sets the output voltage vector V1 at time (T1) at time t00.
Then, at time t01, the output voltage vector V4 is output.
Transition to and output time (T3 / 4), and time t02
At the output voltage vector V3 at time (T2 +
T3 / 2) Output. After outputting the output voltage vector V3, the output voltage vector V4 again transits to the output voltage vector V4 at time t03 and outputs for a time (T3 / 4). After that, the output voltage vectors V1, V4, V3, and V4 are output in this order. .

Similarly, the rotation angle θ of the motor is shown in FIG.
When it exists in the area 2 of the space vector shown in 6, the output voltage vector V3 is first output, then the output voltage vector V5, and then the output voltage vector V2. After the output voltage vector V2 is output, the output voltage vector V5 is output again, and thereafter the output voltage vectors V3, V
Repeat in the order of 5, V2 and V5 output.

As described above, when the control CPU 5 of the inverter control device of the present embodiment wants to make a transition between the output voltage vectors V1 and V3 which are adjacent vectors, the angle with the output voltage vector V3 after the transition is π. [Ra
d] This embodiment reduces the number of resonance inductances for soft switching by outputting different output voltage vectors V4, which are reverse voltage vectors of the output voltage vector V3, between the output voltage vectors V1 and V3. In the inverter circuit of the form, since the simultaneous switching of two or more phases always occurs in the soft switching operation,
When performing time-division PWM control with a space vector, solve the control condition of a three-phase control signal (output voltage vector), which is constrained to be unable to output adjacent vectors in succession, and output adjacent vectors in succession. Performs control equivalent to.

Further, for example, output voltage vectors V3 and V4
Is continuously output, the same effect as the output of the zero vector is obtained. For example, in the example shown in FIG. 8, since the zero vector is a value obtained by subtracting the output voltage vector V4 output time from the output voltage vector V3 output time, T2 + T3 / 2− (T3 / 4 + T3 / 4) = T2, By changing each time depending on the position of the rotation angle θ, PWM control becomes possible. Therefore, similar to the above-mentioned output of adjacent vectors, when performing the time-division PWM control by the space vector, the control condition of the three-phase control signal (output voltage vector) subject to the constraint that the zero vector cannot be output is solved, Performs the same control as outputting a zero vector.

Incidentally, for example, when it is desired to make a transition between the adjacent output voltage vectors V1 and V3, the output voltage vector V4, which is the reverse voltage vector of the output voltage vector V3 after the transition, is output. An output voltage vector V6 that is an inverse voltage vector of the output voltage vector V1 before the transition may be output as the output voltage vector output for a predetermined time between the vectors V1 and V3. In this case, the transition of the output voltage vector is V1->V6-> V
3-> V6, and this is repeated in area 1.

When such output voltage vector control is performed, the switching control of the switching element by the three-phase control signals (Us, Vs, Ws) is performed as shown in FIG. Two-phase simultaneous switching when changing from V1 (1, 0, 0) to output voltage vector V4 (0, 0, 1) (2) Output voltage vector V4 (0, 0, 1) to output voltage vector V3 (1, 3-phase simultaneous switching when changing to 1, 0) (3) 3-phase simultaneous switching (4) output when changing from output voltage vector V3 (1, 1, 0) to output voltage vector V4 (0, 0, 1) When the voltage vector V4 (0, 0, 1) changes to the output voltage vector V1 (1, 0, 0), two-phase or three-phase simultaneous switching always occurs, such as two-phase simultaneous switching.

It is generally known that in the inverter circuit, when PWM control is performed, the efficiency is highest when the duty ratio is 100% because the switching loss is eliminated. However, the conventional time-division PWM
In the control, the output waveform is controlled to output one of the vectors sandwiching the area within one sampling so that the shaping of the output waveform does not become uncontrollable due to the saturation of the PWM duty ratio. For this reason, 100% duty ratio control is not taken into consideration. Further, it is possible to ignore the shaping control of the AC output waveform and perform the control for enabling the time ratio saturation, but the processing time of the control CPU 5 becomes long because of the process for the time ratio saturation, and the sampling time An increase is possible and the PWM frequency may be in the audible range. Therefore, in order to solve this problem, it is preferable that the control CPU 5 further improve the PWM output by time division so as to perform control without increasing efficiency and increasing the sampling time Ts.

Hereinafter, a method of controlling the output voltage vector for suppressing the increase of the sampling time Ts of the control CPU 5 will be specifically described with reference to FIG. Figure 10 (a)
Shows a region in which the six regions of area 1 to area 6 represented by the space vector of the conventional time division PWM control shown in FIG. 16 are redefined in accordance with the new control method. Is an area starting from a point which is advanced by a predetermined angle φ in the rotation direction of the motor from the conventional space vector starting point. In addition, FIG.
FIG. 4 is a diagram showing a method of controlling the output voltage vector in the advance angle region, in which the control CPU 5 starts the area 1 of a new region starting from the point where the rotation angle θ of the motor is advanced by the above-mentioned predetermined angle φ. ', The rotation angle θ of the motor is
While in the new area area 1 ', the output voltage vector V3 existing in the new area area 1'is continuously output.

Such control is shown in FIG.
FIG. 11 shows an example of the case where the inverter circuit of this embodiment is combined with the control for solving the control condition of the three-phase control signal (output voltage vector) which is restricted when performing the time division PWM control by the space vector. And shown in FIG.

FIG. 11 shows that, for example, when the rotation angle θ of the motor moves from the area 6'of a new area to the area 1 ',
At time t06, which is a predetermined time before the boundary between the area 6'and the area 1 ', the output voltage vector V1 in the area 6'which has been output until then is the reverse voltage vector of the output voltage vector V3 in the next area 1'. After transitioning to V4 and outputting the output voltage vector V4 for a predetermined time, at time t07 which is the boundary between the area 6 ′ and the area 1 ′, the transition is made to the output voltage vector V3 in the next area 1 ′.

That is, in the control as shown in FIG.
In the control such that when the output voltage vector is switched, the reverse voltage vector V6 whose angle is π [rad] different from the output voltage vector V1 before switching is output for a predetermined time, and then the output voltage vector V3 after switching is output. The output of the reverse voltage vector V6 is controlled during the output voltage vector V1 before switching, which is a predetermined time before the switching of the new region. Although the above-mentioned transition of the output voltage vector is V1->V4-> V3, the inverse voltage vector V6 of the output voltage V1 may be output between the output voltage vectors V1 and V3.

FIG. 12 shows that, for example, when the rotation angle θ of the motor moves from the area 6'of a new area to the area 1 ',
At time t09 at the boundary between the area 6 ′ and the area 1 ′, the output voltage vector V1 in the area 6 ′ that has been output until then transits to the reverse voltage vector V4 of the output voltage vector V3 in the next area 1 ′, After outputting the output voltage vector V4 for a predetermined time, at time t10, the output voltage vector V4 transits to the output voltage vector V3 in the next area 1 ′.

That is, in the control shown in FIG.
When the output voltage vector is switched, a new region is output in the control in which an inverse voltage vector whose angle is π [rad] different from the output voltage vector V3 after switching is output for a predetermined time, and then the output voltage vector after switching is output. The reverse voltage vector is output only for a predetermined time from the switching point of. The transition of the output voltage vector described above was also V1->V4-> V3, but the output voltage vector V
The reverse voltage vector V6 of the output voltage V1 may be output between 1 and V3. The predetermined angle φ of the advance angle region is, for example, 25 ° to 35 ° in normal operation. When the predetermined angle φ is, for example, about 15 °, field weakening control is performed, and the predetermined angle φ is, for example, about 45 °. Then, the retard angle energization control is performed.

The control using the advance angle region in which the efficiency is not improved and the sampling time Ts is not increased by continuously outputting the output voltage vector existing in the area of the above-mentioned new region is the rotation of the motor. It is advisable to control the output voltage vector by the advance angle region control when the number of revolutions is equal to or higher than a predetermined value. In this way, the motor can be rotated more stably and efficiently by performing the control in the advance angle region after the motor rotation operation is stable when the rotation speed of the motor is equal to or higher than the predetermined rotation speed. .

In the above-described inverter control device of the present embodiment, by controlling the reverse voltage vector output time at the switching timing for switching each of the three phases, an induction motor having a squirrel-cage rotor as a load or When driving a motor capable of generating electricity, such as a DC brushless motor having a permanent magnet rotor, it is possible to control regenerative energy in the motor, and thereby control charging of the power storage device. 13A shows the three-phase motor current during motor driving, FIG. 13B shows the power supply current during motor driving, and FIG. 13C shows the state of the driving signal during motor driving. Each is shown, and according to FIG. 13 (b),
It can be seen that the power supply current is flowing in the positive direction (from the power supply to the motor) and the motor is being driven.

Further, FIG. 14A shows the three-phase motor current during regeneration (charge of the power storage device), FIG. 14B shows the power supply current during regeneration, and FIG. 14C shows the time during regeneration. 14B respectively show the states of the drive signals of the above, and according to FIG. 14B, the power supply current is flowing in the negative direction (from the motor to the power supply), and the power storage device is being charged. In addition, FIG.
By comparing (c) and FIG. 14 (c), regenerative energy is generated by making the output time of the reverse voltage vector represented by the combination of non-zero vectors longer during regeneration than during motor driving. I understand that That is, in the inverter circuit of the present embodiment, when performing the time-division PWM control by the space vector, the three-phase control signal (output In order to solve the control condition of (voltage vector), when outputting a zero vector represented by a combination of non-zero vectors, it is possible to recover regenerative energy efficiently only by controlling the output time of the reverse voltage vector described above. You can

Therefore, by using the inverter circuit shown in this embodiment, the inverter control device for controlling the inverter circuit, the motor capable of generating power, and the power storage device as the power source, the electric motor can be efficiently driven and the power storage device can be operated. EVs (Electric Vehicles) and HEs that run while charging
It is possible to realize a vehicle device such as a V (Hybrid Electric Vehicles).

[0062]

As described above, according to the present invention, in a control device for controlling an inverter circuit in which the number of resonance inductances is reduced as compared with the prior art, simultaneous switching of two or more phases always occurs in soft switching operation. Solved the control condition of the three-phase control signal (output voltage vector) that is restricted when performing the time-division PWM control by the space vector, that is, the zero vector cannot be output or the adjacent vector cannot be output continuously. When outputting the matching output voltage vector, output the zero voltage vector or continuously output the adjacent vector by outputting the reverse voltage vector of either one of the adjacent vectors and then transiting to the adjacent vector. It is possible to perform the same control as that performed.

Further, the maximum output is increased as compared with the case where the PWM control is always performed by performing the control of continuously outputting the constant output voltage vector in the advance angle region when the rotation number of the motor is stable or more. Moreover, since the number of times of switching can be significantly reduced, the effect of improving the efficiency of the inverter circuit can be obtained.

Further, the control device of the present embodiment, when switching the phases of the three-phase motor,
It is possible to adjust the output of the inverter circuit by adjusting the ratio of the output time for outputting the reverse voltage vector of either of them and the output voltage. Further, by making the output time of the reverse voltage vector long, the effect that the regenerative current can flow can be obtained. Therefore, an electric vehicle (EV) driven by a motor or a hybrid vehicle (HEV) driven by a combination of an engine and a motor
When the inverter circuit and the inverter control device of the present embodiment are used in a vehicle device that uses a motor to drive and drive, switching between driving and regeneration can be realized only by adjusting the PWM control duty ratio. is there. Particularly in the HEV, it is possible to realize the zero power control that does not perform regeneration or driving when the vehicle is in a cruise state only by controlling the reverse voltage vector output time. For example, the duty ratio is 50% to 6%.
By adjusting it to 0%, it is possible to prevent regeneration and driving.

Therefore, it is not necessary to perform complicated control, and the control circuit can be simplified and the cost can be reduced.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a waveform diagram showing changes in the waveform of each portion with respect to time of the inverter control device of the same embodiment.

FIG. 3 is a diagram showing an operation in each mode of the inverter circuit according to the same embodiment.

FIG. 4 is a diagram showing an operation of each mode of the inverter circuit according to the same embodiment.

FIG. 5 is a diagram showing an operation of each mode of the inverter circuit according to the same embodiment.

FIG. 6 is a diagram showing an operation in each mode of the inverter circuit according to the same embodiment.

FIG. 7 is a diagram showing a state transition of a three-phase control signal of the inverter control device of the embodiment.

FIG. 8 is a diagram showing a time relationship of control signals of the inverter control device of the embodiment.

FIG. 9 is a diagram showing an output voltage vector and a state transition of each phase in the control of the inverter control device of the embodiment.

FIG. 10 is a diagram showing a new control method of an output voltage vector defining a new region of the inverter control device of the embodiment.

FIG. 11 is a diagram showing an example of output voltage vector switching timing in the new control method for the inverter control device of the embodiment.

FIG. 12 is a diagram showing an example of output voltage vector switching timing in the new control method for the inverter control device of the embodiment.

FIG. 13 is a diagram showing states of a motor current, a power supply current, and a drive signal during driving when a load is a motor in the inverter control device of the embodiment.

FIG. 14 is a diagram showing states of a motor current, a power supply current, and a drive signal at the time of regeneration when the load is a motor in the inverter control device of the embodiment.

FIG. 15 is a circuit diagram showing a conventional soft switching inverter.

FIG. 16 is a diagram showing an output voltage vector of a three-phase inverter with respect to a rotation angle θ of a motor.

FIG. 17 is a diagram showing an example of conventional time division PWM control using a space vector.

[Explanation of symbols]

1 motor 2A Main switching circuit 2B auxiliary switching circuit 2B1 bridge circuit 3 DC power supply (VB) 4 Rotation sensor 5 control CPU 6 Drive signal generating means 7 Resonance current arrival determination means 8 Zero voltage detection means 9 drive circuit Q1-Q6 IGBT D1 to D6 commutation diode C1 to C6 buffer capacitors C9 smoothing capacitor Q7-Q12 IGBT D7-D12 protection diode Lr Resonance inductance Vs1 to Vs6 voltage sensor Is1 to Is4 current sensor

Front page continuation (51) Int.Cl. ⁷ identification code FI theme code (reference) B60R 16/02 670 B60R 16/02 670Z H02M 7/48 H02M 7/48 P W 7/72 7/72 H02P 6/06 H02P 6/02 341J F term (reference) 5H007 AA00 BB01 BB06 CA01 CB02 CB05 CB09 CC05 CC23 DA06 DB13 DC02 DC05 EA05 5H115 PA12 PG04 PI16 PO17 PU10 PU11 PV09 QI04 RB22 TT02 EB02 RR10 EB02 EC10 RR10 EB02 EC13 RR10 EB02 EC13 RR10 EB02 EC10 RR10 EB02 RR10 EB02 EC13 RR01 XA04 5H576 AA01 BB02 BB03 CC02 DD02 DD04 DD07 EE09 EE16 GG02 GG04 GG05 HA04 HB02 JJ03 LL01 LL22 LL24

Claims (8)

[Claims] 1. Six main switching elements connected in a three-phase bridge, and six commutation diodes each connected in parallel between two terminals of the main switching element that are turned on or off by switching control, and A three-phase motor is connected to each of the connection points between the six buffer capacitors and the three sets of the main switching elements forming the respective phases of the three-phase bridge, which are connected in series at two ends of the power source. As the three-phase output terminals for connecting the three connection points of the three pairs of auxiliary switching elements connected in a three-phase bridge to the three-phase output terminals, respectively, and six auxiliary switching elements for passing a current in a single direction. A bridge circuit composed of elements; and a resonance inductance connected to the opposite side of the connection point between the three sets of auxiliary switching elements of the bridge circuit. A resonance type inverter control device for controlling an inverter circuit, which is represented by a three-phase control signal according to a switching control pattern, the angle of which differs by π / 3 [rad] according to the switching control pattern of the inverter circuit. And a three-phase control means for switching and outputting six output voltage vectors, the three-phase control means, when the output voltage vector is switched, the output voltage vector before or after the switching and the π [rad] angle. A reverse-type inverter control device which outputs different reverse voltage vectors for a predetermined time and then outputs the output voltage vector after switching. 2. Six main switching elements connected in a three-phase bridge, and six commutation diodes each connected in parallel between two terminals of the main switching element, which are turned on or off by switching control, A three-phase motor is connected to each of the connection points between the six buffer capacitors and the three sets of the main switching elements forming the respective phases of the three-phase bridge, which are connected in series at two ends of the power source. As the three-phase output terminals for connecting the three connection points of the three pairs of auxiliary switching elements connected in a three-phase bridge to the three-phase output terminals, respectively, and six auxiliary switching elements for passing a current in a single direction. A bridge circuit composed of elements; and a resonance inductance connected to the opposite side of the connection point between the three sets of auxiliary switching elements of the bridge circuit. A resonance type inverter control device for controlling an inverter circuit, which is represented by a three-phase control signal according to a switching control pattern, the angle of which differs by π / 3 [rad] according to the switching control pattern of the inverter circuit. A three-phase control means for switching and outputting six output voltage vectors, and the three-phase control means, for the six regions represented by the output voltage vector,
The advance angle region starting from a point advanced by a predetermined angle in the rotation direction of the motor from the start point of the region is defined, and an optimum single output voltage vector is assigned to each advance angle region, and the rotation angle of the motor is While present in any of the advance regions, the output voltage vector assigned to the advance region is continuously output, and the output of the output voltage vector is stopped for a predetermined time when the advance region is switched. , And the output voltage vector before or after switching and π [r
ad] A resonance type inverter controller characterized by outputting reverse voltage vectors having different angles. 3. The three-phase control means stops the output of the output voltage vector at a predetermined time before the switching time point of the advance angle region, and from the stop time point to the switching time point of the advance angle region, The resonant inverter control device according to claim 2, wherein the reverse voltage vector is output. 4. The three-phase control means stops the output of the output voltage vector at the switching time point of the advance angle region, outputs the reverse voltage vector for a predetermined time from the switching time point, and after the predetermined time period. The resonance type inverter control device according to claim 2, which outputs an output voltage vector in the next advanced angle region. 5. The resonant inverter according to claim 3, wherein the predetermined time is equal to or longer than a time from the start to the end of one switching operation of the auxiliary switching element. Control device. 6. The method according to claim 2, wherein the three-phase control means controls the output voltage vector in the advance region when the rotation speed of the motor is a predetermined rotation speed or more. The resonant inverter control device according to any one of claims. 7. A vehicle comprising: a power storage device that supplies electric energy to an electric motor that drives a drive shaft of the vehicle; and an inverter that adjusts a supply amount of the electric energy supplied from the power storage device to the electric motor. In the device, the inverter includes six main switching elements connected in a three-phase bridge,
Six commutation diodes and six buffer capacitors connected in parallel between the two terminals of the main switching element, which are turned on or off by switching control, and two buffer capacitors connected in series at both ends of the power source. In addition, the connection points between the three sets of main switching elements forming each phase of the three-phase bridge are used as three-phase output terminals for connecting the three-phase motor, and three sets of three-phase bridge-connected auxiliary switching elements A bridge circuit consisting of six auxiliary switching elements that pass a current in a single direction and that connect the respective connection points to each other to the three-phase output terminals; and three sets of auxiliary switching elements of the bridge circuit. An inverter circuit having a resonance inductance connected to the opposite side of the connection point, and a switching control pattern of the inverter circuit. The three-phase control means for switching and outputting six output voltage vectors represented by three-phase control signals corresponding to the switching control pattern, the angles of which differ by π / 3 [rad]. The control means defines, for each of the six areas represented by the output voltage vector, an advance area having a start point that is a point advanced by a predetermined angle in the rotation direction of the motor from the start point of the area, and each advance area. An optimal single output voltage vector is assigned to the angle region, and while the rotation angle of the motor exists in any of the advance regions, the output voltage vector assigned to the advance region is continuously output. When the rotation angle moves to another advance angle region in accordance with the rotation of the motor, the output voltage vector is switched to the output voltage vector assigned to the advance angle region to which the motor has moved, and the output voltage vector is switched when the output voltage vector is switched. Previous output power A reverse voltage vector whose vector and angle are different by π [rad], or an output voltage vector after switching and a reverse voltage vector whose angle is different by π [rad] are output for a predetermined time and then after switching. An inverter control device that outputs an output voltage vector, wherein the inverter charges the power storage device with regenerative energy generated from the electric motor when the reverse voltage vector is output. 8. The inverter outputs the reverse voltage vector when charging the power storage device with regenerative energy generated from the electric motor, for a longer time than the reverse voltage vector when driving the electric motor. The vehicle device according to claim 7.