GB2558376A - A ceramic dual mode microwave resonant filter and a multiplexer including such a filter - Google Patents

A ceramic dual mode microwave resonant filter and a multiplexer including such a filter Download PDF

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Publication number
GB2558376A
GB2558376A GB1717654.6A GB201717654A GB2558376A GB 2558376 A GB2558376 A GB 2558376A GB 201717654 A GB201717654 A GB 201717654A GB 2558376 A GB2558376 A GB 2558376A
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Prior art keywords
filter
resonator
dual mode
mode microwave
ceramic dual
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GB201717654D0 (en
Inventor
Ibbetson David
Ian Mobbs Christopher
Evelyne Guillemette Walker Vanessa
David Rhodes John
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Isotek Microwave Ltd
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Isotek Microwave Ltd
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    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01PWAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00Auxiliary devices
    • H01P1/20Frequency-selective devices, e.g. filters
    • H01P1/201Filters for transverse electromagnetic waves
    • H01P1/205Comb or interdigital filters; Cascaded coaxial cavities
    • H01P1/2056Comb filters or interdigital filters with metallised resonator holes in a dielectric block
    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01PWAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00Auxiliary devices
    • H01P1/20Frequency-selective devices, e.g. filters
    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01PWAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00Auxiliary devices
    • H01P1/20Frequency-selective devices, e.g. filters
    • H01P1/213Frequency-selective devices, e.g. filters combining or separating two or more different frequencies
    • H01P1/2136Frequency-selective devices, e.g. filters combining or separating two or more different frequencies using comb or interdigital filters; using cascaded coaxial cavities

Abstract

A ceramic dual mode microwave resonant filter comprising a microwave resonant cavity defined by an electrically conducting cavity wall; the cavity wall comprising first and second spaced apart end faces and a side wall extending between the end faces, the side wall comprising a ground face and a head face spaced apart from the ground face; a plurality N of TEM mode electrically conducting resonator bodies, each resonator body having an associated electrical length; the resonator bodies being electrically coupled together in a chain extending along the length axis with each resonator body being connected to the next resonator body in the chain by a resonant coupling, each resonant coupling having an associated electrical length; each resonator body extending from the ground face part way to the head face; an electrically insulating spacer layer (e.g. air gap) extending from the head face at least part way to the resonator bodies: a ceramic body arranged within the cavity and surrounding the resonator bodies, the ceramic body filling the remainder of the resonant cavity from the ground face to the spacer layer; the electrical lengths of the resonant couplings being different to the electrical lengths of the resonator bodies.

Description

(54) Title of the Invention: A ceramic dual mode microwave resonant filter and a multiplexer including such a filter Abstract Title: A ceramic dual mode microwave resonant filter with electrically insulating spacer layer (57) A ceramic dual mode microwave resonant filter comprising a microwave resonant cavity defined by an electrically conducting cavity wall; the cavity wall comprising first and second spaced apart end faces and a side wall extending between the end faces, the side wall comprising a ground face and a head face spaced apart from the ground face; a plurality N of TEM mode electrically conducting resonator bodies, each resonator body having an associated electrical length; the resonator bodies being electrically coupled together in a chain extending along the length axis with each resonator body being connected to the next resonator body in the chain by a resonant coupling, each resonant coupling having an associated electrical length; each resonator body extending from the ground face part way to the head face; an electrically insulating spacer layer (e.g. air gap) extending from the head face at least part way to the resonator bodies: a ceramic body arranged within the cavity and surrounding the resonator bodies, the ceramic body filling the remainder of the resonant cavity from the ground face to the spacer layer; the electrical lengths of the resonant couplings being different to the electrical lengths of the resonator bodies.
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Figure 35
A ceramic dual mode microwave resonant filter and a multiplexer including such a filter.
The present invention relates to a ceramic dual mode microwave resonant filter. More particularly, but not exclusively, the present invention relates to a ceramic dual mode microwave resonant filter comprising a plurality of resonator bodies arranged within a ceramic block, the resonator bodies being coupled together in a chain with each resonator body being coupled to the next by a resonant coupling, the electrical lengths of the couplings being different to the electrical lengths of the resonator bodies. The present invention also relates to a multiplexer. More particularly, but not exclusively, the present invention relates to a multiplexer including at least one such ceramic dual mode microwave resonant filter.
Microwave filters typically comprise a plurality of resonator bodies the resonant frequencies of which create the passband of the filter. The non-resonant couplings between the resonators create transmission zeros which define the behaviour of the filter outside the passband. Ideally one desires a large number of such transmission zeros. Microwave filters having N resonators which are non-resonantly coupled together and which have N-1 transmission zeros are known. However, the practical layout of such filters is awkward and multiple couplings to either the source of the load are required. Further, if the filter is a high pass filter at least one of the couplings needs to be of opposite sign to the others. A further problem with such filters is that often a large variation in coupling magnitudes within the filter is required.
The present invention seeks to overcome the problems of the prior art.
Accordingly, in a first aspect, the present invention provides a ceramic dual mode microwave resonant filter comprising a microwave resonant cavity defined by an electrically conducting cavity wall;
the cavity wall comprising first and second spaced apart end faces arranged on a length axis and substantially normal thereto, and a side wall extending between the end faces, the side wall comprising a ground face and a head face spaced apart from the ground face;
a plurality N of TEM mode electrically conducting resonator bodies, each resonator body having an associated electrical length;
the resonator bodies being electrically coupled together in a chain extending along the length axis with each resonator being connected to the next resonator in the chain by a resonant coupling, each resonant coupling having an associated electrical length;
each resonator body extending from the ground face part way to the head face;
an electrically insulating spacer layer extending from the head face at least part way to the resonator bodies;
a ceramic body arranged within the cavity and surrounding the resonator bodies, the ceramic body filling the remainder of the resonant cavity from the ground face to the spacer layer;
the electrical lengths of the resonant couplings being different to the electrical lengths of the resonator bodies.
The ceramic dual mode microwave resonant filter according to the invention generates N-1 transmission zeros from N resonator bodies. The filter is structurally simple to implement without any complex layout issues. Further, no large coupling magnitude variations are required.
Preferably the electrically insulating spacer layer is an air gap.
Preferably the spacer layer extends between the resonator bodies and the head face.
Preferably the filter further comprises an electrically conductive coupling strip arranged within the spacer layer and extending parallel to the length axis.
Preferably the resonator bodies are hollow.
Alternatively the resonator bodies are solid electrically conductive bodies.
Preferably the filter is symmetric about a symmetry plane arranged equidistant from the first and second end faces and parallel thereto.
Preferably N is greater than 2.
Preferably the electrical lengths of the resonant couplings are shorter than the electrical lengths of the resonator bodies.
Preferably the ground face comprises a groove extending in a direction of increasing depth into the cavity, the groove extending in a length direction between at least two, preferably all of the resonator bodies.
Preferably the cross sectional area of the resonator bodies in a plane parallel to the head face decreases in a direction towards the head face.
Preferably the cross sectional area decreases continuously towards the head face.
Alternatively the cross sectional area of the resonator bodies decreases in at least one step towards the head face.
Alternatively the electrical lengths of the resonator bodies are shorter than the electrical lengths of the resonant couplings.
Preferably the cross sectional area of the resonator bodies in a plane parallel to the head face increases in a direction towards the head face.
Preferably the cross sectional area of the resonator bodies increases continuously towards the head face.
Preferably the cross sectional area of the resonator bodies increases in at least one step towards the head face.
Preferably the ground face comprises at least one ridge extending in a direction of increasing height away from the cavity, the ridge extending in a length direction from one resonator body to another resonator body.
Preferably each adjacent pair of resonator bodies has such a ridge extending therebetween.
Preferably at least one of the ridges is of a different height to the others.
In a further aspect of the invention there is provided a multiplexer comprising a plurality of filters, at least one of the filters being a ceramic dual mode microwave filter as claimed in any one of claims 1 to 20.
The present invention will now be described by way of example only and not in any limitative sense with reference to the accompanying drawings in which
Figure 1 shows, in schematic form, the equivalent circuit for a known microwave filter employing non-resonant couplings;
Figure 2 shows, in schematic form, the equivalent circuit for a further embodiment of a known microwave filter employing non-resonant couplings;
Figure 3(a) shows an embodiment of a ceramic dual mode microwave resonant filter according to the invention in perspective view;
Figure 3(b) shows the resonator bodies of the filter of figure 3(a) in perspective view;
Figure 4 shows a first model of a filter according to the invention;
Figure 5 shows the response of the filter of figure 4;
Figure 6 shows a further model of a filter according to the invention;
Figure 7 shows the response of the model of figure 6 behaving as a low pass filter;
Figure 8 shows the response of the model of figure 6 behaving as a high pass filter;
Figure 9 shows a further embodiment of a ceramic dual mode microwave resonant filter according to the invention;
Figure 10 shows the response of the filter of figure 9;
Figure 11 shows a further embodiment of a ceramic dual mode microwave resonant filter according to the invention;
Figure 12 shows the filter of figure 11 from below;
Figure 13 shows a further embodiment of a ceramic dual mode microwave according to the invention;
resonant filter
Figure 14 shows the filter of figure 13 in vertical cross section;
Figure 15 shows the response of the filter of figure 13;
Figure 16 shows a further embodiment of a ceramic dual mode microwave resonant filter according to the invention;
Figure 17 shows the behaviour of the filter of figure 16;
Figure 18 shows a further embodiment of a ceramic dual mode microwave resonant filter according to the invention;
Figure 19 shows a further embodiment of a ceramic dual mode microwave resonant filter according to the invention;
Figure 20 shows a low pass prototype;
Figure 21 shows a target circuit for a filter according to the invention;
Figure 22 shows a filter according to the invention in perspective view;
Figure 23 shows the equivalent low pass prototype network for a filter according to the invention;
Figures 24(a) and 24(b) show the ideal response and location of the transmission zeros without cross coupling for a high pass filter according to the invention;
Figures 25(a) and 25(b) show the ideal response and location of the transmission zeros without cross coupling for a low pass filter according to the invention;
Figures 26(a) and 26(b) show the influence of dual mode coupling on the response and transmission zeros for a high pass filter according to the invention;
Figures 27(a) and 27(b) show the influence of dual mode coupling on the response and transmission zeros for a low pass filter according to the invention;
Figure 28 shoes a desired network response for a filter according to the invention;
Figure 29 shows a modified response combined with the cross coupling;
Figures 30(a) and 30(b) show the response and transmission zeros for a low pass filter according to the invention with a dual mode coupling of a typical value;
Figures 31(a) and 31(b) show the response of a corrected network without and with dual mode coupling;
Figure 32 shows the movement of the transmission zeros under the influence of the dual mode coupling; and,
Figures 33(a) to 35 show equivalent plots to figures 30(a) to 32 for a high pass filter.
Shown in figure 1 is an equivalent circuit 1 for a known microwave filter employing traditional non-resonant cross coupled structures. The equivalent circuit is an asymmetric third degree prototype network with two transmission zeros. The square blocks 2 represent the nonresonant couplings between the resonators.
As can be seen the practical layout of such a structure is complex and requires multiple couplings 2 to either the source or load. Further, if the filter is employed as a high pass filter at least one of the couplings 2 must be of the opposite sign to the others.
Shown in figure 2 is an equivalent circuit 3 for a further known microwave filter employing traditional non-resonant cross coupled structures. The cross coupled form shown in figure 1 can be extended to any degree by adding extra resonators at the bottom of the fold as shown in figure 2. A further problem illustrated by this figure is that for transmission zeros of similar frequencies the couplings 2 indicated by ‘a’, ‘b’, ‘y’ and ‘z’ typically reduce by an order of magnitude for each step down the ladder. For example, if the filter is of degree 8 then the magnitude of the coupling z would be around 1 /1000th of the magnitude of the coupling a.
Shown in figure 3(a) in perspective view is a first embodiment of a ceramic dual mode microwave resonant filter 4 according to the invention. The filter 4 comprises a microwave resonant cavity 5 defined by an electrically conducting cavity wall 6. The cavity wall 6 comprises first and second spaced apart end faces 7,8 arranged on a length axis 9 and substantially normal thereto. A side wall 10 extends between the end faces 7,8. The side wall 10 comprises a ground face 11 and a head face 12 substantially parallel to but spaced apart from the ground face 11. The side wall 10 further comprises support walls 13a extending between the ground face 11 and head face 12 so enclosing the cavity 5.
Extending from the head face 12 towards the ground face 11 is an electrically insulating spacer layer 13. In this embodiment the spacer layer 13 is air. Further arranged within the cavity 5 is a plurality of resonator bodies 14 which are described in detail below. The resonator bodies 14 extend from the ground face 11 part way towards the head face 12. In this embodiment the resonator bodies 14 extend as far as the spacer layer 13.
Also arranged within the cavity 5 is a ceramic body 15. The ceramic body 15 fills the remainder of the cavity 5 from the ground face 11 to the spacer layer 13. The ceramic body 5 surrounds the resonator bodies 14. In this embodiment each resonator body 14 comprises an aperture in the ceramic body 15. The side wall of the aperture is coated with an electrically conductive (preferably metallic) film to define the outer surface of the resonator body 14. In alternative embodiments of the invention the resonator bodies 14 are solid electrically conductive bodies.
The ceramic body 15 is typically magnesium calcium titanate (Mg, Ca) TiO3 along with minor dopants. In an alternative embodiment the ceramic body is a mixture of magnesium silicate and strontium titanate (Mg2SiO4 and SrTiO3). Either MgTiO3 and/or CaTiO3 could be substituted wholly or partially for SrTiO3. Similarly, Ca or Sr could partially replace Mg in the Mg2SiO4 composition.
Figure 3(b) shows the resonator bodies 14 of the filter 4 of figure 3(a). The filter 4 comprises a plurality N (in this case three) resonator bodies 14 arranged within the ceramic body 15. The resonator bodies 14 extend from the ground face 11 (and are in electrical contact with the ground face 11) part way to the head face 12.
The resonator bodies 14 are electrically coupled together in a chain by resonant couplings C formed by the proximity of one resonator body 14 to the next. The chain extends along the length axis 9 as shown. Connected to the first resonator body 14 in the chain is an input signal line 16. The input signal line 16 extends through the ceramic body 15 and through the cavity wall 6 out of the cavity 5. Connected to the last resonator body 14 in the chain is an output signal line 17. The output signal line 17 extends through the ceramic body 15 and through the cavity wall 6 out of the cavity 5.
Each of the resonator bodies 14 has an electrical length 0R. Each resonant coupling has an electrical length 0C. In the absence of the head face the electrical lengths of the resonant couplings would be the same as the electrical lengths of the resonator bodies 14 and so the filter 4 would not pass a signal. However, the capacitive coupling between the resonator bodies 14 and the head face 12 shortens the electrical lengths of the resonant couplings relative to the electrical lengths of the resonator bodies 14 so that they are different. The filter 4 is therefore a low pass filter. More generally, the electrical lengths of the resonator bodies 14 are different to the electrical lengths of the resonant couplings.
A first model of the filter 4 according to the invention is shown in figure 4. Each resonator body 14 is represented by a frequency invariant susceptance 18 in parallel with a capacitor 19. The resonant couplings are represented by N-1 series resonant sections 20, each resonant section 20 comprising a frequency invariant susceptance 21 in parallel with a capacitor 22. Because of the symmetry of the filter 4 both resonant couplings have the same resonant frequency at Wi.
The response of such a filter 4 is shown in figure 5. The two resonant couplings each produce a transmission zero above the passband at the same frequency.
The above model ignores the effect of the microwave resonant cavity 5. This produces a dual mode waveguide coupling running over the resonator bodies 14. The dual mode is excited by the connection of the input and output signal lines 16,17 to their associated resonator bodies 14 and effectively couples the first and last resonator bodies 14 together.
Shown in figure 6 is a further model of the filter 4 including this dual mode waveguide coupling. Wcc is the resonant frequency of the dual mode.
The effect of this dual mode coupling for both low pass and high pass filters 4 according to the invention is shown in figures 7 and 8. These figures show that for the low pass filter 4 what should be a pair of coincident transmission zeros has split. For the high pass filter 4 the zeros have moved into the complex plane. This indicates that there is additional coupling and it is the same sign in the low pass and high pass cases.
It is desirable to be able to control the migration of these transmission zeros in order for them to be in suitable locations with respect to the passband of the filter 4.
Shown in figure 9 is a portion of a further embodiment of a ceramic dual mode microwave resonant filter 4 according to the invention. The ceramic body 15 is not shown. The filter 4 of this embodiment is a low pass filter This embodiment is similar to that of figures 3(a) and 3(b) except it further comprises an electrically conductive coupling strip 23 arranged within the spacer layer 13 and extending parallel to the length axis 9. This coupling strip 23 effectively appears in parallel with the dual mode coupling and can be chosen to reduce the effective dual mode cross coupling to a point where the transmission zeros are in the desired location. The response of such a filter is shown in figure 10.
In the above embodiment the effect of the coupling strip 23 is to reduce the dual mode coupling so that the transmission zeros remain in the desired positions. An alternative approach is to modify the structure of the filter 4 slightly to move the transmission zeros away from the desired positions. The transmission zeros are moved by the correct amount so that when the dual mode coupling is taken into account the transmission zeros move to the desired positions. The embodiments below describe how the filter 4 can be modified in both the low pass and high pass cases.
Shown in figure 11 is a further embodiment of a low pass ceramic dual mode microwave filter 4 according to the invention. The ceramic body 15 is not shown. In this embodiment the ground face 11 comprises a groove 24 as shown which extends along the length axis 9 between the resonator bodies 14. The groove 24 extends in a direction of increasing depth (ie from the mouth of the groove to the base of the groove) into the cavity 5 and so into the space between the resonator bodies 14. This has the effect of shortening the electrical length of the resonant couplings without shortening the electrical length of the resonator bodies 14. By adjusting the dimensions of the groove 24 one can move the positions of the transmission zeros as required so that when the dual mode coupling is taken into account the transmission zeros move to the desired positions.
Figure 12 shows the filter 4 of figure 11 from below showing the groove 24 in the ground face 11. In this embodiment the depth of the groove 24 varies along its length. The groove 24 may be a hollow groove. Alternatively it may be filled with an electrically conductive material.
Shown in figure 13 is a further embodiment of a low pass ceramic dual mode microwave resonant filter 4 according to the invention. In this embodiment the cross sectional area of the resonator bodies 14 in the plane parallel to the head face 12 decreases in a direction from the ground face 11 to the head face 12. This has the effect of increasing the electrical length of the resonator bodies 14 without changing the electrical length of the resonant couplings. By correctly designing the degree of change of cross sectional area along the resonator bodies 14 one can move the transmission zeros as required so that when the dual mode coupling is taken in to account the transmission zeros move to the desired positions.
Figure 14 shows the filter 4 of figure 13 in vertical cross section, including the ceramic body 15. As can be seen the resonator bodies 14 are of different lengths. The central resonator body 14 extends from the ground face 11 to the spacer layer 13. The first and last resonator bodies 14 in the chain extend from the ground face 11 part way to the spacer layer 13. At resonance the resonator bodies 14 are quarter wave TEM mode resonators with a node at the ground face end of the resonator body 14 and an anti-node at the opposite end of the resonator body 14. The first and last resonator bodies 14 in the chain therefore resonate at a different frequency to the center resonator body 14. In contrast to the filter 4 of figure 11 the filter 4 is symmetric about a symmetry plane equidistant from the two end faces 7,8 and parallel thereto. The electrical response of this filter 4 is shown in figure 15.
Shown in figure 16 is a further embodiment of a ceramic dual mode microwave resonant filter 4 according to the invention. The filter 4 of this embodiment is a high pass filter with the electrical lengths of the resonator bodies 14 shorter than the electrical lengths of the resonant couplings. This embodiment is similar to that of figures 13 and 14 except the cross sectional area of the resonator bodies 14 increases in a direction from the ground face 11 towards the head face 12. The electrical behaviour of this filter 4 is shown in figure 17. This has the effect of reducing the electrical lengths of the resonator bodies 14 without changing the electrical lengths of the resonant couplings.
A variant of the filter 4 of figure 16 is shown in figure 18. In this embodiment the cross sectional area of the resonator bodies 14 increases in steps towards the head face 12. Again, this reduces the electrical lengths of the resonator bodies 14 without changing the electrical lengths of the resonant couplings. By correctly designing the increase in cross sectional area along the length of the resonator bodies 14 one can shift the transmission zeros so that when the dual mode coupling is taken into account the transmission zeros move to the desired positions.
A further embodiment of a ceramic dual mode microwave resonant filter 4 according to the invention is shown in figure 19. The filter is a high pass filter. Only the resonator bodies 14 and ground face 11 are shown. In this embodiment the ground face 11 comprises a plurality of ridges 25 between the resonator bodies 14. Each ridge 25 extends in a direction of increasing height away from the cavity 5 as shown. This increases the electrical length of the resonant couplings without changing the resonant length ofthe resonator bodies 14.
The filter 4 has been described above with three or four resonator bodies 14. The structure can be extended to n resonator bodies 14 coupled together. If the coupling lengths are different through the structure then the filter is a degree n structure with n-1 finite transmission zeros at frequencies Wi, w2, ... wn.i.
The electrical synthesis is based on the low-pass prototype of figure 20, a series of n-1 series resonators implementing the finite frequency transmission zeros, the shunt susceptance between the series resonators correctly matches the sections and the final shunt capacitor implements the single transmission zero at infinity.
This structure can be impedance/admittance scaled at any of the nodes without modifying the electrical response, scaling at the source or load will also modify the source and load generator impedance. Scaling at node i can introduce, modify or eliminate any existing shunt capacitance and susceptance at nodes i-1, i and i+1. Thus scaling will potentially introduce shunt capacitance at all nodes. Scaling at any node i will modify the admittance of the series resonances connected to that node but the resonant frequency of the series element remains constant.
The target circuit is n shunt capacitive elements in parallel with shunt susceptance, coupled with n-1 series resonant elements. In this structure the n shunt elements produce the single transmission zero at infinity and the n-1 series elements produce the n-1 finite frequency transmission zeros.
This is the circuit shown in Figure 21.
This final circuit can be implemented in microwave bandpass form as n shunt resonators with n-1 resonant couplings between them.
As the network can be arbitrarily scaled, the scaling may be chosen such that the resonances of the shunt elements are uniform this would result in shunt resonators that would be uniform length with variable frequency (length) couplings.
The response of the structure shown can be defined by a 2-port scattering matrix:
sn(p) s2i(p) si2(p)
S22(P)
The matrix elements are rational polynomials in the complex frequency variable, p.
Ui(p)
Σ/=ο aiPl TCCiP1 s2i(p) = si2(p)
1/=01 cC n/=i(p-/^fc)
Zfyo^p' Cn_1 W
S22O
Σΐ=οάίΡι
Σ/=ο ζ’ίΡί
Where a>k are the desired transmission zero frequencies and the coefficients of the polynomials ab bb ct and dt may be complex.
If the network is lossless then:
(-ΙΓ^Σ/^ρΟ* 522 W Σ/=0^Ρί Σ/=0^Ρί
Where nz is the number of transmission zeros and the * denotes the complex conjugate.
The desired network response can be specified in terms of passband return loss, which defines an approximation for sn(p) and stopband performance which will determine the locations of the transmission zeros ωχ2 -ωη-ι and thus s21(p). From these an input admittance function can be generated from which the network may be extracted.
The filter is an inline structure enclosed within a regular shaped housing as previously described. In addition to the coupling between the resonators there is a waveguide mode propagating down the cavity, of which the cut-off frequency of the mode and resonant frequency of the coupling is determined by the ‘a’ dimension in figure 22.
This mode couples from the input pin to the output pin which effectively couples the first and last resonators of the structure together, the equivalent low-pass prototype network is shown in figure 23.
The additional coupling mode moves the location of the n-1 finite frequency transmission zeros, the zero at infinity is not affected.
The ideal response of the circuit and the location of the transmission zeros is shown without the cross coupling in Figures 24(a), 24(b) and 25(a), 25(b) for typical high-pass and low-pass requirements respectively.
The influence of the dual-mode coupling on these circuits is shown in Figures 26(a), 26(b) and 27(a), 27(b) for the same design with coupling varying from 0 (no coupling) to approximately 1/100 of the internal couplings. The amount of dual mode coupling present in practice is approximately % to a 1/3 of the maximum shown in the plots, i.e. around 1/300 of the value of the main resonant couplings.
The coupling value is resonant well above the filter passband in this example and thus over the filter operating region is approximately uniform, as the coupling is the same mechanism for both the low-pass and high-pass design and is determined by the housing dimensions, the coupling will be of similar magnitude and sign for low-pass and high-pass filters at the same frequency.
The filter response shows what happens under the influence of the dual-mode coupling. The additional coupling is significantly smaller in magnitude compared to the internal couplings in the filter, this is illustrated by the relatively small change to the pass-band response of the filter in both low-pass and high-pass cases. The stopband of the filter is changed significantly, in both cases the transmission zeros are significantly relocated, moving into a complex conjugate pair for a very small amount of dual-mode coupling. The filter stopband rejection is significantly degraded.
In the transmission zero locus plots the * indicates the original (desired) position of the zero for zero dual-mode coupling, the x indicates the final position with increasing dual-mode coupling.
In order for the filter with dual-mode coupling to achieve the desired response it is required to design the filter such that when the dual-mode coupling is present the transmission zeros are re-located to the desired position. As is illustrated by the transmission zero locus plots the direction of transmission zero movement in the high-pass case is the opposite of that in the low-pass case. In the low-pass example shown this requires that the outer pair of transmission zeros are initially positioned closer together and the transmission zero closest to the filter passband starts closer to the passband than the desired final location. In the high-pass example the inner pair of transmission zeros should be located closer together and the outer zero initially positioned further from the filter passband.
The filter synthesis can take account of the additional coupling, using the following process.
In addition to the scattering matrix the desired network response can be characterised with an admittance matrix, Y, as shown in figure 28:
γ= YiiCp) Yiz(p) yzi(p) yzz(p)
Where the relationship between S and Ywhen in a 1Ω system is:
( , = (l ~ -gn(p))(l + ^22(p)) + siz(.p)s2i(.p) 711 P (l + sxl(p))(l + s22(p)) - s12(p)s21(p) ( , =_~2s12(p)_ 712 P (1 + s11(p))(l + s22(p)) - s12(p)s21(p) ( , =_~2s21(p)_ 721 P (1 + s11(p))(l + s22(p)) - s12(p)s21(p) ( . : (1 + -Sll(P))(1 ~ S2Z(PF) + ^12(P)^2l(P) 722 P (l + Sn(p))(l + s22(p)) - s12(p)s21(p)
The network with dual-mode coupling can be represented as a network with a modified response combined with the cross coupling as shown in figure 29:
The admittance matrix of the additional cross-coupling element is:
Cc(p-M) -Cgp-jbig -Cc(p-ja)c) Cc(p-jMc)
The network that must be synthesised for a given dual-mode coupling is
Y' = y'ldp) y'iz(p) y'ziip) y'zz(.p)
Where to maintain the original response yn(p) = y'u(p) + cc(p -M) yiz(p) = y'izCp) - cc(p-j/ yzi(p) = y'zi(p) - cc(p -j/ y22(p) = yWp) + cc(p -)/
The network that is to be extracted will have modified transmission zero locations, y'2i(p)>y'i2(p) are different, and modified matching, y' 22(p) have changed.
To extract the network, the modified transmission zero locations are required, from the equations above it can be seen than the zeros of transmission in s2i(p) are also the zeros of y2i(p)
The complete process for the design is:
• From specification generate the desired s21(p) and s1;L(p) • From s21(p) and sxl(p) generate Y, i.e. yn(p), yi2(pXy2i(p) and y22(p).
• For a given dual-mode coupling, generate Y', i.e. y'n(p), yfrFLy' 21F) and y'22(p) • Determine the modified transmission zero locations from y'12(p) and synthesise the modified network from yin(p), where y/n(p) = y'nCp) y/pfr/p) y'22F) +1
In practical terms this equates to a network with significant movement of the transmission zeros relative to the unmodified network with corresponding changes in the network impedance levels.
This is demonstrated in Figures 30-32 for the low-pass design, for a dual-mode coupling of typical level. Figures 30(a) and 30(b) show the basic initial design under the influence of the dual-mode coupling, the filter stopband performance is significantly distorted. Figures 31(a) and 31(b) show the response of the corrected network with the transmission zeros extracted taking account of the movement introduced by the dual-mode coupling. Figure 32 shows the movement of the transmission zeros under the influence of the dual mode coupling.
The high-pass design can be treated in the same manner, figures 33 - 35 show the correction for a dual-mode coupling of approximately half that of the low-pass case. Figures 33(a) and 33(b) show the behaviour of the uncorrected design with the dual-mode coupling,
Figures 34(a) and 34(b) show the behaviour of the corrected design without and with the dual mode coupling. Figure 35 shows movement of the transmission zeros under the influence of the dual-mode coupling.
Such filters according to the invention typically find application in multiplexers. Such multiplexers typically include one or more filters according to the invention.

Claims (21)

1. A ceramic dual mode microwave resonant filter comprising a microwave resonant cavity defined by an electrically conducting cavity wall;
the cavity wall comprising first and second spaced apart end faces arranged on a length axis and substantially normal thereto, and a side wall extending between the end faces, the side wall comprising a ground face and a head face spaced apart from the ground face;
a plurality N of TEM mode electrically conducting resonator bodies, each resonator body having an associated electrical length;
the resonator bodies being electrically coupled together in a chain extending along the length axis with each resonator body being connected to the next resonator body in the chain by a resonant coupling, each resonant coupling having an associated electrical length;
each resonator body extending from the ground face part way to the head face;
an electrically insulating spacer layer extending from the head face at least part way to the resonator bodies;
a ceramic body arranged within the cavity and surrounding the resonator bodies, the ceramic body filling the remainder of the resonant cavity from the ground face to the spacer layer;
the electrical lengths of the resonant couplings being different to the electrical lengths of the resonator bodies.
2. A ceramic dual mode microwave resonant filter as claimed in claim 1, wherein the electrically insulating spacer layer is an air gap.
3. A ceramic dual mode microwave resonant filter as claimed in either of claimsl or 2, wherein the spacer layer extends between the resonator bodies and the head face.
4. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to
3, further comprising an electrically conductive coupling strip arranged within the spacer layer and extending parallel to the length axis.
5. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to
4, wherein the resonator bodies are hollow.
6. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to 4, wherein the resonator bodies are solid electrically conductive bodies.
7. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to
6, wherein the filter is symmetric about a symmetry plane arranged equidistant from the first and second end faces and parallel thereto.
8. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to
7, wherein N is greater than 2.
9. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to
8, wherein the electrical lengths of the resonant couplings are shorter than the electrical lengths of the resonator bodies.
10. A ceramic dual mode microwave resonant filter as claimed in claim 9, wherein the ground face comprises a groove extending in a direction of increasing depth into the cavity, the groove extending in a length direction between at least two, preferably all of the resonator bodies.
11. A ceramic dual mode microwave resonant filter as claimed in either of claims 9 or 10, wherein the cross sectional area of the resonator bodies in a plane parallel to the head face decreases in a direction towards the head face.
12. A ceramic dual mode microwave resonant filter as claimed in claim 11, wherein the cross sectional area decreases continuously towards the head face.
13. A ceramic dual mode microwave resonant filter as claimed in claim 11, wherein the cross sectional area of the resonator bodies decreases in at least one step towards the head face.
14. A ceramic dual mode microwave resonant filter as claimed in any one of claims 1 to 8, wherein the electrical lengths of the resonator bodies are shorter than the electrical lengths of the resonant couplings.
15. A ceramic dual mode microwave resonant filter as claimed in claim 14, wherein the cross sectional area of the resonator bodies in a plane parallel to the head face increases in a direction towards the head face.
16. A ceramic dual mode microwave resonant filter as claimed in claim 15, wherein the cross sectional area of the resonator bodies increases continuously towards the head face.
17. A ceramic dual mode microwave resonant filter as claimed in claim 15 wherein the cross sectional area of the resonator bodies increases in at least one step towards the head face.
18. A ceramic dual mode microwave resonant filter as claimed in any one of claims 14 to 17, wherein the ground face comprises at least one ridge extending in a direction of increasing height away from the cavity, the ridge extending in a length direction from one resonator body to another resonator body.
19. A ceramic dual mode microwave resonant filter as claimed in claim 18, wherein each adjacent pair of resonator bodies has such a ridge extending therebetween.
20. A ceramic dual mode microwave resonant filter as claimed in claim 19, wherein at least one of the ridges is of a different height to the others.
21. A multiplexer comprising a plurality of filters, at least one of the filters being a ceramic dual mode microwave filter as claimed in any one of claims 1 to 20.
Intellectual
Property
Office
Application No: GB1717654.6 Examiner: Mr Euros Morris
GB1717654.6A 2016-11-09 2017-10-26 A ceramic dual mode microwave resonant filter and a multiplexer including such a filter Pending GB2558376A (en)

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0951089A2 (en) * 1998-04-17 1999-10-20 Murata Manufacturing Co., Ltd. Dielectric filter, dielectric duplexer, mounting structure thereof, and communication device
WO2008051572A1 (en) * 2006-10-27 2008-05-02 Cts Corporation Monoblock rf resonator/filter
US20100141352A1 (en) * 2008-12-09 2010-06-10 Nummerdor Jeffrey J Duplex Filter with Recessed Top Pattern Cavity

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0951089A2 (en) * 1998-04-17 1999-10-20 Murata Manufacturing Co., Ltd. Dielectric filter, dielectric duplexer, mounting structure thereof, and communication device
WO2008051572A1 (en) * 2006-10-27 2008-05-02 Cts Corporation Monoblock rf resonator/filter
US20100141352A1 (en) * 2008-12-09 2010-06-10 Nummerdor Jeffrey J Duplex Filter with Recessed Top Pattern Cavity

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