GB2447997A - Multiplexing a MIMO signal and a non-MIMO (e.g. MISO) signal into an OFDM signal - Google Patents

Abstract

An OFDM transmission system includes multiple transmitters and a plurality of receivers, some of the receivers being MIMO receivers and having multiple parallel receiver stages and some of the receivers being non-MIMO receivers with a single receiver stage. In such a system the transmitters are arranged to transmit OFDM signals which comprise some MIMO portions and some non-MIMO portions; the non-MIMO portions are transmitted using diversity transmission. The non-MIMO receivers are arranged to receive the non-MIMO portions. Preferably the portions are OFDM symbols and the non-MIMO portions are SISO portions transmitted using MISO transmission. The transmitters may use Alamouti mapping (Fig. 6) to transmit OFDM cells, each comprising one OFDM symbol on one carrier. Under this scheme cells are transmitted in pairs. For the second cell of the pair, a first transmitter transmits the complex conjugate of the first (previously transmitted) cell while the second transmitter transmits the inverse of the complex conjugate of the first cell. The pairs of cells may comprise cells from the same symbol, or may comprise cells from the same carrier.

Description

* 244799 OFDM TRANSMISSION SYSTEM, TRANSMITTER AND RECEIVER

BACKGROUND OF THE INVENTION

This invention relates to an OFDM transmission system, and to transmitters and receivers for use in the system It has been proposed that the next-generation terrestrial Digital Video Broadcasting system (DVB-T2) should incorporate the option of multiple-input multiple-output (MIMO) operation. In this configuration, two or more transmitters are used, together with a corresponding number of antennas at the receiver, to deliver a higher bit-rate than would be possible with conventional transmission.

One suggested arrangement is co-sited transmissions on opposite polarisations and a corresponding dual-polarisation receiving antenna; another possibility is to have geographically separated transmissions with multiple directional receiving antennas (see UK Patent Application 0603356.7, 20th February 2006 -not published at the priority date of the present application, "Estimating the channel in Broadcast COFDM-MIMO", inventor Moss, P.N., and BBC R&D White Paper WHP 144, referenced below).

Conversion of domestic reception set-ups to obtain full value from the MIMO transmissions will require at the very least an antenna upgrade, and possibly a new downlead. It is therefore likely that for a considerable transition period MIMO-enabled receivers will co-exist alongside single-antenna receivers receiving conventional or SISO (single-input single-output) transmissions.

To cope with this transition period, it has been suggested that the new standard should offer the possibility of transmitting both MIMO and conventional parts within the same OFDM (orthogonal frequency- division multiplexing) multiplex. MIMO-enabled receivers would be able to decode all of the services, whilst legacy receivers would still be able to decode the services carried in the non-MIMO part. Achieving such backwards compatibility is however not a trivial operation.

Reference may be made to the following published documents by way of

background to the present invention:

1. Digital Video Broadcasting Project, DVB-T2 Call for Technologies, paper no. SB 1644r1, 16 April 2007 (includes generic block diagrams of DVB-T transmitter and receiver). * 2

2. J.D Mitchell, P.N. Moss and M.J. Thorp, "A dual polarisation MIMO broadcast TV system", BBC R&D White Paper WHP 144, December 2006.

3. DAMMANN, A. & KAISER, S., Performance of Low Complex Antenna Diversity Techniques for Mobile OFDM Systems, Proc. 3rd International Workshop on Multi-Carrier Spread-Spectrum, Oberpfaffenhofen, Germany, September 2001.

4. KAISER, S., Spatial transmit diversity techniques for broadband OFDM systems, in Proceedings IEEE Global Telecommunications Conference (GLOBECOM 2000), November 2000, pp. 1824.1828.

5. SM. Alamouti, "A Simple Transmit Diversity Technique for Wireless Communications", IEEE Journal on selected areas of communications, vol. 16 no. 8, October 1998.

SUMMARY OF THE INVENTION

The invention in its various aspects is defined in the appended claims to which reference may now be made. Advantageous features of the invention are set forth in the appendant claims.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will now be described in more detail by way of example and with reference to the drawings, in which: Fig. I is a diagram illustrating a proposed form of multiplexing in the DVB-T2 proposal; Fig. 2 shows modified multiplexing schemes in accordance with this invention and following Alamouti; Fig. 3 shows alternative modified multiplexing schemes in accordance with this invention and following Alamouti but on adjacent carriers rather than symbols; Fig. 4 is a diagram giving examples of phase modification; Fig. 5 is an overall block diagram of a transmitter embodying the invention; Fig. 6 shows a form of the Alamouti mapping block that may be used as block 20 in the transmitter of Fig. 5; FIg. 7 shows a form of the phase modification block that may be used as block 20 in the transmitter of Fig. 5; Fig. 8 is an overall block diagram of a receiver for use with the transmitter of Fig. 5; Fig. 9 shows a form of the equalisation block 124 in the receiver of Fig. 8 in the Alamouti case; and Fig. 10 shows a form of the equalisation block 124 in the receiver of Fig. 8 in the phase rotation case.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS OF THE

INVENTION

Examples of the invention will now be described. These examples illustrate possible forms of coding embodying the invention, and the transmitters and receivers that can transmit and receive such coded signals. In these examples, MIMO transmissions include MISO (multiple-input single-output) signals that can be received on a conventional non-MIMO DVB-T receiver.

This invention is based on COFDM (coded orthogonal frequency-division multiplexing) or at least OFDM. It has a frame structure in which services are FEC-coded (forward error correction coded) individually and then multiplexed together. The multiplexing is applied in time such that a given service is carried on a number of successive OFDM symbols within each frame. A service can begin and end part-way through an OFDM symbol. The bits carried on a particular symbol are grouped together in groups corresponding to QAM (quadrature amplitude modulation) constellations; interleaving is then performed at the bit and constellation levels within the OFDM symbol. Bit interleaving is done on each service.

The invention also preferably includes provision for frequency hopping, with parts of a service carried in different frequencies at different times within a frame.

The part of a service carried on a particular frequency (carrier) in a particular frame is referred to as a "slot". Fig. 1 shows how slots are mapped onto symbols, and they can begin and end part-way through a symbol. For example, slot 2 as shown starts part-way through symbol 3 and ends part-way through symbol 4.

Every service needs to be carried in every frame, otherwise the delivery would be too bursty'. It is not therefore possible to have MIMO-only and SISO-* 4 only frames. Thus the division between MIMO and SISO needs to be done within a frame.

MIMO operation requires that each receiver can obtain, for each OFDM cell' (that is, one symbol on one carrier), a separate estimate of the channel gain h11 from each transmitter, where i is the receiver index and j the transmitter index.

Taking all the receivers together, these form an N x N matrix, which has to be inverted in order to determine the N distinct constellation points transmitted by the N transmitters for that cell.

In this proposed system separate estimates of channel gain are obtained by inverting some of the pilots, in accordance with the above-mentioned UK Patent Application. In the 2 x 2 case (two transmitters, two receivers) the pilots from one transmitter are normal; from the other transmitter half the pilots are normal and half are inverted. The receivers can use these pilots to obtain an estimate of both the sum and the difference of the channel gain for each cell, and from these it can calculate the individual gains by simple addition and subtraction.

MIMO requires a higher pilot density than SISO; in general there need to be N times as many pilots to maintain the same channel estimation bandwidth in both the time and frequency domains as for the SISO case.

Suppose now that for some of the symbols, or for some of the cells within a symbol, it is desired to operate in the non-MIMO mode. One approach is to transmit the same constellation point, x, from both transmitters. The single receiver will therefore receive h00x from one transmitter and h01x from the other; the combined result will be (h00 + h01)x.

It can therefore estimate the required gain (h00 + h01) using only the pilots which were sent non-inverted from both transmitters. The other pilots yield (h00 -h01) which is of no use to the receiver, and from the point of view of the non-MIMO receiver these pilots are a waste of power and bandwidth.

In this case it would be desirable to reduce the number of pilots in the non-MIMO symbols, but pilots are not only used to equalise for the symbols in which they occur. Because of the temporal interpolation, they also contribute to the channel estimate on other symbols. It would be very difficult to design interpolators capable of dealing with an irregular pilot pattern, especially one whose exact nature depended on the configuration of the multiplex at any one time. * 5

There is also the possibility that h00 = -h01 so that the two transmitters interfere destructively. This is quite probable if the transmitters are co-sited or transmit on a single-frequency network (SFN). In this case, since the MIMO receiver receives the two transmissions, it can still deduce both transmitted constellation points, provided that destructive interference does not also occur on the same cell for the other receiving channel. The nori-MIMO receiver on the other hand will be unable to recover the transmitted constellation point. If the relative delay between the transmitters is small, flat fading might occur such that the effect of switching on the second transmitter is to make reception worse.

We have appreciated that the considerations discussed above can be addressed by applying diversity techniques to the non-MIMO parts of the signal.

The following description is given by way of example with reference to 2 x 2 MIMO transmission. Those skilled in the art will appreciate that all of the techniques described below can be extended to higher orders of MIMO and MISO multiplexing. The principles of the techniques are described first and implementational examples are then given subsequently.

Alamouti modulation technique The preferred form of modulation technique in accordance with this invention is a MISO technique. One particularly preferred form of modulation technique is that described by S.M. Alamouti, in "A Simple Transmit Diversity Technique for Wireless Communications", IEEE Journal on selected areas of communications, vol. 16 no. 8, October 1998. We propose that this technique is applied to the non-MIMO parts of the signal.

Alamouti proposes using two transmit antennas. For one cell, one transmitter antenna transmits s0 and the other s1, and on the next cell the first transmits si* and the second transmits s0, where the asterisk indicates complex conjugation. That is, on the second cell, a first one of the two transmitters transmits the complex conjugate of the cell previously transmitted on the second transmitter, while the second transmitter transmits the inverse of the complex conjugate of the cell previously transmitted on the first transmitter. At the receiver the signals are combined by generating both the sum and the difference of the corresponding received signals. The successive cells will normally be successive symbols on the same carrier but in principle may be the same symbol on different carriers. S 6

These techniques are applied to the non-MIMO symbols; the MIMO symbols use normal MIMO techniques.

In the case of Alamouti, constellations are paired up, typically taking two symbols on the same carrier together, and sent twice, once from each transmitter. The phases in which they are transmitted are desirably chosen such that if destructive interference occurs on one symbol, the interference will be constructive on the other symbol. Whatever the channel gains and phases, the two constellations can be recovered at the receiver. The resulting signal-to-noise ratio is as though the signals from the two transmitters combined by "power addition".

In order to recover the two transmitted constellations, the receiver needs to know the complex channel gain from each transmitter individually for each cell, just as in the MIMO case. Fortunately, as discussed above, enough normal and inverted pilots are already being transmitted to allow the receiver to estimate both gains. Although the overhead in lost data rate from the extra pilots has to be accepted, at least the extra pilots are now being put to good use even in the single-channel receiver.

In the standard Alamouti scheme the assumption is made that the channel gains h00 and h01 are the same on the two cells in which a pair of constellations is transmitted. In an OFDM implementation, the two cells in question will normally be successive symbols on a given carrier. This is appropriate since OFDM systems are normally dimensioned such that the channel changes more gradually from one symbol to the next than from one carrier to its neighbour.

However, if there is a significant amount of Doppler shift or spread, there is a possibility that the assumption above will break down and the Alamouti calculation at the receiver will not correctly recover the two transmitted constellations. The result will be cross-talk between the two constellations carried in the pair of cells, combined with mis-equalisation of the wanted constellation. In principle, it would be possible to perform a more complicated calculation given that all four h's are known (h00 and h01 for the two symbols). The necessary hardware may be similar to that required for the matrix inversion in MIMO, in which case this hardware could be re-used.

The proposed frame structure described above and illustrated in Fig. I poses potential problems for the use of the Alamouti technique. If the pairs of S 7 cells are taken from the same carrier on successive symbols, there is the potential problem that the two cells do not belong to the same service because of the time division multiplexing.

Fig. 2(a) illustrates the problem. This shows some 30 carriers for a succession of eight symbols. The cells forming the first two symbols and much of the third belong to one service, the next batch of cells to another and the last two and a half or so symbols to a third. Whilst some cells could be paired up with a partner in the next symbol, there are some cells which do not have a partner; this is indicated on the right-hand side of the figure.

A possible solution is shown in Fig. 2(b), in which the cells are re-arranged. Now the cells (carriers) for each successive pairs of symbols relate to the same service. Some of the cells that were on the third symbol are split evenly between the third and fourth symbols. The multiplexing is performed on pairs of symbols such that a service which finishes during a symbol will be carried partly in that symbol and partly in the next. The cells on a given service now always occur in pairs.

A drawback of this scheme is that a slot generally occupies up to two more symbols than it would otherwise have done. This reduces the time available for switching between frequencies, but since this switching time needs to be kept constant, in practice the effect is to reduce the maximum bit-rate for any one service. For this to work, there needs to be an even number of cells allocated to each slot, but this is fundamental to the Alamouti process.

The effect of symbol interleaving has been omitted here, but it is assumed that the interleaving is the same on each symbol so that pairs of cells remain together after interleaving.

An alternative approach is to arrange the Alamouti pairs on adjacent carriers on the same symbol. This is illustrated in Fig. 3(a). Normally this would be a poor choice, since OFDM systems are generally dimensioned such that channel variation from one carrier to the next is greater than for one symbol to the next on the same carrier. This is because OFDM can tolerate a relatively large delay spread as a fraction of the active symbol period (e.g. one quarter or more depending on the guard interval), but a much smaller Doppler spread as a function of symbol rate (e.g. about 5-10%). However, the parameters being considered for DVB-T2 include very long symbols with very small fractional guard intervals. In this case, the variation from one carrier to the next would be relatively small.

In this case the symbol interleaving needs to be designed so as to keep the Alamouti pairs together. Fig. 3(b) shows the time/frequency grid after such "pair-wise" symbol interleaving. It will be noted that the cells for a given service still occur in neighbouring pairs. One advantage of this is that the symbol interleaver could be different on different symbols.

It is assumed that non-MIMO receivers will be designed in anticipation of the switchover scenario, and will be able to decode the non-MIMO part of the multiplex. However, switching to MIMO will necessitate the use of a denser pilot pattern, as discussed, with a consequent loss of capacity compared to the SISO case. The bit-rate delivered to non-MIMO receivers will be further reduced because they will not be able to decode the MIMO part of the multiplex at all. This leads to a further advantage in using the Alamouti MISO technique in conjunction with MIMO transmissions.

Using a MISO scheme such as Alamouti could go some way to recovering the lost capacity. It will be recalled that the effective SNR (signal-to-noise ratio) on the received constellations corresponds to power addition of the two transmitted signals. Assuming that the original transmitter power is maintained and the new transmitter is of the same power, this could lead to a 3dB improvement in SNR. This can be used to improve reliability of reception for the non-MIMO receiver. Alternatively, a weaker code could be used, allowing some bit-rate to be recovered. Whether the decrease in coding overhead compensates for the increased pilot density will depend on the exact details. However, if the pilot density is low to start with, a small increase in code rate would be sufficient to compensate.

The 3dB benefit assumes that the receiver sees an equal power from the two transmitters. In practice this will depend on the receiving installation and the transmitting network configuration. A good receiving installation would be expected to give significant rejection of the new transmitter, because it would be either on a different polarisation or in a different direction. The benefit is more likely to be seen for indoor, portable antennas which would be less directional and perhaps have worse rejection of the other polarisation. * 9

Alternative modification technique Alamouti is presently seen as the optimum technique in the sense that it gives the same performance as receive diversity using maximum ratio combining.

However, it was explained above that it relies on the channel remaining essentially constant between the two symbols which carry a pair of constellations, and that this might necessitate a change in transmission parameters with a consequent increase in overhead, or an increase in complexity.

An alternative approach is to send the same constellation from both transmitters in each MISO cell, but to modulate the cells differently, i.e. with different modulation formats. An example is to modulate the cells with different phases.

As an example, on half of the carriers the two transmitters can send the constellation point normally, and on the other half the second transmitter sends an inverted constellation (i.e. rotated by 180 degrees). The receiver receives (h00 + h01)x for some carriers and (h -h01)x on others. In what would otherwise be a flat-fading case, only half of the carriers interfere destructively and half constructively. Since the receiver can measure both (h + h01) and (h -h01) from the two sets of pilots, it can equalise the received constellations, assuming it knows the pattern in which the data cells will be inverted and normal.

Fig. 4 shows this diagrammatically. The carriers are represented by arrows, and the angle of the arrows shows the rotation applied to each carrier.

Fig. 4(a) shows the first transmitter, with no rotation applied, whilst Fig. 4(b) shows the alternating inversion on the second transmitter. The two carriers circled will interfere destructively.

This simple example would not be recommended, since in flat fading in the case described half of the carriers will be lost; this would certainly be beyond the capabilities of the forward error correction. Instead some of the cells could be rotated successively by all the multiples of 90 degrees, as shown in Fig. 4(c).

This results in "only" one quarter of the carriers being destroyed by flat fading.

Again, since the receiver can measure (h00 + h01) and (h -h01), it can also deduce h00 and h0, themselves by simple addition and subtraction, allowing it to calculate the composite gains affecting each cell. The principle can be extended further by using the 45 degree steps as well, reducing to one eighth the number of lost carriers for a flat-fading case, as in Fig. 4(d).

In general, in this alternative technique, the transmitters sends x and cx respectively, where c is a unit-magnitude complex number representing the rotation applied to a particular cell. The received signal is therefore (h00 + c.h01)x.

By combining its estimates of h00 and h01 with its prior knowledge of c, the receiver can calculate the transmitted constellation x.

It should be noted that the receiver has more than one set of pilots, and can thus estimate only the individual frequency responses.

This can be compared to conventional cyclic delay techniques. In these techniques, the cyclic delay needs to be kept small compared to the guard interval, for two reasons. Firstly, cyclic delay "uses up" the bandwidth of the frequency interpolator, since the interpolator cannot tell the difference between cyclic delay and genuine delay in the channel. Secondly, receivers often perform time synchronisation based on an impulse response estimate derived from the measured frequency response. Again, the cyclic delay will appear like a true delay, and the receiver might position its FFT window accordingly, so reducing the effective guard interval.

In the present proposal, use can be made of the fact that the receiver can estimate h and h01 separately. The pilots themselves are not rotated, so that the estimates obtained for the h's represented the true transmission channel.

Consequently, the cyclic rotation does not "use up" any of the interpolator bandwidth; furthermore, the impulse response estimates obtained also represent the true delay, allowing optimum positioning of the receiver FFT window.

In the above description, it has been assumed that the rotations are applied in a straight sequence across the carriers. However, there is a danger that a naturally occurring delay will "undo" the rotations and cause flat fading.

This is more likely in a non-co-sited transmitter arrangement, in which the path difference between the signals from the two transmitters varies with location. This problem can be avoided by applying the rotations in a less regular pattern. For example, a different rotation can be applied to each carrier in a pseudo-random sequence, as shown in Fig. 4(e).

With all of these phase-diversity proposals, the sequence of rotation needs to be known to the receiver in advance, since it can not be deduced from the pilots. This is because the rotation pattern is not applied to the pilots, and in any case the rotation is likely to vary too quickly with frequency for the receiver to estimate it from the relatively sparse pilots. The second point is particularly true for a pseudo-random rotation pattern. Some signalling may be required to convey this information to the receiver.

All of these alternative techniques have the disadvantage over the Alamouti technique that a clean channel can look like a 0dB echo but with delay from the point of view of the FEC decoder, and therefore "uses up" a significant amount of its error-correcting capability, increasing the SNR requirement for quasi-error-free operation. This will be mitigated if the receiving antenna provides sufficient rejection of the new transmitter, this is not a problem. On the other hand, this technique does not have the problem of crosstalk resulting from Doppler spread, as suffered by Alamouti.

Exemplaiy implementation -transmitter An exemplary implementation of the invention will now be described. A block diagram of the transmitter (i.e. modulator) 10 is shown in Fig. 5. At the input are the bit-streams corresponding to each of the services in the multiplex. Some services 12, shown at the top, are to be conveyed by MIMO transmission, whilst others 14, shown at the bottom, are for non-MIMO delivery.

The MIMO services are partitioned into two streams, one stream to be carried on each of two transmitters. The detail of the second stream is not shown separately. Channel encoding and interleaving 16 is applied to each stream individually, and the encoded bits are then mapped onto constellations 18. For the non-MIMO services, there is no partitioning, and the channel coding and interleaving 16, and constellation mapping 18, are applied as for the MIMO services.

The resulting non-MIMO constellations are then processed in order to provide two different constellation values for each cell, that is one for each transmitter. Two different options for this processing have been described, namely using Alamouti or phase mapping techniques. The chosen technique is indicated in Fig. 5 by the block 20 marked Alamouti mapping or Phase Modification'. One such block is provided for each non-MIMO service.

The resulting MIMO and processed non-MIMO constellations are then multiplexed together along with pilot values and ancillary information, using time and frequency-division multiplexing 22, as in a conventional OFDM transmitter. A pilot generator 24 generates inverted pilots where necessary on one of the transmitters, as well as any ancillary signalling. The multiplexed constellations * 12 and pilots from the multiplexer 22 are mapped onto OFDM carriers as in a conventional OFDM modulator, and the following stages of inverse FF1 26, guard interval insertion 28, filtering 30, digital-to-analogue (DAC) conversion 34 and RF upconversion 36 are all performed as normal. The output is then sent to the respective transmitting antenna 38.

Alamouti mapping If the Alamouti option is to be used, the block 20 in Fig. 5 is constituted by the circuit 40 shown in Fig. 6. Referring to Fig. 6, incoming constellations at input 42 are grouped into pairs as shown at 44. On the first symbol, two switches 46a and 46b are in the position shown, so that the two constellations are passed unmodified to outputs 54 and the time-and-frequency-division multiplexing blocks 22 of the two transmitters Txl and Tx2. On the second symbol, the switches 46a and 46b are placed in the other position so that the constellations are conjugated in circuit 48, one (only) is inverted in circuit 50, and both are routed to the transmitter that did not carry them on the first symbol. A delay 52 of one symbol is included so that the same cells are carried on successive symbols.

The "form pairs" block 44 will normally contain internal buffering and only produces an output when required. It will be appreciated that this block delivers two constellations at a time at the output, but flalf as often as they arrive at the input, so that the net input and output rates are the same.

This assumes that the Alamouti implementation uses pairs of cells taken from a given carrier on successive symbols. If adjacent carriers on a given symbol are used instead, the symbol delays are replaced by delays corresponding to one carrier.

Phase rotation Fig. 7 shows the circuit 60 that constitutes the block 20 in Fig. 5 in the phase rotation case. A constellation received at input 62 is passed directly to one output 64 and thence to the time/frequency multiplexer 22 of one of the transmitters. In parallel, it is multiplied in a multiplier 66 by a unit-magnitude complex number from a rotation pattern generator 68, causing it to undergo a phase rotation. This is fed an output 70 and thence to the time/frequency multiplexer 22 of the other transmitter, in such a way that the rotated and unmodified versions of the same constellation are carried on the same symbol and carrier from the two transmitters. Exemp!aiy implementation -receiver A MIMO receiver for use with the

transmitter described above will be of conventional MIMO form and have multiple parallel receiver stages, typically two.

Such receivers do not therefore need to be described in detail here.

Fig. 8 shows an overall block diagram of a single-input (i.e. non-MIMO) receiver 100 embodying the invention. The stages of antenna 102, RF front end 104, A-to-D (analogue-to-digital) conversion 106, channel filtering 108, time synchronisation 110, FF1 (fast Fourier transform) 112, and AFC (automatic frequency control) 114 are as in a conventional DVB receiver. There is only a single receiver stage rather than the two stages of the M1MO receiver.

The received cells are then partitioned at 116 into pilots and data.

Furthermore the pilot cells are partitioned into the normal pilots and those pilots that are inverted on one of the transmitters. Interpolation is performed in frequency and time at 118 using any appropriate design of two-dimensional interpolation, so as to obtain an estimate of both h00+ h01 and h00-h01. These estimates are fed to an adder 120 and a subtractor 122 in order to derive estimates of the two channel responses h00 and h01 individually. This process is common to a MIMO receiver and is described in UK Patent Application 0603356.7, referred to above.

The two estimates are then fed together with the data cells to an equalisation block 124. The details of this block depend on which MISO method was employed at the transmitter and are described below.

The output of the equalisation block consists of equalised constellations for the non-MIMO cells. For the MIMO cells, the output is a combination of the two independent constellations transmitted by the two transmitters; these cells are of no use to the non-MIMO receiver.

The subsequent demultiplex block 126 extracts the constellations belonging to the selected non-MIMO stream and passes these on the channel decoding and de-interleaving processes 128, which are conventional; the exact details depend on the particular transmission standard being used. * 14

Alamouti equaliser Fig. 9 shows the circuitry 140 that is in the equalisation block 124 in the case where the Alamouti technique described above is employed. The incoming cells received at input 142 are formed at 144 into the pairs which together carry a pair of constellation points. The block 144 is here assumed to contain the necessary buffering.

The rest of the circuitry of Fig. 9 performs the Alamouti demodulation process by adding and subtracting products of the incoming cells and the complex channel gains, with conjugation where appropriate, in accordance with the Alamouti technique. The result is divided by the sums of the squared magnitudes of the complex channel gains. The equations implemented by the circuitry are: = hr0+h01r hI 1h011 = h1r0+hr 1h001 +1h011 where rO and ri are the pair of received cells, and sO and si are the constellation estimates. The arrangement of multipliers (X), dividers (I), adder (+), subtractor (-), conjugators (Conj.), and circuits for generating the square of the modulus (I. 1)2 is clearly set out in Fig. 9, and the person skilled in the art will not require a detailed textual description of their connections. In each divider the input which is the divisor is indicated by a small circle; the other input is the dividend. The outputs 146 comprising the resulting estimated constellation points are fed to the demultiplexer 126 of Fig. 8. In the drawings the pair of constellation points is shown in parallel; it is assumed that the demultiplexer in this case can take two parallel inputs, but if necessary they could be multiplexed together in the equaliser block 124 before being fed to the demultiplexer 126.

Phase rotation equaliser Fig.iO shows circuitry 160 that can be in the equalisation block 124 if the phase rotation method has been used at the transmitter. For each cell, a rotation pattern generator 162 generates the same unit-magnitude complex number c as the corresponding block in the modulator. This is multiplied in a multiplier 164 by the incoming estimate for h01, and the result is added in an adder 166 to h. The * 15 output of the adder is therefore h00 + ch01, namely the complex gain which the original constellation has undergone by virtue of the phase rotation at the modulator combined with the two transmission channels. The received data cell is therefore divided in divider 168 by this result to give the original constellation point.

This block diagram and the above description describes a zero-forcing equaliser, but alternative equalisers, such as MMSE (minimum mean squared error), could be used instead. * 16

Claims (16)

1. An OFDM transmission system that includes multiple transmitters and a plurality of receivers, some of the receivers being MIMO receivers and having multiple parallel receiver stages and some of the receivers being non-MIMO receivers with a single receiver stage, wherein: the transmitters are arranged to transmit OFDM signals which comprise some MIMO portions and some non-MIMO portions; characterised in that the non-MIMO portions are transmitted using diversity transmission, and the non-MIMO receivers are arranged to receive the non-MIMO portions transmitted by diversity transmission.

2. An ODFM transmission system according to claim 1, in which the portions are OFDM symbols.

3. An OFDM transmission system according to claim I or 2, in which the SISO portions are transmitted using MISO transmission.

4. An OFDM transmission system according to claim 1, 2 or 3, in which the transmitters transmit the cells each comprising one OFDM symbol on one carrier in pairs, wherein, for one cell of the pair, a first one of the two transmitters transmits the complex conjugate of the cell transmitted for the other cell of the pair on the second transmitter, while the second transmitter transmits the inverse of the complex conjugate of the cell transmitted for the other cell of the pair on the first transmitter.

5. An OFDM transmission system according to claim 4, in which the pairs of cells comprise cells from the same symbol.

6. An OFDM transmission system according to claim 4, in which the pairs of cells comprise cells from the same carrier.

7. An OFDM transmission system according to claim 4, 5 or 6, in which the transmission comprises a plurality of slots some of which extend over part of at * 17 least some symbols, and in which the cells of each pair of cells are arranged to be from the same slot.

8. An OFDM transmission system according to claim 1 or 2, in which the non-MIMO portions are transmitted using different modulation formats on each transmitter.

9. An OFOM transmission system according to claim 8, in which the non-MIMO portions are modulated with different phases on each transmitter.

10. An OFDM transmission system according to claim 8, in which some of the non-MIMO portions are modulated with inverted phases on some transmitters.

11. An OFDM transmission system according to claim 8, in which some of the non-MIMO portions are modulated with phases which differ by 90 degrees on some transmitters.

12. An OFDM transmission system according to claim 8, in which some of the non-MIMO portions are modulated with phases which differ by 45 degrees on some transmitters.

13. An OFDM transmission system according to claim 8, in which some of the non-MIMO portions are modulated with substantially random phase the phase being different on each transmitter.

14. An OFDM transmitter for use in the system of claim 1, comprising: means for receiving MIMO services to be transmitted; means for partitioning the received MIMO services into at least two streams; means for channel encoding, interleaving and constellation mapping each partitioned stream to produce a mapped MIMO signal; means for receiving non-MIMO services to be transmitted; means for channel encoding, interleaving and constellation mapping each non-MIMO service to produce a mapped non-MIMO signal; S 18 means for time and frequency division multiplexing the mapped signals and adding pilot signals to the signal to provide a multiplexed signal; and means for applying an inverse FFT function to the multiplexed signal; characterised by means coupled to receive the mapped non-MIMO signal and to modify the signal for diversity transmission prior to time and frequency division multiplexing.

An OFDM receiver for use as a MISO receiver in the system of claim 1, comprising: means for receiving a transmitted signal at an antenna; means for subjecting the received signal to a fast Fourier transform; means for separating pilots from the received signal; means for providing equalisation to the received signal, the equalisation employing diversity techniques.

16. An OFDM receiver according to claim 15, in which the received cells each comprising one OFDM symbol on one carrier are received in pairs.