GB2170668A - Low noise, high frequency oscillator - Google Patents
Low noise, high frequency oscillator Download PDFInfo
- Publication number
- GB2170668A GB2170668A GB08502565A GB8502565A GB2170668A GB 2170668 A GB2170668 A GB 2170668A GB 08502565 A GB08502565 A GB 08502565A GB 8502565 A GB8502565 A GB 8502565A GB 2170668 A GB2170668 A GB 2170668A
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- GB
- United Kingdom
- Prior art keywords
- amplifier
- noise
- ofthe
- power
- quqo
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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- 230000000694 effects Effects 0.000 claims abstract description 5
- 238000010586 diagram Methods 0.000 description 4
- 238000003780 insertion Methods 0.000 description 4
- 230000037431 insertion Effects 0.000 description 4
- 230000010363 phase shift Effects 0.000 description 3
- 230000001419 dependent effect Effects 0.000 description 2
- 241000272470 Circus Species 0.000 description 1
- 101100380504 Schizosaccharomyces pombe (strain 972 / ATCC 24843) atf1 gene Proteins 0.000 description 1
- 238000013459 approach Methods 0.000 description 1
- 230000000295 complement effect Effects 0.000 description 1
- 239000013078 crystal Substances 0.000 description 1
- 229920000136 polysorbate Polymers 0.000 description 1
- 239000007787 solid Substances 0.000 description 1
- 238000001228 spectrum Methods 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
- H03B5/30—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element being electromechanical resonator
- H03B5/32—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element being electromechanical resonator being a piezoelectric resonator
- H03B5/36—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element being electromechanical resonator being a piezoelectric resonator active element in amplifier being semiconductor device
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
- H03B2200/006—Functional aspects of oscillators
- H03B2200/0086—Functional aspects of oscillators relating to the Q factor or damping of the resonant circuit
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
- H03B2200/006—Functional aspects of oscillators
- H03B2200/0088—Reduction of noise
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
- H03B2200/006—Functional aspects of oscillators
- H03B2200/0088—Reduction of noise
- H03B2200/009—Reduction of phase noise
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
- H03B2200/006—Functional aspects of oscillators
- H03B2200/0098—Functional aspects of oscillators having a balanced output signal
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B2202/00—Aspects of oscillators relating to reduction of undesired oscillations
- H03B2202/02—Reduction of undesired oscillations originated from natural noise of the circuit elements of the oscillator
- H03B2202/025—Reduction of undesired oscillations originated from natural noise of the circuit elements of the oscillator the noise being coloured noise, i.e. frequency dependent noise
- H03B2202/027—Reduction of undesired oscillations originated from natural noise of the circuit elements of the oscillator the noise being coloured noise, i.e. frequency dependent noise the noise being essentially proportional to the inverse of the frequency, i.e. the so-called 1/f noise
Abstract
A low noise, high efficiency oscillator having particular application in transmitters and receivers used in communication by wire and/or radio comprises a low output impedance amplifier 10, such as a push-pull switching amplifier, whose noise performance is inversely proportional to the d.c. input power multiplied by the efficiency of the amplifier, and a series resonant circuit (22) coupled in a positive feedback path of the amplifier (10). The output impedance of the amplifier (10) is reduced to provide high power efficiency and to reduce the frequency pulling effect of the load, the ratio QL/Qo of the total oscillator circuit being optimised for low noise, where QL is the loaded Q of the resonant circuit incorporating the input impedance of the amplifier (10) and Qo is the unloaded Q of the resonant circuit. For minimum noise the ratio QL/Qo would be equal to 2/3. <IMAGE>
Description
SPECIFICATION
Low noise, high efficiency oscillator
The present invention relates to a low noise, high efficiency oscillator having particular, but not exclusive, application in transmitters and receivers for communication by radio and over wire links.
A class E high-efficiencytuned power oscillator is knownforexamplefrom an article of that name by J.
Ebert and M. Kazimierczuk in IEEE Journal of Solid
State Circuits, Vol. SC-16, No.2, April 1981 pages 62 to 65. In designing their oscillatorthe authors have been influenced by the aim of getting a high efficiency of the order of 95 percent. However no mention is made of reducing the noise in such high efficiency oscillators.
By reducing the noise in oscillators then the power drain in thetransmitteror receiver is lessforthe same performance or conversely a better performance can be achieved for the same nominal power drain.
An object ofthe present invention is to produce a low noise, high efficiency oscillator.
According to the present invention there is provided an oscillator circuit comprising an amplifier having a low output impedance and a series resonant circuit coupled in a positivefeedbackpath oftheamplifier, wherein the output impedance ofthe amplifier is reduced to provide high power efficiency and to reducethefrequency pulling effect of a load, and wherein the ratio QUOo ofthetotal oscillator circuit is optimised for low noise, QL being the loaded Q ofthe resonator incorporating the input impedance ofthe amplifier and Qo being the unloaded Q ofthe resonator.
In the oscillatorcircuit made in accordance with the invention the oscillator noise performance is proportional to the operating noise figure of the amplifier, and is inversely proportional to the DC input power multiplied by the efficiency ofthe amplifier and inversely proportion to (Qo)2-(QUQo2 -OUOo).
In an embodiment of the present invention the amplifier is a push-pull switching amplifier. By using an amplifier which switches between the power supply rails very little power is dissipated in the switching devices thereby giving high power efficiency.
This combination of an amplifierwith low output impedance together with an understanding ofthe operation of the oscillator circuit makes it possible to obtain a minimum in the sideband noise performance ofthe oscillatorwhile using a minimum of DC input power.
Preferably to obtain minimum noiseQUQo=2/3,the operating noise figure (F) being assumed to be independent of QUQo.
In designing the oscillator circuit made in accordance with the present invention it is desirable to make
Lfm, the ratio of phase noise power in a 1 Hz sideband to total power output, proportional to (Fo/dF)2- [ FkT/((Qo)2-(QUQo)2-(1 -QUOo)PFED) ] , where Fo is the centre frequency
dF is the offset frequency
F is the operating noise figure ofthe amplifier under oscillating conditions
k is Boltzmann's constant
T is temperature in K, and
PFED is the r.f. power in the system which is equal to d.c. power in multiplied by efficiency.
In particularwhen QUQo=2/3then
although the constant 27/32 may vary if noise is
mixed from otherfrequencies.
The present invention will now be explained and
described, by way of example, with reference to the
accompanying drawings, wherein:
Figure 1 is a block diagram of a model of a high
efficiency oscillator,
Figure 2 is a graph of sideband noise power in dbs versus the ratio QUQo,
Figure 3 is a block diagram of an oscillator circuit
made in accordance with the present invention, and
Figure4 is a schematic circuit diagram of an experimental 1 MHz oscillatorwhich demonstrates that it is possible to reduce the noise but which has not been optimised with respect to high efficiency.
In the drawings corresponding reference numerals have been used to indicate similarfeatures.
In describing the present invention it will be shown how the spectrum of a high efficiency oscillator of the type shown in Figure 1 varies with noise figure, amplifier gain, the ratio ofthe loaded Q (QL) to the unloaded Q(Qo), that is QUQo, and the power in the oscillating system. Subsequently the optimum operating conditions for minimum sideband noise in such oscillators will be derived.
Referring to Figure 1, the oscillator circuit includes a non-inverting switching amplifier 10 having first and second inputs 12, 14 respectively, and an output 16. The input 14 is used in the oscillator model to enable one to model the effect of noise. In the illustrated circuit the signals on the first and second inputs 12,14 are summed inside the amplifier but in a practical circuit there need be only one input. A resonantfeedback network 22 connectstheoutput 16 to the input 12. The network 22 is a series LC circuit with an equivalent series loss resistor Rloss which defines Qo,the unloaded Qofthe resonator. In order to obtain a high efficiency the power dissipated in the amplifier 10 is kept to a minimum. The amplifier 10 has low, preferably zero, output impedance.
The input driving force is the noise voltage Vin (2) which is assumed to be addedatthe inputand is dependent on the input impedance Rin ofthe amplifier 10, the source resistance Rloss ofthe resonator seen by the amplifier 10 and the noise figure ofthe amplifier 10. An operating noise figure will be defined which takes all these parameters into account as the noise figure will vary from amplifierto amplifier.
The circuit configuration shown in Figure 1 is very similarto an operational amplifierfeedbackcircuit
The drawing(s) originally filed was (were) informal and the print here reproduced is taken
from a later filed formal copy.
and therefore the voltage transfer characteristic can be derived in a similar way.
Vout = d.(Vin(2) + (1.Vout) -(1) where ss Vout = Vin(1), G is the voltage gain of the amplifier, is the voltage feedback coefficient be- tween points 12 and 16 and Vin (2) is the input noise voltage.
The voltage transfer characteristic is therefore: Vout = c -(2)
Vin 1-3.6 Vy examination of the feedback network 22 the following feedback coefficient can be derived.
By considering only the noise close to carrier where the equation ## < < # can be obeved than:
The transfer characterisitc row @@@:
At resonance Vout/vin is very alrge. the output voltage is defined by the maximum swing capability of the amplifier and the input voltage is noise. As most power is very close to the carrier dF is approximately zero therefore
This effectively saying that at resonance the amplifier gain is equal to thr insertion ioss (G.ss.=1).
The gain of the amplifier is now fibted by the operations concitions.
therefore
ss looking close AGO carrier wnere LL (d#/#o) < < 1 equation (jC) becomes:
This equation does not row describe the sideband noise characteristic very @@@ be carrier if the noise
voltage in this sideband is ever than the total voltage
swing obtainable from the emullsier. This in most circumstances is considerable sister than @@ away from the carrier.
Sideband noise in oscillators is usualy quoted in terms of power and the following it has been assumed that limiting occurs at the input of the amplifier as this is the point where the maximum power is defined by the power supply.
The noise power is usually measured in a 1 Hz
badwidth and the feedback power in a 1Hz alot at any offset be defined as:
he voltage transfer characteristic can now be converted to a power transfer characteristic. The input in a one Hz badwidth is FkT where kT is
the noise power that would have been available at the
input if the source impedence was equal to the input impedance (Rin) and F is the operating noise figure which includes the amplifier parameters under the oscillating operating conditions. This includes such parameters as source impedance seen bythe ampli- ier. The dependence of; with: source impedance wil be discussed later. The square of the input voltage is therefore FkTRin.
It should be noted that the noise voltage generated by the resistor in the tuned circuit is taken into account by the noise figure of the amplifier where the equivalent series loss resistance ofthetuned circuit is the source resistance seen by the amplifier. The important noise is within the bandwidth ofthetuned circuit allowing the tuned circuit to be represented as a resistor over most of the close to carrier performance.
The feedback power in a 1 Hz bandwidth at a particular frequency offset is therefore:
As this theory is a linear theory sideband noise is effectively amplified narrow band noise. To represenil his as a carrier and sideband noise one can consider the signals being a carrier with a small perturbation rotating around a carrier. There aretwo --ectors, one for ihe upper and one forthe lower ach rotating vector can be thought of as two components, the one along the axis of the carrier vector being AM noise and the component orthogonto the carrier vector as phase noise.If the input signal is limited then the AM component would disappear and the phase component would be half of -.he total vaiue shown in equation (14). This assumes that the limiting does not cause extra components uue io mixing which may mean that the equal figure of a 1/2 may be modified.
he total feedback power is defined as FIFE13 then the ratio sideband phase noise to total feedback power can be defined as Lm.
Therefore:
This equation shows that, within the constraints o-.'- OL (d#/#o) < < 1 that Lfm is inversely proportional to
PFED.This is because the absolute value of the ideband power does not vary with the total feedback power; as long as the sideband feedback power is much: sss1 than -he total feedback power, i.e. most or The feedback power should be closer to carrier than he offset interst for the theory to apply. PFED is Am so by the; maximum voltage swing atthe output of the amplifier and the value of Rloss+Rin.
It should be noted that this is the total RF power in
the system excluding the losses in the amplifier.
Therefore PFED=(DC input power to the system)x
EFFICIENCY.
this equation can n now be used to obtain the -ninimum sideband n noise for minimum DC input power.
In optimising equation (16) for minimum phase noise then at n resonance the gain G ofthe amplifier is 1/ssa and as dF is 0 the then equation (3) can be written as as:
The gain G is therefore related to QL/Qo by the relation:
The noise equation can be rewritten as shown below:
I of ind wherethe noise equation is a minimum equation (25) is differentiated with respect QUQo and the differential is made equal to zero.
Is is a minimum when QL/Qo - 2/3 The sideband phase noise to carrier ratio under optimum operating conditions is therefore:
For minimum noise, Qo and/or PFED should be as
large as possible. The constant 27/32 may be mod
ified if the action of limiting causes noise from other
frequencies to be mixed to the operating frequency.
Note that the noise sidebands fall off at 6dB per
octive.
Now let us examine what is required in terms of
amplifier gain and insertion loss of the resonator
element.
From equations (18) and (19) the series loss
resistance of the resonator is equal to twice the input
impedance ofthe amplifier.
Taking equation (20) it can be seen that to satisfy
QUQo = 2/3 that the voltage insertion loss of the
resonator is 1/3 which sets the amplifier gain to 3.
Figure 2 is a graph showing sideband noise power
in decibels versus QUQo.
Examination of figure 2 shows that as QUQo
approaches one (that is the required amplifier gain
tends to infinity) or when QUQo tends to zero (that is the amplifier gain tends to 1) the noise performance is
markedly degraded.
It should be noted thatforthese equations to apply
we have assumed that F and therefore the input noise
voltage is constant and independent ofthe source
impedance i.e. QUQo.
Figure 3 is a block diagram of a low noise, highly
efficient oscillator made in accordance with the
present invention. The oscillator comprises a push
pull low noise switching amplifier 10 having zero or
very low output impedance. A series type of LC, SAW device or crystal resonant circuit or resonator 22 is
coupled between the output 16 and an input 12 ofthe
amplifier 10. If saturation effects cause a large noise
figure then the switching amplifier should be de
signed to avoid saturation. To maintain high efficien
cy the amplifier should still be designed to switch
over most ofthe power supply rail while not going
into saturation.
Optionally an impedancetransformer 24 is coupled
to the amplifier output and another impedance
transformer26 and a phase shift network 28 coupled
between the resonantcircuit 22 and an input of the
switching amplifier 10. The transformers 24 and 26 are amplifier and resonator dependent and in consequence they may not be required.
The unloaded Q (i.e. Qo)ofthe resonator 22 should be as high as possible. If the Series loss resistance Rloss of the tu ned circu it is low compared to the output impedance ofthe amplifier then the impedance transformer 24 should beusedtoensurethat the impedance that the amplifier presents to the resonator 22 is considerably lowerthan the resonator loss resistor.
The impedance transformer 26 would be used at the input ofthe amplifierto ensure optimum loading ofthetuned circuitto obtain QUQo = 2/3. This is achieved by ensuring that the input impedance ofthe amplifier 10 presented to the resonator 22 is half the value of the equivalent series resistance of the resonator 22.
If the noise performance characteristics of the amplifier are known then QUQo can be optimised to incorporatethis.
As a zero phase shift is required around the loop, the phase shift network 28 may need to be incorporated within the feedback network. The open loop phase errorwill rapidly degrade the noise performance. The oscillatorwill still oscillate at the 0 degree point howeverthe insertion loss of the resonator and therefore the gain of the amplifierwill increase. The O ofthe resonatorwill reduce due to the reduction in the phase slope of the resonator.
The switching oscillator must have a quiescent cu rrent to ensure Class A operation during power up.
The experimental 1 M Hz oscillator shown in Figure 4 comprises a switching amplifier 10 having a low output impedance and series resonant feedback circuit 22 coupled in the positive feedback path of the amplifier 10. The amplifier circuit comprises the devices and components ofthetypes and values indicated in Figure 4.
An integrated circuit type CA 3028 comprises a differencing amplifier and operates as a limiting amplifierto fix the voltage swing in the square wave output from the oscillator. The NPN transistor 24 comprises a current source for a low output impedance (3Q) buffer stage formed by complementary transistors 26,28. The quiescent current which is necessary during start-up of the oscillator is provided by the N PN transistor 30. The illustrated oscillator circuit is a low efficiency one but nevertheless has verified experimentally the theory as expressed in the foregoing equations.
Claims (8)
1. An oscillatorcircuitcomprising an amplifier having a low output impedance and a series resonant ci rcu it coupled in a positive feedback path of the amplifier, wherein the output impedance ofthe amplifier is reduced to provide high power efficiency and to reduce the frequency pulling effect of a load, and wherein the ratio QUQo ofthe total oscillator circuit is optimised for low noise, QL being the loaded Q ofthe resonator incorporating the input impedance ofthe amplifier and Qo being the unloaded Q ofthe resonator.
2. An oscillator circuit as claimed in Claim 1, wherein the amplifier is a push-pull switching amplifier.
3. An oscillator circuit as claimed in Claim 1 or 2, wherein QUQo = 2/3 and the operating noise figure (F) isassumedtobeindependentofQUQo.
4. An oscillator circuit as claimed in Claim 1,2 or3, wherein Lfm, the ratio of phase noise power in a 1 Hz sideband to total power output, is proportional to: (Fo/dF)2- [ FkT/((Qo)2-(QUQo92-(1 -QUQo).PFED)j where Fo is the centre frequency
dF is the offsetfrequency F is the operating noise figure
k is Boltzmann's constant
Tisthetemperature in K, and
PFED is the r.f. power in the system which is equal to d.c. power in multiplied by efficiency.
5. An oscillator circuit as claimed in Claim 4,when appended to Claim 3, wherein when QUQo = 2/3
6. An oscillator as claimed in Claim 4, when appended to Claim 3, wherein when QUQo = 2/3
7. An oscillator circuit as claimed in any one of
Claims 1 to 6, wherein the input impedance of the amplifier presented to the resonant circuit is half the value ofthe equivalent series resistance of the resonant circuit.
8. An oscillator circuit constructed and arranged to operate substantially as hereinbefore described with reference to and as shown in the accompanying drawings.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB08502565A GB2170668A (en) | 1985-02-01 | 1985-02-01 | Low noise, high frequency oscillator |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB08502565A GB2170668A (en) | 1985-02-01 | 1985-02-01 | Low noise, high frequency oscillator |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| GB8502565D0 GB8502565D0 (en) | 1985-03-06 |
| GB2170668A true GB2170668A (en) | 1986-08-06 |
Family
ID=10573779
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| GB08502565A Withdrawn GB2170668A (en) | 1985-02-01 | 1985-02-01 | Low noise, high frequency oscillator |
Country Status (1)
| Country | Link |
|---|---|
| GB (1) | GB2170668A (en) |
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| EP0579256A1 (en) * | 1992-07-17 | 1994-01-19 | Murata Manufacturing Co., Ltd. | Active-type band-pass filter |
| WO1999036940A2 (en) * | 1998-01-13 | 1999-07-22 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
| US6137237A (en) * | 1998-01-13 | 2000-10-24 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
| US6313587B1 (en) | 1998-01-13 | 2001-11-06 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
-
1985
- 1985-02-01 GB GB08502565A patent/GB2170668A/en not_active Withdrawn
Cited By (11)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| EP0579256A1 (en) * | 1992-07-17 | 1994-01-19 | Murata Manufacturing Co., Ltd. | Active-type band-pass filter |
| US5379009A (en) * | 1992-07-17 | 1995-01-03 | Murata Manufacturing Co., Ltd. | Active-type band-pass filter |
| WO1999036940A2 (en) * | 1998-01-13 | 1999-07-22 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
| US6137237A (en) * | 1998-01-13 | 2000-10-24 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
| WO1999036940A3 (en) * | 1998-01-13 | 2001-01-04 | Fusion Lighting Inc | High frequency inductive lamp and power oscillator |
| US6225756B1 (en) | 1998-01-13 | 2001-05-01 | Fusion Lighting, Inc. | Power oscillator |
| US6252346B1 (en) | 1998-01-13 | 2001-06-26 | Fusion Lighting, Inc. | Metal matrix composite integrated lamp head |
| US6310443B1 (en) | 1998-01-13 | 2001-10-30 | Fusion Lighting, Inc. | Jacketed lamp bulb envelope |
| US6313587B1 (en) | 1998-01-13 | 2001-11-06 | Fusion Lighting, Inc. | High frequency inductive lamp and power oscillator |
| US6326739B1 (en) | 1998-01-13 | 2001-12-04 | Fusion Lighting, Inc. | Wedding ring shaped excitation coil |
| US6949887B2 (en) | 1998-01-13 | 2005-09-27 | Intel Corporation | High frequency inductive lamp and power oscillator |
Also Published As
| Publication number | Publication date |
|---|---|
| GB8502565D0 (en) | 1985-03-06 |
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Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| WAP | Application withdrawn, taken to be withdrawn or refused ** after publication under section 16(1) |