FR3103335A1 - A method of processing a signal implemented by a receiving device of a first broadcast site to suppress an interfering signal, receiving device and corresponding computer program. - Google Patents

Abstract

TITLE: Signal processing method implemented by a receiving device of a first broadcast site to remove an interfering signal, corresponding receiving device and computer program. The invention relates to a method for processing a signal, implemented by a first device for receiving a first broadcast site, comprising: receiving (221) a combined signal comprising: - an interfering signal, corresponding to a first signal transmitted by a first transmission device of said first broadcasting site, via a transponder, and - at least one desired signal, corresponding to at least a second signal transmitted by a second transmission device of said at least a second site of broadcasting, via said transponder, alignment (222) of said first signal on said combined signal, subtraction (223) of said first signal aligned with said combined signal. According to the invention, the method comprises a preliminary calibration step comprising: - the reception (211) of a reference signal transmitted by said second transmission device, via said transponder, - the measurement (212) of the phase noise associated with the transmission of said reference signal, said alignment (222) of said first signal on said combined signal taking into account said phase noise. Figure for the abstract: Figure 2

Description

Method of processing a signal implemented by a reception device of a first broadcast site to suppress an interfering signal, corresponding reception device and computer program.

1. Field of the invention

The field of the invention is that of digital communications.

More specifically, the invention relates to transmissions between several broadcasting sites, via a transponder.

Even more specifically, the invention proposes a solution making it possible to cancel, or at the very least to reduce, an interfering signal received at the level of one of the broadcasting sites, corresponding to a signal emitted by a transmitter of the broadcasting site, received and retransmitted by the transponder.

The invention finds applications in any transmission system implementing a transponder, for example a satellite, in particular in broadcasting networks according to the DVB-S, DVB-S2, or DVB-S2X standard (in English "Digital Video Broadcasting – Satellite”, in French “digital television broadcasting – satellite”), or other existing or future standards.

2. Prior Art

In order to optimize spectral efficiency for transmissions between different broadcast sites, the “carrier-in-carrier”® technique has been developed.

As illustrated in Figure 1, such a technique allows a couple of modems (modulator-demodulator), communicating via a transponder, to transmit on the same transmission frequency F1 and to receive on the same transmission frequency. reception F2. Such an embodiment is also called “point-to-point” mode.

According to the example of FIG. 1, a first transmission device, comprising a modulator, and a first reception device, comprising a demodulator, located on a first broadcasting site 11 are considered. A second transmission device is also considered. transmission, comprising a modulator, and a second reception device, comprising a demodulator, located on a second broadcasting site 12. The transmission and reception devices of the first broadcasting site, respectively of the second broadcasting site, can be two separate equipment, or else form a single transceiver equipment. The transmission and/or reception devices of each broadcasting site communicate via a transponder 13, for example a satellite.

Such a transponder 13 can therefore receive on the one hand a first signal S1 transmitted by the first transmitting device at the frequency F1, and on the other hand a second signal S2 transmitted by the second transmitting device at the frequency F1.

The transponder 13 can amplify the combination of signals received, and retransmit this combination S1+S2 to the reception devices of each of the broadcasting sites 11, 12 at the frequency F2. This signal is subsequently called a combined signal.

In order to be able to demodulate the second signal transmitted by the transmitting device of the second broadcasting site 12, the receiving device of the first broadcasting site 11 must remove the component of the combined signal received corresponding to the first signal transmitted by the transmitting device of the first broadcast site 11. Symmetrically, in order to be able to demodulate the first signal emitted by the transmission device of the first broadcast site 11, the reception device of the second broadcast site 12 must remove the component of the combined signal received corresponding to the second signal transmitted by the transmission device of the second broadcasting site 12.

To do this, each broadcasting site 11, 12 implements an echo cancellation technique (or “anti-echo”, in English “echo canceller”).

Figure 1B illustrates an example of an echo cancellation technique implemented at the first broadcast site 11.

The first transmission device comprises a modulator 111 which generates the first baseband signal S1. This first signal is transposed to the frequency F1 ("up-converter" 112) and transmitted to the transponder.

The first receiving device receives the combined signal on the frequency F2 coming from the transponder and transposes it to baseband (“down-converter” 114). We denote S1'+S2' the combined signal obtained after passing through the radiofrequency channel.

An estimate 113 of the offset provided to the first signal S1 by the radio frequency channel is also determined, so as to obtain an estimate of the first received signal S1′.

The term offset designates here, and throughout the rest of the document, the deformations in time, frequency, and/or amplitude brought to the first signal S1 by the radiofrequency channel, i.e. by all the elements of the radiofrequency chain (“up- converter” 112, up-channel, transponder 13, down-channel and “down-converter” 114)

The estimate of the first received signal S1' is subtracted 115 from the combined signal S1'+S2', so as to cancel the echo and obtain an estimate of the second signal S2' received by the first reception device.

The second signal S2' can then be demodulated by the demodulator 116 of the first reception device.

However, for the echo cancellation to be effective, it is necessary that the estimation of the offset brought to the first signal S1 by the radio frequency channel be precise.

In particular, in order to correctly estimate the shift brought to the first signal S1 by the radiofrequency channel, it is important to know the static and dynamic deformations brought by the radiofrequency channel. It is therefore necessary to correctly estimate:

• phase and amplitude error and associated variations,
• the frequency offset error and the associated variations,
• the time lag and the associated variations.

However, current techniques do not make it possible to perfectly estimate the offset brought to the first signal S1 by the radiofrequency channel.

There is therefore a need for a new echo cancellation technique that is simple to implement and effective.

3. Disclosure of Invention

The invention proposes a solution allowing at least two modems to communicate via a transponder, in the form of a signal processing method, implemented by a reception device of a first site broadcasting, said first reception device, comprising:

the reception of a combined signal comprising:

an interfering signal, corresponding to a first signal emitted by a transmission device of the first broadcasting site, said first transmission device, intended for at least one reception device of at least a second broadcasting site, said second receiving device, via a transponder, and

at least one desired signal, corresponding to at least a second signal emitted by a transmission device of said at least one second broadcasting site, said second transmission device, intended for the first reception device, via the transponder,

the alignment of the first signal on the combined signal, delivering a first aligned signal,

subtracting the first aligned signal from the combined signal, providing an estimate of said at least one desired signal.

According to the invention, such a method comprises a preliminary calibration step comprising:

the reception of a reference signal transmitted by the second transmitting device, via the transponder,

measuring the phase noise associated with the transmission of the reference signal.

Moreover, according to the invention, the alignment of the first signal on the combined signal takes phase noise into account.

Thus, the proposed solution makes it possible to improve the estimation of the shift brought by the radiofrequency channel to one of the transmitted signals by taking into account the phase noise measured in the transmission system. The "alignment" aims to correct this shift, i.e. the distortions in time, frequency, and/or amplitude brought by the radiofrequency channel.

In this way, the "interfering" component of the received combined signal can be more accurately estimated, and suppressed, or at the very least reduced, before demodulation.

Indeed, the inventors have demonstrated that phase noise has an impact on echo cancellation performance.

It is recalled that phase noise is a short-term instability, coming mainly from the oscillators of a radiofrequency chain, and corresponds to rapid variations (> 1 Hz) in frequency. Phase noise is actually a slight frequency modulation of the carrier.

Since the phase noise varies little over time, it can be considered that the initial measurement of the phase noise, implemented for example during the installation of the transmission/reception devices, is representative of the phase noise of the chain. radio frequency between the first broadcast site and the second broadcast site, via the transponder. Indeed, the phase noise is mainly related to the quality of the reception head (LNB converter or "Low Noise Block") of the reception device and changes little during operation.

According to a particular embodiment, the proposed solution allows at least two modulator-demodulators to transmit on the same frequency F1 and to receive on the same frequency F2, which allows a gain in terms of spectral efficiency.

In addition, we note that it can be difficult to extract the phase noise from the received signal, knowing that other physical phenomena add degradations, such as Gaussian noise, intermodulations linked to non-linearities of the radiofrequency chain, etc

Thus, according to a particular embodiment, the reference signal is a sinusoid.

In this way, when the first transmitting device transmits a reference signal of the sinusoidal type (“sine” or CW mode, for “continuous wave”), the second receiving device can make a measurement of the phase noise of any the radiofrequency chain, and vice versa.

An advantage of this mode is that it makes it possible to highlight the reference signal from the Gaussian reception noise. Moreover, this mode is not sensitive to the non-linearities of the system.

The invention therefore proposes a clever solution for measuring phase noise.

In another embodiment, the invention relates to a receiving device of a corresponding receiving site.

Such a device for receiving a signal from a first broadcasting site, called the first reception device, comprises:

receiving means configured to receive a combined signal comprising:

an interfering signal, corresponding to a first signal emitted by a transmission device of the first broadcasting site, said first transmission device, intended for at least one reception device of at least a second broadcasting site, said second receiving device, via a transponder, and

at least one desired signal, corresponding to at least a second signal emitted by a transmission device of said at least one second broadcasting site, said second transmission device, intended for the first reception device, via said transponder,

alignment means configured to align the first signal with the combined signal, delivering an aligned first signal,

subtraction means configured to subtract the aligned first signal from the combined signal, delivering an estimate of said at least one desired signal.

According to the invention, the reception means being configured to receive beforehand a reference signal transmitted by the second transmission device, via the transponder, the first reception device also comprises phase noise measurement means configured to measure the phase noise associated with the transmission of the reference signal, and the alignment means are configured to align the first signal with the combined signal taking into account the phase noise.

Such a reception device is in particular suitable for implementing the method for processing a signal described above. It can in particular be integrated into a transmission device of the distribution site. This reception device may of course include the different characteristics relating to the process for processing a signal according to the invention, which may be combined or taken separately.

In particular, such a device comprises at least one processor, operationally coupled to a memory, configured to implement the different steps of the signal processing method according to the invention.

The invention also relates to one or more computer programs comprising instructions for the implementation of a signal processing method as described above when this or these programs are executed by at least one processor.

The invention also relates to an information medium readable by a computer, and comprising instructions of a computer program as mentioned above.

4. List of Figures

Other characteristics and advantages of the invention will appear more clearly on reading the following description of a particular embodiment, given by way of a simple illustrative and non-limiting example, and the appended drawings, among which:

Figures 1A and 1B illustrate the general principle of an echo cancellation technique in a point-to-point mode;

FIG. 2 illustrates the main steps implemented by a reception device of a first broadcast site according to one embodiment of the invention;

Figure 3 shows the general principle of the alignment step of Figure 2;

FIG. 4 presents a block diagram of the frequency correction according to an embodiment of the invention;

FIG. 5 presents a block diagram of the correction in time, phase and/or power according to an embodiment of the invention;

Figure 6 illustrates the phase noise mask according to one embodiment of the invention;

FIG. 7 illustrates an example of implementation of a reception device on an FPGA-type programmable circuit;

Figure 8 illustrates an example architecture for the FIFO of Figure 7;

Figure 9 illustrates an example architecture for the NCO of Figure 7;

Figure 10 illustrates an example architecture for the correlator of Figure 7;

Figure 11 illustrates an example architecture for the frequency correction module of Figure 7;

FIG. 12 illustrates an example of architecture for the time, phase and/or amplitude correction module of FIG. 7;

FIG. 13 presents the simplified structure of a reception device according to a particular embodiment;

FIG. 14 illustrates the implementation of the invention in a point-to-multipoint mode.

5. Description of an embodiment of the invention

5.1 General principle

The invention is placed in the context of communications between at least two broadcasting sites, via a transponder.

The general principle of the invention is based on taking into account the phase noise associated with the radio frequency channel ("upconverter", uplink channel, transponder, downlink channel, "downconverter"), to optimize the performance of a cancellation technique echo implemented at the level of at least one of the broadcast sites.

FIG. 2 illustrates the main steps implemented by a reception device of a first broadcasting site, called first reception device, in a communication system comprising two broadcasting sites, such as that illustrated in FIG. 1A (point mode -medium rare). Note that, symmetrically, the same steps can be implemented by a reception device of the second broadcast site. It can also be noted that the proposed solution can be applied to the point-to-multipoint mode, as described below.

As illustrated in FIG. 2, the first receiving device first implements a calibration step 21, comprising:

• the reception 211 of a reference signal transmitted by a transmission device belonging to the second broadcasting site, said second transmission device, via the transponder,
• the measurement 212 of the phase noise associated with the transmission of the reference signal.

This calibration step can be implemented once during the installation of the transmission/reception devices, or repeated. It can be implemented periodically or on an ad hoc basis, for example following a change in the propagation channel between the two broadcasting sites, via the transponder.

Once the phase noise associated with the transmission of the reference signal is measured, the first receiving device can process useful signals.

Thus, the first receiving device implements:

• the reception 221 of a combined signal comprising:
  • an interfering signal, corresponding to a first useful signal emitted by a transmission device of the first broadcasting site, said first transmission device, intended for the reception device of the second broadcasting site, said second reception device, via a transponder, and
  • a desired signal, corresponding to a second signal transmitted by the second transmitting device, intended for the first receiving device, via the transponder,
• the alignment 222 of the first signal on the combined signal, delivering a first aligned signal, taking into account the measured phase noise,
• subtracting 223 the aligned first signal from the combined signal, yielding an estimate of the desired signal.

5.2 Example of implementation

The general principle of the alignment step 222 of FIG. 2 is described below, in relation to FIG.

As previously indicated, the first receiving device receives (221) a combined signal Sc comprising an interfering signal and the desired signal. The first receiving device also knows the first useful signal S1 transmitted by the first transmitting device.

The first receiving device then seeks to align (222) the first signal S1 with the combined signal Sc. The first aligned signal S1' can then be subtracted from the combined signal Sc, so as to obtain an estimate of the desired signal S2'.

According to at least one embodiment, the alignment 222 implements a first synchronization step 31, called coarse synchronization.

In particular, such a coarse synchronization step implements:

• a correlation between the combined signal Sc and the first signal S1 offset in time and/or in frequency, for at least two offset values in time and/or in frequency,
• a determination of a correlation peak,
• an estimate of a rough shift in time and/or frequency between the combined signal and the first signal.

For example, for each offset in time and/or frequency, the value of the amplitude of the correlation between the combined signal Sc and the first signal S1 offset in time and/or frequency is stored in a table. Then the table of amplitude values as a function of time and/or frequency is analyzed in order to detect the maximum value, corresponding to a correlation peak between the combined signal Sc and the first signal S1. The time and/or frequency values associated with this correlation peak make it possible to determine a coarse offset in time and/or in frequency between the combined signal and the first signal.

The correlation is expressed for example in the following form:

with L corresponding to the length of the correlation.

As an example, the following parameters are chosen for coarse synchronization 31:

• the length of the correlation is chosen to be greater than 1000 samples, for example equal to 10000 to have better precision,
• the frequency shift increment step is equal to 10 kHz,
• the frequency offset test range is +/- 1MHz,
• the time shift increment step is equal to 10 ns (depending on the transmission speed, in English “baudrate”, here for a baudrate > 50MHz),
• the time shift test range is 5ms.

This coarse synchronization step 31 can be implemented once during start-up, or else repeated, for example following the detection of a loss of synchronization.

Optionally, several iterations of the coarse synchronization algorithm in time and/or frequency can be implemented, by refining the search range of the time and/or frequency offset in order to have an increment step of the offset in frequency less than 10 kHz, for example of the order of 100 Hz.

It is noted that, according to a particular embodiment, the first signal carrying a header not obtained from a scrambling sequence (header not "scrambled"), the correlation step is deactivated during the reception of the header of the first signal.

According to another particular embodiment, the first signal is constructed from a scrambling sequence distinct from that used to construct the second signal.

These two variants, which can be implemented independently or jointly, make it possible to limit the risk of false detections, in particular when the signals S1 and S2 emitted by each of the emission devices resemble each other / are correlated.

For example, if we place ourselves in the context of the DVB-S2 standard, such a header is of type “PLHEADER” and the scrambling sequence of type “PL Scrambler”.

Once the coarse shift in time and/or frequency between the combined signal and the first signal has been determined, fine synchronization algorithms based on frequency servoing using a digital phase-locked loop (DPLL) and /or a recursive adaptive filter (FIR) can be implemented to follow the variations of the combined signal and to maintain the alignment of the first signal on the combined signal.

Thus, according to at least one embodiment of the invention, the alignment 222 also implements:

• a frequency correction of the first signal, implementing a phase locked loop 32,
• a time, phase and/or gain (i.e. amplitude) correction of the first frequency-corrected signal, implementing adaptive filtering 33.

Figure 4 shows a block diagram of the frequency correction, used to fine-tune frequency synchronization, and track and correct frequency offsets between the first signal and the combined signal.

According to the example illustrated in Figure 4, the frequency correction implements at least one iteration of the following steps:

• frequency shift 41 of the first signal S1 according to a setpoint ,
• conjugate complex multiplication 42 of the combined signal Sc with the first signal S1 shifted in frequency according to the setpoint ,
• low-pass filtering 43 of the result of the multiplication,
• estimation 44 of the phase difference between the combined signal Sc and the first signal shifted in frequency from the signal obtained at the output of the low-pass filtering,
• error correction 45 delivering a setpoint update.

In particular, the frequency shift step is implemented by an NCO oscillator (in English “numerically-controlled oscillator”, in French “oscillateur contré par entrée numérique”).

For the first iteration (initialization), the setpoint corresponds to the frequency offset obtained at the output of the coarse synchronization step 31. For the following iterations, the setpoint is that obtained at the output of error correction 45.

The low-pass filtering 43 for example implements a low-pass filter with an adjustable cut-off frequency. In particular, such a low-pass filter makes it possible to filter the phase noise by allowing frequencies below a cut-off frequency determined during the calibration step to pass. As explained below, such a cutoff frequency is determined from the phase noise of the radiofrequency chain.

The conjugate complex multiplication 42, associated with the low-pass filtering 43, makes it possible in particular to carry out a correlation between the combined signal and the first transmitted signal delayed, then corrected in frequency.

According to the example illustrated in FIG. 4, the estimate 44 of the phase difference between the combined signal Sc and the first frequency-shifted signal implements an arctangent calculation, and the error correction 45 implements a corrector of the type "Proportional, Integral, Derivative" (PID) in order to update the setpoint Δf. According to a particular embodiment, the coefficients of the PID corrector also depend on the cutoff frequency determined during the calibration step.

By way of example, seeking to obtain a residual error of the order of 1 Hz when the frequency locking algorithm has converged, the following parameters are chosen for the phase-locked loop 32:

• PID proportional gain: ,
• PID integral gain: ,
• gain derived from PID: .

In particular, the inventors have observed that the residual noise of the phase-locked loop degrades the performance of the adaptive filter. It is therefore sought, according to at least one embodiment, to optimize the coefficients of the phase-locked loop in order to reduce this level of residual noise.

After the frequency correction, or simultaneously with the frequency correction, a correction in time, phase and/or gain (amplitude) of the first signal is implemented.

FIG. 5 presents a block diagram of the correction in time, phase and/or power of the first signal, implementing an adaptive filtering 33. Such an adaptive filtering makes it possible to closely follow the variations in time, in phase and/or in power.

For example, such an adaptive filtering 33 implements an algorithm belonging to the group comprising:

• an LMS type algorithm (in English “Least Mean Square Error”, in French algorithm for minimizing the mean square error),
• an NLMS type algorithm (in English “Normalized Least Mean Square Error”, in French algorithm for minimizing the normalized mean square error”),
• an RLS type algorithm (in English “Recursive Least Square”, in French recursive least squares algorithm).

According to the example illustrated in figure 5, the correction in time, phase and/or gain implements a mean square error minimization type algorithm, implementing at least one iteration of the following steps:

• adaptive filtering of the first frequency-corrected signal, implementing an adaptive filter 51 with finite impulse response, delivering said first aligned signal,
• update 52 of the coefficients of the adaptive filter,

the updating of the coefficients being implemented after the step 223 of subtracting the first aligned signal from the combined signal, delivering an estimate of the desired signal (E=S2'), and multiplying the estimate of the desired signal by a parameter convergence .

Noting:

S1 the first signal transmitted by the first transmitting device (interfering signal to be suppressed by the first receiving device),

S2 the signal transmitted by the second transmitting device (desired signal to be demodulated by the first receiving device,

Sc the combined signal received by the first reception device, carrying the first signal S1 and the second signal S2 deformed by the propagation channel (including in particular the transponder),

Y the signal at the output of the adaptive filter 51;

E the error signal, corresponding to the desired signal S2',

we have:

, with b a Gaussian white noise and H corresponding to the time response of the propagation channel,

, with W corresponding to the coefficients of the adaptive filter 51,

If we consider an algorithm of the LMS or NLMS type, the coefficients of the adaptive filter 51 can be expressed in the following form:

, with μ corresponding to the adaptation step, also called convergence parameter.

If we consider an RLS type algorithm, the coefficients of the adaptive filter 51 can be expressed in the following form:

with

,

corresponding to the adaptation step.

By way of example, the following values can be chosen for the adaptation step:

• for the LMS algorithm: ;
• for the NLMS algorithm: ;
• for the RLS algorithm, .

In particular, an algorithm of the RLS type can be favored for a short length of the adaptive filter, for example less than 100 coefficients.

According to a particular embodiment, the convergence parameter is determined from the residual phase noise obtained during the calibration step, as described below.

5. 3 calibration

A description is given below, in relation to FIG. 6, of the measurement of the phase noise implemented during the calibration step 21, and its use. Indeed, as indicated previously, the calibration step 21 makes it possible to determine a cut-off frequency for the low-pass filtering 43 and for the error correction 45, used for the frequency slaving, as well as a noise of residual phase used to determine the convergence parameter of the adaptive filter.

1. Determination of phase noise

It is considered for example that the reference signal S2 _ref emitted by the second emission device during the calibration step is a sinusoid.

In this case, the phase noise φ(t) can be expressed as:

with:

V0 the amplitude of the sinusoid,

F2 the frequency of the sinusoid

φ(t) the instantaneous phase variation, also called phase noise

The spectrum of such a signal must therefore show a line at frequency F2, with amplitude V0. The dispersion around this line represents the phase noise template.

On reception of the reference signal, the first reception device can determine a mask 61 of the phase noise, centered on the carrier frequency used by the downlink transponder. It is recalled that, as the reference signal passes through the transponder, it is transmitted on a frequency F1 by the second transmitting device, but received on a frequency F2 by the first receiving device.

For example, the determination of a phase noise mask implements the measurement of the power of the reference signal received on at least one frequency belonging to a narrow frequency band (i.e. of the order of a few kilohertz) centered on the carrier frequency used by the downlink transponder. In other words, different measurements of the power of the reference signal received by the first reception device (611, 612, 613, etc.) are carried out for different frequencies close to the frequency F2.

It can be seen in Figure 6 that the template 61 makes it possible to break down the narrow frequency band into two parts:

• a first part 62 corresponding to the frequency band situated around the central frequency F2, corresponding to a higher power of the received signal. These are fairly slow variations that can be compensated;
• a second part 63, remote from the central frequency F2, corresponding to the residual phase noise. This second part corresponds to the phase noise that an adaptive filter, used to compensate for variations in time, phase and/or amplitude of the received signal, can compensate. These instantaneous variations have an impact on the responsiveness of the recursive algorithm.

It is then possible to estimate a cutoff frequency Fc associated with the first part 62 of the template 61 which can be applied to a low-pass filter to filter the phase noise and to optimize the performance of the system. Indeed, this first part corresponds to the phase noise that the frequency slaving algorithm, used to compensate for the frequency variations of the received signal, can compensate.

For example, the cutoff frequency is estimated from the standard deviation associated with measurements of the power of the reference signal received on said narrow band. Indeed, even if the phase noise template does not correspond exactly to a Gaussian distribution, the inventors have shown that a calculation of the standard deviation makes it possible to obtain a correct estimate of the cutoff frequency Fc.

The cut-off frequency thus determined, for example of the order of a few hundred hertz, can then be used by the frequency slaving algorithm making it possible to compensate for the variations in frequency of the signal received.

It is also possible to estimate the residual phase noise associated with the second part 63 of the template 61, associated with the reference signal received outside the frequency band defined by the carrier frequency used by said downlink transponder plus or minus the frequency of cut (+/- Fc and +/- 1MHz of the center frequency F2).

For example, the residual phase noise measurement (residual_PN, in dB) is performed by measuring the power of the carrier at frequency F2, denoted Power_CW, and comparing it with the power of the signal in the narrow band around the frequency F2, , denoted Power_residual:

Residual_PN = 10*log10(Power_CW / Power_Residual)

The residual phase noise thus determined, for example of the order of -30dB to -50dB, can then be used to adjust the parameters of the adaptive filter used to compensate for variations in time, phase, and/or amplitude of the received signal.

1. Use of phase noise

Once the cutoff frequency and the residual phase noise have been determined, it is possible to use these parameters to adjust the frequency servoing implemented by the phase-locked loop 32 and to adjust the coefficients of the adaptive filter 51 described previously.

More specifically, the inventors have demonstrated that the algorithm which makes it possible to perform frequency slaving must be more or less fast depending on the phase noise.

Thus, according to at least one embodiment, the proposed algorithm is able to follow the slow variations of the phase noise and to filter the instantaneous variations to optimize the frequency control.

To do this, a cutoff frequency associated with the first part 62 of the template 61 is determined, as explained above, and the low-pass filter 43 (i.e. phase filter) of the phase-locked loop 222 is adjusted with this cutoff frequency value.

Recall that the phase filter is an exponential weighted average function expressed as:

Or:

is a filter parameter ( ) For example ;

is the complex conjugate product of the combined signal with the transmitted signal delayed and then corrected in frequency.

For a continuous system, the Laplace transform of such a transfer function is written:

By expressing the filter parameter as , Or is the maximum filtering constant (<<1) and k is an adjustment parameter included in the interval , it can be shown that, knowing the cutoff frequency of the phase filter, the adjustment parameter k can be written:

with:

where Fc is the cutoff frequency of the phase filter and is the sampling frequency.

Thus, the determination of the cutoff frequency Fc from the phase noise makes it possible to determine the adjustment parameter k itself used to determine the filter parameter of the low-pass filter 43.

Also to optimize the frequency control, the coefficients of the PID error corrector 45 can be determined according to the cut-off frequency.

By way of example, we consider that the coefficients , And are initialized to values , And , set for a nominal cut-off frequency of 300 Hz.

To maintain the PID error corrector setting regardless of the phase filter cutoff frequency, the coefficients , And are modified according to a band factor β such that:

with:

The inventors have also demonstrated that the recursive algorithm based on an adaptive filter can be optimized by taking phase noise into account.

Indeed, as indicated above, if we consider an algorithm of the LMS or NLMS type, the coefficients of the adaptive filter 51 can be expressed in the following form:

, with corresponding to the adaptation step.

The larger this adaptation step, the faster the algorithm is to converge but the greater the residual noise. In order to get better performance from the adaptive filtering algorithm, the phase noise value can enter into the calculation of the convergence parameter according to the following formula:

,

with M a constant allowing to weight the measurement of the phase noise on the convergence parameter . For example, the constant M is of the order of 0.8.

It is also possible to express the parameter as a function of the phase noise if an RLS type algorithm is considered.

5.4 Example implementation

Presented below, in relation to FIG. 7, is an example of implementation of a reception device according to at least one embodiment of the invention, according to which the echo cancellation technique is implemented on a programmable circuit of the FPGA type (in English “Field Programmable Gate Arrays”).

For the sake of simplifying the description, we always place ourselves in the context of a communication system comprising two broadcasting sites, such as that illustrated in FIG. 1A, and more precisely at the level of the reception device of the first broadcasting site.

The complex samples corresponding to the combined signal received Sc (coming from the analog-digital converter 71) and the complex samples corresponding to the first signal S1 coming from the modulator are first of all interpolated (decimation filters 721, 722) at twice the frequency of sampling ( ). For example, IQ_COMBINE denotes the interpolated complex samples corresponding to the received combined signal Sc and IQ_INTF the complex samples corresponding to the first signal S1.

It is then sought to remove the interfering signal (S1 received) from the combined signal Sc.

The suppression of the interfering signal from the combined signal Sc implements according to this example two processing chains:

• a chain 73 for processing the data stream, at the frequency , operating in real time,
• a chain 74 for searching for corrections in frequency, time, phase and/or amplitude, which can operate by block of data.

According to the example illustrated in FIG. 7, the correction search chain 74 comprises a controller 741 making it possible to manage the sequence of coarse synchronization, frequency slaving and adaptive filtering algorithms, and to check their convergence.

At the end of the suppression of the interfering signal from the combined signal Sc, an estimate of the second signal S'2 is obtained, at the frequency . The signal S'2 obtained can be interpolated (interpolation filter 75) at the frequency , before being converted into an analog signal in an analog-digital conversion module 76 and transmitted to the demodulator 77 of the first reception device. According to this implementation, the demodulation function is therefore not implemented in the FPGA. The second estimated signal S2' is thus reconverted into a radiofrequency signal before entering the demodulator.

In particular, an external memory 78, for example of the DDR (“Double Data Rate”) type, can be added. A DDR 79 driver as well as FIFOs internal to the FPGA at the write and read level can be associated with this memory.

The two processing chains 73 and 74 are described below in more detail.

The data stream processing chain 73 makes it possible to determine the offset to be applied to the first signal emitted by the transmission device of the first broadcasting site in order to be able to align it with the combined signal and then subtract it from the combined signal.

According to the example illustrated in figure 7, the first signal S1 is processed by different modules:

• a FIFO 731, allowing to delay the complex samples of the signal S1, for example up to 280ms. We note for example IQ_delay the complex samples at the output of the FIFO 731;
• an NCO 732, to provide frequency correction to delayed complex samples. We denote for example IQ_NCO the complex samples at the output of the NCO 732;
• a complex filter 733, making it possible to provide phase and/or amplitude correction to the delayed and frequency-corrected complex samples. We denote for example IQ_FIR the complex samples at the output of the filter 733;
• a subtractor 734 making it possible to subtract the first aligned signal (IQ_FIR samples) from the combined signal (IQ_COMBINE samples). We note for example IQ_DES the complex samples at the output of the subtractor 734.

Figure 8 illustrates an example architecture for FIFO 731.

Such a FIFO 731 comprises for example means making it possible to shift the complex samples by an integer value, as well as means making it possible to shift the complex samples by a fractional value, in order to precisely follow the variations in time.

For example, the means making it possible to shift the complex samples by an integer value implement a memory 7311 (FIFO IN, FIFO OUT). The means making it possible to shift the complex samples by a fractional value implement an interpolation filter, for example of the Lagrange type of the ^{5th minimum} order.

Note that the integer shift implemented by memory 7311 and the fractional shift implemented by Lagrange filter 7312 must be synchronized.

Figure 9 illustrates an example architecture for the NCO 732.

Such an NCO 732 comprises for example a phase accumulator 91, a phase to sine/cosine conversion 92, then a complex multiplication 93 for the frequency shift.

For example, phase to sine/cosine conversion 92 implements a memory (LUT table) or a Cordic (in English “COordinate Rotation DIgital Computer”, in French “numerical calculation by rotation of coordinates”).

The complex filter 733 is for example a finite impulse response filter with at least 15 coefficients, making it possible to correct the amplitude, the phase and/or the residual error in time. The coefficients can be updated almost instantaneously in order to follow the rapid variations of the combined signal.

The subtractor 734 allows for its part a subtraction of complex samples. It does not use the resources of the FPGA in terms of DSP (in English “Digital Signal Processing”) and BRAM (in English “Block Random Access Memory”).

The chain 74 for searching for corrections in frequency, time, phase and/or amplitude makes it possible to estimate the shifts to be made to the signal S1 to align it with the combined signal Sc.

According to the example illustrated in FIG. 7, the processing chain 74 implements three different modules:

• a correlator 741, making it possible to estimate a coarse offset in time and/or in frequency;
• a frequency correction module 742, making it possible to follow the variations in frequency;
• a time, phase and/or amplitude correction module 743, making it possible to adjust the coefficients of the filter 733 to quickly follow the variations in time, phase and/or amplitude.

The algorithms implemented by these different modules can be managed by a 744 controller, implementing for example a state machine making it possible to clock the sequences and to check the attachment and convergence of the algorithms.

Figure 10 illustrates an example architecture for the 741 correlator.

Such a correlator 741 takes as input the complex samples of the combined signal (IQ_COMBINE), as well as the complex samples of the first signal S1 after passing through the FIFO 731 and the NCO 732 (IQ_NCO).

The objective is to synchronize these two signals in time and/or frequency, in order to be able to follow the variations linked in particular to the movement of the satellite-type transponder.

The correlator 741 implements a first step of coarse synchronization in time and/or in frequency (reference 31 in relation to FIG. 3), making it possible to shift in time and/or in frequency the first signal thanks to the detection of a correlation peak. This first synchronization step is based on a central correlation 101c, i.e. comparing a complex sample of the combined signal at time t (IQ_COMBINE(t)) with a complex sample of the first signal S1 after passing through the FIFO 731 and the NCO 732 at the same time t (IQ_NCO(t)). The result of the central correlation can be stored in a memory 102c.

Optionally, the correlator 742 can implement a second synchronization step, called fine synchronization.

This second synchronization step is based firstly on a "before" correlation 101u (or "under" correlation), i.e. comparing a complex sample of the combined signal at time t -1 (IQ_COMBINE(t-1)) with a complex sample of the first signal S1 after passing through FIFO 731 and NCO 732 at time t (IQ_NCO(t)), and on the other hand on an “after” correlation 101o (or “over” correlation), i.e. comparing a complex sample of the combined signal at time t+1 (IQ_COMBINE(t+1)) with a complex sample of the first signal S1 after passing through FIFO 731 and NCO 732 at time t (IQ_NCO( t)).

The "before" correlation result may be stored in a memory 102u and the "after" correlation result may be stored in a memory 102o.

The difference obtained between the "before" correlation and the "after" correlation is transmitted to the controller 744, which can thus update the integer and fractional values of the time delay and supply them to the FIFO 731. This difference therefore makes it possible to adjust the fine shift and adjust the Lagrange filter.

Controller 744 can also provide NCO 732 with the setpoint update.

During the second synchronization step, the central correlation makes it possible to verify that the algorithm is still locked on the correlation peak detected during the first synchronization step.

Figure 11 illustrates an example architecture for the 742 frequency correction module.

The fine frequency correction, as well as the monitoring of the variation of the frequency, is based on a digital PLL comprising four modules in addition to the NCO 732. These modules implement the steps of figure 4. The same references are therefore used as those of figure 4:

• a conjugate complex multiplication 42 between the complex samples of the combined signal (IQ_COMBINE) and the complex samples of the first signal S1 after passing through the FIFO 731 and the NCO 732 (IQ_NCO);
• a module 43 for low frequency filtering of the result of the multiplication, making it possible to limit the overall calculation noise. For example, the cutoff frequency is between 100Hz and 10kHz, with a nominal value at 300Hz);
• an arctangent calculation 44 of the result of the filtering, making it possible to estimate the phase difference between the combined signal and the first signal S1 after passing through the FIFO 731 and the NCO 732;
• a PID type error correction module 45 making it possible to correct the phase shift and to update the setpoint .

The controller 744 can update the smoothing parameter of the low-pass filter 43 and the coefficients of the PID 45, and supply the NCO 732 with the setpoint thus updated.

FIG. 12 illustrates an example of architecture for the time, phase and/or amplitude correction module 743, based on an LMS type algorithm.

The LMS algorithm is based on a calculation of the coefficients of the filter 733 from a minimization of the quadratic error. Such an algorithm takes as input the complex samples of the first signal S1 after passing through the FIFO 731 and the NCO 732 (IQ_NCO) and the complex samples obtained after subtracting the first aligned signal from the combined signal (IQ_DES), corresponding to the estimation of the desired signal.

The complex samples of the estimate of the desired signal IQ_DES (corresponding for the algorithm to the error) are multiplied 121 by the convergence parameter . The result is then multiplied by the samples IQ_NCO of the first signal S1 corrected in time and frequency to be able to update the coefficients W of the filter 733.

Returning to FIG. 7, the decimation 721, 722 and interpolation 75 filters are described below.

The first decimation filter 721, which makes it possible to transform the complex samples of the combined signal coming from the ADC 71 from a fixed frequency to a frequency at dependent on the baud rate can be chosen sufficiently selective to avoid noise aliasing in the useful band.

The second decimation filter 722 which takes the complex samples from the first transmitting device (internal modulator) requires little constraint because the signal is much cleaner (since the first transmitting and receiving device are located on the same site of broadcast, and possibly belong to the same equipment). It can settle for a lower order compared to the first decimation filter 721.

Finally, the interpolation filter 75 of to the fixed frequency of the DAC 76 has little constraint because the signal leaving the DAC 76 can be directly demodulated by the demodulator 77, for example an S2X demodulation chip.

According to at least one embodiment, implemented for example in the reception device illustrated in FIG. 7, the main synchronization, frequency correction and adaptive filtering steps implement the following specificities:

for synchronization:

• use of separate scrambling sequences for different modems;
• disabling correlation during the header of the first signal (interfering signal);
• minimizing the number of correlations by adapting the time/frequency adjustment step;
• once the coarse synchronization has been achieved, the correlation can be used to follow the slow variations of the transponder by comparing the “before” correlation at time t-1 and the “after” correlation at time t+1;
• an offset management system in the external DDR memory is provided so that the echo is always between t-1 and t+1. The management of the shift in the memory can be done at the same time as a shift of the coefficients of the correlator;

for frequency correction:

• use of a low pass filter before the arctangent function in order to limit the artefacts of this function;
• adjusting the parameters of the phase locked loop according to the cutoff frequency of the low pass filter determined from the measurement of the phase noise;

for adaptive filtering:

• use of a finite impulse response filter;
• use of an LMS algorithm with a limited filter length (e.g. 15 coefficients);
• the LMS algorithm can work in blocks in order to limit the resources needed for all parallel calculations. The block size can be reduced to make the algorithm more responsive. This is a trade-off between response time and FPGA resources.

5. 5 Receiving device

An exemplary implementation of the invention with an FPGA has been described above.

FIG. 13 more generally illustrates the simplified structure of a reception device according to one embodiment of the invention.

As illustrated in FIG. 13, a first reception device of a first broadcasting site comprises a memory 131 (comprising for example a buffer memory) and a processing unit 132 (equipped for example with at least one processor, FPGA, or DSP), controlled or pre-programmed by an application or a computer program 133 implementing the signal processing method according to one embodiment of the invention.

On initialization, the code instructions of the computer program 133 are for example loaded into a RAM memory before being executed by the processing unit 132. The processing unit 132 implements the steps of the method of processing described above, according to the instructions of the computer program 133.

To do this, according to one embodiment, the processing unit 132 is configured to:

receive at least one reference signal emitted by at least one second emission device from a second broadcasting site, via the transponder,

measure the phase noise associated with the transmission of the reference signal,

receive a combined signal comprising:

an interfering signal, corresponding to a first signal transmitted by a first transmitting device of the first broadcasting site, said first transmitting device, intended for at least one receiving device of the second broadcasting site, said second transmitting device reception, via the transponder, and

at least one desired signal, corresponding to a second signal transmitted by the second transmitting device, intended for the first receiving device, via the transponder,

align the first signal on the combined signal taking into account the phase noise, delivering a first aligned signal,

subtracting the aligned first signal from the combined signal, providing an estimate of said at least one desired signal.

5.6 Variants

A point-to-point mode has been described above implementing two broadcasting sites, each comprising a transmitting device and a receiving device. In particular, each broadcast site implements a reception device as described above. It is therefore a symmetrical "full-duplex" transmission system, where the two modems, which receive the combination of the two signals (S1 transmitted by the first transmission device and S2 transmitted by the second transmission device ), extract the signal they want to demodulate (S2 for the first receiving device and S1 for the second receiving device) by subtracting the transmitted signal (S1 for the first transmitting device and S2 for the second transmitting device ).

If the suppression is perfect and if the two modems transmit on exactly the same band, 50% of the spectral band can be used for other applications.

Alternatively, the proposed solution can be applied in point-to-multipoint mode.

In the point-to-multipoint mode, a main modem (also called “Hub”), located on a main broadcast site, supplies a multitude of secondary modems (also called “end-user”). The transmission remains a “full duplex” transmission, but asymmetrical.

As for the point-to-point mode, it is possible to gain in spectral efficiency by using the same transmission frequencies for the different transmitting devices and the same receiving frequencies for the different receiving devices.

FIG. 14 illustrates the architecture of such a point-to-multipoint mode by simplifying it to a main modem 141 communicating with three secondary modems 142, 143, 144 via a transponder 145. The application case typical is of the order of a few tens to a few hundred modems.

In the point-to-multipoint mode, it is generally possible for the secondary modems to directly demodulate the signal transmitted by the main modem, because its carrier Cm has a higher power than all the other carriers.

Thus the reception device according to at least one embodiment can only be implemented at the level of the main broadcast site for this point-to-multipoint mode, without being implemented at the level of the secondary broadcast sites.

According to other variants, in order to improve the operation of the echo cancellation technique, it is also possible:

• to add a gain amplifier (AGC) at the input of the reception device to limit false detections, and/or
• to limit the power difference between the carriers of the signals transmitted in the point-to-point mode, and/or
• to limit the bandwidth difference between the carriers of the transmitted signals and/or
• to operate in the linear range of the transponder amplifier.

Claims (16)

Process for processing a signal, implemented by a reception device of a first broadcasting site, said first reception device, comprising:
receiving (221) a combined signal comprising:
- an interfering signal, corresponding to a first signal transmitted by a transmitting device of said first broadcasting site, said first transmitting device, intended for at least one receiving device of at least one second broadcasting site, said second receiving device, via a transponder, and
- at least one desired signal, corresponding to at least a second signal transmitted by a transmitting device of said at least one second broadcasting site, said second transmitting device, to said first receiving device, via said transponder,
aligning (222) said first signal with said combined signal, providing an aligned first signal,
subtracting (223) said aligned first signal from said combined signal, yielding an estimate of said at least one desired signal,
characterized in that said method comprises a prior calibration step comprising:
- the reception (211) of a reference signal transmitted by said second transmitting device, via said transponder,
- the measurement (212) of the phase noise associated with the transmission of said reference signal,
and in that said alignment (222) of said first signal to said combined signal accounts for said phase noise. Method according to Claim 1, characterized in that the said measurement of the phase noise implements:
- the determination of a phase noise template, centered on the carrier frequency used by said downlink transponder,
- estimating a cutoff frequency associated with said template. Method according to Claim 2, characterized in that the determination of a phase noise mask implements the measurement of the power of the reference signal received on at least one frequency belonging to a narrow frequency band centered on the carrier frequency used by said downlink transponder. Method according to claim 3, characterized in that said cut-off frequency is estimated from the standard deviation associated with said measurements of the power of the reference signal received on said narrow band. Method according to any one of Claims 2 to 4, characterized in that the said measurement of the phase noise also implements the estimation of a residual phase noise, associated with the said reference signal received outside the frequency band defined by the carrier frequency used by said downlink transponder plus or minus the cutoff frequency. Method according to any one of Claims 1 to 5, characterized in that the said alignment implements a first synchronization step, called coarse synchronization, comprising:
- a correlation between the combined signal and the first time-shifted and/or frequency-shifted signal, for at least two time-shifted and/or frequency-shifted values,
- a determination of a correlation peak,
- an estimate of a rough offset in time and/or in frequency between the combined signal and the first signal. Method according to claim 6, characterized in that, said first signal carrying a header not obtained from a scrambling sequence, said correlation is deactivated during reception of said header. Method according to any one of Claims 1 to 7, characterized in that the said first signal is constructed from a scrambling sequence distinct from that used to construct the said at least one second signal. Method according to any one of Claims 1 to 8, characterized in that the said alignment also implements:
- a frequency correction of said first signal, implementing a phase-locked loop (32),
- a time, phase and/or gain correction of said first frequency-corrected signal, implementing adaptive filtering (33). Method according to claim 9, characterized in that said frequency correction implements at least one iteration of the following steps:
- frequency shift (41) of the first signal according to a setpoint ,
- conjugate complex multiplication (42) of said combined signal with the first signal shifted in frequency according to the instruction ,
- low-pass filtering (43) of the result of the multiplication,
- estimation of the phase shift (44) between said combined signal and the first signal shifted in frequency from the signal obtained at the output of the low-pass filtering,
- error correction (45) delivering an instruction update. Method according to Claim 10, characterized in that the said low-pass filtering implements a low-pass filter allowing frequencies below the said cut-off frequency to pass according to any one of Claims 2 to 4. Method according to either of Claims 10 and 11, characterized in that the said error correction implements a corrector of the "Proportional, Integral, Derivative" (PID) type, the coefficients of which depend on the cut-off frequency according to the any of claims 2 to 4. Method according to any one of Claims 9 to 12, characterized in that the said correction in time, phase and/or gain implements an algorithm of the minimization of the quadratic error type, implementing at least one iteration of the following steps :
- adaptive filtering of the first frequency-corrected signal, implementing an adaptive filter, delivering said first aligned signal,
- updating of the coefficients of said adaptive filter,
said updating of said coefficients being implemented after said step of subtracting said first aligned signal from said combined signal, and multiplying said estimate of said at least one desired signal by a convergence parameter . Method according to Claim 13, characterized in that the said convergence parameter is determined from the said residual phase noise according to Claim 5. Computer program comprising instructions for implementing a method according to any one of Claims 1 to 14 when this program is executed by a processor. Device for receiving a signal from a first broadcasting site, said first reception device, comprising:
receiving means configured to receive a combined signal comprising:
- an interfering signal, corresponding to a first signal transmitted by a transmitting device of said first broadcasting site, said first transmitting device, intended for at least one receiving device of at least one second broadcasting site, said second receiving device, via a transponder, and
- at least one desired signal, corresponding to at least a second signal transmitted by a transmitting device of said at least one second broadcasting site, said second transmitting device, to said first receiving device, via said transponder,
alignment means configured to align said first signal with said combined signal, delivering a first aligned signal,
subtraction means configured to subtract said first aligned signal from said combined signal, delivering an estimate of said at least one desired signal,
characterized in that, said receiving means being configured to receive beforehand a reference signal transmitted by said second transmitting device, via said transponder, said first receiving device also comprises phase noise measuring means configured to measure the phase noise associated with the transmission of said reference signal,
and in that said means for aligning is configured to align said first signal with said combined signal taking into account said phase noise.