EP4716987A2 - Control of inertia-less high-frequency isolated converters - Google Patents

Abstract

Systems and methods for controlling power converter system. The system comprises a DC bridge electrically coupled to a DC power source. The system further comprises an AC bridge. The system further comprises a transformer comprising a primary winding interfacing with the DC bridge and a secondary winding interfacing with the AC bridge. The system also comprises a controller configured to control the AC and DC bridges to convert a DC voltage from the DC power source to an AC voltage at an output of the AC bridge. The controller is configured to control the DC bridge to provide an input voltage to the primary winding of the transformer. The input voltage comprises a positive voltage portion and a negative voltage portion. The controller causes the AC bridge to flip a polarity of the output.

Description

CONTROL OF INERTIAL-LESS HIGH-FREQUENCY ISOLATED CONVERTERS

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application claims the priority benefit of U.S. Provisional Patent Application Serial No. 63/526,796, filed July 14, 2023, which is hereby incorporated by reference in its entirety.

BACKGROUND OF THE DISCLOSURE

1. FIELD OF THE INVENTION

[0002] The present invention relates generally to systems and methods, and more particularly to systems for transferring energy between a DC source and an AC source.

2. DESCRIPTION OF RELATED ART

[0003] Rapidly declining costs of PV solar, energy storage, and power converters have fueled an exponential global growth in the deployment of distributed energy resources (DERs). In most cases, the DERs have a primary energy resource that is characterized by a DC voltage but is then connected to the AC grid on the other side. A typical DC-to-AC conversion system comprises a power electronic building block to achieve AC voltage with desired magnitude and frequency. In most of the scenarios, such conversion applications also require an isolation stage to either provide connection of different voltage levels or as a galvanic isolation element to decouple faults. Such a need is conventionally met with a low-frequency bulky transformer connected to the output of the AC stage, as shown in FIG. 1A.

[0004] The state-of-the-art solution to address the DC/AC conversion without using a bulky 60 Hz transformer involves the utilization of multi-stage power conversion stages with a high- frequency link obtained with a dual active bridge (DAB) DC/DC converter as shown in FIG. IB followed by a conventional DC/AC bridge. Further, the DAB converter uses a finite leakage inductor (Llk), with power transfer achieved through the control of phase angle difference between the two high-frequency AC waveforms across Llk. This involves careful consideration of Llk value and closed-loop control over the required phase shift for the desired power transfer. However, such a configuration, due to the multiple power conversion stages and intermediate energy storage, suffers from poor dynamic performance, as well as high cost and size of the system. [0005] Recently, with the availability of fast-switching devices such as Silicon Carbide MOSFETs (SiC-MOSFET), single-stage DC/AC conversion has been achieved with a DC side full bride and AC side cycloconverter separated by high-frequency isolation, but once again using a finite leakage inductor. The principle of operation for this configuration is similar to the DAB. However, the phase shift operation with the cycloconverter involves a closed-loop control with complex realtime computation as an integral part for accurate power transfer. Other approaches with the single- stage DC/AC conversion are achieved with the Soft-Switching Solid-State-Transformer (S4T), which utilizes a flyback principle to achieve single-stage power transfer. However, for the S4T, the transformer and the zero-voltage switching (ZVS) resonant switch become limiting factors to its scalability. The current-source characteristics of the S4T also require reverse blocking switches, which are realized with Silicon IGBTs, and Silicon Carbide diodes connected in series. Further, the S4T still uses the magnetizing inductance of the transformer as an energy transfer element leading to finite inertia between the DC/AC conversion.

[0006] Some of the afore-discussed technologies realize single-stage DC/AC power conversion but require either a complex control mechanism or comprise additional energy storage elements, that represent “inertia” and can slow down the overall controllability and dynamic performance of the plant. Particularly for cost-sensitive applications such as low-cost energy access portals, solar microinverters, and residential EV chargers, among others, the existing approaches have been constrained by the cost and size impact of these storage elements. Accordingly, there is a need for improved power converters that address one or more of the disadvantages of conventional systems.

SUMMARY OF THE INVENTION

[0007] Briefly described, according to exemplary embodiments of the present invention, systems and methods of an innovative system for control of power converts.

[0008] In an exemplary embodiment, a power converter system comprises a DC bridge electrically coupled to a DC power source. The power converter may further comprise an AC bridge. The power converter may further comprise a transformer comprising a primary winding interfacing with the DC bridge and a secondary winding interfacing with the AC bridge. The power converter may also comprise a controller configured to control the AC and DC bridges to convert a DC voltage from the DC power source to an AC voltage at an output of the AC bridge. The controller may be further configured to control the DC bridge to provide an input voltage to the primary winding of the transformer. The input voltage may comprise a positive voltage portion and a negative voltage portion. The controller may be further configured to cause the AC bridge to flip a polarity of the output when the primary winding receives one of a positive voltage portion or the negative voltage portion of the input voltage to maintain a desired polarity at the output of the AC bridge.

[0009] In various embodiments, the output of the AC bridge may comprise an inductive or inductive-capacitive filtering circuit configured to provide a sinusoidal output.

[0010] In various embodiments, the input voltage to the primary winding may be a quasi-square wave. The positive voltage portion may be a positive voltage pulse, and the negative voltage portion may be a negative voltage pulse.

[0011] In various embodiments, the DC bridge may comprise four single direction switches.

[0012] In one or more embodiments, the AC bridge may comprise four bidirectional switches.

[0013] In various embodiments, the input voltage to the primary winding may further comprise a zero-voltage portion between the positive voltage portion and the negative voltage portion.

[0014] In various embodiments, the controller may be configured to change a current switch state of the bidirectional switches when the voltage of the secondary winding is in the zero-voltage portion of the input voltage.

[0015] In one or more embodiments, the AC voltage may be a single-phase AC voltage.

[0016] In various embodiments, the controller may be further configured to generate a gating signal to control the bidirectional switches.

[0017] In various embodiments, the controller may be configured to generate the gating signal, at least in part, by generating an error signal that is a difference between a terminal voltage at the output of the AC bridge and a reference voltage.

[0018] In various embodiments, the controller may be further configured to generate the gating signal, at least in part, by integrating the error signal.

[0019] In various embodiments, the gating signal may be further based, at least in part, on a determination of whether the integrated error signal is positive or negative.

[0020] In various embodiments, the AC bridge may comprise six bidirectional switches.

[0021] In various embodiments, the AC voltage may be a three-phase AC voltage.

[0022] In various embodiments, the controller may be further configured to generate a gating signal to control the bidirectional switches.

[0023] In one or more embodiments, the controller may be configured to generate the gating signal by performing the following steps: obtaining a reference AC voltage vector, selecting, based on the reference vector, two line-voltage errors, inputting the two selected line-voltage errors to two integrators, selecting, based on outputs of the two integrators, a vector such that both line-voltage errors are minimized, and applying the vector until an output of one of the two integrators becomes less than zero.

[0024] In one or more embodiments, the transformer may comprise a leakage inductance less than 1% of a magnetizing inductance of the transformer.

[0025] In various embodiments, the transformer may comprise a leakage inductance of less than 0.1% of a magnetizing inductance of the transformer.

[0026] In various embodiments, the AC bridge may further comprise a clamping circuit configured to limit voltage spikes across switches of the AC bridge.

[0027] In various embodiments, the controller may be configured to alter the output of the AC bridge by utilizing pulse width modulation.

[0028] In various embodiments, the pulse width modulation may be sigma-delta modulation.

[0029] In various embodiments, the input voltage may be a square wave.

[0030] In one or more embodiments, the input voltage may be a quasi-square wave.

[0031] In various embodiments, the AC bridge may not be connected in shunt with an energy storage capacitor.

[0032] In various embodiments, the DC voltage from the DC power source may comprise a first magnitude. The AC voltage at the output of the AC bridge may comprise a second magnitude greater than the first magnitude.

[0033] In one or more embodiments, the converter may further comprise at least one active damping auxiliary circuit connected in parallel with the AC bridge. The at least one active damping circuit may be configured to damp oscillations that occur due to excitation of parasitic elements of the converter. The at least one damping circuit may comprise a damping resistor in series with a bi-directional switch.

[0034] In one or more embodiments, the converter may further comprise at least one active damping auxiliary circuit connected in series with the AC bridge. The at least one active damping circuit may be configured to damp oscillations that occur due to excitation of parasitic elements of the converter. The at least one damping circuit may comprise a damping resistor in parallel with a bi-directional switch.

[0035] These and other aspects, features, and benefits of the claimed invention(s) will become apparent from the following detailed written description of the preferred embodiments and aspects taken in conjunction with the following drawings, although variations and modifications thereto may be affected without departing from the spirit and scope of the novel concepts of the disclosure. BRIEF DESCRIPTION OF THE DRAWINGS

[0036] Implementations, features, and aspects of the disclosed technology are described in detail herein and are considered a part of the claimed disclosed technology. Other implementations, features, and aspects can be understood with reference to the following detailed description, accompanying drawings, and claims. Wherever possible, the same reference numbers are used throughout the drawings to refer to the same or like members of an embodiment. Reference will now be made to the accompanying figures and flow diagrams, which are not necessarily drawn to scale.

[0037] FIG. 1A provides a schematic illustrates of a conventional DC to AC power converter system known in the prior art.

[0038] FIG. IB provides a schematic illustrates of a conventional DC to AC power converter system known in the prior art.

[0039] FIG. 1C provides a schematic illustrates of a conventional DC to AC power converter system known in the prior art.

[0040] FIG. 2A provides a schematic illustration of a power source in accordance with various embodiments of the present disclosure.

[0041] FIG. 2B provides a schematic illustration of a power source in accordance with various embodiments of the present disclosure.

[0042] FIG. 2C provides a schematic illustration of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0043] FIG. 2D provides a schematic illustration of a portion of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0044] FIG. 2E provides a graphical illustration of exemplary primary and secondary transformer winding voltages, transformer current, and AC bridge output voltage of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0045] FIG. 2F provides a graphical illustration of exemplary voltage output of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0046] FIG. 3A provides a schematic illustration of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0047] FIG. 3B provides a schematic illustration of a portion of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0048] FIG. 3C provides a schematic illustration of a portion of an exemplary power converter system in accordance with various embodiments of the present disclosure. [0049] FIG. 4 provides a graphical illustration of exemplary waveforms of single-phase AC converter in accordance with various embodiments of the present disclosure.

[0050] FIG. 5A provides a schematic illustration of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0051] FIG. 5B provides a schematic illustration of a cycle control scheme of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0052] FIG. 5C provides a graphical illustration of exemplary waveforms of three-phase AC converter in accordance with various embodiments of the present disclosure.

[0053] FIG. 6A provides a schematic illustration of an exemplary power converter system in accordance with various embodiments of the present disclosure.

[0054] FIG. 6B provides a schematic illustration of an exemplary power converter system in accordance with various embodiments of the present disclosure.

DETAILED DESCRIPTION OF THE INVENTION

[0055] Although preferred exemplary embodiments of the disclosure are explained in detail, it is to be understood that other exemplary embodiments are contemplated. Accordingly, it is not intended that the disclosure is limited in its scope to the details of construction and arrangement of components set forth in the following description or illustrated in the drawings. The disclosure is capable of other exemplary embodiments and of being practiced or carried out in various ways. Also, in describing the preferred exemplary embodiments, specific terminology will be resorted to for the sake of clarity.

[0056] To facilitate an understanding of the principles and features of the present disclosure, various illustrative embodiments are explained below. The components, steps, and materials described hereinafter as making up various elements of the embodiments disclosed herein are intended to be illustrative and not restrictive. Many suitable components, steps, and materials that would perform the same or similar functions as the components, steps, and materials described herein are intended to be embraced within the scope of the disclosure. Such other components, steps, and materials not described herein can include, but are not limited to, similar components or steps that are developed after development of the embodiments disclosed herein.

[0057] As used in the specification and the appended claims, the singular forms “a,” “an” and “the” include plural referents unless the context clearly dictates otherwise.

[0058] Also, in describing the preferred exemplary embodiments, terminology will be resorted to for the sake of clarity. It is intended that each term contemplates its broadest meaning as understood by those skilled in the art and includes all technical equivalents which operate in a similar manner to accomplish a similar purpose.

[0059] Ranges can be expressed herein as from “about” or “approximately” one particular value and/or to “about” or “approximately” another particular value. When such a range is expressed, another exemplary embodiment includes from the one particular value and/or to the other particular value.

[0060] Similarly, as used herein, “substantially free” of something, or “substantially pure”, and like characterizations, can include both being “at least substantially free” of something, or “at least substantially pure”, and being “completely free” of something, or “completely pure”.

[0061] By “comprising” or “containing” or “including” is meant that at least the named compound, member, particle, or method step is present in the composition or article or method, but does not exclude the presence of other compounds, materials, particles, method steps, even if the other such compounds, material, particles, method steps have the same function as what is named.

[0062] Mention of one or more method steps does not preclude the presence of additional method steps or intervening method steps between those steps expressly identified. Similarly, it is also to be understood that the mention of one or more components in a device or system does not preclude the presence of additional components or intervening components between those components expressly identified.

[0063] The materials described as making up the various members of the invention are intended to be illustrative and not restrictive. Many suitable materials that would perform the same or a similar function as the materials described herein are intended to be embraced within the scope of the invention. Such other materials not described herein can include, but are not limited to, for example, materials that are developed after the time of the development of the invention.

[0064] Reference will now be made in detail to exemplary embodiments of the disclosed technology, examples of which are illustrated in the accompanying drawings and disclosed herein. Wherever convenient, the same references numbers will be used throughout the drawings to refer to the same or like parts.

[0065] FIGS. 1A-1C illustrate exemplary configuration of prior art direct current (DC) to alternating current (AC) converter systems in accordance with various embodiments. An example DC to AC converter system 10 comprises a power electronic building block to achieve AC voltage with desired magnitude and frequency. In most of the scenarios, such conversion applications also require an isolation stage to either provide connection of different voltage levels or as a galvanic isolation element to decouple faults. In various embodiments, as depicted in FIG. 1 A, conventional converter systems comprise a low frequency transformer 16 connected to the output of the finished DC to AC stage 14. Other converter systems 10, as depicted in FIG. IB, remove the bulk low- frequency transformer by utilizing a multi-stage power conversion stage with a high-frequency link 18 obtained with a dual active bridge (DAB) DC/DC converter 20 followed by a conventional DC/AC bridge 14. Additionally, the DAB converter system uses a finite leakage inductor (Lik), with power transfer achieved through the control of phase angle difference between the two high- frequency AC waveforms across Li . This involves careful consideration of Lik value and closed- loop control over the required phase shift for the desired power transfer. However, such converter systems, due to the multiple power conversion stages and intermediate energy storage, suffers from poor dynamic performance, as well as high cost and size of the system.

[0066] In other embodiments, as depicted in FIG. 1C, use single-stage DC/AC conversion. The DC/AC conversion is achieved with a DC side full bride 14 and AC side cycloconverter 22 separated by high-frequency isolation 18. These systems may utilize fast-switching devices such as Silicon Carbide MOSFETs (SiC-MOSFET) to assist with the conversion. However, these systems use a finite leakage inductor. Additionally, the phase shift operation with the cycloconverter involves a closed-loop control with complex real-time computation as an integral part for accurate power transfer. As a consequence, the conventional systems described above require either a complex control mechanism or comprise additional energy storage elements, that represent 'inertia' and can slow down the overall controllability and dynamic performance of the plant. Particularly for cost-sensitive applications such as low-cost energy access portals, solar microinverters, residential electrical vehicle (EV) chargers, etc., the existing approaches have been constrained by the cost and size impact of these storage elements.

[0067] This disclosure presents new systems and methods for achieving an isolated DC to AC conversion with nearly zero (ultra-low) inertia operation for the isolated high-frequency converters. As shown in FIGS. 2A-D, an exemplary high-frequency inertia-less isolated converter (IIC) comprises DC 210 and AC 220 bridges and a high-frequency transformer 230. The DC bridge 210 can comprise four single direction switches 212A-D, which can be any switches known in the art, including, but not limited to, MOSFET switches. The DC bridge 210 can be controlled to act as a simple high-frequency (fsw=l/Tsw) quasi-square wave or square wave generator, which can be achieved with a full bridge or three-level inverter configuration, which is well known in the prior art. Further, in order to provide bi-directional power transfer with DC or AC output, the AC side bridge 220 can comprise a bidirectional switch-based bridge, comprising four bidirectional switches 222A-D. [0068] At the heart of the proposed configuration is a simple and low-cost control of the AC bridge 220 that can simultaneously satisfy the control of DC or low-frequency AC waveforms on the output 240 of the AC bridge while also ensuring no DC or low-frequency flux in the high-frequency transformer windings. For proper operation, it can be important that an ultra-low leakage inductance (Llk=0) high-frequency transformer be used between the DC and AC bridges to ensure fidelity between the waveforms on both sides of the transformer. In various embodiments, the transformer comprises a leakage inductance less than 1%, 0. l%-0.5%, or less than 0.1% of a magnetizing inductance of the at least one transformer. Such ultra-low Llk values can be easily achieved with Coaxial Winding Power Transformers (CWT) or with coaxial cable wound transformers. As used herein, “high-frequency transformer” refers to a galvanically isolated power transformer typically operated above 10 kHz, and typically between 20-40 kHz depending on the rated current and voltage for the transformer. Nonetheless, a clamping circuit 250, as shown in FIG. 2D with locally managed energy dissipation, can be added on the transformer side connection of the AC bridge 220 to avoid any occurrence of overvoltage on the bridges due to any remnant energy in Llk.

[0069] With the ultra-low Llk value and the bidirectional switches in the AC side bridge 220 of the IIC, the configuration provides one of the simplest possible means to achieve high-frequency isolated DC to AC conversion.

[0070] More importantly, if such a converter is used with a current limited DC source, it can provide the lowest cost implementation of a practical DC/AC inverter, including high-frequency isolation. Two exemplary cases utilizing DC-to-low-frequency ff /Tf) square-wave AC conversion are shown in FIGS. 2C and 3A. As can be seen, the input DC source can be a PV panel with inherent current limiting capability, or a battery with a current limiting converter, providing a low-cost and simplest realization of isolated low-frequency ff=\/Tf) square-wave AC supply for cost-sensitive applications such as for energy access.

[0071] As mentioned earlier, the typical control of the AC side bridge 220 involves the application of a gating signal in such a manner that it realizes the desired output waveform (e.g., the low- frequency square wave, e.g., about 50-60 Hz) while at the same time ensuring that the transformer does not saturate. This can utilize “flipping logic,” which can be used to ensure that both conditions are simultaneously met. Such operation ensures that the transformer winding is not saturated and operates with essentially zero DC flux over every switching cycle. FIGS. 2A-F shows the circuit schematic and representative waveforms for the proposed IIC, including gating pulses for the DC 210 and AC 220 side bridge switches for both positive and negative output voltage generation. [0072] It can be seen from FIGS. 2E that for positive output voltage generation, the AC bridge is operated to flip the input during the negative pulse while passing the positive pulse without flipping. On the other hand, for negative output voltage generation, the AC bridge is operated to always flip its input during the positive pulse while passing the negative pulse without flipping. This flipping mechanism is like the 'unfolding' concept known in the literature. However, the utilization of this concept with the ultra- low Llk offers a new operating principle of the inertia-less configuration providing management of converter parasitic inductances and capacitances, resulting in a simple control for high-frequency isolated DC to DC/AC conversion with net-zero DC flux over each switching cycle.

[0073] In a typical application with an inductive filter or inductive load on the output side, an intentional overlap duration can be introduced in the DC 210 and AC 220 side bridges to protect the switches against overvoltage peaks. For this, an intentional short-duration (1-2% of Tsw) zerostate is introduced in the DC 210 and AC 220 bridges during the positive (negative) to negative (positive) voltage level change (yielding quasi- square-wave operation). This overlap duration ensures safe commutation of output current Io in the AC side switches 222A-D. Further, the introduced zero-state allows zero-voltage soft-switching for the AC bridge switches 222A-D, where these switches can be switched with zero voltage. Moreover, the overlap duration also helps to drain the current in Llk towards zero before it reverses its polarity in the second half of the switching cycle (Tsw). The representative waveforms for such an operation are depicted in Fig. 2F.

[0074] It is to be noted that the above-discussed operation of IIC achieves a low- frequency square wave output with the desired value of frequency but without any control over the output voltage magnitude, wherein the current limiting voltage-current characteristics on the input enables its operation even under overload or short-circuit conditions. In order to realize a controlled output voltage, an additional modulation strategy can be incorporated in such a manner that the fundamental operation of the bridge to achieve zero DC flux over each switching cycle remains intact. This has typically been achieved with pulse width modulation (PWM) schemes, such as sinusoidal PWM, which can be used to obtain the controlled output voltage. However, utilization of such a scheme, where the pulse widths are not synchronized with the square wave switching, can lead to DC or low- frequency flux in the transformer over a switching cycle, even when flipping logic is used. It may be possible to reduce such flux to zero over a few cycles either by controlling the DC bridge 210 to synchronize with the AC bridge 220 or performing real-time computations over each switching cycle to reduce the DC flux to zero. However, such implementations may involve a closed-loop approach with complex real-time computations. Further, the requirement of pulses with nonuniform widths introduces an excessively higher number of switching events. This makes the overall implementation of the inertia-less converter impractical with the conventional PWM schemes.

[0075] Discrete Pulse Control of Inertia-less High-frequency Isolated Converters

[0076] In some embodiments of the present disclosure, the output voltage control of the abovediscussed IIC can be achieved by utilizing a discrete pulse modulation control strategy, such as sigma-delta modulation, but integrated with the flipping logic to eliminate transformer core saturation. The utilization of discrete pulse control enables the delivery of an integral number of discrete pulses to the output, such that fine control of the output voltage spectral components is realized. The discrete pulse modulation control acts on the instantaneous error between the reference and terminal voltage across the AC bridge 220 before connecting to any passive element. This terminal voltage at the AC bridge output 240 can either be sensed or obtained using the previously applied switching states in the controller itself. The integrated error output can be passed to the AC bridge in the form of switching states to reduce the error. Further, the flipping logic can impress the right polarity on the output while ensuring that DC flux is not present. It can also eliminate issues of partial pulses from PWM operation, causing a low-frequency flux in the transformer. Also, the DC bridge 210 can be controlled to act as a simple quasi-square wave (comprising a positive voltage portion, a negative voltage portion, and a zero-voltage portion therebetween) or square wave generator, as discussed earlier. However, their combination basically led to achieving controlled output voltage with bidirectional power transfer capability of the IIC using simpler control and net zero DC flux across the high-frequency transformer. This can ensure an open-loop inertia-less power transfer.

[0077] FIG. 3A shows a typical circuit schematic for the IIC realizing DC to single-phase controlled AC conversion with the LC filtering stage 255 comprising Lf and Cf providing AC sinusoidal output. It is to be noted that, unlike the conventional DAB configurations, the AC bridge 220 of the IIC can directly connect to the AC side through an inductive element without any intermediate passive element or energy storage element. The earlier discussed overlap with short- duration zero states in DC 210 and AC 220 side bridges ensures safe commutation of output current Io even with the presence of an inductive element on the output of the AC bridge 220. Nonetheless, an additional clamping circuit 250 (See FIGS. 2D & 3C) on the output side of the AC bridge 220 can be added to provide an additional layer of protection against any missing overlap event.

[0078] An exemplary control scheme block diagram for the DC to single-phase controlled AC using the sigma-delta modulation is shown in FIG. 3C for the single -phase bridge shown in FIG. 3A. Herein, the input signals v_ref and v_ab are two real-time analog signals which involve the reference output voltage and the AC bridge 220 output voltage signals. These signals provide an error signal. The error signal is then applied to an integrator, whose output is then applied to the AC bridge switches 222A-D through the earlier discussed flipping logic. In this way, the error between the reference voltage and the output voltage can be minimized in real-time with net zero flux across the high-frequency transformer 230 over the switching cycle. Finally, the gating signals for the AC bridge 220 can be synchronized with the high-frequency link zero crossings to ensure discrete pulses on the output side.

[0079] The representative waveforms for the operation of the DC to single-phase AC IIC are shown in FIG. 4. As can be seen, the integrator output of the sigma-delta modulation drives the error towards zero at each sample instant defined by the sample and hold element defining the maximum allowed switching frequency (fsw=l/Tsw).

[0080] The direct extension of the afore-discussed concept for the three-phase AC side bridge can be realized with the use of a three-integrator-based control scheme, where each integrator receives the error in each phase output. While this can work well for a three-phase four-wire system, for a three-phase three-wire system, it may not be possible to independently control all the three-phase errors. This may lead to the presence of low-frequency errors and the possibility of saturation of one of the three integrals leading to distorted output voltage. In order to address this issue, a two integral-based discrete pulse modulation control can be utilized for the considered DC/three-phase AC IIC. Such realization can be conceptually possible with the utilization of well-known synchronous frame transformation of three-phase quantities leading to only two quantities to control for the balanced three-phase system. However, such implementation can lead to complex control involving synchronization loops and requires information on frequency, which may not always be known.

[0081] In order to address this issue, this disclosure presents a two integral-based discrete pulse modulation for the three-phase bridge shown in FIG. 5A. The three-phase configuration comprises an LCL filtering stage 510 with Lf, Cf, and Lf passive filtering elements. A block diagram of an exemplary control scheme is shown in FIG. 5B. One of the main objectives for the presented control scheme is to ensure that, unlike in the three integral-based system, both integrators correct the error in real-time at all instants. This is achieved using an 'error selector' logic, which selects two-line voltage errors based on the location of the reference vector. For this, the space vector plane is divided into six sectors (I to VI) defined by the line voltage error vectors (eab, ebc, eca, eba, ecb, and eac). [0082] Considering an exemplary case when the reference vector (V*) is in sector T. In this scenario, the error selector block applies errors e_ab and e to the integrator input. The outputs of these integrators are utilized to select a suitable vector among VI, V2, V6, and VO or V7 such that both errors are converged. If both integrator outputs are greater than zero, vector V 1 will be selected to minimize both errors. In this scenario, vector V 1 will be applied till the moment any of the integrator outputs changes becomes less than zero. In a similar manner, depending on the status of integrator outputs and the location of the reference vector, the suitable vector is selected at each instant, which minimizes the corresponding errors. In this way, at each instant, both the error integrators are operational to minimize the errors without saturation. Finally, the selected vectors are applied with the earlier discussed flipping logic to ensure the application of a similar waveform as that of the DC side across the AC side transformer terminals modified by the turns ratio, achieving the simple, low-cost, inertia-less operation with essentially zero DC flux over every switching cycle.

[0083] The representative waveforms for the operation of the DC to three-phase AC IIC are shown in FIG. 5C. As can be seen, both the integrator outputs are driving the input errors toward zero. Further, depending on the position of the reference vector, suitable error vectors are selected by the error selector logic to be applied to the integrator inputs. Finally, the vector selection is carried out, and suitable switching states are selected with the flipping logic by considering the status of the transformer's secondary side voltage. The obtained three-phase voltages are free from any distortion, which is achieved by acting on each of the integrators simultaneously in real-time. In this manner, with just two integrators and without utilizing any transformation, three-phase AC voltage is obtained by maintaining zero phase shift. This provides an inertia-less operation of the converter.

[0084] Utilization of Damping Auxiliary Circuits

[0085] Hardware embodiments of the IIC can expose a plethora of parasitic components associated with the HF transformer (HFT), switching devices, magnetics, power board traces/planes, terminations, and wiring, among others. More specifically, the transformer leakage inductance and the equivalent lumped capacitance of the DC and AC bridges are of special interest, as they tend to dominate the high-frequency oscillations (above hundreds of kHz) that occur throughout specific switching transitions. An example of such oscillations for an IIC switching cell, shown in FIG. 2C, is demonstrated in FIG. 6C. In this particular switching transition, the DC bridge is shorting the LV-side of the HFT, and all the bidirectional switches of the AC bridge are triggered ON. Subsequently, when the AC bridge transitions to its next switching state by opening two of its diagonal switches, the energy trapped in the parasitic elements triggers the undesired oscillations showcased in FIG. 6C. When undamped, the voltage and current oscillations depicted in the waveforms can degrade the performance of the IIC converter (e.g., additional component stress and losses).

[0086] In light of this, the IIC can be equipped with an active damping auxiliary circuit 260, as displayed in FIGS. 6A-B. Herein, an additional bidirectional switch (S_d) with a damping resistor R_d (e.g., 100-150 Ohm) can be connected in series (FIG. 6A) and/or shunt (FIG. 6B) with the HV- side (AC) terminals of the HFT. In both configurations, the switch can be only activated for a small period of time (<lus) to insert R_d in the circuit and damp the oscillations. Moreover, S_d can be operated with a variable duty cycle to insert a controlled equivalent resistance into the circuit. Lastly, the right column figures of FIG. 6C show the performance of the active damping circuit and its capability to reduce parasitics-related oscillations when implemented in its series configuration. For such implementation, the peak voltage and current values can be drastically reduced when the damping is activated.

[0087] While certain embodiments of the disclosed technology have been described in connection with what is presently considered to be the most practical embodiments, it is to be understood that the disclosed technology is not to be limited to the disclosed embodiments, but on the contrary, is intended to cover various modifications and equivalent arrangements included within the scope of the appended claims. Although specific terms are employed herein, they are used in a generic and descriptive sense only and not for purposes of limitation.

[0088] This written description uses examples to disclose certain embodiments of the disclosed technology, including the best mode, and also to enable any person skilled in the art to practice certain embodiments of the disclosed technology, including making and using any devices or systems and performing any incorporated methods. The patentable scope of certain embodiments of the disclosed technology is defined in the claims, and may include other examples that occur to those skilled in the art. Such other examples are intended to be within the scope of the claims if they have structural elements that do not differ from the literal language of the claims, or if they include equivalent structural elements with insubstantial differences from the literal language of the claims.

Claims

WHAT IS CLAIMED IS:

1. A power converter, comprising: a DC bridge electrically coupled to a DC power source; an AC bridge; a transformer comprising a primary winding interfacing with the DC bridge and a secondary winding interfacing with the AC bridge; and a controller configured to control the AC and DC bridges to convert a DC voltage from the DC power source to an AC voltage at an output of the AC bridge, wherein the controller is further configured to control the DC bridge to provide an input voltage to the primary winding of the transformer, the input voltage comprising a positive voltage portion and a negative voltage portion, wherein the controller is further configured to cause the AC bridge to flip a polarity of the output when the primary winding receives one of a positive voltage portion or the negative voltage portion of the input voltage to maintain a desired polarity at the output of the AC bridge.

2. The power converter of claim 1, wherein the output of the AC bridge comprises an inductive-capacitive filtering circuit configured to provide a sinusoidal output.

3. The power converter of claim 1, wherein the input voltage is a quasi-square wave, the positive voltage portion is a positive voltage pulse, and the negative voltage portion is a negative voltage pulse.

4. The power converter of claim 1, wherein the DC bridge comprises four single direction switches.

5. The power converter of claim 1, wherein the AC bridge comprises four bidirectional switches.

6. The power converter of claim 5, wherein the input voltage further comprises a zero-voltage portion between the positive voltage portion and the negative voltage portion.

7. The power converter of claim 6, wherein the controller is configured to change a current switch state of the bidirectional switches when a voltage of the secondary winding is in the zerovoltage portion of the input voltage.

8. The power converter of claim 5, wherein the AC voltage is a single-phase AC voltage.

9. The power converter of claim 5, wherein the controller is further configured to generate a gating signal to control the bidirectional switches.

10. The power converter of claim 8, wherein the controller is configured to generate the gating signal, at least in part, by generating an error signal that is a difference between a terminal voltage at the output of the AC bridge and a reference voltage.

11. The power converter of claim 9, wherein the controller is further configured to generate the gating signal, at least in part, by integrating the error signal.

12. The power converter of claim 10, wherein the gating signal is further based, at least in part, on a determination of whether the integrated error signal is positive or negative.

13. The power converter of claim 1, wherein the AC bridge comprises six bidirectional switches.

14. The power converter of claim 13, wherein the AC voltage is a three-phase AC voltage.

15. The power converter of claim 14, wherein the controller is further configured to generate a gating signal to control the bidirectional switches.

16. The power converter of claim 15, wherein the controller is configured to generate the gating signal by performing the following steps: obtaining a reference AC voltage vector; selecting, based on the reference vector, two line-voltage errors; inputting the two selected line-voltage errors to two integrators; selecting, based on outputs of the two integrators, a vector such that both line-voltage errors are minimized; applying the vector until an output of one of the two integrators becomes less than zero.

17. The power converter of claim 1 , wherein the transformer has a leakage inductance less than 1% of a magnetizing inductance of the transformer.

18. The power converter of claim 1, wherein the transformer has a leakage inductance of less than 0.1% of a magnetizing inductance of the transformer.

19. The power converter of claim 1 , wherein the transformer has a leakage inductance of 0. 1 %- 0.5% of a magnetizing inductance of the transformer

20. The power converter of claim 1, wherein the AC bridge further comprises a clamping circuit configured to limit voltage spikes across switches of the AC bridge.

21. The power converter of claim 1, wherein the controller is configured to alter the output of the AC bridge by utilizing pulse width modulation.

22. The power converter of claim 21, wherein the pulse width modulation is sigma-delta modulation.

23. The power converter of claim 1, wherein the input voltage is a square wave.

24. The power converter of claim 1, wherein the input voltage is a quasi-square wave.

25. The power converter of claim 1, wherein the AC bridge is not connected in shunt with an energy storage capacitor.

26. The power converter of claim 1, wherein the DC voltage from the DC power source has a first magnitude and wherein the AC voltage at the output of the AC bridge has a second magnitude greater than the first magnitude.

27. The power converter of claim 1, further comprising at least one active damping auxiliary circuit connected in parallel with the AC bridge, the at least one active damping circuit configured to damp oscillations that occur due to excitation of parasitic elements of the converter, the at least one damping circuit comprising a damping resistor in series with a bidirectional switch.

28. The power converter of claim 1, further comprising at least one active damping auxiliary circuit connected in series with the AC bridge, the at least one active damping circuit configured to damp oscillations that occur due to excitation of parasitic elements of the converter, the at least one damping circuit comprising a damping resistor in parallel with a bidirectional switch.