EP4666813A1

EP4666813A1 - Converter for a light-emitting diode load

Info

Publication number: EP4666813A1
Application number: EP24722030.4A
Authority: EP
Inventors: Dietmar Klien
Original assignee: Tridonic GmbH and Co KG
Current assignee: Tridonic GmbH and Co KG
Priority date: 2023-04-27
Filing date: 2024-04-26
Publication date: 2025-12-24
Also published as: EP4456671A1; WO2024223798A1

Abstract

A converter is proposed for supplying a LED load, the converter comprising a PFC circuit (11) supplying a further switched converter stage or being the only switched converter of a single stage converter, wherein the PFC circuit (11) comprises a flyback converter with a power switch (112) and an active clamp circuit (120, 121) having a clamp switch (120), wherein the switch-on time period of the power switch (112) is determined by a peak-control of the flyback current, i.e., the power switch (112) is switched-off as soon as it is sensed that the raising flyback current reaches a given threshold.

Description

CONVERTER FOR A LIGHT-EMITTING DIODE LOAD

Technical Field

The present disclosure relates to lighting technology, and in particular, to a converter for a lightemitting diode (LED) load, and a luminaire based thereon.

Background Art

Power factor correction (PFC) circuits attempt to maximize a power factor of a connected load by drawing from an alternating current (AC) power supply an AC current that corresponds in amplitude curve and phase to the supplied AC voltage, thereby minimizing a Total Harmonic Distortion (THD).

Such PFC circuits typically make use of peak-current control, since an average-current control needs a more complicate current sensing. However, also calculating the on-time instead of the peak-control is known in the art.

Due to parasitic effects in the components of a PFC circuit, it turns out that a peak-current control results in a non-zero voltage plateau across the EMI capacitor around the zero-crossings of the supplied AC voltage, which ultimately results in a higher THD.

Integrated circuits (ICs) for analog peak-current control do not avoid or reduce said plateau, whereas ICs for digital peak-current control typically increase an on-time of a power switch of the PFC circuit close to the zero-crossings via calculations or table-based loop-up. A parameterization of such solutions is difficult because of their relatively complicated design, and for higher switching frequencies multiple look-up tables may be needed.

Summary It is an object to overcome the above-mentioned and other drawbacks. The objective is achieved by the embodiments as defined by the appended independent claims. Preferred embodiments are set forth in the dependent claims and in the following description and drawings.

The invention proposes a converter with a PFC circuit. The PFC circuit comprises a flyback converter with active clamp (active clamp flyback converter ACF). The switch-on time period of the flyback switch is thereby determined by a peak-control of the flyback current, i.e., the flyback switch is switched-off as soon as it is sensed that the raising flyback current reaches a given threshold.

An Active Clamp Flyback (ACF) converter is a type of power converter topology that combines the features of a conventional flyback converter with an active clamp circuit. The active clamp circuit in an ACF converter helps to improve efficiency, reduce voltage stresses on components, and enhance the overall performance of the converter.

An Active Clamp Flyback converter has the function of a conventional flyback converter, i.e., energy is stored in the primary winding of the transformer during the switch-on time and transferred to the secondary winding during the switch-off time. Further, an active clamp circuit is added to the flyback converter topology. The active clamp circuit comprises an additional actively controlled switch (clamping switch) and a clamp capacitor. The active clamp circuit provides a path for the energy stored in the transformer's leakage inductance to be transferred back to the input voltage source, rather than e.g., dissipating it as heat in a passive clamp circuit.

Preferably the ACF is operated in the BCM mode, i.e., the flyback switch is switched on as soon as the sensed primary side current has fallen to zero.

A first aspect of the present disclosure relates to a converter for an LED load. The converter comprises a power factor correction, PFC, circuit. The PFC circuit comprises a series connection of an inductance and a power switch; and a control circuit. The control circuit is configured to regulate an output voltage of the PFC circuit in accordance with a current through the series connection and a peak current envelope for the same; and to offset the peak current envelope by a direct current, DC, component that depends on an output power of the PFC circuit. The peak current envelope may depend on a control variable of the control circuit. The control variable may depend on a differential of the output voltage and its reference variable, and on a multiplicative fraction of a rectified input voltage of the PFC circuit.

The DC component may depend on the control variable of the control circuit, a constant coefficient of the control variable, and a constant offset of the DC component.

The constant coefficient and the constant offset may be derived from simulations or measurements at a minimum value and a maximum value of the output power of the PFC circuit.

The control circuit may comprise a proportional integral, PI, controller, being configured to provide the control variable of the control circuit in dependence of the differential of the output voltage and its reference variable.

The control circuit may comprise a pulse width modulation, PWM, signal generator, being configured to provide a first PWM control signal for the power switch in accordance with the control variable, the current through the series connection and the peak current envelope for the same.

An on-time of the first PWM control signal may depend on a given minimum on-time, on a given maximum on-time, and preferably on an instant of on-time expiration defined by the event that the current through the series connection exceeds the peak current envelope for the same.

The PFC circuit may comprise a Boost converter; and the first PWM control signal may be in accordance with one of: a discontinuous conduction mode, DCM, and a borderline conduction mode, BCM.

The PFC circuit may comprise a Flyback converter.

The Flyback converter may comprise a further series connection of a clamp switch and a clamp capacitance in parallel with the inductance. The PWM signal generator may further be configured to provide a second PWM control signal for the clamp switch.

An on-time of the second PWM control signal may depend on a given on-time.

The on-times of the first and second PWM control signals may be mutually exclusive in accordance with a dead-time depending on a given minimum dead-time, and on an instant of dead-time expiration when a time derivative of a drain-source voltage of a last active switch of the power switch and the clamp switch becomes zero.

A second aspect of the present disclosure relates to a luminaire, comprising an LED load and a converter of the first aspect.

Advantageous Effects

If a peak-current control is used, adding a DC component to the sinusoidal peak-current envelope improves an AC current waveform close to zero-crossing, but if the DC component is constant, the THD at low output power is adverse.

A DC component that depends on the output power of the PFC circuit provides a simple control scheme for good THD over a wide output power range.

The DC component proposed herein comprises a constant coefficient and a constant offset wherein both constants are derived from simulations or measurements at a minimum value and a maximum value of the output power of the PFC circuit. Notably, the constants themselves are independent of the output voltage of the PFC stage but do capture the dependency of the DC component of the output voltage by a simple parameterization. The parameterized DC component avoids or at least reduces the non-zero voltage plateau across the EMI capacitor around the zero-crossings of the supplied AC voltage for the whole output power range, ultimately resulting in a low THD.

In contrast to digital approaches which typically increase the on-time of the power switch of the PFC circuit close to the zero-crossings to correct the discharge of the EMI capacitor, adding a DC component / offset that depends on the output power but involves constant values that are independent of the output power is very simple and robust and independent of a switching frequency of the PFC circuit.

In turn, the switching frequency may be increased and a size of passive components may be reduced.

The technical effects and advantages described above equally apply to the converter and to the luminaire comprising the same.

Brief Description of Drawings

The above-described aspects and implementations will now be explained with reference to the accompanying drawings, in which the same or similar reference numerals designate the same or similar elements.

The features of these aspects and implementations may be combined with each other unless specifically stated otherwise.

The drawings are to be regarded as being schematic representations, and elements illustrated in the drawings are not necessarily shown to scale. Rather, the various elements are represented such that their function and general purpose become apparent to those skilled in the art.

FIG. 1 illustrates a luminaire in accordance with the present disclosure;

FIG. 2 illustrates a converter in accordance with the present disclosure; FIG. 3 illustrates a PFC circuit in accordance with the present disclosure;

FIG. 4 illustrates a drive circuit for the clamp switch 120;

FIG. 5 illustrates a state diagram of the PFC circuit in accordance with the present disclosure; and

FIG. 6 illustrates a various electrical quantities of the PFC circuit in accordance with the present disclosure.

Detailed Descriptions of Drawings

FIG. 1 illustrates a luminaire 3 in accordance with the present disclosure.

The luminaire 3 comprises an LED load 2, R₁ and a converter 1 for supplying the LED load 2, R₁ off output terminals of the converter.

The LED load 2, R₁ may comprise one or more LEDs in series and/or in parallel, depending on the lighting scenario.

FIG. 2 illustrates a converter 1 in accordance with the present disclosure.

The converter 1 comprises a PFC circuit/stage 11 and may further comprise a DC-DC converter/stage 12.

FIG. 3 illustrates a PFC circuit 11 in accordance with the present disclosure. PFC may refer to a maximization of a power factor of a connected load by drawing from an AC power supply an AC current that essentially corresponds in amplitude curve and phase to the supplied AC voltage, thereby minimizing a THD.

A power factor of an AC circuit may refer to a ratio of a real power absorbed by a connected load to an apparent power flowing in the circuit.

A THD may refer to a ratio of a sum of powers of all harmonic frequency components to a power of the fundamental frequency.

In accordance with FIG. 3, the PFC circuit 11 comprises a Flyback converter. Alternatively, a Boost converter (not shown) being operated in accordance with modes of operation such as a discontinuous conduction mode (DCM) or a borderline conduction mode (BCM) may be deployed as well.

The PFC circuit 11 comprises a series connection 111, 112 of an inductance 111 and a power switch 112.

In accordance with FIG. 3, the inductance 111 may be modelled as a parallel circuit of a magnetizing inductance L2 and a primary inductance L4 of an ideal transformer, for example.

The power switch 112 may in particular comprise a power semiconductor switch, such as a power metal-oxide-semiconductor field-effect transistor (MOSFET).

The PFC circuit 11 further comprises a control circuit 113 controlling the switching operation of the switch 112 and of a clamp switch, see further below.

The control circuit 113 is configured to regulate an output voltage 114, V_out of the PFC circuit 11 in accordance with a current 115, / through the series connection 111, 112 and a peak current envelope 116, 1 for the same. In accordance with FIG. 3, the output voltage 114, V_out may be tapped off at the LED load 2, R₁ being connected to an output port of the PFC circuit 11, the current 115, / through the series connection 111, 112 may be captured by means of a ground-referenced current sensing resistor Rio in series to the power switch 112, and the peak current envelope 116, 1 for the current 115, 7 may be provided by a ground-referenced resistive voltage divider R2, R3 in parallel to an EMI capacitor C₄.

The control circuit 113 may comprise a proportional integral (PI) controller 1131 and a PWM signal generator 1132.

PWM may refer to a continual on-off modulation of an electrical quantity, such as a voltage, by a switch, in accordance with a particular mode of operation of the superordinate circuit. For example, a PFC circuit may involve a PWM in accordance with modes of operation such as the DCM, preferably the BCM or a continuous conduction mode (CCM).

The PWM signal generator 1132 may be configured to provide a first PWM control signal for the power switch 112, the duty cycle of the PWM control signal being in accordance with a control variable 119, 1? of the control circuit 113, the current 115, 1 through the series connection 111, 112 and the peak current envelope 116, 1 for the same.

The PI controller 1131 may be configured to provide the control variable 119, 1? of the control circuit 113 in dependence of a differential V_out — V_{out re}f of the output voltage 114, V_out and its reference variable V_{out re}f .

The peak current envelope 116, 1 = C ■ | V_bus | ■ R(V_0Ut — V_{out re}f') may depend on the control variable 119, 1? of the control circuit 113 and on a multiplicative fraction C of a rectified input voltage C ■ | V_bus | of the PFC circuit 11.

In accordance with FIG. 3, the multiplicative fraction C may correspond to a given ratio — —

R2+R3 defined by the above-mentioned resistive voltage divider R2, R3. The control circuit 113 is further configured to offset the peak current envelope 116, I by a DC component 117, I_DC that depends on an output power of the PFC circuit 11.

The DC component 117, I_DC = k ■ R(V_0Ut — V_{out re}^ + d may depend on the control variable 119, R of the control circuit 113, a constant coefficient k of the control variable 119, /?, and a constant offset d of the DC component 117, I_DC.

The constant coefficient k and the constant offset d may be derived from simulations or measurements at a minimum value and a maximum value of the output power of the PFC circuit 11, thereby capturing the dependency of the DC component 117, I_DC on the output power of the PFC circuit 11 over the whole output power range.

The Flyback converter may comprise a further series connection 120, 121 of a clamp switch 120 and a clamp capacitance 121 in parallel with the inductance 111, thereby ending up with an active clamp Flyback (ACF) converter.

The PWM signal generator 1132 may further be configured to provide a second PWM control signal for the clamp switch 120.

The floating electric reference potential of the clamp switch 120 may require a so-called high-side drive circuit, such as the one illustrated in FIG. 4 being operable in a transforming manner.

More specifically, FIG. 4 illustrates an exemplary implementation of a drive circuit 123 for the clamp switch 120 including an ancillary transformer 131, 132, which may be connected to the PWM signal generator 1132 (see FIG. 3) and configured to transform a drive current Isw2 in terms of its electric potential.

As indicated in FIG. 4, the (transformed) drive current Isw2 includes alternating trains of positive current pulses and negative current pulses. The clamp switch 120 is operable in accordance with the transformed drive current Isw2. To this end, the drive circuit 123 may further comprise a circuitry for conditioning of the transformed drive current Isw2 into a control signal Vsw2 of the clamp switch 120.

The conditioning circuitry is configured to convert each train of positive current pulses into a positive voltage pulse of the control signal Vsw2, wherein a time duration of the positive voltage pulse corresponds to a time duration of the underlying train of positive current pulses.

The conditioning circuitry is further configured to convert each train of negative current pulses into a zero line voltage of the control signal Vsw2, wherein a time duration of the zero line voltage corresponds to a time duration of the underlying train of negative current pulses.

FIG. 5 illustrates a state diagram of the PFC circuit 11 in accordance with the present disclosure.

As previously explained, as the PFC circuit 1 comprises an ACF converter, the PWM signal generator 1132 may be configured to provide a first PWM control signal for the power switch 112 and may further be configured to provide a second PWM control signal for the clamp switch 120.

In accordance with states 41 to 43 in FIG. 5, an on-time of the first PWM control signal may depend on a given minimum on-time t_0N __LS __min, on a given maximum on-time toN_LS_max, ^and on an instant of on-time expiration when the sensed current 115, / through the series connection 111, 112 exceeds the peak current envelope 116, I for the same. This corresponds to a peak-current control of the power switch 112.

In accordance with states 45 to 46 in FIG. 5, an on-time of the second PWM control signal may depend on a given on-time t_0N __HS. This corresponds to an active clamping of the inductance 111, thereby supplying the energy stored in a leakage inductance (not shown) of the inductance 111 to the LED load 2, R₁ and improving a converter efficiency. In addition, a peak voltage across the power switch 112 during its turn-off events may be reduced significantly, resulting in a lower voltage rating of the power switch 112. In accordance with states 43 to 45 and states 46 to 41 in FIG. 5, the on-times of the first and second PWM control signals may be mutually exclusive in accordance with a dead-time depending on a given minimum dead-time t_dead __min, and on an instant of dead-time expiration when a time derivative of a drain-source voltage V_ds of a last active switch of the power switch 112 and the clamp switch 120 becomes zero. More specifically, in states 43 to 45 the power switch 112 was last active, whereas in states 46 to 41 the clamp switch 120 was last active. Note that a dead time when the power switch 112 was last active may be different from a dead time when the clamp switch 120 was last active.

FIG. 6 illustrates a various electrical quantities of the PFC circuit 11 in accordance with the present disclosure.

A top diagram of FIG. 6 shows a control variable 119, I? of the control circuit 113 and a DC component 117, I_DC of the peak current envelope 116, 1 vs. LED load 2, P_X1, whereas a bottom diagram of FIG. 6 depicts the regulated output voltage 114, V_out and an output power 118, P of the PFC circuit 11 vs. LED load 2, P_X1.

As can be seen, the output power 118, P = declines inversely proportional to the LED load 2, Rn

P₁₁₅ as may be expected in connection with the regulated output voltage 114, V_out.

The control variable 119, P shows a similar decline vs. the LED load 2, R^.

The DC component 117, I_DC depending on the control variable 119, P rises with increasing LED load 2, P_X1 and approaches an asymptotic value. It may be established that the DC component 117, I_DC varies with / depends on LED load 2, P₁₁₅ or alternatively on the output power 118, P.

In accordance with FIG. 6, an increase in the output power 118, P therefore coincides with a decrease in the DC component 117, I_DC, and vice versa.

Claims

1. Converter for supplying a LED load, the converter comprising a PFC circuit (11) supplying a further switched converter stage or being the only switched converter of a single stage converter, wherein the PFC circuit (11) comprises a flyback converter with a power switch (112) and an active clamp circuit (120, 121) having a clamp switch (120), wherein a control circuit such as e.g., an ASIC or a microcontroller is arranged to control the switch-on time period ofthe power switch (112) by a peak-control ofthe flyback current, i.e., the power switch (112) is switched-off as soon as it is sensed that the raising flyback current reaches a given threshold.

2. The converter (1 ) of claim 1 , wherein the PFC circuit (11), comprises: a series connection (111, 112) of an inductance (111) and said power switch (112); and a control circuit (113), being configured to regulate an output voltage (114, V_out) of the PFC circuit (11) in accordance with a current (115, /) through the series connection (111, 112) and a peak current envelope (116, I) for the same; and offset the peak current envelope (116, 1) by a direct current, DC, component (117, I_DC) that depends on an output power of the PFC circuit (11).

3. The converter (1) of claim 2, the peak current envelope (116, 1 = C ■ depending

- on a control variable (119, 1?) of the control circuit (113), the control variable (119, 7?) depending on a differential (V_out — V_Out_ref) °f the output voltage (114, V_out) and its reference variable (V_{out ref}), and

- on a multiplicative fraction of a rectified input voltage (C ■ IV^ |) of the PFC circuit (11).

4. The converter (1) of claim 3, the DC component (117, I_DC = k ■ R(V_0Ut — V_{out re}^ + d) depending

- on the control variable (119, R) of the control circuit (113),

- on a constant coefficient (k) of the control variable (119, R), and

- on a constant offset (d) of the DC component (117, I_DC).

5. The converter (1) of claim 4, the constant coefficient (k) and the constant offset (d) being derived from simulations or measurements at a minimum value and a maximum value of the output power of the PFC circuit (11).

6. The converter (1) of any one of the claims 3 to 5, the control circuit (113) comprising a proportional integral, PI, controller (1131), being configured to provide the control variable (119, R) of the control circuit (113) in dependence of the differential (V_out ^— out_ref ) °f th^e output voltage (114, V_out) and its reference variable (V_{out re}f ).

7. The converter (1) of any one of the claims 3 to 6, the control circuit (113) comprising a pulse width modulation, PWM, signal generator (1132), being configured to provide a first PWM control signal for the power switch (112) in accordance with the control variable (119, R), the current (115, 1) through the series connection (111, 112) and the peak current envelope (116, 1) for the same.

8. The converter (1) of claim 7, an on-time of the first PWM control signal depending

- on a given minimum on-time (toN_LS_min),

- on a given maximum on-time (t_{0N LS max}), and

- on an instant of on-time expiration when the current (115, 1) through the series connection (111, 112) exceeds the peak current envelope (116, 1) for the same.

9. The converter (1) of claim any of the preceding claims, the Flyback converter comprising a further series connection (120, 121) of a clamp switch (120) and a clamp capacitance (121) in parallel with the inductance (111).

10. The converter of any of the preceding claims, wherein the flyback converter is operated in the DCM or BCM mode.

11. The converter (1) of any of the preceding claims, a PWM signal generator (1132) being configured to provide a second PWM control signal for the clamp switch (120).

12. The converter (1) of claim 11, an on-time of the second PWM control signal depending

- on a given on-time (t_{0N HS}).

13. The converter (1) of claim 11, the on-times of the first and second PWM control signals being mutually exclusive in accordance with a dead-time depending

- on a given minimum dead-time (t_dead __mt_n), and

- on an instant of dead-time expiration when a time derivative of a drain-source voltage of a last active switch of the power switch (112) and the clamp switch (120) becomes zero.

14. A luminaire (3), comprising an LED load (2); and a converter (1) of any of the preceding claims for the LED load (2).