EP0098801B1 - Line with divided low-pass filter - Google Patents

Line with divided low-pass filter Download PDF

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Publication number
EP0098801B1
EP0098801B1 EP83810289A EP83810289A EP0098801B1 EP 0098801 B1 EP0098801 B1 EP 0098801B1 EP 83810289 A EP83810289 A EP 83810289A EP 83810289 A EP83810289 A EP 83810289A EP 0098801 B1 EP0098801 B1 EP 0098801B1
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EP
European Patent Office
Prior art keywords
line
characterized
sections
line according
line section
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Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired
Application number
EP83810289A
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German (de)
French (fr)
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EP0098801A3 (en
EP0098801A2 (en
Inventor
Jean-Joseph Max
Arvind Shah
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Feller AG
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Feller AG
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Priority to CH4021/82 priority Critical
Priority to CH4021/82A priority patent/CH656738A5/en
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Publication of EP0098801A2 publication Critical patent/EP0098801A2/en
Publication of EP0098801A3 publication Critical patent/EP0098801A3/en
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Publication of EP0098801B1 publication Critical patent/EP0098801B1/en
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    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01PWAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00Auxiliary devices
    • H01P1/20Frequency-selective devices, e.g. filters
    • H01P1/201Filters for transverse electromagnetic waves
    • H01P1/202Coaxial filters
    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01BCABLES; CONDUCTORS; INSULATORS; SELECTION OF MATERIALS FOR THEIR CONDUCTIVE, INSULATING OR DIELECTRIC PROPERTIES
    • H01B11/00Communication cables or conductors
    • H01B11/02Cables with twisted pairs or quads
    • H01B11/12Arrangements for exhibiting specific transmission characteristics
    • HELECTRICITY
    • H01BASIC ELECTRIC ELEMENTS
    • H01BCABLES; CONDUCTORS; INSULATORS; SELECTION OF MATERIALS FOR THEIR CONDUCTIVE, INSULATING OR DIELECTRIC PROPERTIES
    • H01B11/00Communication cables or conductors
    • H01B11/18Coaxial cables; Analogous cables having more than one inner conductor within a common outer conductor
    • H01B11/1895Particular features or applications

Description

  • The invention relates to an electrical line with at least one distributed low-pass filter for suppressing high-frequency interference signals on the line.
  • Known interference filters with discrete circuit elements, which are optionally ohmic, capacitive and inductive, have the disadvantage that the parasitic inductances connected to their capacitive circuit elements or the parasitic capacitances connected to their inductive switching elements give rise to undesirable resonances in the region of higher frequencies.
  • From the magazine IEEE Transactions on Electromagnetic Compatibility, January 1964, pages 55 to 61, further from the magazine Proceedings of the IEEE, January 1979, pages 159 to 163, and from DE-A-29 39 616 are shielded electrical cables with at least a distributed low-pass filter known as a noise filter. In the first-mentioned literature reference a coaxial transmission line is described which comprises one or more line sections with a magnetic material, e.g., between the central conductor and the outer shield. a ferrite material, as lossy insulating material. A similar coaxial interference protection filter provided with a magnetic ceramic material, which is mainly proposed as a feedthrough filter, is described in the second-mentioned literature reference. DE-A-29 39 616 describes a lossy electrical cable in which at least one conductive element in connection with an absorbent mixture at least partially surrounding the conductor has a composite structure, namely a core formed by a thread or a fiber and a conductive coating, such that the element has a high resistance with good mechanical properties.
  • The known distributed low-pass or interference protection filters have the disadvantages that they must be subject to high magnetic losses, dielectric losses or line losses in the insulating material, since such losses alone have their low-pass effect, and that they have a complicated structure, which is not only theirs Manufacturing, but also their universal applicability difficult.
  • The object of the present invention is to provide an electrical line of the type mentioned at the beginning, the distributed low-pass filter of which has a low cut-off frequency and, for signal frequencies up to the highest frequency range, high attenuation without noticeable resonance phenomena, and with a simple structure, neither on the use of materials with high loss factors is still dependent on great lengths.
  • According to the invention, the line has the features stated in the characterizing part of patent claim 1.
  • The combination according to the invention of reflections on both sides of a line section of different impedance and of dielectric losses and / or skin effect losses in this line section enables a mutual increase in the two damping effects for higher frequencies. On the one hand, multiple, in the best case almost total reflections of the signals of higher frequency and thus longer path lengths for these signals result on the end sides of the line section of different impedance, and on the other hand, the losses in this line section are also increased due to the greater equivalent path length of the lossy line section. Furthermore, by suitable selection of the dielectric in the lossy line section, i.e. whose dielectric constants, a relatively low cut-off frequency of the low-pass filter and at the same time high frequencies of resonances, in particular the lowest of the resonances that occur. In addition, a line with a line section, or, to increase the interference filter effect, with several successive line sections of different impedance and higher dielectric losses or skin effect losses can be produced in a relatively simple manner and practically any length, so that the present line as an interference filter, which allows low-frequency or direct-current electrical current to pass through without noticeable damping, but has high damping for high-frequency currents and can be used universally.
  • Exemplary embodiments of the subject matter of the invention are explained below with reference to the drawings. Show it:
    • 1 is a schematic representation of a basic line according to the invention with a lossy line section of different impedance,
    • FIG. 2 shows a schematic illustration of the signal reflections on the end sides of the line section of different impedance from FIG. 1
    • 3 shows the exemplary course of a supplied unit voltage jump signal at the end of the line section of different impedance from FIG. 1,
    • 4 shows the exemplary course of the filter attenuation for a line according to FIG. 1,
    • 5 and FIG. 6 a cut-away view of a two-wire or three-wire coaxial cable for the practical realization of the line according to the invention
    • Fig. 7 is a cut view of a Power and distribution rail for the practical implementation of the line according to the invention, and
    • 8 is a partial view of a coaxial cable with several line sections of different impedance,
    • 9 shows a line with two discrete inductivities having an equivalent wave impedance,
    • 10a shows a line with a discrete inductance and a discrete capacitor, both of which have an equivalent wave impedance,
    • Fig. 10b is an illustration of the line of Fig. 10a as a line with changing wave impedance, and
    • 11 shows a section through the cable of a line, the losses of which are based on the skin effect.
  • 1 shows schematically a coaxial line 1, which in a manner known per se has a conductor 2, an outer shield 3 and an insulating material or dielectric 4, not shown, located between the conductor 2 and the outer shield 3. The line 1 has a first and a third line section 5 and 6, both of which have as characteristic data an impedance Z o and a loss factor tg δ 0 , which in the present example is zero (loss-free line section). In between a line section 7 is provided, the impedance Z 1 and is very different from Z o , which has a relative dielectric constant ε r and a loss factor tg 8 1 , and the length of which is List.
  • If a signal 8, which is shown in FIG. 1 for example as a unit voltage jump signal and which propagates in line section 5 of impedance Z o , reaches point A of line 1, namely the beginning of line section 7, at which whose impedance suddenly takes the value Z 1, part of the signal is reflected, while the other part propagates in the line section 7. At point B of line 1, namely the end of line section 7, at which the impedance suddenly returns to the value z o , there is a further reflection of part of the transmitted signal, the other part of which propagates in line section 6. The reflected part of the signal, which preferably makes up almost all of the remaining signal, is sent back to point A, where again a nearly total reflection occurs. A multiple reflection of the signal components thus takes place in the line section 7, which has a different impedance than the adjacent line sections 5 and 6, as is shown in more detail in FIG. 2.
  • 2 shows the reflected or continuous portions of the unit jump signal 8 reaching point A to line section 7 as a function of time t. Here, the respective amplitudes for the individual reflected or continuous signal components are specified by means of the reflection factor, where:
    • ζ = (Z 0 -Z 1 ) / (Z 0 + Z 1 ) reflection factor of Z, towards Zo
    • 1- ζ = 2Z 1 / (Z 0 + Z 1 ) transmission factor from Z o towards Z :.
  • The prerequisite is that only the TEM mode of line 1 is taken into account.
  • The signal components appearing in time at the transition of line section 7 with impedance Z, to subsequent line section 6 with impedance Z o and then transmitted in line section 6 form a step-shaped curve, the signal amplitude of the first stage being 1-ζ 2 , that of second stage (1-ζ 2 ) ζ 2 etc., this in the event that the line section 7 is not subject to dielectric losses. Such an output signal curve for the unit step signal 8 is shown in broken lines in FIG. 3.
  • In the case of dielectric losses in line section 7, ie tgδ 1 ≠ 0, the signal curve shown in FIG. 3 in solid line results in line section 6. It can be seen from this that a pronounced low-pass effect is achieved due to the multiple reflections and the dielectric losses of line section 7 , as will be illustrated below with reference to FIG. 4. This low-pass effect is based on the fact that not only does a small part of the binary jump signal 8 entering the line section 7 of different impedance have to go back and forth several times over this line section before it can build up a noticeable voltage at the output of the line section 7, but that the effect of the dielectric losses in this line section can also be increased, because the “equivalent length” of the line section is multiplied by a factor which is essentially inversely proportional to the very small deviation of the reflection factor ζ from 1. This equivalent length is defined as the mean path length that a pulse-shaped wave must travel through on the same line section when it goes back and forth several times until half of it comes out of the line section in question.
  • As already mentioned, resonances occur in the line section 7 with the different impedance Z 1 at higher frequencies, which are fundamentally undesirable. It can now be seen that the amplitudes of such resonances can be significantly reduced by the effect of the dielectric losses of the line section 7 or the resonances can even be suppressed.
  • 4 shows the calculated and experimentally confirmed course of the filter attenuation for a line according to FIG. 1, the Attenuation A in dB and the frequency f with respect to the cut-off frequency f 3dB are plotted for 3 dB attenuation on a logarithmic scale.
  • It can therefore be seen from FIG. 4 that in a first region 10 of the filter curve, due to the reflections explained, attenuation occurs with a slope of approximately 20 dB per decade of frequency. In the subsequent area 11 of the filter curve, if there were no dielectric losses in the line section 7, high resonance peaks 12 would occur, which, however, only appear as weak increases 13 thanks to the dielectric losses mentioned. In the last area 14 of the filter curve, which can be above 1 GHz, the attenuation has an even higher slope because the dielectric losses predominate there.
  • Mathematically, it can be shown that the total attenuation, expressed in dB, is made up of three elements, provided that Z 0 > Z 1 :
    • a) from a first link determined by the reflections, which is given by
      • + 20.log [1 / (1-ζ 2 )]

      where ζ means the reflection factor already mentioned,
    • b) from a second member, determined by the dielectric losses, which is given by
      • + 867. π. f. T d . tgδ 1

      where f is the frequency, T d is the delay of the line section 7, and tgδ 1 is the loss factor of the line section 7,
    • c) from a third member, determined by the resonances, which is given by
      • - 20.log (| F |)

      where F is a variable dependent on the frequency f, the loss factor tg5 1 and the delay T d , the absolute value of which is> 1. This third link is negative, ie it reduces the damping.
  • Here, the delay T d = L / v, the product of the length L of the line section 7 and the inverse reproductive speed 1 / v in this section.
  • Thus, the reflections caused by the different impedance in the conductor section 7 determine the filter steepness and, as will be explained below, the cut-off frequency of the low-pass filter, while the dielectric losses of the line section 7 increase the frequency with an extinction or at least a strong attenuation of the resonances caused by the reflections and then a stronger weakening in the direction of higher frequencies is effected.
  • The cut-off frequency of this low-pass filter is given by
    • f 3 dB = (1-ζ) /2π.T d .
  • The frequency of the nth resonant is given by
    • f rn = n / 2.T d .
  • Since on the one hand the limit frequency is as low as possible and on the other hand the frequency of the first resonance (n = 1) is to be as high as possible, an optimum cannot be achieved by choosing a certain delay T d , ie the length L of the line section or the propagation speed v in the line section since both f 3dB and f m are proportional I / Td - A high ratio f rn to f 3dB can therefore only be achieved via the reflection factor ζ, which should be as close as possible to one.
  • The reflection factor ζ depends on the one hand on a change in the dielectric constant ε r and on the other hand on a change in the geometry of the line at the ends of the line section 7. Since the dielectric constant can only be changed to a relatively small extent due to the material, it is advantageous to bring about a considerable increase in the ratio of the frequency f rn of the first resonance to the cut-off frequency f 3dB in that, in addition to the length L of the line section, the two other dimensions, ie the Cross dimensions are changed, for example the diameter of a cable-shaped line.
  • In order to achieve the different impedances Z o and Z 1 of the line sections 5, 6 and 7 for the line 1 shown in FIG. 1, those with different relative dielectric constants can be used for the insulating materials 4 of these line sections. Above all, namely with regard to the aforementioned definition of the low frequency filter's limit frequency by different dielectric constants of the line sections 5, 6 or 7, the line geometry along the line 1 can also be changed, for example by changing the diameter of the insulating material 4. The loss factor tg5 1 of the line section 7 should be sufficiently high considering the damping of the resonances. However, special measures in the choice of materials, such as magnetic materials, are by no means necessary. In addition, the entire line 1, ie also in line sections 5 and 6, can have the same loss angle tgö. Suitable insulating materials for the lossy line section 7 with different impedance Z 1 are, for example, polyethylene with tg 8 between 0.02 and 0.2 or polyvinylidene fluoride (PVDF) with tg 8 between 0.1 and 0.2 in the frequency range from 0.5 to Called 200 MHz.
  • Depending on the application, the line 1 shown only schematically in FIG. 1 can have different embodiments, three examples of which are shown in FIGS. In the cut views, only one of the line sections 5, 6 and 7 of FIG. 1 is shown.
  • For using the line as a line noise filter for electrical and electronic devices, for example, the embodiment of a multi-core, shielded connection cable according to FIGS. 5 and 6 is suitable. FIG. 5 shows a two-wire line with two conductors 15, each of which is surrounded by an insulating material 16 of a certain diameter and certain dielectric properties. A separate metallic shield 17 envelops each insulating material 16. Furthermore, a plastic protective jacket 18 is provided. 6 shows a similar arrangement with three conductors 15, but in which a shield 19 for the three insulating materials 16 of all three conductors 15 is common. The embodiment according to FIG. 5 is also suitable for applications as an anti-parasitic signal or data line, while the embodiment according to FIG. 6 is also particularly suitable for use as an anti-parasitic mains cable for building and house installations.
  • The present line can also have the embodiment of a current or distribution rail for the supply inside or outside electrical and electronic devices, as shown in FIG. 7. Two busbars 20, which are provided with connecting lugs 21, are embedded in an insulating material 22 of certain dimensions and certain dielectric properties. The insulating material 22 is enclosed by a shielding metal housing 23 which is open on the underside and which is provided with a larger number of connecting lugs 24 and is surrounded by a plastic protective jacket 25.
  • It is advantageous to provide several lossy line sections of different impedance along the line in order to increase the filter effect, instead of a single line section 7 according to FIG. 1. Such a development is shown schematically on a coaxial cable in FIG. 8, with the shielding and the protective jacket for clarity are omitted. This cable has a central conductor 26 and a plurality of line sections 27, 28, 29, 30 etc. consisting of insulating material, the corresponding impedances Z 1 , Z 2 , Z 3 , Z 4 etc. and corresponding lengths Lt, L 2 , L 3 , L 4 etc. have. It can also be seen that the line sections 27, 28, 29, 30 have different diameters. The dielectric constants of the insulating materials of these line sections and their loss angles are also different in the general case. In practice, however, it will often be expedient to design every second section in the same way with regard to its diameter and with regard to the dielectric constant and the loss angle of its insulating material. The lengths L 1 to L 4 can, however, differ from one another in order to avoid any cumbersome accumulation of minor disturbing effects of the reflections. Practically, the lengths L 1 to L4 as well as the length L according to
  • Fig. 1 have values between about 1 cm and 500 cm, so that in the case of small lengths, the present line also takes the form of a discrete interference filter component for electrical and electronic devices, e.g. for mounting on a circuit board.
  • In such a simplified cascade arrangement, in which, with reference to FIG. 1, a line section with the impedance Z o is followed by a line section with the impedance Z 1 and the loss factor tg δ 1 , this is followed by a line section with the impedance Z o and thereon again a line section with the impedance Z 1 and the loss factor tg 8 1 follows, etc., multiply the aforementioned attenuators a) and b) by the number of lossy line sections Z 1 , so that the filter effect is greatly increased.
  • In the previously described exemplary embodiments of the subject matter of the invention, it has been assumed that the distributed low-pass filter is effective, i.e. at any frequency, has uniformly distributed impedances and loss elements along the line sections, but no discrete elements. If one looks at the behavior of any electrical components in relation to very fast pulses or high frequencies, one sees that in the sense of the location "discrete" circuit elements such as inductors and capacitors no longer exist, but that there are only elements that are distributed in a regular or irregular manner Has.
  • Therefore, if a discrete inductance is connected to the ends of a line section with a certain wave impedance, the damping curve of this arrangement for the higher frequencies to be damped must be viewed from the point of view that the inductors are distributed elements whose impedance is a function of the coordinate between a starting point and the end of the inductance.
  • An approximation of such an impedance can be obtained by taking only the average value, which is called the equivalent wave impedance. Thus, the above-mentioned arrangement is a line having a first line section with an equivalent impedance Z ä q, a second conduit section having a characteristic impedance Z and a third line section with an equivalent impedance Z eq. There is therefore a line with discontinuously changing wave impedances, the frequency-dependent attenuation of which can be calculated by reflections at the points of changing wave impedance, as in the previous exemplary embodiments.
  • Approximate values of the mean equivalent wave impedance for inductors (L) and capacitors (C) are given by the relationships
    • Z a q (L) = Lv / l or Z aq (C) = I / Cv

    where 1 is the length of the respective line section and v is the propagation speed dependent on the insulating material. In the case of an inductance L, the length I is equal to the existing wire length, while in the case of a capacitor the length 1 is its total length if it is wound, or its mean length if it is not wound.
  • FIG. 9 shows an exemplary embodiment of the electrical line according to the invention, in which one line section has a discrete inductance 31, a second line section is formed by a coaxial cable 32 and a third line section has a further discrete inductance 33, the second line section having a wave impedance Z and the adjacent line sections have equivalent wave impedances Z ä q and Z ' ä q different from Z.
  • 10a shows a similar design of a line, but in which the corresponding third line section has a capacitor 34. In terms of impedance, this configuration corresponds to the line shown in FIG. 10b, the line sections of which have the equivalent wave impedance Z a q (L), the wave impedance Z and the equivalent wave impedance Lgq (C). The capacitor 34 plays the same role as an open stub. As indicated in FIGS. 10a and 10b, the entire line can consist of several, alternately successive line sections of the type described.
  • As an alternative to the dielectric losses described and also provided in the exemplary embodiments according to FIGS. 9 and 10, the known skin effect, which is effective at higher frequencies, can be used to generate losses in a simple manner, which strongly dampen the resonances occurring as a result of the signal reflections and also effect the desired filter attenuation of the present line for the maximum frequency range (FIG. 4). The measure for generating frequency-dependent losses due to the skin effect is that the conductor of the line has an inner conductor part (or a core) with high electrical conductivity in order to transmit the relatively low frequencies up to a few thousand hertz including the direct current without loss. The inner conductor part has a coating or a surface layer which has a lower electrical conductivity or is even semiconducting, in which the currents of higher frequency flow due to the skin effect. Since this coating is a poor conductor, the current-conducting layer or skin becomes even thinner at higher and very high frequencies than in the case of a conductor made entirely of a highly conductive material, so that the current conduction is further deteriorated, i.e. the losses that occur as a result of the skin effect are substantially greater.
  • Dielectric losses increase in proportion to the frequency, but losses due to the skin effect only increase with the square root of the frequency. However, since, as mentioned below, the aforementioned coating can have a significantly lower electrical conductivity than, for example, copper, the skin effect losses which can be achieved are sufficient to obtain the desired filter damping.
  • 11 shows the section through a corresponding cable-shaped line. An inner conductor part 35 consists of an electrically highly conductive material, for example copper with a specific electrical resistance of 1.7 μQ.cm. The inner conductor part 35 has a thin surface layer 36 made of a poorly conducting metal, for example
    • Antimony (spec.el.resistance 42 µQ.cm)
    • Bismuth (specific electrical resistance 120 µQ cm)
    • Nichromo (spec.el.resistance 100 µQ.cm)
    • Manganese (spec.el.resistance 70 µQ.cm).
  • The surface layer can also consist of a semiconducting material, preferably of copper (I) oxide Cu 2 0.
  • A layer 37 of an insulating material adjoins the surface layer 36, which in turn is encased by an outer conductor provided as a shield with high electrical conductivity, for example also made of copper. This simple design of the line maintains the properties of the central conductor, which conducts the signals of relatively low frequencies, while at the same time strongly attenuating the signals of higher and highest frequencies.
  • The inner conductor part 35 can also be provided with a plurality of outer, thin layers of a poorly conducting material lying on top of one another, the specific resistance of the layers increasing towards the outside. This ensures that the current penetrates into the poorly conducting outer conductor part at high frequencies.
  • Of course, it is also possible to combine the dielectric losses described above with the skin effect losses, namely by appropriate selection of the insulating material and the covering material of the central conductor.

Claims (12)

1. An electric line with at least one distributed low-pass filter for suppressing higher-frequency interference signals present on the line, characterized in that over at least one section (L) of the line the wave impedance of the line (Zo) ha - a different value (Z1) in comparison with the wave impedance of the neighbouring line sections (Zo) or in comparison with the equivalent wave impedance of a neighbouring discrete element, so as to create reflections of the interference signals at the two ends (A, B) of the line section concerned, at which ends the wave impedance changes, and that this line section is affected by substartial dielectric losses and/or skin effect losses, so as to attenuate the resonances and higher frequencies arising from the reflections.
2. A line according to claim 1, characterized in that it comprises at least three successive line sections (5, 7, 6) which have differing wave impedances (Zo, Zi, Zo), at least one (7) of these sections being affected by substantial dielectric losses and/or skin effect losses.
3. A line according to claim 1, characterized in that it comprises several successive pairs of line sections of varying wave impedances, such that along the line a line section with one wave impedance is positioned next to a line section with the other wave impedance, at least one of the line sections of each pair being affected by substantial dielectric losses and/or skin effect losses.
4. A line according to any one of claims 1 to 3, characterized in that it is coupled at least at one end to at least one discrete element (31 ,33).
5. A line according to any one of claims 1 to 4, characterized in that it comprises at least one conductor, an insulating material enclosing the latter and a shield at least partially covering the. insulating material.
6. A line according to claim 5, characterized in that it is constructed as a single- or multicore cable.
7. A line according to claim 5, characterized in that it is constructed as a contact- or distribution rail arrangement (Fig. 7).
8. A line according to claim 5, characterized in that it is constructed as a microfilter by thick- or thin-film technology.
9. A line according to any one of claims 5 to 8, characterized in that the insulating material of at least one of the line sections has a different dielectric constant from the insulating material of the neighbouring line sections.
10. A line according to any one of claims 5 to 9, characterized in that at least one of the line sections has different geometric dimensions from the neighbouring line sections, for example a different length and/or a different insulating material diameter.
11. A line according to any one of claims 5 to 10, characterized in that the conductor consists of an inner conductor partand at least one outer layer located thereon, the specific electric resistance of which outer layer is greater, for example more than ten times greater, than that of the inner conductor part.
12. A line according to claim 12, characterized in that the inner conductor part is provided with several superposed outer layers all of which have a greater specific electric resistance than the inner conductor part and the inner layer of which has the lowest and the outer layer of which, located on the surface of the conductor, has the greatest specific electric resistance.
EP83810289A 1982-07-01 1983-06-29 Line with divided low-pass filter Expired EP0098801B1 (en)

Priority Applications (2)

Application Number Priority Date Filing Date Title
CH4021/82 1982-07-01
CH4021/82A CH656738A5 (en) 1982-07-01 1982-07-01 LINE distributed LOW PASS.

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
AT83810289T AT24983T (en) 1982-07-01 1983-06-29 Pipe with distributed low-pass filter.

Publications (3)

Publication Number Publication Date
EP0098801A2 EP0098801A2 (en) 1984-01-18
EP0098801A3 EP0098801A3 (en) 1984-07-18
EP0098801B1 true EP0098801B1 (en) 1987-01-14

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EP83810289A Expired EP0098801B1 (en) 1982-07-01 1983-06-29 Line with divided low-pass filter

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US (1) US4683450A (en)
EP (1) EP0098801B1 (en)
AT (1) AT24983T (en)
CH (1) CH656738A5 (en)
DE (1) DE3369228D1 (en)

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Also Published As

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EP0098801A3 (en) 1984-07-18
AT24983T (en) 1987-01-15
DE3369228D1 (en) 1987-02-19
EP0098801A2 (en) 1984-01-18
US4683450A (en) 1987-07-28
CH656738A5 (en) 1986-07-15

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