EP0072788A1  Method and apparatus for controlling an ac induction motor  Google Patents
Method and apparatus for controlling an ac induction motorInfo
 Publication number
 EP0072788A1 EP0072788A1 EP80902058A EP80902058A EP0072788A1 EP 0072788 A1 EP0072788 A1 EP 0072788A1 EP 80902058 A EP80902058 A EP 80902058A EP 80902058 A EP80902058 A EP 80902058A EP 0072788 A1 EP0072788 A1 EP 0072788A1
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 European Patent Office
 Prior art keywords
 stator
 signal
 representing
 signals
 motor
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 230000001276 controlling effects Effects 0 title claims description 10
 238000004804 winding Methods 0 claims description 27
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Classifications

 H—ELECTRICITY
 H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
 H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMOELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
 H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
 H02P21/06—Rotor flux based control involving the use of rotor position or rotor speed sensors
Abstract
Description
METHOD AND APPARATUS FOR CONTROLLING AN AC INDUCTION MOTOR
Technical Field This invention relates to a method and an apparatus for controlling an AC induction motor.
Background Art For a long time there has been a need for induction motor control systems. An induction motor is, as a rule, smaller, lighter and cheaper than a DC motor for the same power, but considerably more difficult to control, especially at low speed. Since the induction motor and especially the squirrel cage induction motor, requires less service and is more reliable than the DC motor, the induction motor is preferred to the DC motor if a suitable control system exists.
For this reason it is desirable to obtain a control system, which gives the induction motor equally good control possibilities compared to the DC motor with full torque at all speeds down to standstill, and which furthermore is cheap and simple.
The control of an induction motor is complicated because of the fact that both the field current (for stator magnetization) and the rotor current (for torque generation) are fed through the same winding. Furthermore separate amplifiers are required for all motor phases. Present conventional control systems for induction rotors are suitable for control within a large speed range, but not for speeds close to zero. These control systems operate with simultaneous variation of frequency and amplitude of the control voltages. The Swedish patent No. 334.671 discloses a method which improves the control possibility at low speeds, including zero speed. Especially the response time on control signals is shortened, which makes the induction motor suitable for closed loop control systems. However, the control system has technical limitations. It is not possible to obtain full torque and short reaction time simultaneously.
The "Siemens" company has developed a control system named "Transvektor". Here the control problems are solved in a theoretically correct way. In general, for all kinds of process control it is necessary to measure the state of the process before the optimal control signals can be calculated. This system uses measuring probes in the motor' s airgap for measuring the stator magnetization. Based on these measured values the control signals are calculated for the stator magnetic field as well as for the motor torque. This control system is said to give the motor a performance equivalent to the DC motor's both in static and dynamic respect. A great disadvantage of the Siemens system is the need for special measuring probes in the motor. Furthermore the calculation of control signals is so complex that the system is hardly of interest for small motors because of the costs. The long signal chains, including filtering of the measuring signals, may be expected to recuce the system's speed of response.
Disclosure of Invention An object of the present invention is to create a new method and a corresponding apparatus for control of an AC induction motor, which eliminates the known disadvantages of prior systems. An induction motor controlled according to the method of the invention will have the same performance as a controlled DC motor, statically as well as dynamically. Maximum torque is obtained at all speeds and the reaction time on control signals is minimized. A normal standard motor can be used without any need of modification or measuring equipment on the motor. However, also an alternate embodiment, including special measuring windings in the motor, is disclosed. The circuits for signal processing are simple and cheap compared to the power amplifiers, which are of conventional design. This opens applications at all motors, from the smallest to the biggest. The motor can be operated in four quadrants with active drive or brake in both directions of rotation. The motor can be used in open systems as well as in closed control loops. Any outer control loop is designed conventionally according to known control theory.
According to the invention, a method is disclosed for controlling an AC induction motor comprising a stator, a rotor and at least two phase windings, within its whole speed range down to and including zero speed in both directions, in dependence of control signals. The control signals are fed to a vector rotating means, the output signals of which are used for generating individual supply signals to each motor phase winding. Signals corresponding to the magnetic field of the motor are fed back to the control system. The magnetic field signals above and down to a predetermined frequency are derived by integrating a signal sensing the magnetizing voltage of the stator, and below said frequency, by means of a signal representing the stator current. The signal representing the magnetizing voltage of the stator is obtained directly from a measuring winding of the stator, positioned adjacent tie airgap between the stator and the rotor of the motor, or by reconstruction, by measuring the stator voltage and the stator current and calculating according to the formula E = V_{o}  I_{o} × R_{S}. The signal representing the current of the stator is obtained directly by measuring the stator current or by reconstruction, by measuring the stator voltage and processing the measured signal by multiplication by a factor 1/R_{S}. The derived integrated signal is passed through a high pass filter and to a sunnier and the derived signal representing the stator current is passed through a low pass filter and to the summer, whereby the signal at the output of the summer represents the magnetic field at all frequencies.
Furthermore, an apparatus is disclosed for controlling the induction motor. The apparatus comprises measuring and calculation means corresponding to at least two phase windings of the motor for generating said magnetic field signals. Said means including a first calculation channel for generating said magnetic field signals above and down to a predetermined frequency, and a second calculation channel for generating said magnetic field signals below said frequency. The first calculation channel includes an integrator for integrating the signal representing the magnetizing voltage of the stator and the second calculation channel is supplied with a signal representing the stator current.
The apparatus includes a measuring winding for directly measuring the magnetizing voltage of the stator, or a member for measuring the stator voltage and a member for measuring the stator current and subsequent reconstructing a signal representing the magnetizing voltage. Moreover, the apparatus includes a member for measuring the stator voltage for reconstructing a signal representing t.ne stator current at low frequencies, or a member for measuring stator current. Brief Description of Drawings The above and other features of the invention will be fully understood from the following detailed description and the accompanying drawings, in which: Fig. 1 is a schematic diagram showing the symbol of a resolver, which is used as a building block in the following figures.
Fig. 2 is a block diagram showing how the resolver may be realized.
Fig. 3 is a schematic diagram, which defines the coordinate system used in the resolver.
Fig. 4 is a schematic diagram of a twophase induction motor. Fig. 5 is a schematic diagram of the equivalent circuit of one phase in the induction motor.
Figs. 68 are block diagrams of known systems for speed control of an induction motor.
Fig. 9 is a block diagram of an apparatus according to the invention for speed control of an induction motor.
Fig. 10 is an alternative block diagram similar to Fig. 9. Fig. 11 is a block diagram of a portion of Fig. 10, a so called quadrature oscillator.
Fig. 12 is a schematic diagram of a geometric model of the amplitude control in the quadrature oscillator.
Fig. 13 is a block diagram of a complete quadrature oscillator including amplitude control. Fig. 14 is a block diagram of a circuit for calculation of the motor's inductive current in one phase.
Fig. 15 is an alternative block diagram of a circuit for calculation of the motor's inductive current in one phase.
Fig. 16 is a block diagram of a control apparatus according to the invention adapted to a threephase motor.
Fig. 17 is a schematic diagram of the geometric relations for a coordinate transformation according to Fig. 16.
In the Figures, components performing the same operation have been given the same identification numbers. In Figures and formulas the symbol "s" is used for the Laplaceoperator. With this symbol, "1/s" indicates an integrator.
Best Mode of Carrying Out the Invention and Industrial Applicability Fig. 1 shows the symbol for a resolver 1. This is a cevice normally used in circuits for trigonometric calculations. An input signal vector i s rotated an angl e θ and obtains the new val ue . The l ength of the vector is not changed. The fol lowing trigonometric relations are val id:
Fig. 2 shows how the resolver may be constructed with four multipliers 2a2d, and two summers 3a, 3b.
Fig. 3 defines the coordinate system used in the resolver and in the following specification. The Xaxis and the Yaxis are stationary in relation to the motor stator. Positive rotation is counterclockwise from the Xaxis. To make the following block diagrams clearer, it is pointed out that the input vector to a control apparatus according to the invention is composed of variable DCvalues (control signals). This vector is not rotating. The rotation angle θ is almost synchronized with the rotor of the induction motor. Therefore the output vector will, be rotating,, with x_{2} and y_{2} as AC values.
Fig. 4 is a simplified schematic diagram of a twophase induction motor. V_{1}, V_{2} are supply voltages and B_{1}, B_{2} the generated magnetic field components. Only those rotor windings with maximum coupling to the stator windings are shown. I_{1}, I_{2} are the induced currents in the rotor windings shown.
The following (idealized) relations are valid:
V1 = V_{o} sin wt B_{1}= B_{o} cos wt I_{1} = I_{o} sin wt V_{2} = V_{o} cos wt B_{2 =} sin wt I_{2 =} I_{o} cos wt
w = signal frequency (rad/s).
The magnetic field from one phase cooperates with the rotor current from the other phase. This is of fundamental importance for the operation of the motor. Thus, current I_{1} generates torque together with magnetic field B_{2}. In the same way I_{2} generates torque together with magnetic field B_{1}. As can be understood from the formulas, the cooperating quantities have the same phase angle. This is necessary for maximum torque generation. Fig. 5 shows the equivalent circuit of one phase of the induction motor. The circuit includes only the primary components from the theoretical model. The broken lines represent the airgap between stator and rotor. Components to the left of the broken lines represent the stator, and components to the right represent the rotor. Stator resistance is R_{s}, stator inductance is L and rotor resistance is R_{r}. Supply voltage is V_{o} . Total current is I_{o}, inductive stator current is I_{s} and rotor current is I_{r}. Motor torque is proportional to the rotor current. The magnetizing voltage E is transformed from the stator to the rotor through the airgap. A counter electromotive voltage U is induced in the rotor winding. U is proportional to the motor speed.
It is often desirable to operate the motor with a constant magnetic field amplitude. The magnetic field is generated by the inductive stator current I_{s}, which should have a constant amplitude at all frequencies. Thus, it is required that the magnetizing voltage E increases proportionally to the frequency. In order to avoid weakening of the magnetic field it is important that the control apparatus complies with the requirement. From the following specification it will be evident that conventional control systems that do not take the motor load into account do not comply with this requirement.
The motor can be controlled only be means of the supply voltage V_{o} (or the total current I_{o}). Thus, the important voltage E cannot be directly controlled. The stator current I_{s} and the rotor current I_{r} give a voltage drop in the stator resistance R_{s}, making a difference between the voltages V_{o} and E. This voltage drop is different for different motor loads, since the rotor current I_{r} is proportional to the load. The voltage drop cannot be compensated by a fixed quantity. Lack of rotor current compensation may halve the stator magnetic field at low frequencies and thus the generated torque. The problem with compensation for voltage drop in stator resistance R_{s} exists only at low frequencies. At higher frequencies the voltages V_{0} and E are much larger than the voltage drop, which thus can be neglected.
The idealized relations in Fig. 4 show that the magnetic field and the rotor current always cooperate with correct mutual phase angle. This favourable condition is also valid for the more realistic motor model in Fig. 5. There may be a phase angle difference between the voltages V_{o} and E, but it is the voltage E that generates current for stator magnetization as well as for rotor current. Thus, the magnetic field and the rotor current will always, and automatically cooperate with correct mutual phase angle. This is valid at all frequencies and for all shapes of the supply voltage.
Fig. 6 is a block diagram of a commonly known circuit for speed control of a twophase induction motor 7. Frequency and amplitude of the two supply voltages V_{1} , V_{2} to the motor are simultaneously varied. The motor is controlled by control signal S_{1} according to a predetermined relationship. Power amplifiers 6 drive the motor.
The control signal S_{1} is fed to an amplitude circuit 4, which basically gives a linear function of the control signal but with a contribution at lower frequencies, which compensates for the stator resistance voltage drop. In addition the signal S_{1} controls an oscillator 5 (VCO), which generates a frequency proportional to the control signal. The oscillator generates control voltages for the resolver 1. The basic design of Fig. 6 is used in most socalled thyristor controls. The circuit may be realized in a simpler way than in Fig. 6. A complete resolver is not necessary. Fig. 6 has been shown for easy comparison with the following figures.
Fig. 7 is a block diagram of an improved version of Fig. 6. This system, which is described in the Swedish patent No. 334.671, comprises two separate control signals S_{1} and So to the resolver instead of the nonlinear amplitude circuit 4 (Fig. 6). The control signal S_{2} is fixed and intended to compensate for the stator resistance voltage drop. The control signal S_{1} is variable and intended to supply the magnetizing voltage E. A solution according to Fig. 7 gives the control system a radically improved dynamic performance at low speeds. The improvement is due to the fact that control signal S_{1} is always transformed to the rotor with correct phase angle for generating torque with the shortest possible time delay. The system can be said to control frequency and amplitude as well as phase angle of the motor supply voltages.
If only static and not dynamic performance is considered, the two solutions according to Fig. 6 and Fig. 7 are equivalent. The sum of the two control signals S_{1} and S_{2} to the system in Fig. 7 gives the same amplitude as the output signal from the nonlinear circuit 4 in Fig. 6.
Neither the system in Fig. 6 nor that in Fig. 7 compensates for varying motor load. Thus, the magnetic field might be weakened at low frequencies, as described in connection with Fig. 5.
Fig. 8 is a block diagram of the central operation of Siemens' control system "Transvektor". The basic structure of Fig. 7 is main tained. The main difference is that oscillator 5 has been replaced by a socalled vectorfilter 8, and that the motor has been equipped with measuring probes 9 in the airgap. The two probes measure the components B_{1}, B_{2} of the magnetic airgap field. The vector filter filters the signals and calculates the amplitude B_{o} and the direction angle θ of the magnetic field. The direction angle θ is represented by signals for sin θ and cos θ , which control the resolver 1 in the same way as before. According to Siemens "a field orientation is achieved". This means that the control vector (S_{1}, S_{2}) is rotated the same angle as the magnetic field and controls the motor in an optimal way.
Control signal S_{2} represents the desired amplitude of the magnetic field and is compared to the measured value B of the magnetic field. The difference signal is fed from the comparator block 10, eventually through a control amplifier 11, to the resolver 1.
This system maintains the desired amplitude of the magnetic field independent of the motor load.
Alternative methods for measuring the magnetic field are described in a paper "Verfahren der Felderfassung bei der Regelung Stromrichtergespeister Asynchronmaschinen" by Felix Blaschke and Klaus Bohm (IFAC SYMPOSIUM on Control in Power Electronics and Electrical Drives, Dϋsseldorf 1977 Vol. 1 pp. 635649). By measuring at least two available motor parameters, such as stator current, stator voltage, rotor angle, rotor speed, and calculating signals representing the motor magnetic field, it is possible to generate necessary control signals for. the "Transvektor" system.
Thus it is proposed to generate signals representing the magnetic field from measured values of stator current and stator voltage. This method works well at higher motor speeds. However, since the calculation includes an integrator, the result becomes unsatisfactory at lower speeds, especially at zero speed, because of the integrator drift.
Fig. 9 is a block diagram of an apparatus for the new motor control method according to the invention. The basic structure of Fig. 8 is maintained. However, a new type of measuring signals, representing the magnetic field components B_{1} , B_{2}, is used. Measuring and calculation blocks 12, which will be described in detail, are connected to each of the two phases. The signals B_{1}, B_{2} from these blocks, representing the magnetic field, have such a high quality that the socalled vectorfilter 8 (Fig. 8) is unnecessary and has been omitted . The si gnal s B_{1} , B_{2} are di rectly connected to the resol ver 1 as feedback s ignal s , and to a cal culati ng bl ock 13 , which cal cul ates the ampl i tude of the magnetic fi el d B_{o} accordi ng to the formul a
In real practice in a control system, the calculating block 13 may be simplified by eliminating the square root. Instead of controlling the magnetic field B it is possible to control B_{2o}.
The signal S_{1} controls the motor torque.
The signal S_{2} controls the magnetic field in the same way as in
Fig. 8, indicating the comparator block 10 and the control amplifier
11. The control systems of Figs. 6, 7, 8 and 9 have all the same purpose which is to "convert" the AC induction motor to a DC motor. It is natural that the systems are similar since they are intended to control the same type of motor. To facilitate a comparison all block diagrams have been shown in a similar way. The systems of Figs. 8 and 9 have one common feature. The motor takes an active part in the control system. The feedback signals B_{1}, B_{2}, representing the amplitude and direction of the magnetic field, control the resolver which generates the AC supply voltages for the power amplifiers of the motor. However, the system is not easily analyzed in the way it is shown.
Fig. 10 is a block diagram of exactly the same control system as in Fig. 9. Only the way in which it is drawn is different. The motor 7 with power amplifiers 6 and measuring and calculating blocks 12 is now replaced by two blocks 14, one for each motor phase. Each block 14 contains an integrator 23, the function of which will be described below. The resolver 1 has been shown in details with the multipliers 2a2d and the summers 3a, 3b.
Fig. 11 shows only the central parts of the control system in Fig. 10. The circuits and signals for amplitude stabilization have been omitted. Thus, only two multipliers 2c and 2d from the resolver are shown. The negative input to the summer 3a is now represented by an inverter 15.
The system in Fig. 11 corresponds exactly to a wellknown electronic device, a socalled quadrature oscillator. The two integrators 23 are connected in a closed loop. The phase shift around the loop is 360° (90° in each integrator and 180° in the inverter 15). Control theory states that such a system will selfoscillate at that frequency where the loop gain is unity (1). The signal S_{1} controls the loop gain with the aid of the multipliers 2c and 2d, and thus the oscillator frequency.
The output signals B_{1}, B_{2} from the oscillator are sinusoidal with 90° phase shift.
B_{1} = B_{o} sin θ B_{2} = B_{o} cos θ
Fig. 12 shows the two signals B_{1}, B_{2} in a rectangular coordinate system. The two signals form together a rotating vector with the length B_{o}. Furthermore the Figure illustrates the geometric relations for the amplitude stabilization. The amplitude difference ε (in the same direction as the vector B_{o}) may be resolved in components along the axes B_{1}, B_{2}.
ΔB_{1} = ε sin e θ ΔB_{2} = ε cos θ
Fig. 13 shows the complete quadrature oscillator including the circuit for amplitude stabilization. The two different components of the amplitude difference are fed back, each to its own integrator 23 in the block 14, and work in such a way that the amplitude is corrected.
The block 14 in Figs. 10, 11 and 12 has been said to contain an integrator. The transfer function of the block can be calculated, slightly simplified, if the rotor influence is neglected. This makes no principal difference. The block diagram in Fig. 5 gives the following equation (rotor neglected):
Because of R_{s} the integration decreases at low frequencies, but the amplitude stabilization according to Fig. 13 compensates for this.
The direct feedback signal B_{2} from the output of block 14 and back to the input of same block 14, through the multiplier 2b, constitutes a positive feedback, which compensates for the influence of R_{s}. The control signal from block 11 to the multiplier 2b adjusts the degree of feedback to exactly the required value. The same relations are valid for the other half of the circuit with feedback signal B_{1}.
The control amplifier 11 is designed according to control theory, with proportional and eventually integrating operation.
It follows from the theoretical motor model in Fig. 5, that the rotor current I_{r} gives a voltage drop in the stator resistance R_{s}. This voltage drop decreases the magnetizing voltage E, thus decreasing the integration of inductive current I_{s}. This gives a "lower gain" in the measuring and calculating block 12 (Fig. 9) and thus a lower quadrature oscillator frequency. In this way the frequency of the control system is influenced by the motor load. At high motor load the frequency decreases, giving time for the magnetic field to build up to the correct amplitude. The systems according to Figs. 13 and 9 are identical.
In connection with Fig. 9 a measuring and calculating block 12 was introduced into the block diagram. The block 12 is designed to measure those motor quantities which are easily accessible, and use these measured values to calculate (reconstruct) the amplitude and direction of the magnetic stator field. The results of the calculations are the signals B_{1} , B_{2}.
A twophase induction motor requires two blocks 12, one block in each phase. For accuracy and symmetry reasons one block 12 per phase is preferred in multiphase motors. In that case the signals must be combined to form the required signals B_{1} and B_{2}. The magnetic field vector is completely described by only two components.
The magnetic field in the stator is created by the inductive stator current I_{s}. The strength of the magnetic field is proportional to the current I_{s}. Thus the current I_{s} can be used as a measure of the magnetic field strength. (Later it is described how the nonlinearity of the magnetic circuit can be taken into account.)
Fig. 14 shows a first embodiment of the measuring and calculating block 12 for one motorphase. It is not possible to measure the current I_{s} directly but the current has to be calculated from other quantities. Those quantities, which are most easily accessible, are the supply voltage V_{Q} and the total current I_{o}. The theoretical motor model according to Fig. 5 gives the following equation: where R_{s} = stator resistance L = stator i nductance s = Laplace operator E = magnetizing voltage
Assuming that the motor parameters R_{s} and L are known, the current I_{s} can be calculated from the measures values of V_{o} and I_{o} . It is of special interest to notice that the calculation is correct independent of the value of rotor current I_{r}. On the other hand the accuracy is influenced by errors in the theoretical motor model and by variations in the motor parameters.
The calculated value I'_{s}, multiplied by a suitable constant can be used as a measure of the magnetic field strength in each motorphase. Fig. 14 shows how the calculations are performed by the apparatus within block 12, operating according to formula (I).
A measures value V'_{0} of the supply voltage V_{o} is connected from the motor winding to a summer 21. A measured value I_{o} ' of the total current I_{o} is obtained from a current measuring device 17, which is connected to a multiplier 22 where it is multiplied by the stator resistance R_{s} . The multiplier output is connected to an inverting input of the summer 21, the output of which is V_{o} '  R_{s} × I_{o}'. This is the calculated magnetizing voltage E'. The summer is connected to integrator 23, v/here the signal E' is integrated and multiplied by the constant 1/L (L = stator inductance). Thus; the output of integrator 23 is a calculated value I'_{s} of the inductive stator current, according to formula (I).
The integrating action according to formula (I) creates problems at low frequencies because of integrator drift. In order to eliminate this problem, a high pass filter 24, e.g. with transfer function (s/w_{o} )/(1 + s/w_{o} ) , is connected to the integrator output. In practice the integrator 23 and the high pass filter 24 are combined to a coπroon filter or block with the following transfer function: This eliminates the integrator problem, since the combination deducts the block to an ordinary low pass filter.
The theoretical model with two separate blocks is used only in order to explain the theory. It is the integrating operation of the block above the cutoff frequency that is of primary interest. Since low frequency signals are blocked by the high pass filter, an additional calculated signal I'_{s} must be found, which represents the low frequency current in the stator winding. At those low frequencies in question, the magnetizing voltage E and thus the rotor current I_{r} are so small that they can be neglected. This means that the total current I_{o} is a good approximation of the inductive current I_{s}. A measured value of I_{o} can be used instead of I_{s}.
Thus, the current measuring device 17 is connected to the first input of a summer 25. The second input of the summer is connected to the output from the high pass filter 24. At low frequencies the signal I_{o}' at the fi rst input of the summer 25 is a good approximation of the inductive current I_{s}. The signal I_{o}' must not influence the output signal from block 12 at higher frequencies. Therefore a low pass filter 26, e.g. with the transfer function 1/(1 + s/w_{o}), is connected in the signal path for I_{o}' before the summer 25.
With this method the measuring and calculating block 12 generates two separately reconstructed signals I_{s}' representing the inductive current I_{s} in the stator winding. One signal passes through a high pass filter and one signal through a low pass filter, and then the signals are added. The output of the summer will always at all frequencies, deliver a reconstructed value I_{s}' which is a good approximation of the inductive current I_{s} in the stator winding. The use of a high pass filter and a low pass filter with transfer function according to the given formulas gives an output from the summer which is frequency independent:
The output from the summer 25 is connected to a multiplier 27 where the signal is multiplied by a suitable constant scaling factor, which results in a signal B that represents one component of the magnetic field vector B_{1}, B_{2}.
Although it has not been demonstrated here it is possible to increase the accuracy of the calculating circuit by introducing a correction for the nonlinear magnetization curve of the iron. The magnetic permeability of the iron is depending on the magnetic field strength. This has influence on the inductance L of the integrator 23 and the constant scaling factor of the multiplier 27.
The reconstructed signal B represents the magnetic field in the motor stator. If desirable, a correction term can be introduced in order to make B represent the airgap field between rotor and stator or the magnetic field in the rotor.
Fig. 15 shows a second version of the measuring and calculating block 12 for one motorphase. According to the invention this is an alternative method for calculating the inductive current I_{s} in one stator winding.
A separate measuring winding 20, with magnetic coupling to the stator winding L, generates a measured value E' of the magnetizing voltage E. This measuring is based on the actual field strength variations in the iron and takes into account the nonlinear magnetization curve of the iron. According to formula (I) a signal I'_{s}, representing the inductive stator current, is calculated:
Fig. 15 shows how the calculations are performed by the apparatus within block 12. The measuring winding 20, giving the measured value E' of the magnetizing voltage, is directly connected to the integrator 23. The signal is integrated and multiplied by the constant 1/L. The integrator output delivers a calculated value I_{s}', representing inductive stator current, which is connected to the summer 25. As was the case in Fig. 14 a high pass filter is connected in the signal path in order to block low frequency signals. At low frequencies, as in Fig. 14, an additional calculated signal I_{s}' for the inductive stator current is required, and the method from Fig. 14 may be used. Fig. 15 shows an alternative calculating apparatus for the current I_{o} at low frequencies, v/here I_{o} is an approximation of the inductivecurrent I_{s}. At those low frequencies, which are concerned, the current is defined basically by the stator resistance R_{s}. Then the following approximate formul a is val id:
where V = supply voltage.
Thus, only a measured value of the supply voltage is required in order to calculate the total current I_{o} at low frequencies, and to use this value as a measure of the inductive current I_{s}. This simplified, and less accurate method is of special interest in connection with the use of a measuring winding 20 in the motor, because the current measuring device 17 can be completely eliminated. The calculating and measuring block 12 has a multiplier 28 with the transfer function 1/R_{S} connected in the signal path before the summer 25. This is in order to calculate a signal I_{o}' representing the total current at low frequencies according to formula (III), also representing the inductive stator current I_{s}'. A measured signal V '_{o} of the supply voltage is connected to the multiplier 28, the output of which delivers a signal I_{o}' to the summer 25. At low frequencies I_{o}' is a good representation of the stator inductive current. As in Fig. 14 a low pass filter is connected in the signal path before the summer 25. The output from the summer 25 is connected to a multiplier 27 where the signal is multiplied by a suitable constant scaling factor, which results in a signal B that represents one component of the magnetic field vector B_{1} , B_{2}.
Of course the described methods for calculation of I_{s}' at high and low frequencies can be combined in several ways. The combination giving the best motor control might be to use the measuring winding 20 according to Fig. 15 for calculation of I_{s}' at high frequencies and the measuring method according to Fig. 14 at low frequencies. This is because these measurements are relatively temperature independent. The described measurement methods may be improved in details in order to obtain higher accuracy. The largest error source is the temperature variation in the motor, giving resistance variations in the windings. This can be compensated by automatic temperature or resistance measuring devices and correction circuits.
It is not necessary to place the measuring windings exactly in conformity with the phase windings. Alternative placings are possible, if the calculation circuits take the geometric relations into account. The high pass filter and the low pass filter have been described as socalled Butterworth filters with the same cutoff frequency, because this gives a frequencyindependent transfer function after the summer. This means that in the frequency region around the cutoff frequency both current signals from the two calculation circuits are contributing to the output signal and that the contributions from the two circuits are equal at the cutoff frequency. However, the filter design is not critical. Other filter types may be used and the filters may have different cutoff frequencies. Also, the summer 25 may be replaced by a selector switch selecting the largest of the two signals, or other suitable device.
According to the invention, it is essential that the calculation or generation of a signal B representing the magnetic field is made in a conventional way at frequencies above a certain frequency and that the calculations below that frequency are performed by a second calculating circuit, not using integrating operation. This second calculating circuit is more approximata than the first calculating circuit and the approximations become greater with increasing frequency. Thus , i t is preferrable to choose the socalled cutoff frequency, below which the second calculation circuit takes over, as low as possible with respect to the first calculating circuit. This cutoff frequency should be below approximately 5 Hz and above approximately 0.01 Hz. A preferred value is between approximately 0.1 Hz 1 Hz. Of course the selection of cutoff frequency is depending on the actual motor and the required performance.
All figures in this description have referred to a twophase motor. Normal induction motors have three phases. However, the control system is easily converted to control a motor with any number of phases.
Fig. 16 shows as an example how the control system according to Fig. 9 has been converted to a threephase motor. Two new blocks 18 and 19 for coordinate transformation have been added to the circuit. Block 18 converts the two output signals V_{1}, V_{2} from resolver 1 to three control signals V_{R}, V_{S}, V_{T}, which are connected to the power amplifiers 6. The convertion is made according to conventional geometric transformations.
Fig. 17 shows the geometric relations. The following formulas are given as an example of possible transformations:
V_{R} = V_{1}
Block 19 converts the reconstructed signals B_{R}, B_{S}, B_{T} from the measuring and calculating devices 12 from threephase to twophase, giving the signals B_{1} and B_{2}. The following formulas are given as an example of possible transformations:
The control system according to the invention may be realized with analogue as well as with digital components, for example with a micro computer. Especially in digital designs, the possibility exists to write alternative mathematical formulas for the calculation to be performed by the control system, still within the scope of the invention.
The control system according to the invention may be used with the power amplifiers 6 designed as voltage sources or current sources or any other design. In all cases the necessary measuring signals are available.
Normally this type of control system is used for speed control of the motor. However, it is possible to control other quantities, such as the motor torque or the motor position.
When controlling normal DC motors it is usual to control the rotor current as well as the field current, depending on the application. As an example it is possible to decrease the field current in order to run the motor at elevated speed. All such control methods are possible for the induction motor with a control system according to the invention. The control system in Fig. 9 is designed for the "normal" application with constant magnetic field strength in the induction motor. In this case control signal S_{2} is constant, and the feedback loop controls B_{o} to a constant value. If signal S_{2,} and thus field strength B_{o}, is varied, a slightly more complex system is required. The signals B_{1},B_{2}, which are used as feedback signals to the resolver, must be "normal ised" to a constant scaling factor, preferably according to the following formulas:
This requires an additional calculating block, not shown in Fig. 9, in the signal path for B_{1}, B_{2} respectively. It is possible, within the scope of the invention, to improve the accuracy of the control system by extending the theoretical motor model in Fig. 5. The first improvement should be to add leakage reactances in series with the stator and rotor reistances. These reactances should be added in Fig. 5, as well as in the measuring and calculating blocks 12 (Figs. 14, 15).
Claims
Priority Applications (1)
Application Number  Priority Date  Filing Date  Title 

PCT/EP1980/000122 WO1982001628A1 (en)  19801030  19801030  Method and apparatus for controlling an ac induction motor 
Publications (1)
Publication Number  Publication Date 

EP0072788A1 true EP0072788A1 (en)  19830302 
Family
ID=8164811
Family Applications (1)
Application Number  Title  Priority Date  Filing Date 

EP80902058A Withdrawn EP0072788A1 (en)  19801030  19801030  Method and apparatus for controlling an ac induction motor 
Country Status (7)
Country  Link 

EP (1)  EP0072788A1 (en) 
JP (1)  JPS57501658A (en) 
AU (1)  AU6399880A (en) 
DK (1)  DK279882A (en) 
FI (1)  FI824002A0 (en) 
NO (1)  NO822101L (en) 
WO (1)  WO1982001628A1 (en) 
Families Citing this family (1)
Publication number  Priority date  Publication date  Assignee  Title 

SE9000497L (en) *  19900212  19910813  Ragnar Joensson  Foerfarande and apparatus foer controlling an induction motor by indirect maetning of luftgapsspaenningen 

1980
 19801030 JP JP55502473A patent/JPS57501658A/ja active Pending
 19801030 WO PCT/EP1980/000122 patent/WO1982001628A1/en not_active Application Discontinuation
 19801030 EP EP80902058A patent/EP0072788A1/en not_active Withdrawn
 19801030 AU AU63998/80A patent/AU6399880A/en not_active Abandoned

1982
 19820622 DK DK279882A patent/DK279882A/en not_active Application Discontinuation
 19820623 NO NO822101A patent/NO822101L/en unknown
 19821122 FI FI824002A patent/FI824002A0/en not_active Application Discontinuation
NonPatent Citations (1)
Title 

See references of WO8201628A1 * 
Also Published As
Publication number  Publication date 

JPS57501658A (en)  19820909 
FI824002A (en)  
FI824002L (en)  19821122 
WO1982001628A1 (en)  19820513 
DK279882A (en)  19820622 
FI824002A0 (en)  19821122 
NO822101L (en)  19820623 
FI824002D0 (en)  
AU6399880A (en)  19820521 
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