DE2552536A1 - ELECTRICAL PROTECTIVE SWITCHING DEVICE - Google Patents

Description

DIPL.-INQ. KLAUS NEUBECKER 2552536

Patent attorney 4 Düsseldorf 1 Schadowplatz 9

Dr.-Ing. Ernst Strafmann

Patent attorney

Düsseldorf, Nov. 21, 1975 45,373
7576

Westinghouse Electric Corporation
Pittsburgh / y V. St. A.

Electrical circuit breaker

The invention relates to an electrical protective switching device, which is to be used in particular to protect AC networks.

In known measuring devices for electrical quantities, the information about the current and about the operating power of the relay circuit derived from two current transformers. For some operating states, at least one was Transformer designed so that it saturates within the expected current ranges. With other well-known Facilities a current transformer was used in combination with a separate power supply.

In the case of static, time-delayed relay circuits of a known type, it is common to provide curve shaping devices, in order to obtain relay properties that match the properties of the previously widely used electromechanical Bodies are similar. Curve shaping arrangements are shown, for example, in U.S. Patents 3,496,417 and 3 544 846 of the applicant.

609823/0299

Telephone (O211) 32 08 58 telegrams Custopat

ORIGINAL

The object of the invention is to provide an electrical protective switching device, in particular to provide a device responsive to an electrical quantity which only has a low Transformer loading and accurate network information signals supplies which are linearly dependent on the currents in the network over a wide range.

The task is solved according to the features of the main to saying.

The invention thus comprises a device which reacts to an electrical variable, which consists of a transformer which has an input winding which is excited by an AC network. The output winding of the Transformer supplies a first and a second electrical quantity, the first electrical quantity exciting an energy storage device. The second electrical quantity possesses a value which is determined by the value of the electrical quantity of the alternating current network. Furthermore are Control means provided which control the output windings such that the first and second electrical Quantities are switched off one after the other during half a period of the network alternating current from the output winding, to replace the energy consumed by the energy storage device during the previous half cycle and to to provide a copy of the current from the AC network.

The advantage of this arrangement is that the burden of the transformer is very low and the display is linear with great accuracy over a wide range.

The invention is explained below with reference to in the drawings illustrated embodiments explained in more detail.

609823/0299

It shows

Fig. 1 is a block diagram of an electrical variable responsive device according to the invention for Protection of a three-phase AC network;

Fig. 2 is a schematic diagram for explaining the in 1 shows the input circuit shown in block form;

Fig. 3 is a schematic diagram for explaining the AC-DC conversion circuit and the time delay circuit shown in Figure 1 only in block form;

Fig. 4 is a simple logarithmic representation of the species and manner in which the waveform is using the desired inverse time reaction curve an overlay process is achieved;

Fig. 5 is a schematic diagram for explaining that used in the time delay circuit Rc network;

6 shows a schematic illustration to explain the instantaneous comparison circuit and the trigger circuit, which are shown in Figure 1 only as blocks;

7 shows a schematic illustration for explanation another embodiment of the one shown in FIG Input circuit;

Fig. 8 is a schematic illustration for explanation yet another embodiment of the in FIG input circuit shown; and

6 0 9 8 2 3/0 2-8-9

-A-

9 shows a schematic illustration for explanation an input overvoltage protection circuit and a power supply using a transformer with a single secondary winding.

The invention is directed to a new device responsive to an electrical quantity for an alternating current network to protect. In a particularly favorable embodiment a transformer is used to alternately charge a power supply and provide a control variable whose Value is proportional to the value of the electrical quantity to be monitored in the AC network. A detector circuit is excited by the control variable and provides a time-delayed output signal that, with regard to the value the control variable has an inverse time characteristic. The time delay circuit includes RC network devices and digital counters to a desired time inverse Shape reaction curve.

Fig. 1 illustrates a measuring device for an electrical variable or a protective relay, which is connected to the electrical to be protected Network is connected. This electrical network can be of any type that has a state can assume to which the measuring device is to react. For purposes of illustration, assume that the network is a is three-phase AC network operating at a frequency of 60 Hz and through the phase lines L1, L2 and L3 is shown. These phase lines carry alternating current from a suitable source through a circuit breaker 4 with a trip coil 6 to a load. The isolating switch 4 comprises several separate line contacts 8, 10 and 12, which are closed when the circuit breaker 4 is closed and open when the circuit breaker 4 is open. An excitement the tripping coil 6 when the isolating switch 4 is closed leads to a tripping or opening of the isolating switch. It it should be expressly pointed out that the invention, the

B 0 9 8 2 3/0 2- ^ 9

is described in connection with a multi-phase network is applied as well to a single-phase network can be. The protective switching device according to the invention can e.g. B. can be used to calculate the total current either in a single-phase network or the total current in a monitor multiphase network, depending on the design of the circuit that winds the output of the sensing windings the current transformers combined.

According to a preferred embodiment of the invention, a current relay is sensitive to the amount of through phase currents flowing through the phase lines L1, L2 and L3 and is responsive to the highest current value flowing through phase lines L1, L2 and L3. When the height of this current Exceeds a predetermined value during a time interval which depends on the current level, energizes the relay the trip coil 6 to switch the circuit breaker 4 either essentially immediately or after a predetermined time delay, which depends on the current level to open.

As explained in FIG. 2, the input circuit comprises a plurality of essentially identical current transformers CT1, CT2 and CT3, the primary windings of which are individually excited by the line currents of the phase lines L1, L2 and L3. Each current transformer or converter has a first output winding 104, referred to as the power winding, and a second output winding 106, referred to as the information winding. Both output windings are traversed by the same magnetic flux and the number of turns N ₁ in each power winding is less than the number of turns N _{2 in} each information winding.

For the sake of simplicity, only the operation of the current transformer CT1 with its associated circuit will be described in detail. All three transformers and their associated circuits operate in a similar manner to provide first and second output _currents I Q1 and I _Q2 , respectively

609823/0299

are proportional to the line current, which is the corresponding Transformer energized.

The primary winding 102 of the current transformer CT1 becomes accordingly excited by the line current of the phase line L1. The power winding 104 and the information winding 106 are respectively connected to the input terminals of full wave bridge rectifiers RE108 and RE110, the output terminals 112-114 and 116-118, respectively. Connections 112 and 116 are connected to a common line which is grounded as shown. The connections 114 and 118 are with a first and a second input terminal of a control circuit 120 connected.

The control circuit 120 comprises a voltage operated device, the one according to the illustrated embodiment Zener diode Z122 represents. Furthermore, the control circuit 120 comprises a first switching device reproduced as a pnp transistor T124, several diodes D126, D128, D130, D132 and D134, a second switching device shown as a thyristor SCR136, a capacitor C137 and several resistors R138, R140, R142, R144 and R146. Control circuit 120 provides sequential operation of power winding 104 and information winding 106.

In the following, the term "thyristor" is intended to to describe any switching device having a first or anode terminal, a second or cathode terminal and a has third or control terminal. After electrical conduction has entered this facility, the electrical Do not stop conduction unless the current flowing into the anode falls and remains below a predetermined level Value for at least a minimum time period, which is to be referred to as the minimum switch-off time.

After each zero crossing of the line current in the phase line L1 becomes a balance of the primary amp turns

609823/0299

2552538

_ 7 -

first achieved by the current flow I ₀₁ through the power output winding 104. During this time, the information output winding 106 remains open and inoperative due to the non-conduction or de-energized state of the SCR136 thyristor, which acts as an open switch.

The current I ₀₁ is rectified by the rectifier RE108 and used to charge a power supply 148. A portion of the current I _Q1 flows between the terminals 112 and 114 through a diode D126 and an energy storage device or capacitance C150 to produce a voltage V at terminal 152 which is positive with respect to ground. A second portion of the rectified current I ₀₁ flows between the output terminals 112 and 114 through a diode D128 and an energy storage device or capacitance C162 to provide a second power supply circuit which has a substantially ripple-free regulated voltage V _R between ground and an output terminal 156 supplies. The second power supply terminal 154 is used for supplying low power consumption devices. The supply circuit 154 for a reference voltage is parallel to the capacitor C150 and comprises the series connection of a resistor R158 and a rheostat R160, which is connected in parallel to the capacitor C162. The wiper of the rheostat R160 is connected to connection 156. A filter capacitor C164 is located between the wiper of the rheostat R160 and ground.

If the voltage between terminal 114 and ground is below the breakdown voltage V _{2122 of} a temperature-compensated Zener diode Z122, there is no significant current Iq ₄ . flow into the emitter of transistor T124 or through resistor R138. While the rectified current I _Q1 capacitors C150 and C162 Kon the loads, the voltage across the capacitors C150 and C162, until it reaches a size equal to V ^ _122, which is the regulated output voltage V. In addition to preventing the capacitors C150 and C162 from discharging, the diodes D126 and D128 still deliver

609823/0299

a temperature compensation for the regulated voltages as the diodes D126 and D128 tend to reduce the temperature effects of the base-emitter junction of the transistor T124 to compensate.

When the voltage level across capacitors C150 and C162 essentially reaches V_ ₁₂₂ , Zener diode Z122 breaks down and conducts current through resistor R138. This reduces the potential at the base of transistor T124, so that a base current then begins to flow and transistor T124 generates a current between emitter and collector and thus through diode D130 and resistor R142 in order to excite the control electrode of thyristor SCR136. In the embodiment shown in FIG. 2, a resistor R144 is connected to capacitors C166 and C168 in such a way that the thyristor SCR136 is prevented from being excited by noise signals possibly occurring at the control electrode or by a temperature-dependent leakage current at the anode.

The excitation of the control electrode of the thyristor SCR136 causes it to conduct and closes a current path for the rectified current I _Q2 between the output terminals 116-118, via the anode and cathode of the thyristor SCR136, the resistor R146 and the grounded line. This generates a voltage across the resistor R146, the magnitude of which is equal to (I ₀₂ ) (R146).

Since the same magnetic flux passes through both output windings 104 and 106, the ratio of the voltages of windings 104 and 106 must be the same as the turns ratio N ₁ to N ₂ . When the thyristor SCR136 conducts, the voltage across the terminals 116 and 118 will be essentially the same as that across the resistor 146 and the. _{The voltage level at the output terminals} 112 and 114 is determined by the approximate level of V A122 plus the emitter-base

60982.3 / 0

Transition voltage of transistor T124 to the value (I _n ,,) (R146) (N ₁ )
—— s - fall off.

The values of BL / N ₂ and the resistance of R146 as well as the maximum expected value of I _Q2 must be selected in this way

(I ₀₂ ) (R146) (N ₁ ) the fact that the size of - - to be expected for all -

the value of the line current in the phase line L1 never _{exceeds the value of V 2122} plus the emitter-base voltage of the transistor T124. A suitable, but not critical, guide value for the measuring device explained is that at which the value of I _{Q2 is} limited to the value that corresponds to the maximum line current value in the phase lines L1, L2 and L3, which is 40 times the value minimum input current. The minimum draw current is defined as the minimum value of the line current that will produce a minimal magnitude draw voltage that will actuate the draw comparator circuit 402. Limiter devices, which are to be explained in more detail below, were also used to limit the size of I ₂ for line currents that exceed 40 times the value of the absorption value.

When thyristor SCR136 is energized, the level of the voltage across resistor R146 is greater than the voltage between output terminals 112 and 114, thereby reverse biasing diode D130. The magnitude of the voltage between the terminals 112 and 114 thus falls below the voltage V ₂₁₂₂ and the current through the Zener diode Z122 ends. The voltage across terminals 112 and 114 will drop below the voltage across capacitors C120 and C162, but no discharge current will occur because of the reverse biased diodes D126 and DI28. Conducting thyristor SCR136 thus effectively opens the loop of winding 104.

The control electrode of the thyristor SCR136 loses its effectiveness on the thyristor device after one time

609823/0299

rectified current I _Q2 begins to flow between the anode and cathode, and the SCR136 thyristor remains energized until the line current in the phase line L1 reaches its next zero crossing. After de-energizing the thyristor SCR136, the circuit of the information winding 106 is opened. At this time, the primary ampere-turns balance is achieved solely by power winding 104 and the magnitude of the voltage at output terminals 112 and 114 jumps to a magnitude determined by the principle of the ampere-turns of windings 102 and 104. Power winding 104 in turn operates to charge capacitors CI50 and C162 in the manner described above.

The rectified current Iq ₂ > more passes through the resistor R146 when the thyristor SCR136 is energized, provides a suitable information voltage, regardless of whether the signal is processed in the form of the rms value, the peak value or as an average value. The voltages across the resistors R170 and R172 accordingly supply information voltages which are proportional to the currents in the phase lines L2 and L3, respectively.

The power supply 148 is recharged during each initial portion of each half cycle of the phase line current and thereafter supplies information voltage during the remainder of the half cycle. The duration of each half cycle of the current in the phase line L1, in which the power winding 104 and the information winding 106 are each active, depends on the time it takes to recharge the voltages across the capacitors C150 and C162 to a value, which is equal to V ₂₁₂₂ i ^st · If the information voltages are processed in the form of the rms value or in averaged form, the accuracy of the processed information voltages changes inversely with the maximum charging time of the capacitors C150 and C162, the circuit components should therefore be selected so that the charging the power supply 148 takes place in such a short time,

6 09823/0299

2552538

as the resilience of the network to be monitored still allows. For peak signal detection it is only necessary limit the charging time to less than 90 ° of the sinusoidal phase current. The size of resistors R146, R17O and R172 should be as low as it is the desired sensitivity still allows to alleviate the burden. A suitable value is e.g. B. 50 ohms.

The multi-phase overcurrent relay of Fig. 2 reacts to the highest value of the line currents in the phase lines L1, L2 and L3. A high select circuit 174 provides a first Maximum selection voltage signal that only points to the highest value of the three generated across resistors R146, R17O and R172 Voltages reacts, with this circuit 174 diodes D175, D176 and D177, each between the ungrounded Connections of resistors R146, R170 and R172 as well as are connected to a common output conductor 178 which is connected to the AC-DC conversion circuit 2OO connected is. The voltage across one of the resistors R146, R170 or R172, which has the highest value, determines the Size of the voltage of the conductor 178. Two of the three diodes D175, D176 and D177 are biased in the opposite direction and thus be blocked.

The high select circuit 174 provides a second high select voltage, which is similar to the first high select voltage, this circuit having diodes D181, D182 and DI83, each between the ungrounded ends of resistors R146, R170 and R172 and one common Output conductor 184 is connected to that with a limiter circuit 186 is connected, whereby a second maximum selection voltage signal of the aforementioned limiter is fed.

Limiting circuit 168 connects conductor 184 to ground, in parallel with resistors R146, R170 and R172. A voltage regulating device Z188, which is represented as a Zener diode

609823/0299

is set, breaks down when the second high select voltage assumes a value that corresponds to more than 40 times the input current. If the device breaks through Z188, is a thyristor SCR19O, whose cathode is connected to the ground potential lying common line is connected, made conductive or energized. The shunt circuit 186 becomes thus render the entire input circuit ineffective by essentially shunting all currents, which is normally through resistors R146, R170 and R172 flow. The resistor R192 is provided to the current by the control electrode of the thyristor SCR19O when the Zener diode Z188 is in its conductive state. Resistor R194 and capacitor C196 are connected in such a way that they prevent the thyristor SCR19O from being excited by noise signals on its control electrode.

A second type of limiting device is achieved in that the current transformers CTI, CT2 and CT3 in the Manner that they saturate at a certain line current, so that the maximum level of the voltage between the terminals 112 and 114 is limited to a value which is lower than the voltage level V "plus the Emitter-base voltage of transistor T124 if one or several of the windings 106 have their respective charging resistances Excite R146, R170 and R172.

As shown in Figure 3, conductor 178 provides the first maximum select voltage to a peak voltage averaging circuit 200, which may take any form, but in the illustrated embodiment consists of a filter network consists, which comprises a filter capacitor C2O2 and a resistor R2O4 connected in parallel therewith. The circuit 200 excites the output conductor 202 with a DC voltage, whose size is proportional to the mean values of the peak values of the first highest selection voltage. The head 202 of the averaging circuit 200 is provided with a time delay measuring circuit 400 and a circuit 500 for detection associated with an immediate trip.

6 0 9823/0 2 * 8

- 13 time delay circuit

The time delay circuit 400, shown in greater detail in FIG. 3, includes a capture circuit 402, a digital counting circuit 406, an RC reset delay network 410, an interface circuit 412, and a monostable multivibrator circuit 414. After the level of the DC voltage from the conversion circuit 200 exceeds its minimum predetermined recording value, after a predetermined time interval TD the counting circuit supplies a control signal in order to excite the triggering circuit 600, either via the time-delayed comparison circuit 400 or via the directly measuring circuit 500, depending on the size of the pick-up current that supplies the pick-up voltage signal. The duration of the time delay TD provided by the circuit 400 varies inversely with the magnitude of the DC input voltage, with the exception of a fixed minimum delay time, which is given by the fact that the multivibrator 414 takes a certain time to change its state. As will be explained in more detail, the time delay circuit is capable of providing an output voltage corresponding to any desired inverse time response curve, depending on the values of R and C used in the delay network 410 and on the setting of the counting circuit 406.

The recording circuit 402 suppresses the operation of the time delay circuit 400 and resets it, as soon as the magnitude of the DC input voltage of the averaging circuit 200 is less than the predetermined input value. The recording circuit 402 triggers the operation of the Circuit 400, however, off when the input voltage exceeds the minimum value. The recording circuit 402 comprises a comparison circuit 408, the negative input terminal of which is connected to the conversion circuit 200, and is excited by this while his positive connection with

609823/0299

terminal 146 of reference voltage circuit 154 is connected. The voltage V ₇ determines the minimum input voltage and can be adjusted with the potentiometer (rheostat) R160. When the voltage output of circuit 200 is less than the level of V _R , comparator 406 will not draw current from supply terminal 152 and output conductor 409 will be maintained at substantially the voltage V _g . When the pickup voltage _{exceeds the value V R} , the comparator 408 effectively connects the conductor to ground and the counters 444, 446 and 447 are placed in a state in which they count the multivibrator 414 pulses. The first unit of a monolithic MC33O2P integrated circuit manufactured by Motorola, Inc., which comprises four identical comparison units, which is arranged in a dual plastic casing provided with fourteen connections, is suitable as the recording comparator 408. The second, third and fourth units of the monolithic building block MC33O2P are expediently used elsewhere, e.g. B. used for the circuits 400 and 500.

The inverse time-response curve is first formed by a passive network 410, which has resistance elements and capacitive elements, the interface arrangement 412 and a monostable multivibrator 414. The capacitive elements are de-energized each time the output voltage at terminal 436 exceeds the voltage V_. The transistors T414, T416, T418, T420 and T422 are shunted across these capacitive elements and their base terminals are connected to the output of the multivibrator via the transistor T422. Each time the multivibrator 414 is activated, the transistors become conductive and the capacitive elements are discharged, as a result of which the network 410 enables cyclical operation and the counters 444, 446 and 447 switch on.

Two time characteristics are provided by the pulse generator circuit 404, the time delay TD1, which is determined by the

609823/0299

RC network 410 is provided and is a function of the magnitude of the phase line current and the fixed time delay TD2 of the monostable multivibrator 414.

The resetting delay network 410 comprises several parallel branches. Each branch includes capacitors so that the charging of the capacitor of each branch takes place at a different time rate, which in FIG. 4 is on a semi-logarithmic basis Network is shown and to the Zext stream relationships 426, 428, 430 and 432 leads. Curve 424 shows the inverse time stream relationship obtained from network 210 when the network is suddenly excited by the DC voltage signal proportional to the highest peak value of the through phase currents flowing through the phase lines L1, L2 and L3 is. It can be seen that curve 424 is the sum of the time relationships 426 to 432 is. The time required for the size of the equivalent voltage response of the RC network 410 increases from zero to a predetermined value is therefore inversely proportional to the highest peak value of the phase line currents mentioned.

Of course, although four branches are shown in Figure 3, any number of branches may be used. Each branch of the illustrated RC network 410 includes two resistors connected between an input terminal 434 and an output terminal 436 are cascaded while a capacitor is connected to one point between the two Resistors is connected in parallel to the ground potential.

5 is intended to explain the RC network 410 in more detail. It is assumed that initially all the capacitive elements are de-energized and the voltage V_ _N between the input terminal 434 and ground is suddenly increased to the minimum consumption value or a higher value. The branch capacitors will charge at their respective charging rates. During this time, the voltage V is determined by the magnitude of the fault current (which for a given fault

6098 2 3/0299

with a fixed size), whereby a charging current ¹ RCA flows ^{to the capacitors} . A voltage V _R is maintained at terminal 436 of the conversion circuit 200. It can be seen that the amplifier 413 will attempt to maintain the potential V _n at its input terminal 439. A second component of the charging current I _n ^ _n flows to the network 410 from the output terminal 438 through the zener diode Z44O and the diode D442. When the capacitors of the network 410 reach their critical charge, the charge component I _{RCB will be} zero. However, since the voltage at terminal 434 will necessarily be greater, current will continue to flow to charge the capacitors and the voltage at terminal 436 will rise sufficiently to cause the operational amplifier to decrease the voltage at its output terminal 436 in an effort to maintain the voltage level V_ on its negative input terminal and thus the voltage level on terminal 436.

The voltage at terminal 434 is denoted by V, the voltage at terminal 436 by V _R , the voltage at the ungrounded terminal of the capacitor as V ", the resistance value of the resistor between the capacitor and the terminal as RJ, the resistance value of the resistor between the Capacitor and terminal 436 with R ₂ , the magnitude of the current through the capacitor from terminal 436 with 1 _ηΓιτ . and if the direction of current to the capacitor from terminals 434 and 436 is used, the following mathematical formula results for the current I _RCB as a function of time, whereby a derivation is possible either directly or using the LaPlace transformation:

RCB

R ₁ R ₂ C

^V IN R ₁₊ R ₂

1-e

R ₁ R ₂ C

6 0 9 8 2 3/0 2-Θ-9

By adding the currents I _RCB for each RC branch and setting the sum equal to zero, the time t at which the network 410 operates the operational amplifier 413 in the interface 412 can be determined.

The presence of the zener diode Z44O serves to raise the voltage at the output terminal 438 of the operational amplifier 413 sufficiently above the voltage V _R to prevent the monostable multivibrator 414 from operating during the charging period of the network 410.

As soon as the repeater 413 responds to the network 410, the potential of the connection 438 of the network 410 is almost instantaneously lowered in value in order to supply an operating signal for triggering the monostable multivibrator 414 deliver a positive pulse to the counter 444 and on after a specified time interval has elapsed the transistor T422 supplies. When transistor T422 conducts, it causes transistors T414, T416, T418 and T420 to be closed conduct and discharge the capacitors of network 410. When the capacitors discharge, the potential drops of the negative input terminal 436 and is held at the voltage level V. At the end of the time delay interval TD2 of the multivibrator 414 will return this to its low output state and cause the transistors T414-T422 stop conducting and network 410 will repeat its timing operation.

The monostable multivibrator 414 also provides its positive going pulse to the counting circuit 406, which in turn emits an output voltage signal when receiving a predetermined number of multivibrator pulse signals. Of the monolithic timing circuit NE / SE 555 provides a suitable monostable multivibrator 414 and is used by the Signetics Corporation.

609823/0299

As shown in FIG. 3, the counting circuit includes 406 the counters 444, 446 and 447, as well as connecting means 448 to the three counting circuits in appropriate sequence cascade to connect with each other. In order to obtain greater accuracy, it is desirable to have a large Number of counts from network 410 with a relatively low RC time constant. A suitable area for an RC time constant of network 410 is 0.5 ms to 5 ms. A suitable counting range for the counter 444 is between 500 and 4000. This relationship determines the timescale reference point. A binary counter 444 is particularly suitable, e.g. B. the binary counter / divider available from RGA Corporation CD4O4OAE, the one at the negative transition of the pulse of the monostable multivibrator progresses. The second comparison unit 448 of the circuit already described MC33O2P (the first unit of which was used for the comparator used) has its negative input terminal connected to a selection switch 452, while the output of the Amplifier is at the count input terminal of the decade counter 446. The binary counter is arranged to have a Provides output pulse for a certain desired number of operations of the multivibrator. The number you want is different for each connection of switch 452. The counter 444 is returned to its initial or starting state by the positive signal, which is normally from the comparator 408 is supplied to it and is caused to count when the output of the comparator decreases.

The counters 446 and 447 may be in the form of decade counters, with the number of output pulses being binary Counters 444 required to operate the trip circuit can be selected. These counters 446 and 447 collectively select the one of the plurality of timing curves that controls actuation of the trigger circuit 600.

The ten-position selector switches 450 and 451 are respectively connected in such a way to the decade counters 446 and 447,

609823/0 29 9

that for the purpose of a more precise selection 100 different time curves result which are based on the time reference curve. When the counters 446 and 447 each have a positive output pulse their respective selector switches 450 and 451, both diodes D462 and D464 are biased in the opposite direction, to operate the comparator 454, which may form the third unit of the MC33O2P device, which is also the comparator 408 and 448 included. The decade counters can be of the type sold under the designation CD4O17AE by RCA Corporation is delivered.

The time current relationships 426-432 can be selected in such a way that a trip curve 424 results, which can take an almost unlimited number of forms, in order to simulate any existing inverse time-current curve of conventional electromechanical overcurrent relays, such as the characteristic of the CO relay, currently manufactured and sold by the applicant. The values of the required resistors and capacitors can be determined by setting the number of time-current relationships on the curve to be simulated and at the same time setting the time-current relationship for each of the RC branches of the network 410 according to the above equation for Ιρ £ _{Β is} solved.

immediate triggering circuit

Immediate trip circuit 500 provides an output voltage signal to trip the circuit breaker 4 immediately after the magnitude of the current in any of the Phase lines L1, L2 or L3 and thus the DC voltage of the conversion circuit 200 has a predetermined value exceeds.

As shown in Figure 6, the immediate trip circuit 500 includes voltage dividing resistors R5O2, R5O4 and a rheostat (potentiometer) R5O6, a filter con-

609823/0299

capacitor C5O8, a resistor R510, a comparator 512 (this can be the fourth unit of the MC33O2P device already mentioned), and a Zener diode Z514 for overvoltage protection. When the level of the voltage across capacitor C508 reaches a magnitude _{equal to V R} , the output of comparator 512 provides a negative going signal to trip circuit 600. The wiper of potentiometer R506 adjusts the magnitude of the DC voltage from the conversion circuit 200, which is required in order to actuate the comparison circuit 500 which triggers the immediate trigger.

Trip circuit

A suitable trip circuit 600 for exciting the trip coil 6 of the circuit breaker 4 due to the negative going Output from either the time delay circuit 400 or the immediate trigger comparison circuit 500 is shown in FIG. 6. The trigger circuit 600 includes a display circuit 602, a Circuit breaker actuation circuit 604 and a display reset circuit 606.

A voltage drop at the output of the directly triggering circuit 500 causes a current to flow from the comparator 51.2 through the resistor R616 and a Zener diode Z614 into the base of a pnp transistor T6O8. This base current causes transistor T608 to raise the potential at terminal 618 to essentially the potential of V "+. A current then flows through three separate lines 620, 622 and 624.

The current in line 620 flows through a diode D626, a resistor R628 and through the comparator 512 to ground. This feedback loop keeps comparator 512 energized, until the line current in the phase lines L1, L2 and L3 is interrupted by the disconnector 4.

.60 9 823/0 2 * 9

The current on line 622 flows through diode D630 and the circuit breaker actuation circuit 604 where it is passed through a resistor R632, a diode D634 and into the control electrode of a thyristor SCR636 flows. The energized thyristor SCR636 closes a current path from the positive in the conductive state Connection 638 of the station battery 640 via the diode D642, the thyristor SCR636, the diode D644, which are normally closed Contacts 52a of the circuit breaker 4, the trip coil 6 to the negative terminal 646 of the battery 640. The trip coil 6 triggers disconnector 4 when energized. The capacitors C648 and C65O and the resistor R652 are provided, to prevent the thyristor SCR636 through Noise or leakage currents is excited. The contacts 52a are arranged on the circuit breaker 4 and are opened, when the circuit breaker 4 trips, so that the current through the coil 6 and the thyristor SCR636 is interrupted.

The circuit breaker actuation circuit 604 further includes a resistor R654 which has a negative voltage-resistance characteristic to absorb momentary spikes at terminal 638 of battery 640, further includes a discharge circuit 656, which consists of a resistor R658 and a diode D66O, around the stored in the trip coil 6 to discharge inductive energy, if it is not already consumed when the circuit breaker 4 closes again.

The display circuit 602 will only provide an display after the SCR636 thyristor has been energized. The display of a Immediate tripping is indicated by the flow of current in conductor 624 through diode 662, an electrically resettable indicator 664, a switch 666 with two positions (reset position for the display device and setting position), diodes D668, Coil 6, battery 640, diode D642, thyristor SCR636 and common Manifold 667, which is connected to ground, is generated.

The indicator 664 may be comprised of a light reflective electromagnetic status indicator that includes a

609823/0 299

has entered memory, whereby its display state is maintained even when the energizing current is increased by opening the contacts 52a. The display device will therefore continue the display after a single excitation, until the display device is reset. A suitable indicator 664 is from Ferranti-Packard Ltd. available in the stores.

The operation of the trip circuit 600 due to a negative The signal traveling from the time delay circuit 400 is essentially similar. A voltage drop on the Output of comparator 454 causes a base drive current for pnp transistor 670 through resistor R678 and the Zener diode Z676. The base current causes transistor T67O to conduct and increases the potential at point 680 essentially the potential V_ +. A current flows through conductors 682 and 684.

The current of conductor 682 flows through diode D686, conductor 622, resistor R632 and diode D634 and causes thyristor SCR636 to conduct and energize trip coil 6 in the manner already described. Current in conductor 684 flows between V _g + and the grounded line through diode D688, an electrically resettable indicator 689, selector switch 666, diode D668, contacts 52a, coil 6, battery 640, diode D643, and thyristor SCR636 . The display device 689 is essentially the same as the device 664 and also remains in the actuated state until reset, once the device has been actuated.

The Zener diodes Z614 and Z676 were provided to avoid an undesirable To prevent tripping of the circuit breaker 4 during the initial period of relay operation when the voltage between terminal 152 of power supply terminal 148 and ground has not yet reached the normal operating value V "+ Has. The breakdown voltages of the Zener diodes Z614 and Z676

609823/0299

should be high enough to prevent current flow during this period.

A conductor 690 is connected to conductor 682 so that the limit circuit 186 and the one operating the circuit breaker SCR636 thyristor can be energized at the same time. The limit circuit 186 prevents the failure of circuit components due to high voltages during the period during which the line current is caused by the opening of the circuit breaker contacts 8, 1O and 12 is interrupted. Currents flow between terminal 680 and ground through diode D686, line 682, line 690, a resistor R691, a diode D692 and the control electrode of the thyristor SCR190. When the thyristor SCR190 conducts, it connects the conductor 184 to ground, whereby the load resistors R146, R170 and R172 are bridged will.

to reset indicators 664 and 689 is a reset circuit 606 is provided which includes a resistor R693 and a capacitor C694 drawn from the battery 640 is loaded. When switch 666 is in the reset position, capacitor C694 overdischarges the display devices 664 and 689, the resistors R695 and R696, a Zener diode Z698 and the common line, which is connected to ground potential. The Zener diode Z698 has a breakdown voltage high enough to generate a To prevent power conduction through the busbar, except during the reset operation of the trip circuit 600,

It is conceivable that the DC voltage signal from the conversion circuit 200 continues to increase after it has actuated the time delay circuit 400 until it reaches a size which is sufficient to also actuate the comparison circuit 500, which is triggered immediately. So that this does not occur, is a diode D699 between the negative input terminal of the Comparator 512 and the common connection of the resistor

609823/0299

Stand R678 and the Zener diode Z676 provided. If the When delayed comparison circuit 454 emits its negative going signal, the cathode of diode D699 comes on a potential sufficiently low to permit operation of comparator 512 of the immediate trip circuit 500 and the subsequent making of the transistor T6O8 conductive and the excitation of the immediate triggering indicating Facility 664 to prevent. Therefore, the display circuit 602 is able to display whether the circuit is or the circuit 500 originally triggered the trigger circuit 600.

7 illustrates another embodiment 700 of the input circuitry 100 of FIG. 2. To simplify the explanation, only the one with the phase transformer CT1 connected device, while the circuits associated with the phase transformers CT2 and CT3 essentially are identical. The SCR136 thyristor and associated circuit and two diodes of the rectifier bridge RE110 are through SCR7O2 and SCR7O4 thyristors and associated circuitry, the capacitors C7O6, C7O8 and C710 and resistor R712 include, replaced.

The modified input circuit 700 ensures that the information winding 106 of the transformer CT1 after each Zero crossing of the line current in the phase line L1 is rendered ineffective. As has already been explained, the thyristor SCR136 shown in Fig. 2 is designed so that it switches off when the rectified current at its anode drops to a predetermined size and there for at least a minimum switch-off time remains. If the first derivative of the anode current curve with respect to time is such is that the current does not stay below the predetermined level for a minimum turn-off time, the thyristor will SCR136 never de-energized. However, if a pair of thyristors SCR7O2 and SCR7O4 instead of the rectifier pair in the Bridge circuit RE110 of Fig. 2 is used, as in

The RE710 bridge circuit is shown to everyone Thyristor has an alternating potential at its anode. When the SCR7O2 thyristor is energized, it provides a current path between the information winding 106 and the bus, which is connected to ground potential, through the Tyhristor SCR7O2 and resistor R146 if the upper terminal the bridge RE710 is positive. Similarly, the SCR7O4 energized thyristor provides a current path between the information winding 106 and the grounded bus through thyristor SCR7O4 and resistor R146. The operation of the input circuit 700 is similar in other respects to the input circuit of FIG. 1 and its operation will result from the corresponding description.

8 illustrates another embodiment 800 of the input circuitry 100. The input circuit 800 includes a dual regulated line supply for the delivery of positive and negative regulated voltage sources for the detection circuit 804 which requires these voltages. The input circuit 800 also provides an AC information signal, that is not rectified and on the line current through the phase lines L1, L2 and L3 reacted. To simplify the explanation, only one phase of the three-phase arrangement is shown for the following description.

The current transformer CT4 has a secondary power winding 806, a secondary information winding 808 and a primary winding 810. The primary winding 810 is excited according to the line current in the phase line L1. The number of turns N _{2 of} the information winding 808 is greater than the number of turns N. of the power winding 806. A center tap 812 on the power winding 806 is connected to the common line.

To simplify the explanation, it is assumed that the current in the phase line L1 has just passed through zero

609823/0299

and that the power winding 806 is effective and the information winding 808 as a result of the two directional thyristor diodes is not effective. Line current flows in phase line L1 in a direction that is the upper terminal of the winding makes positive with regard to the lower connection (this part of the wave is referred to below as the positive half-wave there is a charging current for the capacitor C816 between the grounded connection of the capacitor C816 and the grounded center tap point of winding 806 across the capacitor C816, diodes D818 and D82O and the lower one Half of the power winding 806. Current for charging the Capacitor C826 flows through the top between the grounded terminals of winding 806 and capacitor C826 Half of winding 806, diodes D822 and D824, and capacitor C826. During the opposite or "negative" Half-cycle of the line current in the phase conductor L1, a charge current flows into the capacitors C816 and C824 from power winding 806 through diodes D828 and D83O instead of diodes D82O and D822. In other words, while the "positive" half-cycle is the capacitor C816 of the lower half of winding 806 and capacitor C826 of the top half of winding 806 is charged. During the "negative" half-cycle, the relationship between the capacitor and the winding is reversed. The current running in the two halves of the winding 806 flows may or may not be the same, since it is only necessary that the entire ampere-turns in winding 806 are equal to the total ampere-turns in winding 810, and that the division of the current on the two halves is only determined by the state of charge of capacitors C816 and C824.

A control circuit. 814 is provided to control the voltage supply regulate and control the sequential operation of the secondary windings 806 and 808. The control circuit 840 includes a voltage regulator Z832 such as. for example a Zener diode, a switching device T834 such as an npn transistor, a capacitor C836,

603823/0299

Resistors R838, R84O and R842 and a bidirectional thyristor device 844. The term "bidirectional thyristor" is intended to mean any switching device which a first or anode connection, a second or cathode connection and a third or control connection and the following Features: A current flowing from the control terminal enables the device to have current in both Directions between the anode and cathode connection. Once the conduction has begun, the device can no longer be de-energized until the one between the anode and Cathode flowing current falls to a predetermined value for at least a predetermined period of time and remains there.

While capacitors 816 and 826 are in the above described When charging wisely, they will eventually reach a predetermined voltage value equal to the breakdown voltage the zener diode is Z832. When this occurs, a current is drawn into the base through the Zener diode Z832, the resistor R84O of the transistor T834 flow. The transistor T834 becomes conductive and current is passed through the control electrode of the bidirectional thyristor assembly 844, resistor R842, the transistor T834, the diode D82O or D822 and through one half of the power winding 806 flow. The conduction of the transistor T834 can either be in the "positive" or "negative" half-wave of the transformer.

The diodes D818 and D824 provide temperature compensation for the transistor T834 and thus continue to regulate the voltage across the capacitors C816 and C826. The diode D818 provides isolation for capacitor C816 from the leakage current of the transistor T834 or the Zener diode Z832.

The energized bidirectional thyristor device, hereinafter also called triac 844, makes the information winding 808 effective in the conductive state by creating a current path for current through the Information winding 808, the triac 844 and a charging resistor R846 is created. This in turn induces one

609823/0299

Voltage drop across power winding 806 and effectively terminates current flow therethrough in the already connected with Fig. 2 described manner. The difference between the embodiments shown in FIG. 2 and FIG. 8 lies in that the voltage across the charging resistor R846 provides an electrical quantity which is related to the line current in the Phase line L1 reacts, which changes polarity.

The triac device 844 de-excites each half-cycle of the line current in a manner similar to that of the thyristor SCR136 according to FIG the embodiment of Fig. 2, whereby the information development 808 is opened and ineffective and as a result of which the power - winding 806 becomes effective, - around the capacitors C816 and C826 to reload every half-wave.

Operation with saturable current transformers

9 illustrates an embodiment of the invention at which the current transformer CT5 has a single secondary winding. As shown is the turns to core ratio of transformer CT5 such that the core saturates within the anticipated network current size range. Again, for the sake of simplicity of explanation, only one phase of the input circuit is shown.

The saturable current transformer CT5 has a primary winding 902 excited by a magnitude proportional to the current in phase conductor L1, as well as a secondary winding 904. The secondary winding excites the series circuit of resistors R9O8 and R9O9 via a full-wave rectifier RE9O6. A power supply 148 is provided through a control circuit 910 and an input high voltage protection circuit 912 is connected to an output of the rectifier RE9O6 in shunt with the resistors R9O8 and R9O9. Diodes D914 and D915 prevent any reverse flow of power from the power supply circuit 148. The transformer

60 9 823/0 2-9-9

CT5 is designed in such a way that it saturates at a current which corresponds to twice the size of the network current (2PU). Instead, it can saturate at any other desired level.

To simplify the explanation, it is assumed that the line current in the phase line L1 has just passed through zero and that the energy storage devices C150 and C164 of the power supply 148 are initially de-energized. The energization of the primary winding 902 causes current to flow in the secondary winding 904 by the full wave rectifier RE9O6 is rectified. Output terminals 924 and 926 of rectifier RE9O6 are between the ground bus and a conductor 928 connected.

Initially (with capacitors C150 and C164 discharged) the impedance of power supply 148 is less than the total impedance of resistors R9O8 and R9O9 and essentially all of the rectified secondary current flows from terminal 926 through conductor 928 to power supply 148. When the line current is below 2PU , a current flows during one or more cycles to charge the power supply 148. The number of cycles required depends on the size of the line current and the capacitance of capacitors C150 and C164. Eventually, the voltage across the capacitors (and across the series resistors R9O8 and R9O9) will reach the desired value to provide outputs V _g + and V _R. As soon as this occurs, the control circuit 910 is actuated and causes current to flow between the conductor 928 and the grounded bus through the resistor R940 and the diode Z936.

The control circuit 910 comprises npn transistors T93O, T932 and T934 as well as the Zener diodes Z936 for voltage regulation and resistors R938, R94O and R942. Initially, transistor T93O conducts to provide base current to transistor 932, woas by this a low impedance path between

609823/0299

the conductor 928 and the power supply 148 to provide the flow of charging current in the manner described above. By arranging the transistors in the manner shown, a current flow through the transistor T932 is due of a small current flowing into the base of transistor T93O. When the magnitude of the voltage of the power supply 148 reaches the desired value, the potential is line 928 high enough for zener diode Z936 to break down and conduct current through resistor R94O. this leads to to a current flow in the base of the transistor T934 and brings it to conduct.

When the transistor T934 conducts and the potential at the base of the transistor T93O is reduced, the conduction is reduced of transistor T932 and a state of equilibrium is in the way achieved that the current through the transistor T932 drops to a size that is necessary to the desired Maintain voltage value of the power supply 916, as it is determined by the Zener diode Z936.

When the chain of power supply 916 compared with the current supplied by the transformer CT5 is low, there is a main current flow through the load resistors R908 and R9O9. Resistors R9O8 and R9O9 provide a voltage divider circuit that their relative sizes depend on the relative size of the voltage of the power supply 148 and the desired output voltage range of the peak voltage averaging circuit 200 depend. The voltage across resistor R909 provides the information signal for the high select circuit 174 for supplying an information signal to the peak voltage averaging circuit based on the magnitude of the highest line current in any of the phase lines L1, L2 and L3 is responsive.

The Zener diode Z936 is a temperature compensated device and regulates the voltage of the power supply 916. This is because the diodes D914 and D915 are added

609823/0299

tend to eliminate the temperature effect of the base-emitter junction of transistor T938.

The input overvoltage protection circuit 912 prevents breakdown of the circuit components due to excessive voltages that may be caused by high leakage currents. The protection circuit 912 comprises a zener diode Z946, npn transistors T948 and T95O and resistors R952, R954 and R956. A second series path is provided by transistors T948 and T95O, which connect conductor 928 to the grounded bus when zener diode Z946 conducts. A suitable size for the Zener breakdown voltage ^V Z946 ^{can be 20} ° ^V.

The use of saturating current transformers, which will saturate over substantially all of their normal operating range, will reduce the stress on the current transformer, while at the same time an inverse time-current response is achieved for current input ranges that are far lie above the saturation current level. In this regard, it should be remembered that if the saturation of the current supplied to the transformer is reached in a shorter period of time. Because the height of the current pulse in the secondary winding depends on the increase in the transformer core magnetic flux (and thus the input current) the peak value of the voltage across the charging resistor will increase when the level of the input current increases. In the area below saturation there is a straight line relationship and the secondary current becomes -the primary current during the whole Follow half wave. If the level of the input current increases above the core saturation value, the current will increase as it increases Input currents flow, but only during part of each half-cycle, reducing the load on the primary-phase line during the time periods when no current flows in the secondary winding is greatly reduced or eliminated entirely.

609823/0299

The operation of the current transformers in the saturation range provides a constant rms current output and therefore such an arrangement uses the peak values to generate the Acquire input current values in the saturation range.

Although the use of saturating current transformers has been described in connection with the apparatus of FIG. Such a transformer can of course also be used in the construction with double secondary winding which the windings are used one after the other in each half-wave. If a construction with two secondary windings is used, the transformer should only saturate during the time periods when the size sensing surges to be used.

Patent claims:

6 0 9 8 2 3/0

Claims (30)

F a f e η t a n 's ρ r ü c h e: Electrical protective switching device, which is opposite to a electrical parameters is sensitive, characterized by a first transformer (CT1 in Fig. 1,2 and 7; CT4 in Figure 8; CT5 in Fig. 9), which has an input winding (L1) which is excited by an AC network and output winding means (NI, N2 in FIGS. 2, 7 and 8; 900 in FIG. 9) which have a first and a second electrical quantity (101 or 102) deliver, wherein the first electrical quantity represents an energy source for an energy storage device (148) and the second electrical quantity has a value which is different from the value of the electrical parameter in the AC network is determined, further characterized by control devices (120 in Fig. 1 and 2; 700 in Figure 7; 800 in Figure 8; 910 in Fig. 9) showing the output winding devices in the manner control that the first and the second electrical quantity successively during a half cycle of the alternating current of the network are switched to that of the energy storage device (148) during the previous Half-wave to replace consumed energy and to duplicate the current of the AC network to deliver. 2. Electrical protective switching device according to claim 1, characterized in that the output winding devices have first and second output windings (N1, N2) which are traversed by the same magnetic flux, the first output winding being the first electrical Size and the second output winding supplies the second electrical size. 3. Electrical protective switching device according to claim 1 or 2, characterized in that the control devices a first (T124 in Figs. 2 and 7; T834 in Fig. 8) 609823/0299 and second (SCR136 in Fig. 2; SCR7O2 in Fig. 7; 844 in Fig. 8) comprises switching devices, the first switching devices being capable of the first output winding ineffective and the second output winding effective, and wherein the second switching device is able to make the second output winding ineffective and the first output winding effective. 4. Electrical protective switching device according to claim 3, characterized in that the first switching device has semiconductor elements (Z122 in FIGS. 2 and 7; Z832 in FIG. 8), a resistor (R146 or R846) and at least one electronic switch (T124 in FIG and 7y T834 in Fig. 8), the semiconductor elements being connected to electrically shunt the energy storage device (148) to limit the maximum voltage level across the storage device to a substantially predetermined value and to energize the switching device when the predetermined maximum voltage level above the storage device (148) is reached, wherein the switching device can assume an excited and a de-excited state and is connected to the second output winding (N ₂ ) and the resistor (R146 or R846) in such a way that the second Output winding (N ₃ ) effective and the first output winding (N _- .) Is ineffective when the switching device is in the excited state. 5. Electrical protective switching device according to claim 4, characterized in that the resistor (R146 or R846) is connected to the second output winding (N ₃ ) so that the second electrical variable generates a voltage across the resistor when the second output winding is made effective . 609823/0299 6. Electrical protective switching device according to claim 4 or 5, characterized in that the voltage level • across the first output winding (N-) when the first output winding (N ₁ ) is inactive and the second output winding (N2) is active is essentially equal to N ../ N_ times the voltage across the resistor (R146; R846), where the first winding (N ₁ ) turns and the second winding (N ₂ ) turns and N _{1 is} less than N ₂ . 7. Electrical protective switching device according to claim 6, characterized in that the resistor (R146; R846) is chosen so large that for all foreseeable values of the electrical parameters of the AC network of Nj / N ₂ - £ ache value of the voltage across the resistor Has magnitude that is less than the magnitude of the predetermined maximum voltage across the energy storage device (148). 8. Electrical protective switching device according to one of the preceding Claims 4 to 7, characterized in that the second switching device devices (SCRl36 in Fig. 2, SCR7O2 in Fig. 7) for de-energizing the switching device when the value of the second electrical quantity falls below the predetermined value. 9. Electrical protective switching device according to claim 8, characterized in that the device (SCR136 or SCR7O2) is a thyristor to de-energize the switch. 10. Electrical protective switching device according to one of the preceding Claims 4 to 9, characterized in that the semiconductor devices have at least one Zener diode (Z122 in Figs. 2 and 7; Z832 in Fig. 8). 609823/0299 11. Electrical protective switching device according to claim 1, 2 or 3 / characterized in that the energy storage device (148; 800) a first capacitor (C816 in Fig. 8) and a second capacitor (C826) and diode connection means (D818, D824), the first capacitor having a negative regulated voltage supply and the second capacitor forms a positive regulated voltage supply and the diode connection means connect the capacitors to the winding devices in such a way that the capacitors be charged at the same time, but in the opposite direction. 12. Electrical protective switching device according to one of the preceding Claims 1 to 10, characterized in that the energy storage device (148; 800) at least a capacitor (C150 in Figs. 2 and 7; C816 in Fig. 8) having. * 13. Electrical protective switching device according to one of the preceding Claims 1 to 12, characterized by rectifier devices (RE108, 110 in Fig. 2, 7; D82O-D83O in Figure 8; RE9O6 in Fig. 9), which is the first rectify electrical quantity and through a measuring circuit device (400 in FIGS. 1, 3 and 9; 804 in FIG. 8) for supplying an electrical quantity when the value the second electrical quantity exceeds a predetermined value. 14. Electrical protective switching device according to claim 13, characterized in that the measuring switching device includes an activation circuit (402 in Fig. 3), pulse generating devices (414) and digital counting means (406), the circuit activation means suppress and de-energize the measuring switching device, if not the value of the first electrical Size exceeds a predetermined intake value, 6 0 9 8 2 3/0 2-9-9 and wherein the pulse generating means are pulse shaped Supply voltage signals to the digital counting device, the digital counting devices being a time-delayed Output signal when receiving a predetermined number of the pulse-shaped voltage signals. 15. Electrical protective switching device according to claim 14, characterized in that the measuring switching devices further comprise an instantaneous comparing circuit (500 in Figures 1 and 6) to provide an output signal that is not time delayed when the value of the second electrical quantity exceeds a predetermined value which is greater than the time-delay pick-up value. 16. Electrical protective switching device according to claim 15, characterized by a trip circuit (600 in Figs. 1 and 6) for interrupting the flow of the electrical Parameters in the AC network, the trigger circuit ■ either by the instantaneous Output signal or is excited by the delayed output signal. 17. Electrical protective switching device according to claim 16, characterized in that the trigger circuit Indicators (602 in Fig. 6) includes which only come into operation when the flow of the electrical Parameter in the AC network was interrupted, the display devices still continue indicate which of the output signals was first picked up by the trip circuit, with the Display devices comprising at least one electrically resettable display (689) and blocking devices (666) consists of the display device until it is electrically reset for further display after a one-time Causes excitement. 609823/0299 18. Electrical protective switching device according to claim 13, if this is dependent on claims 4 to 10, characterized by limiter devices (186 in FIG. 1 and 2) to make the measuring circuit device insensitive, when the magnitude of the current in the AC network exceeds a predetermined value. 19. Electrical protective switching device according to claim 18, characterized in that the current transformer contains a core (CT5 in Fig. 9) made of magnetically saturable material and the limiter devices contain predetermined values for the resistors and for N ₁ and N ₂ in such a way that the Core in connection with the predetermined values saturates when the current of the AC network exceeds a predetermined value. 20. Electrical protective switching device according to claim 18, characterized in that the limiter devices (186) a shunt circuit (Z188 in Fig. 2) for shunting of the second output current from the resistor (146) when the current in the AC network is the exceeds a predetermined value. 21. Electrical protective switching device according to claim 20, characterized in that the shunt circuit a voltage-sensitive switching device (Z188) and Connection devices (D181) comprises, which the voltage-sensitive Connect switching device in parallel with the resistor (146) so that when the voltage across the resistor (146) exceeds a predetermined value, the second output current from the resistor (146) is shunted through the switching device. 22. Electrical protective switching device according to claim 21, . characterized in that the voltage-sensitive switching device is a Zener diode (Z188), a 609823/0299 Thyristor (SCR190) and connection devices (C196, R192, R194) includes the Zener diode (Z188) with the Connect thyristor (SCR19O) in such a way that the second Output current flows through the thyristor (SCR196) when the voltage level across the zener diode (Z188) is Breakdown voltage exceeds. 23. Electrical protective switching device according to claim 14, if this is dependent on claims 4 to 10, characterized in that the measuring switching device (400; 804) has a predetermined time delay (TD) in introduces the output of the digital counter (406) after the magnitude of the generated across the resistor Input voltage exceeds a predetermined intake value. 24. Electrical protective switching device according to claim 23, characterized in that the pulse generating devices .A monostable multivibrator (414) and triggering devices (412), the triggering devices trigger a series of voltage pulses, each of a size sufficient to generate the to excite monostable multivibrator, the monostable multivibrator due to the series of triggered Voltage pulses supplies a series of rectangular pulse-shaped voltages to the digital counter (406), each having a predetermined pulse width (TD2). 25. Electrical protective switching device according to claim 24, characterized in that the tripping devices comprises an RC network (410) of resistance and capacitance elements, also resetting devices (422) for the momentary de-excitation of all capacitive elements, further connecting devices (200), which the RC Connect the network (410) to the resistor (R146) and to the monostable multivibrator (414) , the release device 609823/0299 connections (412) require a predetermined time period (TD1) after the momentary de-excitation of the capacitive elements in order to generate an RC network voltage signal,
the value of which is large enough to excite the monostable multivibrator, the time delay (TD1) changing inversely with the value of the DC input voltage. 26. Electrical protective switching device according to claim 25,
characterized in that the RC network (410) comprises parallel branches (T414-T42O), each branch including resistance and capacitance elements, the connection means connecting the branches being of such a type
are arranged so that each branch supplies a predetermined branch output voltage which has an inverse time characteristic, the sum of the branch voltages being equal to the RC network voltage signal. 27. Electrical protective switching device according to claims 23
to 25, characterized in that the digital counting means (406) comprises a precision time-scale selector (450, 451, 452) capable of changing the respective number of rectangular pulse-shaped voltage signals required to generate the causing digital counter (406) to provide an electrical output. 28. Electrical protective switching device according to claim 25,
characterized in that the predetermined time delay (TD) of the measuring switching devices corresponds to the value C equals (TDI + TD2), where C is a predetermined constant is. 29. Electrical protective switching device according to claim 15,
characterized in that the input voltage generated across the resistor (146) has a magnitude equal to
the highest value of the phase currents in the AC network reacts, and that a plurality of transformers 6 09823/0 2-9-9 gates (CT2 and / or CT3 in Figs. 1 and 2) that are the same the first transformer (CT1) are provided / a transformer being provided for each phase current, further a maximum value selection circuit (174 in 1 and 2) connected to the second output winding devices of each transformer, in order to deliver such an electrical quantity that is only sensitive to the highest value of the phase currents in the network, and a conversion circuit (200 in Figs. 1 and 3) which has the electrical magnitude with the highest Value is converted into a DC voltage in order to feed it to the measuring switching device (400; 804). 30. Electrical protective switching device according to claim 29, characterized in that the instantaneous triggering circuit is connected such that it is on electrical output signal provides when the highest value of the phase currents in the network a predetermined Value exceeds. ES / hs 3 609823/0299 Blank page