DE10303267A1 - Control process for brushless three phase motor system works on fifty percent cycle ratio - Google Patents

Control process for brushless three phase motor system works on fifty percent cycle ratio

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Publication number
DE10303267A1
DE10303267A1 DE10303267A DE10303267A DE10303267A1 DE 10303267 A1 DE10303267 A1 DE 10303267A1 DE 10303267 A DE10303267 A DE 10303267A DE 10303267 A DE10303267 A DE 10303267A DE 10303267 A1 DE10303267 A1 DE 10303267A1
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DE
Germany
Prior art keywords
switch
current
phase
duty cycle
bridge
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
DE10303267A
Other languages
German (de)
Inventor
Hirohide Satoh
Sigeki Furuta
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Denso Corp
Original Assignee
Denso Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority to JP2002018712 priority Critical
Priority to JP2002348533A priority patent/JP4062074B2/en
Application filed by Denso Corp filed Critical Denso Corp
Publication of DE10303267A1 publication Critical patent/DE10303267A1/en
Withdrawn legal-status Critical Current

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Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14Electronic commutators
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/08Arrangements for controlling the speed or torque of a single motor
    • H02P6/085Arrangements for controlling the speed or torque of a single motor in a bridge configuration

Abstract

A control process for a brushless three phase motor (1) system (10) has an inverter with upper (21-23) and lower (24-26) bridges. One side of the inverter is switched by a signal synchronous with a rotor angle in a 50% cycle ratio and the other side with pulse width modulation when a bridge switch is open. A control process for a brushless three phase motor (1) system (10) has an inverter (20) with three phase bridge arms each with upper (21-23) and lower (24-26) side switches. One side of the inverter is switched by a three phase signal, synchronous with a rotor angle in a cycle ratio of 50%. The other side of each bridge is switched with a pulse width modulation signal during a period in which the switch on the one side of the bridge is not conducting. Independent claims are also included for a control device and for a motor system.

Description

    BACKGROUND OF THE INVENTION 1. Field of the Invention
  • The invention relates to a brushless 3-phase motor system with one brushless 3-phase motor or a synchronous machine and a controller, and More particularly, relates to a method and controller for controlling the brushless 3-phase motor to make it work based on a 50% duty cycle to let.
  • 2. Description of the prior art
  • Three phase brushless DC motors are well known. These engines are operated under a well known converter control, in which a DC voltage is converted to AC voltage by a pair of PWM (Pulse width modulation) signals upper and lower switching elements (MOSFETs (metal oxide Semiconductor field effect transistors) or IGBTs (isolated gate bipolar transistors)) each of three bridge branches of a 3-phase converter are created around this to cause a switching operation to be performed. Two of well known ones Control schemes of a duty cycle of the PWM signals consist of 120 degrees Duty cycle schemes in which the duty cycle period of each bridge branch is the same with 2π / 3, corresponding to an electrical angle, and a 180 degree Duty cycle diagram in which the duty cycle period of each bridge branch π in electrical degrees.
  • In the 180 degree duty cycle, two of the three Windings or bridge branches are continuously in an ON state, the current on two ON-state windings or bridge branches distributed. This reduces in advantageously the loss of resistance and heat. Since that too is 180 degrees Duty cycle scheme in terms of the duty cycle period is longer than 120 degrees Duty cycle scheme, the 180 degree duty cycle scheme can have a larger output deliver compared to the 120 degree duty cycle, even in a high speed operating range in which an electromotive counterforce thereby occurs by having sufficient current from the DC power supply or Battery is fed to the engine. The torque of the brushless 3-phase motor is controlled by the PWM signals that are sent to the switch elements of the 3-phase Converter.
  • To get a target torque from the brushless 3-phase motor, the pulse duty cycles of the PWM signals are usually according to one Feedback controlled to achieve that the difference between the Target torque and the detected torque is zero by the Torque current of the brushless 3-phase motor is detected.
  • As a current detection method in brushless 3-phase motor systems is a Method well known in which the current from each of the three phases of the brushless 3- Phase motor is detected, a method is known in which the current is detected which is by the lower switch element of each bridge branch in the Converter circuit flows, and a simple process in which the converter circuit flowing current is detected.
  • However, in the conventional 180 degree duty cycle scheme, the top ones and the lower switch elements turned ON and OFF at the same time, with the result has that the switching losses are larger and consequently larger switch element drivers are needed. Supported for mass production of 180 degree duty cycle Brushless 3-phase motors for vehicles require it Circuit configuration and / or the control method of the controller of the motor system simplify and / or reduce energy consumption.
  • What is needed is a control method for a brushless 3-phase motor, which reduce switching losses with a simplified circuit configuration can.
  • What is needed is also a control unit for a brushless one 3-phase motor, which has a reduced switching loss and a simplified Has circuit configuration.
  • What is also needed is a brushless 3-phase motor system, which has a reduced switching loss and a simplified circuit configuration.
  • SUMMARY OF THE INVENTION
  • According to one aspect of the invention, a method for controlling a brushless 3-phase motor created. The brushless 3-phase motor is in one Motor system included, which also contains a converter circuit with 3-phase Bridge branches. Each bridge branch comprises an upper-side switch element and one underside switch element. The process comprises the following steps: Switching of the switch elements on one side of the converter circuit under Use of first 3-phase signals with an angular position of a rotor are synchronized and have a duty cycle of 50%; and switching the Switch elements on the other side for each bridge branch with one Pulse width modulation signal (PWM) during a period in which the switch element on the one side of the bridge branch is not conductive.
  • In a preferred embodiment, the respective forward directional currents of the switch elements measured on one side. It will Duty cycle of the PWM signal based on a predetermined threshold and one Sum of measured forward currents determined to be a maximum current or Limit the amount of heat that is permitted for the switch element on the other side are.
  • The switch elements on one side can either be from the top Switch elements or the underside switch elements exist.
  • According to another aspect of the invention, an apparatus for controlling a brushless 3-phase motor created. The brushless 3-phase motor contains a rotor and a device for outputting information, the one Specify the angular position of the rotor. The device comprises a converter circuit with 3-phase Bridge branches, each bridge branch having a top switch element and a has switch element on the underside; one based on the information on the Switching the switch elements of the converter circuit located on one side operable device, wherein first 3-phase signals are used, which are synchronous with the angular position of the rotor and a duty cycle of essentially 50% exhibit; a device that is provided for each bridge branch to one measure the first current of the switch element on one side of the bridge arm; and a facility based on not only the information but also on the basis of the measured first currents for the 3-phase bridge branches works with using the switch element on the other side for each bridge branch a pulse width modulation (PWM) signal during a period which the switch element on one side of the bridge arm is not conductive.
  • According to another aspect of the invention, an engine system includes one brushless 3-phase motor and the device described above is provided.
  • BRIEF DESCRIPTION OF THE DRAWINGS
  • Further objects and advantages of the present invention result from the following description of preferred embodiments of the invention as set forth in are illustrated in the accompanying drawings, in which:
  • Fig. 1 is a schematic diagram of the invention showing an arrangement of a brushless 3-phase synchronous machine system for use in starting an engine and generating electricity;
  • FIG. 2 is a schematic circuit diagram showing an arrangement of the controller 3 of FIG. 1 according to an illustrative embodiment of the invention;
  • Fig. 3 is a timing diagram% (ie 180 degrees drive operation) is the waveforms of the pulse width modulation signals (PWM) and the 3-phase winding currents in the Tastverhältnisbetrieb according to 50;
  • FIGS. 4A and 4B are diagrams showing the manner in which the currents in each of the 6 sub-cycles flow, forming a cycle or revolution of the rotor;
  • Fig. 5A is a flowchart showing an initial setting operation, which is executed by a computer 30 at a time of engine start;
  • FIG. 5B is a flow chart showing a control or regulating operation of the engine 1, which is adapted to a target torque;
  • Fig. 6 is a circuit diagram partially showing an arrangement of a motor controller unit in which the winding currents are detected using MOSFETs with the aid of a current sensor electrode according to a modified embodiment of the present invention;
  • Fig. 7 is a circuit diagram illustrating another embodiment of the circuit 50 a of Fig. 6; and
  • FIGS. 8A and 8B are circuit diagrams partially illustrating a 3-phase motor controller unit in which the winding currents are detected on the basis of currents flowing through the top switch elements of the bridge arms of the converter circuit.
  • In all drawings are the same elements if they are more than one figure are shown with the same reference numerals.
  • DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
  • Fig. 1 is a schematic diagram showing an arrangement of a brushless synchronous 3-phase machine and system according although for use in starting an engine and generating electricity to the invention. In FIG. 1, the brushless 3-phase synchronous machine system 10 comprises a wound rotor-synchronous machine or brushless 3-phase motor 1 and a motor controller unit 2 for controlling and driving the brushless 3-phase motor 1 .
  • The brushless 3-phase motor 1 comprises a rotor with a U-phase winding 11 , a V-phase winding 12 and a W-phase winding 13 , which are connected in a star configuration; a field magnet winding 14 ; a field magnet current controller 15 for supplying a desired current to the field magnet winding 14 ; and a resolver or rotor angle sensor 16 which has three-phase windings and which provides 3-phase signals Au, Av and Aw indicating the angular position of the rotor.
  • The field magnetic current controller 15 has the same construction as the regulator of conventional AC dynamos. In a machine start operation, electric current is fully applied to the field magnet current controller 15 . In the generation of electricity, the controller 15 performs feedback control of the current flowing through the field magnet winding 14 to make the voltage between the terminals of a DC power supply source, not shown, constant in the same manner as in the case of the regulator of conventional AC machines.
  • The motor controller unit 2 comprises a 3-phase converter 20 for driving the motor 1 ; and a controller 3 for controlling the 3-phase converter 20 using the output signals of the angle sensor 16 and a torque command issued from the outside.
  • The converter 20 comprises three bridge branches for three phases and a smoothing capacitor 6 . A U-phase bridge arm includes an upper circuit breaker 21 , a lower circuit breaker 24 and a winding current measuring resistor 27 which are connected in series, and the connection point between the circuit breakers 21 and 24 is connected to the U winding 11 . A V-phase bridge arm comprises an upper circuit breaker 22 , a lower circuit breaker 25 and a winding current measuring resistor 28 which are connected in series, the connection point between the circuit breakers 22 and 25 being connected to the V winding 12 . A W-phase bridge arm comprises an upper circuit breaker 23 , a lower circuit breaker 26 and a winding current measuring resistor 29 which are connected in series, the connection point between the circuit breakers 23 and 26 being connected to the W winding 13 . Each of the power switches 21 to 26 comprises a power MOSFET (metal oxide semiconductor field effect transistor) and a free-wheeling diode (D) which is connected in anti-parallel to the power FET. The diodes (D) can either be parasitic free-wheeling diodes or dedicated diodes. The upper switch end of each bridge arm is connected to a positive line LH of a battery power supply source, not shown, and the resistance of each bridge arm is connected to a negative line LL of the battery power supply source, not shown, which has a battery voltage across the smoothing capacitor 6 and the positive Line LH and the negative line LL feeds.
  • The smoothing capacitor 6 , which has a large capacitance, is connected between the output terminals of the DC power supply source, not shown, to reduce adverse effects caused by a surge voltage by the switching operation of the inverter 20 on the DC power supply source, not shown, and the radiation suppress electromagnetic interference signals.
  • In the specific embodiment of Fig. 1, the lower resistance resistors are used as current detector elements 27 to 29 . However, any suitable elements such as Hall devices can be used for the current detector elements 27 to 39 .
  • The controller 3 is energized only during a period according to an ON state of the ignition switch and during a period of starting the engine after an idle stop. During such an energy supply period, the controller 3 supplies the gate voltages VU u , VV u , VW u , VU L , VV L and VW L in gate electrodes of the switch elements 21 to 26, respectively. The gate voltages VU u , VV u and VW u have a pulse duty factor of 50% (ie logically "1" for 180 degrees of a rotor rotation angle) and differ in phase by 2π / 3 from each other. The gate voltages VU L , VV L and VW L are inverted versions of the respective voltages VU u , VV u and VW u .
  • FIG. 2 shows a schematic circuit diagram showing an arrangement of the controller 3 of FIG. 1 in accordance with an illustrative embodiment of the present invention. In Fig. 2, controller 3 includes a microcomputer 30 which enables controller 3 to operate in response to operating conditions or conditions; Amplifiers 31 to 33 , the input terminals of which are connected to the respective connection points between the lower switch elements and the current measuring resistors of the converter circuit 20 ; Diodes (D ') whose anodes are connected to the respective output terminals of amplifiers 31 to 33 ; and an adder 34 having three input terminals connected to the respective cathode terminals of the diodes (D '); a comparator 35 whose positive (+) input terminal is supplied with a threshold voltage Vth (determined by the computer 30 ) and whose negative (-) input terminal is connected to the output terminal of the adder 34 ; Comparators 36 to 38 , the negative input terminals of which are supplied with a reference voltage V0 (determined by the computer 30 ) and the positive input terminals of which are connected to the respective output terminals Au, Av and Aw of the angle sensor 2 , AND gates 39 to 41 with two inputs , of which a first input terminal is connected to the output of the comparator 35 and the second input terminal of which is connected to the respective output terminal Su, Sv and Sw of the comparators 36 to 38 ; Power amplifiers 43 , 45 and 47 of the non-inverting output type, the input terminals of which are connected to the respective output terminals of the AND gates 39 to 41 and the outputs VU u , VV u and VW u of the respective gate electrodes of the U, V and W phase upper Switch elements 21 to 23 are connected; and inverting output type power amplifiers 42 , 44 and 46 whose input terminals are connected to the respective output terminals Su, Sv and Sw of the comparators 36 to 38 and whose outputs VU L , VV L and VW L are connected to the respective gate terminals of the U-, V - And W phases lower switch elements 24 to 26 are connected.
  • The microcomputer 30 may consist of any suitable single-chip / multi-chip special / finished microcomputer which includes a CPU (central processing unit), not shown, a ROM (read-only memory), a RAM (random access memory), various IO (input and output) circuits, etc., as is well known in the art.
  • The operation of the 3-phase synchronous machine system 10 configured in this way is described below. In response to control commands or signals or making a decision based on detected information, the microcomputer 30 performs an operation according to an electricity generation operation (as a dynamo) in an ordinary running state, an electric drive operation (as a motor) for starting the engine, and an electric drive -Operation (as a motor) for delivering a target torque which is specified by a torque command which is output for the torque support.
  • In the electricity generating operation as a dynamo in the ordinary running state, the switch elements 21 to 26 are kept OFF to effect 3-phase full-wave rectification via the free-wheeling diodes (D) contained in the switch elements 21 to 26 and so on control is also performed to converge the voltage between the terminals of an unillustrated power supply source (or battery) to a predetermined value in the same manner as that performed by the regulator of the conventional AC machines. Such an electricity generation operation is well known and therefore will not be explained in detail.
  • The electric drive operation as a motor is described in detail below described.
  • Electric drive operation to start the machine
  • In the electric drive operation to start the engine, the computer 30 initially performs a setting operation in accordance with the flowchart shown in FIG. 5A. When the user operates the ignition switch, the subroutine of Fig. 5A is called. In Fig. 5A, the computer 30 first winding current controller 15 a to cause a maximum current flowing through the field magnet winding 14 in the step 60. The computer 30 sets the threshold voltage Vth to a value corresponding to the maximum allowable current of the switch elements 21 to 26 , which is done at step 62 , and then returns. The inverter circuit 20 is then activated to start an electric drive operation.
  • Then the angle sensor 16 outputs 3-phase sine wave voltages Au, Av and Aw, which indicate the angular position of the rotor. The voltages Au, Av and Aw are fed to the comparators 36 to 38 which compare the respective voltages with the reference voltage V0 in order to convert the respective voltages into 3-phase pulse signals Su, Sv and Sw which are synchronous with the angular positions of the respective magnetic poles of the rotor vary and which have a pulse duty cycle of 50% and a phase difference of 120 degrees to each other.
  • The voltages Vu, Vv and Vw, which are detected by the current measuring resistors 27 to 29 and which are proportional to the respective currents which flow through the lower switch elements for the U, V and W phases, are given by the respective amplifiers 31 to 33 are rectified by the respective diodes (D ') and are added in the adder 34 to form a voltage Vs which is proportional to the sum of the detected voltages Vu, Vv and Vw and corresponding to the sum of the currents flowing through the lower switch elements 24 to 26 . It should be noted that the diodes (D ') can be replaced by full wave rectifiers or absolute value generators.
  • The comparator 35 compares the voltage Vs with the threshold voltage Vth. The threshold voltage Vth is determined based on the maximum allowable current of the switch elements 21 to 26 . The comparator 35 outputs a high level voltage when the adder 34 outputs Vs which is smaller than the threshold voltage Vth, and otherwise outputs a low level voltage.
  • The 3-phase pulse signals Su, Sv and Sw output from the comparators 36 to 38 are power amplified and are inverted by the power amplifiers 42 , 44 and 46 from the inverting output to become gate control signals VU L , VV L and VW L to be applied to the gate terminals of the lower switch elements 24 to 26, respectively. Also, the 3-phase pulse signals Su, Sv and Sw are supplied to the power amplifiers 43 , 45 and 47 of the non-inverting output type through the AND gates 39 to 41 . The output voltages VU u , VV u and VW u of the power amplifiers 43 , 45 and 47 of the non-inverting output type are applied to the gate electrodes of the upper switch elements 21 to 23, respectively.
  • If the output of the comparator 35 is at the high level or the voltage Vs is less than the threshold voltage Vth or if the sum of the currents flowing through the lower switch elements 24 to 26 is less than the maximum permissible current value, the result Power amplifiers 43 , 45 and 47 amplified versions of the signals Su, Sv and Sw as signals VU u , VV u and VW u, respectively. In other words, if no over-current flows in the windings, the upper switch elements of the inverter circuit 20 3-phase pulse signals logic and the negative controlled 21 to 23 and the lower switch elements 24 to 26 by the positive logic 3-phase pulse signals, the duty cycle of a of 50% and differ in phase by 120 degrees.
  • When the output of the comparator 35 is low or the voltage Vs exceeds the threshold voltage Vth, that is, when the sum of the currents flowing through the lower switch elements 24 to 26 exceeds the maximum allowable current value, the power amplifiers 43 , 45 give and 47 the low level as signals VU u , VV u and VW u regardless of the output levels Su, Sv and Sw of the comparators 36 to 38 to cause the upper switch elements 21 to 23 to go to the OFF state. Thus, the lower switch elements 24 to 26 (in this specific example) are controlled by the pulse signals with a 50% duty cycle, but the upper switch elements ( 21 , 22 , 23 in this specific example) of each bridge branch during a PWM period. are controlled in which the lower switch element of the bridge arm is non-conductive to prevent an overcurrent from flowing.
  • The switch elements (the upper switch elements 21 to 23 in this specific embodiment) that are PWM controlled are called "switch elements of the PWM controlled side". The switch elements (the lower switch elements 24 to 26 in this specific embodiment) which are controlled by the pulse signals with a duty cycle of 50% are referred to as "switch elements of the 180 degree duty cycle controlled side" hereinafter.
  • Fig. 3 shows a timing chart representing the following shapes:
    3-phase positive logic and negative logic pulse width modulation (PWM) signals
    VU U (for element 21 ), VU L (for element 24 ),
    VV U (for element 22 ), VV L (for element 25 ),
    VW U (for element 23 ) and VW L (for element 26 ), and 3-phase winding currents
    iu,
    iv, and
    iw
    when operating at 50% duty cycle (ie, 180 degree duty cycle (or conduction) operation according to the invention. Fig. 4 is a diagram illustrating how currents flow in each of 6 sub-cycles, one cycle of revolution form of the rotor. in Fig. 3 is a period from a rotor cycle or revolution (2 π) in 6 sub-cycles T1 divided to T6. Each of the periods are shown with dots, is a period in which the relevant switching element ( 4 indicated by a solid black circle in the left column of Fig. 4) ON or conductive under control according to a 50% duty cycle based on the angular position signals Au, Av and Aw from an angle sensor (or resolver) 16. Each of the periods, which is represented by vertical stripes, is a period in which the relevant switch element (which is marked by a cross-in-a-circle symbol) is ON or conductive, namely below the PWM control based on both the angular position signals Au, Av and Aw and the winding currents iu, iv and iw, which are measured by using the current measuring resistors 27 to 29 . In the other periods, which are shown as a low level, the respective relevant switch elements (which are marked with white filled circles) are OFF or non-conductive. Also, in each of the graphs of the 3-phase winding currents iu, iv, and iw, the positive sections drawn on the upward arrow side indicate that the phase winding currents flow out of the respective windings, but the negative sections that follow on The arrow side pointing below indicates that the phase winding currents flow into the respective windings. As can be seen from Figure 3, when the upper switch element is ON in a particular phase, the current flows from the upper switch element into the winding of the phase; and when the lower switch element is ON in a certain phase, the current flows from the winding of the phase to the lower switch element.
  • The winding currents iu, iv and iw will now be explained with reference to FIGS. 3 and 4. It is assumed that the duty cycles of the switch elements (in this example 21 to 23) of the PWM-controlled side are constant (ie always 50%) and that the differences in the counter electromotive forces in the 3-phase windings are neglected. Then the 3-phase windings are always in a configuration in which one winding and the other winding, which are connected in parallel, are connected in series. In other words, if only one of the upper switch elements 21 to 23 is turned ON, two of the lower switch elements 24 to 26 (which belong to the phases other than that of the only one upper switch element in the ON state) are turned ON, which in the sub - Cycles T2, T4 and T6 occur. When two of the upper switch elements 21 to 23 are ON, only one of the lower switch elements 24 to 26 (of a phase other than the phases of the two upper switch elements in the ON state) is in the ON state, which is the case with the sub-cycles T1, T3 and T5 takes place. Since the 3-phase windings have substantially the same impedance, the current that flows between the power supply source, not shown, and the converter circuit 20 does not vary very much.
  • For example, if the duty cycle of the switch elements 25 in the sub-cycle T1 is the same as that of the switch elements 25 and 26 in the sub-cycle T2 and if the differences in the counter electromotive forces in the 3-phase windings are neglected, the total currents must flow through the current measuring resistors 27 to 29 , remain unchanged (i ampere in the specific example in FIG. 4) in the sub-cycles T1 and T2. This enables the low-frequency fluctuations in the current caused by the switching operations to be suppressed based on the current measurements by the resistors 27 to 29 .
  • Although the total currents flowing through the current measuring resistors 27 to 29 remain unchanged through the sub-cycles T1 and T2, the current flowing through the V-phase winding 12 in the sub-cycle T1, for example, is twice the current, which flows through the V-phase winding 12 in the sub-cycle T2. Thus the 3-phase winding currents iu, iv and iw are pseudo sine waves, of which the amplitude consists of four levels.
  • In order to adapt the waveforms of the winding currents iu, iv and iw even more to a sine wave, the pulse duty cycle value of the PWM controller can be changed in each of the sub-cycles T1 to T6. More specifically, in a conductive period (or the ON state) (p) of each of the upper (ie, the PWM controlled side) switch elements 21 to 23, the duty cycle of the central π / 3 period can be, for example, twice the duty cycle of the first π / 3 period and the last p / 3 period. Another way to approximate the winding currents iu, iv and iw more to a sine wave is to change the duty cycle to multiple values in each conductive period (π) (or ON state).
  • As described above, the PWM control is achieved based on the sum of the rectified components, the full-way rectified components or the absolute values of the currents detected by the current measuring resistors 27 to 29 , which advantageously protects the switch elements 21 to 26 allows.
  • Electromotive operation for torque support
  • FIG. 5B is a flow diagram illustrating an electric-motor operation for providing a target torque, which is specified by a torque command from an unillustrated ECU (electronic control unit) is output for the torque support. In response to receiving a torque command, the computer 30 determines a target current value It corresponding to the target torque, which is done in step 70 .
  • In step 72 , the computer 30 calculates the mean value Ia of the sums of the detected currents (the detected actual voltages Vu, Vv and Vw), which is done in step 72 . It should be noted that in this case it is preferable to calculate the sum of the detected voltages Vu, Vv and Vw without rectifying the detected voltages Vu, Vv and Vw, for example by short-circuiting each diode (D '). This makes it possible to get a sum that does not include free-running currents of the non-conductive (or OFF) switch elements on the 180 degree duty cycle side.
  • It should also be noted that PWM control is achieved in this case by providing a pulse carrier signal of a constant carrier frequency (which in this specific example ranges from 15 to 20 kHz) to each of the switch elements ( 21 to 23 ) of the PWM controlled one Page is created. At step 74 , the computer 30 sets the duty cycle of the PWM signals to such a value that the average value Ia converges towards the target current value It. More specifically, when the average value Ia is smaller than the value It, the duty cycle of the PWM signals is increased by a predetermined amount. Otherwise the pulse duty factor of the PWM signals is reduced by a predetermined amount. Steps 72 and 74 are repeated in the same manner.
  • During the operation described above, the 3-phase currents flowing from the converter switch 20 to the brushless 3-phase motor 1 can be set to a desired value.
  • At the start of electromotive machine operation, the PWM control for suppressing an overcurrent is carried out to prevent the switch elements 21 to 26 from failing. However, setting the threshold voltage Vth also enables a lower value to limit the winding currents Iu, Iv and Iw to a value which is lower than the maximum permissible current. Although the duty cycle of the PWM signals increases or decreases by a predetermined amount as in the example explained above, it is possible to increase or decrease the duty cycle in response to a difference between the mean value Ia and the target current value It.
  • Fig. 6 shows a circuit diagram which partially reproduces an arrangement of a motor controller unit in which the winding currents using MOSFETs with a current sensor electrode terminal are detected, and indeed in accordance with a modification of the present invention. The motor controller unit, which is partially shown in Fig. 6, is identical to that of Fig. 1 with the exception that in Fig. 6, the converter circuit 20 and the circuit block 50 by a converter circuit 20 a and a circuit block 50 a have been replaced.
  • The converter circuit 20 a is identical to the circuit 20 with the exception that in Fig. 6 the current measuring resistors 27 to 29 have been removed and the MOSFETs 24 to 26 have been replaced by MOSFETs 24 a to 26 a which have a current sensor electrode connection. Each of the MOSFETs 24 a to 26 a comprises a main transistor 241 with a pair of electrodes S1 and D1 and with a current sensing resistor 242 with a pair of electrodes S2 and D2. The electrodes D1 and D2 are jointly formed in a semiconductor substrate. The electrode S1 is made much wider than the electrode S2, which represents the current sensor electrode connection. Such MOSFETs with a current sensing electrode connection are well known in the art.
  • Since the current sensor electrode connection is connected to a current sensing electrode S2, which is arranged adjacent to the electrodes S1 of the main transistor 241 , a smaller current (in the case of 24 a Iu) flows in proportion to the current (in the case of 24 a this is α Iu) flowing through the main transistor 241 through the current sensor electrode terminal in the form of a current mirror output current.
  • The circuit block 50 a is identical to the circuit block 50 with the exception that in Fig. 6 the series connected diode D 'and the amplifier 31 have been replaced by a current-to-voltage converter (CVC) 51 ; the series connected diode D 'and the amplifier 32 were replaced with a CVC 52 ; and the series connection of the diode D 'and the amplifier 33 was replaced by a CVC 63 . Each of the CVCs 51 to 53 includes an operational amplifier 511 and a resistor 512 connected between an output terminal and an inverting input terminal of the operational amplifier 511 . The current sensor electrodes (S2) of the MOSFETs 24 a to 26 a are connected to the respective inverting input connections of the current-to-voltage converter 51 to 53 .
  • The combination of a MOSFET (such as 24 a to 26 a) with a current sensor electrode connection and a CVC (such as 51 to 53 ) makes it possible to detect the current that flows through the MOSFET and consequently varies in the direction of flow, as in will be described in detail below.
  • If electrodes S1 and S2 serve as source electrodes or as electron injection electrodes (ie the current of the phase flows out of the winding), then it can be considered that the inverting input terminal of the operational amplifier, such as amplifier 51 , is essentially open Ground potential or ground potential: Accordingly, it can be assumed that electrodes S1 and S2 are at the same potential (or are at ground potential or ground potential), with the result that the source-gate voltages of transistors 241 and 242 are equal to one another are. As a result, a current that is 1 / α times the current flowing through the main transistor 241 flows through the transistor 242 . Here, α means a constant that is greater than 1.
  • Similarly, when electrodes D1 and D2 serve as source electrodes or as electron injection electrodes (ie, the current of the phase flows into the winding), the source gate voltages of transistors 241 and 242 are equal to each other since electrodes D1 and D2 have an identical potential exhibit. As a result, a current that is 1 / α times the current flowing through the main transistor 241 flows through the transistor 242 .
  • Thus, in any case, the current mirror output currents Iu, Iv and Iw of the are current sensing electrode terminals of the MOSFETs 24 a is converted to 26a by the CVCs 51 to 53 into voltages R0Iu, R0Iv and R0Iw proportional to the currents Iu, Iv and Iw. R0 is the resistance value of resistor 512 . In this way, each of the CVCs 51 to 53 provides a reliable value proportional to the current Iu, Iv or Iw flowing through the switch element regardless of the direction of the current.
  • The output voltages R0Iu, R0Iv and R0Iw from the CVCs 51 to 53 are added up by the adder 34 to obtain the sum of the voltages R0Iu, R0Iv and R0Iw. The adder 34 includes, for example, an operational amplifier 341 whose non-inverting input terminal is grounded; three input resistors 342 to 344 , one end of which is connected to the respective output of CVCs 51 to 53 , and the other end of which is connected to the inverting input terminal of operational amplifier 341 ; and a feedback resistor 348 connected between the output terminal and the inverting input terminal of the operational amplifier 341 .
  • Fig. 7 is a circuit diagram illustrating another embodiment of the circuit 50 a of Fig. 6. In Fig. 7, a circuit block 50 b is identical to that according to 50 a of Fig. 6, except that the CVCs 52 and 53 have been removed, the current sensor outputs Iu, Iv and Iw of the switch elements 24 a to 26 a to that inverting input terminal of CVC 51 ; and the adder 34 was replaced by an amplifier 34 a.
  • Since all of the current sensor outputs Iu, Iv and Iw of the switch elements 24 to flow a to 26 a by the feedback resistor 512 of CVC 51, the output terminal of the CVC 51 provides a voltage Vo that is proportional to the sum (.sigma..sub.i) of the currents Iu, Iv and Iw, where ΣI = Iu + Iv + Iw. Therefore, an amplification of the output voltage Vo of the CVC 51 using the amplifier 34 a with a suitable amplification provides a suitable scaled sum of the winding currents iu, iv and iw. According to this embodiment, more precise current detection is possible without an error due to fluctuations in the properties of the feedback resistors 512 and the operational amplifiers 511 that form the CVCs 51 to 53 .
  • , In which the winding currents are detected on the basis of the currents FIGS. 8A and 8B are circuit diagrams showing part of a 3-phase motor controller unit, flowing through the top switch elements of the bridge arms of the converter circuit. Fig. 8A shows a converter circuit 20 a. Fig. 8B shows a controller 3 a for use with the inverter circuit 20 a. Controller 3 a of FIG. 8B is identical to controller 3 of FIG. 2 with the exception that in FIG. 8B the output connections of elements 42 to 47 with the respective gate electrodes of switch elements 21 , 24 , 22 , 25 , 23 and 26 of Fig. 8A were each connected.
  • By the inverter circuit A and the controller 3 are configured in such a manner 20 a, the upper switch elements 21-phase control signals 3 can be controlled, which, based on the angular position signals Au Av and Aw of the angle sensor (or resolver) 16 to 23 by and therefore have a duty cycle of 50%. The lower switch elements 24 to 26 are controlled by the 3-phase PWM signals, which are based on both the angular position signals Au, Av and Aw and the winding currents iu, iv and iw, which are measured using the current measuring resistors 27 to 29 .
  • According to this embodiment, it becomes possible to reduce the switching operations of the upper switch elements 21 to 23 , which operations require gate voltages that are much higher than those used for the switching operations of the lower switch elements 24 to 26 . This enables reductions in the size and energy consumption of the switch element drivers 82 , 84 and 86 for the converter circuit 20 a.
  • In the above description, the current flowing through a switch element flows through, to be interpreted as containing the current flowing through a free-flowing diode flows through, which is connected antiparallel to the switch element. Although each switching MOSFET has a free-wheeling diode in the usual configuration contains a dedicated free-wheeling diode to a switching MOSFET to be added. This can reduce the current through the embedded free-flowing diode flows.
  • The foregoing description only illustrates the principles of Invention. The embodiments described above and modified versions can be combined in many ways, but without the Leave the scope of the invention.
  • Many widely different embodiments of the present Invention can be constructed without departing from the scope of the invention. It is noted that the present invention is not limited to the specific Embodiments that are described in the description and by the appended claims is defined within its scope.

Claims (15)

1. A method for controlling a brushless 3-phase motor ( 1 ) with a motor system ( 10 ) which contains, inter alia, an inverter circuit ( 20 ) with 3-phase bridge branches, each bridge branch having an upper-side switch element ( 21 , 22 , 23 ) and an underside switch element ( 24 , 25 , 26 ), the method comprising the following steps:
Switching the switch elements on one side of the converter circuit using first 3-phase signals which are synchronous with an angular position of a rotor and have a duty cycle of essentially 50%, and
Switching the switch element on the other side of each bridge branch with a pulse width modulation signal (PWM) during a period in which the switch element on one side of the bridge branch is not conductive.
2. The method of claim 1, further comprising the following steps:
Measuring respective first currents of the switch elements on one side; and
Determining a duty cycle of the PWM signal based on a predetermined threshold and a sum of the measured first currents in order to limit a second current flowing through the switch element on the other side of each bridge branch to or below a maximum current or a quantity of heat, which is approved for the switch element on the other side.
3. The method according to claim 1 or 2, wherein the switch elements of one side form the switch elements on the upper side.
4. The method according to claim 1 or 2, wherein the switch elements of one side form the switch elements on the lower side.
5. Device ( 2 ) for controlling a brushless 3-phase motor ( 1 ), which contains a rotor and a device ( 16 ) for outputting information indicating an angular position of the rotor, which device ( 2 ) has the following:
A converter circuit ( 20 ) with 3-phase bridge branches, each bridge branch having an upper-side switch element and a lower-side switch element;
means ( 36 to 38 ) operating on the basis of the information for switching the switch elements of one side of the converter circuit using first 3-phase signals which are synchronous with the angular position of the rotor and have a duty cycle of substantially 50%;
means ( 27 to 29 , 31 to 33 or 242 , 51 to 53 ) provided for each of the bridge branches to measure a first current of the switch element of one side of the bridge branch; and
means ( 39 to 41 ) which operates on the basis of not only the information but also uses the measured first currents for the 3-phase bridge branches to switch the switch element of the other side for each bridge branch with a pulse width modulation (PWM) signal during to switch a period in which the switch element of one side of the bridge arm is not conductive.
6. The apparatus of claim 5, wherein the means for switching the switch element of the other side includes:
Means for determining a duty cycle of the PWM signal based on a predetermined threshold and a sum of the measured first currents to limit a second current flowing through the switch element on the other side of each bridge arm, at or below a maximum current or amount of heat that is permissible for the switch element on the other side.
7. The apparatus of claim 6, wherein the means for determining the duty cycle of the PWM signal includes:
A device for setting the duty cycle to 50% as long as the sum is below the predetermined threshold.
8. The apparatus of claim 7, wherein the means for determining the duty cycle of the PWM signal includes:
Means for reducing the duty cycle by a predetermined amount in the event that the sum is equal to or greater than the predetermined threshold.
9. The apparatus of claim 7, wherein the means for determining a duty cycle of the PWM signal includes:
Means for reducing the duty cycle in response to a difference between the sum and the predetermined threshold in a case that the sum is equal to or greater than the predetermined threshold.
10. Apparatus according to any one of claims 5 to 7, wherein the means for measuring the first current includes:
A circuit element which is connected in series with the switch element on one side in order to cause a voltage drop.
11. Apparatus according to any one of claims 5 to 9, wherein the means for measuring the first current includes:
A circuit arrangement which, together with the switch element on one side, represents a current mirror with a connection through which a third current flows, which is less than and is proportional to the first current of the switch element on the one side; and
means for measuring the first current from the third current for each bridge branch.
12. The apparatus of claim 11, wherein the switch elements from one side Metal oxide semiconductor field effect transistors (MOSFETs), which are the Circuit arrangement included and the connection as a current sensor electrode exhibit.
13. Device according to one of claims 5 to 12, wherein the switch elements of one Side consist of the top switch elements.
14. Device according to one of claims 5 to 12, wherein the switch elements of one Side consist of the underside switch elements.
15. Motor system with a brushless 3-phase motor and a device after one of claims 5 to 15.
DE10303267A 2002-01-28 2003-01-28 Control process for brushless three phase motor system works on fifty percent cycle ratio Withdrawn DE10303267A1 (en)

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