DE102011102340B4 - Method for increasing the signal energy in a receiver and differential correlator, main processing arrangement and signal transit time measuring system therefor - Google Patents

Abstract

A method of increasing signal energy in a receiver - wherein the receiver receives a code spread signal modulated onto a carrier. Wherein the code and the carrier each have a phase, wherein the receiver tracks the phase of the code and the phase of the carrier, wherein the receiver for the tracking of the phase and for tracking the code correlates the received signal with a predefined code, wherein in correlating internal correlator values are generated, characterized in that - the receiver additionally receives data from which the receiver forms reference correlator values, - the receiver forms differences between the internal correlator values and reference correlator values

Description

The invention relates to a method for increasing the signal energy in a receiver. The receiver is preferably a satellite navigation receiver. In the method, the available signal energy is increased for improved estimation of the relevant radionavigation parameters such as the carrier phase, Doppler frequency and pseudo range. These parameters can thus also be used for stronger shading of the signal for positioning. This is accomplished by subtracting correlator values from an external source and internally generated correlator values.

In satellite navigation is problematic that usually a direct line of sight from the receiver to the satellite is necessary. Both the availability of direct line-of-sight and propagation-related or system-related sources of error mean that the position becomes inaccurate or a signal can not be found or tracked at all.

If z. B. reflects the signal, the position is corrupted. There are algorithms that try to calculate the direct signal.

If the signal, for example due to heavy rain, leaves or even walls or ceilings in a building shadowed, the receiver loses the signal and a position can not be calculated.

Highly sensitive receivers try z. B., these problems by local sensors (such as inertial navigation systems INS (Inertial Navigation System)) to fix.

The technology used in highly sensitive Global Navigation Satellite System (GNSS) receivers focuses on acquisition and tracking in poor signal-to-noise ratios. Decisive here is the duration of the coherent and non-coherent integration in the correlation between the received signal and the reference signal.

Correlation in this description is to compare a reference signal, such as a reference signal. As a code or a portion of a signal with a received signal, understood, the signals are usually offset in time to each other.

The integration can be accomplished digitally, for example, by multiplying by a sample interval of sampled values of the signal with a stored signal and adding the products within a predefined integration time interval. For example, if the signals are based on the same pseudorandom code and the bit sequences (chips) of the two signals are the same length, the correlator value becomes maximum as soon as there is a match in time. The correlator value goes to zero for a pseudo-random code when z. B. the time shift is at least one chip. However, due to superimposed thermal noise, the correlator values deviate from the theoretical values under ideal conditions, i. H. pure signals without noise, from.

In coherent integration, the result obtained at the end of the integration time interval is used directly for the following processing stages in the receiver. That is, unlike non-coherent integration, it is not cached and not correlated with a subsequent correlation of the next integration time interval with z. As squaring and addition connected.

The correlation can be both in the time domain by methods using z. B. matched filter or optimal filter (in English "Matched Filter") or a large number of correlators and in the frequency domain, z. B. be implemented by FFT (Fast Fourier Transformation) methods. These methods make it possible to acquire the signal under bad signal conditions in a short time. Typically, it is impossible to receive the data modulated on the signal.

The so-called assistance procedure (in English "Assisted GNSS (Global Navigation Satellite System)") helps to remedy this situation. In this case, a GNSS receiver receives information for calculating the satellite orbits via a data connection (eg via GSM (Global System for Mobile Communications), UMTS (Universal Mobile Telecommunications System), WLAN (Wireless Local Area Network), etc.).

It is also possible that coarse position data (with an accuracy of, for example, in the range of 1 km) and time are also transmitted. Based on this information, the receiver can restrict the search space, which is spread over the code phase and frequency, in the signal acquisition.

In order to achieve the highest possible sensitivity, the duration of the coherent integration, ie the time interval over which is integrated, must be selected as high as possible. As a rule, the coherent integration time is limited by the data rate, since a data bit change breaks the coherence. There are approaches to solve this problem. As you know the data bits vividly speaking, the data modulation can undo and you get a data bit-free signal, z. B. transmit the data bits in real time to the receiver. The data bits can thus be applied to increase the duration of the coherent integration.

An improvement in GNSS signal tracking is achieved according to the prior art by support by means of external sensors which measure the motion dynamics of the receiver. Nowadays common methods realize this either at the position level, at the level of the so-called raw data or even at the loop level: The dynamics are measured by external sensors and used to control the frequencies of the Numerically Controlled Oscillators (NCOs).

The method of loop support is in 1 outlined. The received signal ( 102 ) is used with the frequency compensation ( 106 ) converted to baseband. In this description, a baseband signal is understood to mean a signal which has no carrier frequency components or which only has residual components which remain from the mixture due to two frequencies which are not exactly the same.

The signal is multiplied by the sine or cosine of the frequency of the frequency change determined by a PLL (Phase Locked Loop) / FLL (Frequency Locked Loop) filter. Then the result from the frequency compensation ( 106 ) multiplied by the locally generated reference signal and coherently integrated ( 108 ). The reference signal ( 128 ) is typically generated in 3 versions: early, on time and late. The result of the integration ( 108 ) with the punctual signal is forwarded to the PLL / FLL signal tracking system ( 110 . 112 . 114 . 104 . 106 . 108 ). The remaining signals are forwarded to the DLL (Delay Locked Loop) signal tracking system ( 118 . 120 . 122 . 126 . 128 . 108 ). Both systems use so-called discriminators ( 110 . 118 ), which estimate the carrier phase (PLL) / carrier frequency (FLL) and code phase (DLL) phases, respectively. The result is filtered (F _{PLL / FLL (s)} ( 112 ) and F _{DLL (s)} ( 120 )) and scales (L _PLL ( 114 ) and L _DLL ( 122 )). The output from the external sensors (eg INS, Inertial Navigation System) ( 130 ) is additionally used to determine the valid frequency offset for the next integration interval. It is important that the frequency offset during the integration remains constant.

The sensitivity of a receiver is determined to a significant degree by the duration of the coherent integration. The maximum settable coherent integration time is in turn limited, among other things, by the dynamic movement of the receiver or variations in the receiver clock. In 2A and 2 B the problem is sketched. 2A shows the search space or the integration paths for the acquisition. The vertical axis ( 202 ) indicates the applied frequency shift and the horizontal axis ( 204 ) indicates the code phase. Typically, an integral over the code phase is calculated for a given frequency shift. This can be done either in the time domain (such as massive parallel processing in or in the frequency domain.) The required accuracy of the frequency shift during integration is inversely proportional to the coherent integration time, which means that the longer the integration time, the more accurate the frequency shift must be If integration times in the range of 1 second are to be achieved, the movement of the recipient plays an increasing role 2 B shown. The search space or the coherent integration path must be adapted dynamically during the integration. This means that the common assumption about the constancy of the frequency shift or the code Doppler is no longer valid. Would this assumption in the situation as in 2 B applied, the result would be greatly reduced. This problem applies to both the acquisition and the signal tracking.

One prior art approach for compensating for at least the acceleration and the higher order dynamic terms on the line-of-sight during coherent integration is to subdivide the (extended) integration interval into sub-intervals, where the frequency adjusted for each sub-interval. The determination of the frequencies takes place, for example, via INS.

The integration interval is also limited by the lack of stability of the receiver clocks. The control loops hitherto used for positioning (DLL, FLL, PLL), Kalman filters or vector tracking methods can only work with short coherent integration times (eg 20 ms) without ultrastable oscillators (ie no OCXO, Oven-Controlled Crystal Oscillators).

To increase the accuracy, the so-called differential GNSS can be used. Here, for example, according to the prior art, measurements of an accurately measured, high-quality reference station in the form of position data or code or phase measurements for the individual lines of sight are sent to the "moving"("rover") station. If the distance between the two stations is not too great (eg up to 50 km), the deviations of the reference measurements from the theoretical measurements can be applied to the rover, which can cause common errors such as: As ionospheric, tropospheric delays or satellite clock errors are extinguished. Non-common errors such as multipath or receiver clock errors can not be eliminated. Likewise, the correction has an effect on the accuracy, but not on the tracking behavior (tracking behavior). That is, longer integration intervals to increase the receiver sensitivity are not possible with the inclusion of reference data according to the prior art.

The term rover is understood here to mean, as a rule, moving user receiver, as a counterpart and unlike the fixed reference station. It should be noted that the rover does not actually have to move.

Differential GNSS is also used in RTK (Real Time Kinematics) surveying in geodesy, where the reference stations are provided by the surveying and mapping authorities of the federal states. With RTK one reaches position accuracies in the centimeter or even sub-centimeter range.

The Stöber, C., Anghilieri, M et al. "IpexSR: A Real-Time Multi-Frequency Software GNSS Receiver", IEEE Conferences, 2010, p. 407-416 shows a prior art in which a highly sensitive indoor software receiver has both "Assisted GNSS", i. H. Receive and store external navigation data, as well as apply Differential GNSS by receiving correction data according to the Radio Technical Commission for Maritime Services (RTCM) standard or by receiving European Geostationary Navigation Overlay Service (EGNOS) data to more quickly obtain a position solution and to obtain increased positional accuracy.

The object of the invention is to provide a method and an arrangement in order to increase the available energy in a receiving channel and thus also the sensitivity and accuracy of a GNSS receiver.

The object is achieved by a method according to claim 1. The object is further achieved by a differential correlator according to claim 8, a signal processing arrangement according to claim 9 and a signal delay measurement system according to claim 10.

Embodiments of the invention are illustrated in the figures and are explained in more detail below.

Show it

1 a phase tracking arrangement with external sensor according to the prior art

2 an illustration of the search space of code shift and Doppler shift according to the prior art,

3 a basic constellation according to an embodiment of the invention,

4 a receiver and a reference receiver according to an embodiment of the invention,

5 a method according to an embodiment of the invention,

6 another method according to an embodiment of the invention,

7 an arrangement according to an embodiment of the invention, and

8th Correlator values according to an embodiment of the invention

3 shows the constellation that is a prerequisite for the inventive method. Several channels 302 . 304 , send out signals that z. B. each have a carrier frequency and a modulated code. For example, in satellites navigation, several satellites are available which emit signals in a code division multiple access (CDMA) mode where the satellites are distinguished by pseudo random noise codes (PRN codes). on carriers in the range of z. B. 1 to 3 GHz can be modulated. For example, BPSK (Binary Phase Shift Keying), QPSK QuadraturePhase Shift Keying, Quadrature Phase Shift Keying) or other types of modulation are used as modulation types. In some satellite navigation systems, such. Although GLONASS (Globalnaja Nawigazionnaja Sputnikovaja Sistema, Global Navigation Satellite System) uses a FDMA (Frequency Division Multiple Access) method, the signal is nevertheless modulated with a PRN code, as is essential to the invention. The signals are from the stations 306 and 308 receive. station 306 is a user station, the z. B. is mobile and hereinafter referred to as "Rover". station 308 is a reference station. The reference station 308 measures, like the Rover, the transit times of the signals. Since the reference station knows its position exactly, it can determine data about deviations of the measurements from the theoretical values and as a correction to the rover ( 306 ). The corrections can z. B. individual lines of sight, the position, or derived from the measurements parameters such. B. ionosphere parameters. The reference station 308 but also measurements, such. B. Send phase information or signal strength to the rover. Furthermore, the reference station can also send navigation data.

According to one embodiment of the invention, as in 5 1, a method is provided for increasing the signal energy in a receiver, wherein the receiver receives a signal spread with a code ( 502 ) modulated onto a carrier, the code having a code phase and the carrier having a carrier phase, wherein the receiver tracks the code phase and the carrier phase, the receiver for tracking the phase and for tracking the code the received signal with a predefined code correlates and where correlator internal correlator values are generated ( 504 ). The receiver additionally receives data ( 512 ), from which it forms reference correlator values ( 514 ) and forms differences between the internal correlator values and reference correlator values ( 516 ).

According to a further embodiment of the invention, the differences are used for code phase determination and / or carrier phase determination.

The method will be explained in more detail below.

A standard tracking channel simulates a GNSS signal using a DLL (Delay Locked Loop) and a FLL (Frequency Locked Loop) that have low bandwidth (eg, <1 Hz). These tracking loops are stable and will track the signal even if it is slightly degraded (eg due to shading by trees). Alternatively, the DLL / FLL tracking block may be replaced by a vector tracking unit that estimates the user status vector (position, velocity, clock error, and drift) rather than estimating the individual line of sight parameters. This vector tracking block can be supported by an inertially derived trajectory (eg, "strapdown" or "dead-reckoning"). Regardless of which tracking method is used, the tracking loops produce correlator values that form the starting point of the method of the invention.

Strapdown means vehicle-mounted acceleration sensors. As a rule, they consist of three accelerometers and three rotors. Since strap-down systems are firmly mounted on the carrier platform, a gimbal suspension is eliminated.

The term dead reckoning or dead reckoning is understood to mean a location determination in which direction, speed and time are measured at specific epochs. The speed and direction are assumed to be constant between the fairs.

Under Epoch in the satellite navigation is a time, for. As the timing of a signal delay measurement, understood.

The internal correlator values are obtained by correlating the sampled digital signal with a reference code stored or generated in the receiver, such as a pseudorandom code. When correlating, the code and the digital signal z. B. multiplied at short intervals and added the results of each multiplication. Corresponds to the scanned in the If the code contained in the signal is a reference code, the amount of added values essentially depends on the timing of the two codes. The correlator values are used with methods known to those skilled in the art in order to keep the two codes in time by means of a control loop (DLL) and to track the carrier phase in a second control loop (FLL or PLL). The control loops can be coupled with each other or by means of external information, eg. B. inertial sensors are supported. In a vector tracking unit -. For example, similar to the position determination using the least squares method, information from all the channels for position determination is brought together and from this the errors of the individual line of sight are estimated (residuals). These errors can then be used as loop feedback to minimize the errors by adjusting the position and thereby correcting or tracking the relative timing deviation of the reference code and signal for correlation. In addition, the phases for z. B. the position solution can be better appreciated.

Under error here is often the deviation of a measurement z. B. a distance from the actual value, for which deviation in addition to actual measurement errors mainly physical phenomena, such as deflection of an electromagnetic wave in the ionosphere, are causal. If the error refers to a distance of a line of sight in the position solution of an overdetermined equation, one speaks in the satellite navigation also of residuals.

The correlator values calculated in the correlation of the vector tracking unit and the DLL / FLL are referred to as "internal correlator values" in this specification. In contrast, the receiver calculates reference correlator values from the data received via an interface to a reference receiver. These can, according to an embodiment of the invention z. B. are obtained from carrier observations that have been sent, for example, from the reference station in the form of a RINEX file to the rover, which stores this file and reads the carrier observations.

RINEX denotes a standard standard in satellite navigation, which describes a file format for observation data and navigation data. The use of RINEX as a reference station input format and output format is only given as an example here and can be achieved by other formats such. RTCM, BINEX, RTCA.

According to one embodiment of the invention, the receiver forms simple differences between the internal correlator values and reference correlator values.

Difference is understood here, as generally in this description, in particular also the product of the correlator value with the complex conjugate reference correlator value, as described, for example, in US Pat. Is formed at the differential correlators to obtain the difference of the correlator phases, as explained below.

Compiling differences at the correlator level is a bit tricky, as there are different ways to define carrier phases, and because the time relationship of the data is important.

Single and double difference observations (code and carrier) are well known concepts in precise positioning to cancel common mode errors. Differential correlators extend this concept to the level of correlator values. In this case, the same common mode errors are extinguished, the signal dynamics being drastically reduced. Then longer filter times can be used to reduce noise and remove multipath contributions.

φ^k'rov

Rover-Trägerphase zu Satellit k [rad]

φ^k,ref

Referenz Trägerphase zu Satellit k [rad]

Δφ^k

Einfach-Differenz-Trägerphase [rad]

Generally, single differences of the phase observations are defined by an equation of the following kind: Δφ ^k (t ^k ) = φ ^{k, rov} (t ^k ) - φ ^{k, ref} (t ^k ) (1)

φ ^k'rov

Rover carrier phase to satellite k [rad]

φ ^{k, ref}

Reference carrier phase to satellite k [rad]

Δφ ^k

Single difference carrier phase [rad]

Carrier phases are read from a RINEX file or similar source. The epoch t ^k refers generally to the rover time scale, but if reference station values to other epochs are available, then these temporal errors can be tolerated, if satellite positions are considered correctly. This is usually done in many RTK (Real Time Kinematic) software packages.

Since the reference station essentially contains a receiver, the term reference receiver is used synonymously in this description.

α(t)

Amplitudenfunktion (hier nicht relevant)

φ^k'NCO

Rover Trägerphase zu Satellit k [rad], basierend auf dem NCO-Messwert während der Verwendung interner FLL-Nachführung

P^k,rov

Rover pünktlicher Korrelator (komplexwertig), der die Differenz zwischen der internen FLL-Nachführung und dem wahren empfangenen Signal aufnimmt

To form correlator differences, a slightly different approach is needed. First, the rover tracking channel outputs a carrier phase measurement based on the internal NCO (Numerically Controlled Oscillator) controlled by an FLL. The internal tracking is not necessarily locked to the received carrier phase and the difference between the received and the internal carrier phase is included in the punctual correlator: exp {iφk ^{, rov} (t)} = | α (t) | exp {iφk ^{, NCO} (t)} Pk ^{, rov} (t) (2)

α (t)

Amplitude function (not relevant here)

φ ^k'NCO

Rover carrier phase to satellite k [rad], based on the NCO reading while using internal FLL tracking

P ^{k, rov}

Rover punctual correlator (complex-valued), which takes the difference between the internal FLL tracking and the true received signal

d

Ausgestrahltes Navigationsdatenbit (falls vorhanden)

ΔP

Einfach-Differenzkorrelator

Therefore, a simple difference correlator is written as .DELTA.P ^k (t ^k) = exp {Iø ^{k, NCO} (t)} P ^{k, rov} (t ^k) exp {-iφ ^{k, ref} (t ^k)} d (tk / sent) (3)

d

Radiated navigation data bit (if available)

.DELTA.P

Easy-Differenzkorrelator

The simple difference correlator has a lower signal dynamics than the undifferentiated correlator because the dynamics caused by z. As the satellite clock error, the satellite orbit or atmospheric propagation delays eliminated by the difference formation or reduced. Therefore, the simple difference correlator can be more gradually integrated or processed with smaller filter bandwidths.

A data stripping process uses the "sent time" for the correlator value P ^{k, rov.} It is assumed that the internal receiver time to the true Global Positioning System (GPS) time is better than e.g. B. +/- 1 ms is controlled, and the same applies to the reference receiver. Then one can simply take t ^k (which is a rover time) and use it as a reference station time to extract the reference station carrier phase. Later, this process is reversed, compensating for any time errors in the distance. The satellite speed and acceleration will be accurate enough with this little time error.

Alternatively t ^k can be replaced by an estimate of the true GPS time to minimize Synchronizsationsfehler on.

According to one embodiment of the invention, the reference correlator values are obtained by interpolation of the carrier phase of the reference station.

t^k

Korrelatorepoche (z. B. 50 Hz)

Tⁿ

Niedrig-Raten (z. B. 1 Hz) Referenzdatenepochen

The calculation of (3) requires the evaluation of the carrier phase of the reference station at a rate of the correlator values (eg 50 Hz). The phase itself is typically available at a lower rate (eg 1 Hz). Therefore, a suitable interpolation method has been designed. In a first step, the closest but most recent reference station epoch is located:

t ^k

Correlator epoch (eg 50 Hz)

T ⁿ

Low rates (eg 1 Hz) Reference data epochs

Thereafter, the carrier phase data at T ⁿ , T ^n-1 , T ^{n + 1} are used to define a quadratic polynomial and perform the interpolation. The interpolation scheme is z. In a way designed to give a continuous interpolation result; a change of the control point only occurs if t ^k = T ⁿ and then both sets give identical results.

According to one embodiment of the invention, however, other transmission methods in non-real time or in real time such. As a direct radio connection, a radio network connection or an optical connection possible. The transmission can z. For example, via GSM (Global System for Mobile Communications), UMTS (Universal Mobile Telecommunications System), WLAN (Wireless Local Area Network), WiFi, Adhoc networks, wired, etc. The data can also be buffered in memory cells.

It should be noted that also the reference correlator values are subject to errors. With regard to the rover, these errors have to a large extent the same cause, such as: As satellite clock error or ionosphere and troposphere, and thus are the same or at least similar in magnitude and direction. About the difference correlator these errors are at least partially eliminated.

According to one embodiment of the invention, the receiver has means for converting reference correlator values from the reference antenna position to another geographical position.

The deviation of the reference correlator values from the rover correlator values increases as the geometric distance between the reference antenna and the rover antenna increases. This increases the remaining errors in the differential correlators, which generally results in shorter integration times. In part, the geometric dependence of the errors is well known and reference correlator values determined for a given antenna position can be counted to a different location (near the rover). For example, purely geometrical dependency can be taken into account by determining the distance difference between actual reference antenna positions and assumed reference antenna positions to the satellite and multiplying the distance difference expressed in units of the carrier phase wavelength in the form of a complex exponential with the reference correlator value. Similar corrections can also be obtained from standard models of the troposphere and ionosphere.

According to one embodiment of the invention, the receiver forms double differences between the internal correlator values and reference correlator values.

The double difference method makes satellite differences between two different single difference correlators. The reference satellite gets index 1. It is typically the satellite with the highest elevation.

t^k

Weitere Satellitenepoche in [s], wenn ein Korrelatorwert für den Satellit k verfügbar ist

∇ΔP

Doppeldifferenzkorrelator

l

Referenzsatellitindex

k

Satellitenindex des weiteren Satelliten

und hängt von den zwei Epochen der beiden eingebundenen Einfach-Differenzkorrelatoren ab. Typischerweise können sie als die zeitlich nähesten Korrelatorwerte gewählt werden.The double difference correlator is written as

t ^k

Additional satellite epoch in [s], if a correlator value for the satellite k is available

∇ΔP

Doppeldifferenzkorrelator

l

Reference satellite Index

k

Satellite index of the further satellite

and depends on the two epochs of the two involved simple difference correlators. Typically, they can be chosen as the closest temporal correlator values.

The double difference correlator has a lower signal dynamics than the simple difference correlator, since the dynamics caused by z. B. the Empängeruhrfehler or rover movement by subtraction eliminated or reduced. Therefore, the double difference correlator can be more gradually integrated or processed with smaller filter bandwidths.

4 shows an inventive arrangement in which a high-frequency signal, eg. B. a CDMA signal from an analog receiving unit 422 Recipient 420 is received, is converted from analog to digtal and the DLL / FLL or the vector loop 428 is supplied.

• – einen Korrelator 428 zur Erzeugung interner Korrelatorwerte durch Korrelation, wobei bei der Korrelation ein intern zur Verfügung stehender Code 426 mit einem empfangenen Signal 440 korreliert wird, wobei das empfangene Signal 440 einen Träger und einen auf den Träger modulierten Code aufweist,
• – eine Empfangsschnittstelle 424 zum Empfangen von Trägerphasenwerten, wobei die Signalverarbeitungsanordnung
• – eine Referenzkorrelatorwerteerzeugungseinheit 430 zur Erzeugung von Referenzkorrelatorwerten aus den Trägerphasen aufweist, und
• – einen Differenzkorrelator 432 aufweist, der die Differenz zwischen den Referenzkorrelatorwerten und den internen Korrelatorwerten bildet.

As in further 4 In accordance with one embodiment of the invention, a signal processing arrangement is provided, comprising

• - a correlator 428 for generating internal correlator values by correlation, wherein in the correlation an internally available code 426 with a received signal 440 is correlated with the received signal 440 has a carrier and a code modulated on the carrier,
• - a reception interface 424 for receiving carrier phase values, the signal processing arrangement
• A reference correlator value generation unit 430 for generating reference correlator values from the carrier phases, and
• - a difference correlator 432 which is the difference between the reference correlator values and the internal correlator values.

It should be pointed out once again that differential also means, in particular, the product of the correlator value with the complex conjugate reference correlator value, as described, for example, in US Pat. B. in the differential correlators 432 is formed to obtain the difference of the correlator phases.

It should also be noted that the operations described in the digital blocks in FIG 4 that is, all blocks except the analog receiving unit R22, for each received signal, are performed, depending on the computing capacity of the receiver. The signals differ here z. In CDMA code, frequency, etc.

According to one embodiment of the invention, the differential correlator forms 432 simple and / or double differences, as already described above.

According to an embodiment of the invention, the differences are with a filter 434 filtered and a code phase determination unit 436 and / or carrier phase determination unit forwarded.

The filter 434 may be an adaptive spectral filter or a cost minimization filter according to one embodiment. Through the filter z. B. filtered spectral components to z. B. multipath and eliminate existing Doppler shifts. The filter 434 will be explained later in this description.

In order to obtain the rover carrier phase observations, according to one embodiment of the invention, as in FIG 6 shown, performed the following steps:
For the correlator values the phase discriminator (eg atan2) is formed ( 750 ). The phase discriminator 750 represents the wound carrier phase.

"Wound carrier phase (English wrapped carrier phase) means here that when a phase value of a progressive phase of 360 ° is reached, the phase begins again at 0 °, or has a value range from 0 ° to 360 °. With an atan2 discriminator, this range can also extend over -180 ° ... + 180 °, which can be mapped back to the range 0 ° ... 360 ° when adding 360 ° to the negative values. In the case of an unwrapped phase, on the other hand, the phase values of a progressive phase are continuously increased; H. it can also assume values over 360 °, or over 180 ° in the aforementioned atan2 example.

The wound carrier phase is in 750 settled. A polynomial adaptation 752 smooths the completed phase between two output epochs. Then the polynomial at the previous edition epoch is evaluated ( 754 . 756 ) and the result is compared with the previous carrier phase output ( 754 ). Ideally, the difference is very close to an integer value and this integer value can be used to resolve the carrier phase ambiguity for the current epoch. If the fractional fraction of this difference is greater than z. B. 0.25 cycles, a cycle slip is explained. This process is called carrier phase continuation. Using the resolved ambiguity, the polynomial is evaluated at the current output epoch ( 756 ). This represents the carrier phase estimate. If supporting information 718 (S. 7 ) (eg, if this channel is part of an integrated GPS / INS system), then the inertial or geometry based phase offset is added to the carrier phase pseudorange. The procedure is performed repeatedly for all tracked GNSS signals independently and the carrier phase estimates are written to a RINEX file or in a different format, such as a RINEX file. B. RTCM, BINEX supplied to the next processing stage. This can, for. B. determine the position of the rover antenna.

• – der Referenzempfänger und der Empfänger örtlich voneineander getrennt sind
• – der Referenzempfänger Mittel aufweist, Signallaufzeiten eines elektromagnetischen Signals zu messen,
• – wobei das elektromagnetische Signal ein hochfrequentes Signal ist
• – wobei auf das elektromagnetische Signal mit einem Code moduliert ist
• – wobei der Referenzempfänger Mittel aufweist, um Trägerphasen zu messen
• – wobei der Referenzempfänger Mittel aufweist, um die Werte der gemessenen Trägerphasen auszusenden
• – wobei der Empfänger Mittel aufweist, um die vom Referenzempfänger gesendeten Trägerphasenwerte zu empfangen
• – wobei der Empfänger Mittel aufweist, um das elektromagnetische Signal zu empfangen und mit einem intern zur Verfügung gestellten Code zu korrelieren, so dass interne Korrelatorwerte erzeugt werden, wobei
• – der Empfänger Mittel aufweist, um aus den vom Referenzempfänger gesendeten Trägerphasen Referenzkorrelatorwerte zu erzeugen
• – der Empfänger einen Differenzkorrelator aufweist, der die Differenz zwischen den internen Korrelatorwerten und den Referenzkorrelatorwerten bildet

According to one embodiment of the invention, a signal transit time measurement system having a reference receiver and a receiver is provided, wherein

• - The reference receiver and the receiver are physically separated from each other
• The reference receiver has means to measure signal propagation times of an electromagnetic signal,
• - wherein the electromagnetic signal is a high-frequency signal
• - wherein the electromagnetic signal is modulated with a code
• - wherein the reference receiver has means to measure carrier phases
• - wherein the reference receiver comprises means for transmitting the values of the measured carrier phases
• - wherein the receiver comprises means for receiving the carrier phase values transmitted by the reference receiver
• - wherein the receiver comprises means for receiving the electromagnetic signal and correlating with an internally provided code, so that internal correlator values are generated, wherein
• - The receiver comprises means for generating reference correlator values from the carrier phase transmitted by the reference receiver
• - The receiver has a differential correlator, which forms the difference between the internal correlator values and the reference correlator values

The reference receiver has at least one carrier phase measuring unit to measure carrier phases and a transmitting interface to transmit the measured values. The transmission interface can be specified here arbitrarily, z. B. according to a wireless protocol, WLAN, etc. Preferably, the reference receiver will have multiple channels, so that multiple signals can be measured simultaneously. The signals can z. B. come from different navigation satellites and also belong to different navigation systems, such as. For example, GPS, GALILEO, SBAS, EGNOS, GLONASS, COMPASS, etc. or terrestrial stations, such. B. pseudo-satellites (English pseudolites).

The reference receiver may be, for example, an ordinary GNSS receiver that can output readings in any form. For example, it would be possible that the transmission interface is an Ethernet interface, and the receiver ZB according to the Rinex protocol via an Internet Protocol connection (LAN, WLAN) at regular intervals data via UDP, TCP / IP or file transfer to the receiver sends, which has a corresponding counter-interface.

According to an embodiment of the invention, the received signal is modulated with a data bit, and the receiver removes the data bit before forming the differences. For this purpose, a data stripping process uses the "sent time" for the correlator value P ^{k, rov} . In satellite navigation systems, navigation messages with a low symbol rate of e.g. B. 50 Hz or 100 Hz associated with the pseudo-random code. By way of example, in a two-phase modulation (BPSK), which knows the states 0 ° and 180 °, the data bit can cause the pseudo-random code to be inverted as long as the data bit z. B. is a "1". In the case of a phase modulation, this can, for example, in turn cause the carrier phase to be rotated by 180 ° with respect to the carrier phase without data bits. Since this phase rotation is not contained in the reference carrier phase, this data bit must be taken into account, ie removed, during the subtraction. The data bit can also be in another place, such. B. in the FLL or after subtraction are eliminated.

According to one embodiment of the invention, the differences are filtered and the filtered differences used for code phase determination and / or carrier phase determination. The filtering primarily removes any remaining Doppler shifts and multipath signals, as described in more detail below.

According to an embodiment of the invention, the differences are filtered with an adaptive filter. In general, the adaptive filter F has the form Q (t _μ ) = F {P (t _μ )} (6)

and converts the raw correlator values P into filtered correlator values Q. The filter can work with non-differentiated values, single differences or double differences.

• – Transformation der Differenzen in den Frequenzbereich
• – Bestimmung der Dopplerverschiebung in Richtung der Sichtlinie
• – Adaptive Dopplerfilterung
• – Rücktransformation in den Zeitbereich
• – Ausgabe der gefilterten Differenzen

According to an embodiment of the invention, in filtering the differences using the adaptive filter, the following steps are performed:

• - Transformation of the differences in the frequency domain
• - Determination of the Doppler shift in the direction of the line of sight
• - Adaptive Doppler filtering
• - Back transformation into the time domain
• - Output of the filtered differences

The estimate in the frequency domain first multiplies the correlator values with the appropriate window function h (eg a Hamming window) P '(t _μ ) = P (t _μ ) h (t _μ ) (7)

Then, the vector length is increased to obtain a better frequency resolution and the Fourier transform is calculated. P ~ (f) = FFT {P '(f)) (8)

In the next step, the dominant spectral component is determined as

Spectral components that are beyond a certain threshold are considered noise or meter-path and are deleted in the frequency domain:

The threshold value γ is typically 99%.

Finally, the filtered correlator values are obtained via an inverse Fourier transform.

ΔQ^k

Gefilterter Einfach-Differenzkorrelator

F

Adaptives Filter

Δη^k

Abgewickelte Phase des gefilterten Einfach-Differenzkorrelators [rad]

According to an embodiment of the invention, the adaptive filter is applied to the single difference correlator values and the phase of the filtered correlator values is unwound to obtain non-differentiated carrier phase output data from the single difference correlator values: ΔQ ^k (t) = F {ΔP ^k (t)} = | ΔQ ^k (t) | exp {iΔη ^k (t)} (11)

ΔQ ^k

Filtered simple difference correlator

F

Adaptive filter

Δη ^k

Unwound phase of the filtered simple difference correlator [rad]

Finally, the unwrapped phase is added to the phase of the reference station, obtaining the new, non-differentiated Rover carrier phase, which is then z. For example, writing to a RINEX file: φ ^{k, rov} (t) = Δη ^k (t) + φ ^{k, ref} (t) (12)

Alternatively, the simple difference correlator values can be used directly for further evaluation, such as, for example, B. are used to determine the position.

∇ΔQ^k,l

Gefilterter Doppel-Differenzkorrelator

F

Adaptives Filter

Δη^k,l

Abgewickelte Phase des gefilterten Doppeldifferenzkorrelators [rad]

According to one embodiment of the invention, in the case of double differences, the adaptive filter is applied to the double difference correlator values and the resulting phase is unwound. ∇ΔQ ^{k, l} (t) = F {∇ΔP ^{k, l} (t)} = | ∇ΔQ ^{k, l} (t)} | exp {i∇Δη ^{k, l} (t)} (13)

∇ΔQ ^{k, l}

Filtered double difference correlator

F

Adaptive filter

Δη ^{k, l}

Completed phase of the filtered double difference correlator [rad]

In order to obtain correction values for the phase observation, single-difference observations are needed, whereby finally, as already described above, non-differentiated observations are obtained. The recovery of single difference from the double difference phase is not straightforward because the receiver clock error has been completely canceled out during the process. In the following two methods are proposed to reintroduce the receiver clock error.

According to one embodiment of the invention, the recovery of single difference from the double difference phase is accomplished by an inversion based on the strongest satellite. If a strong satellite is available, ie, the reference satellite, its carrier phase can be estimated using the simple difference approach or non-differentiated data. Then one obtains the single difference carrier phase of the other satellites by adding the double difference to the reference single difference. The non-differentiated observations are finally obtained by adding the carrier phase of the reference station: φ ^{k, rov} (t ^k) = ∇Δη ^{k, l} (t ^k, t ^l) + Δη ^l (t ^l) + φ ^{k, ref} (t ^k) (14)

Please note that this calculation only has to be calculated for the RINEX output period.

According to one embodiment of the invention, the filtered differences include punctual, early and late differences. The filtered differences are processed in a code discriminator and pseudoranges are calculated from the results of the code discriminator.

R^l,rov

Code-Pseudoentfernung des Rovers für den Referenzsatelliten

λ

Träger-Wellenlänge [m]

According to another embodiment of the invention, the recovery of single difference from the double difference phase is done by a pseudorange based inversion. If simple or undifferentiated tracking can not be used for the reference satellite, then one can choose to derive the carrier phase of the reference satellite from the code pseudorange. The non-differentiated carrier phase of all satellites is given by:

R ^{l, rov}

Code pseudorange of the rover for the reference satellite

λ

Carrier wavelength [m]

Please note that this calculation only has to be calculated for the RINEX output period.

Preferably, the RTK (Real-Time Kinematics) positioning is performed with double difference observations. Any error introduced into the non-differentiated carrier phase values due to the use of the code pseudorange via (15) is canceled out. If the positioning algorithm works with non-differentiated carrier phases, errors introduced by (15) will affect the carrier-phase-based receiver clock estimate. The artificially introduced receiver ^{clock error} is quite high and is directly related to the code pseudorange accuracy of R ^{l, rov} .

Alternatively, the double difference correlator values can be used directly for further evaluation such as. B. are used to determine the position.

E

Früh-/Spätkorrelatorwert

α

Früh-/Spätindex; bezieht sich auf den Codephasenoffset oder Korrelatorabstand

l

Referenzsatellitenindex

k

Satellitenindex des betrachteten Satelliten

According to one embodiment of the invention, the spectral adaptive filter can also be used for code tracking. The idea is to form double difference correlators, but instead of using 4 punctual correlator values (reference station / rover and two satellites), the rover correlator value of the considered satellite (ie, not the reference satellite) is given an early or late correlator value (or multiple correlator value). replaced. Therefore, the double difference early / late correlator is written as: ∇ΔE k / α (t ^k) = exp {Iø ^k, k E} ^NCO, rov / a (t ^k) exp {-iφ ^{k, ref}} d (tk / sent) × exp {-iφ ^{l, NCO}} P l, rov / a (t ^k) exp {Iø ^{l, ref}} d (tl / sent) (16)

e

Early / Spätkorrelatorwert

α

Early / late index; refers to the code phase offset or correlator distance

l

Reference satellite Index

k

Satellite index of the considered satellite

The early / late correlator values are formed over the same interval comprising a bundle of correlator values as the prompt correlator values.

In a further step, the adaptive frequency domain filter is used with the dominant frequency taken from the punctual double difference correlator value. All early / late correlators are then filtered with the same dominant frequency values. This filtering clears multipath posts in the same way as multipath is deleted from the punctual double difference value.

Finally, the filtered correlator values are separately added for all values of the code phase offsets over the burst period and used to estimate the code phase. This can be done either using non-parametric methods, such as. "Narrow Correlator" or more complicated multipath parameter estimation methods.

According to a further embodiment of the invention, the filtering is performed with a cost minimization filter.

φ

Trägerphasenoffset [rad]

f

Dopplerfrequenz [Hz]

α

Beschleunigung [Hz/s]

The cost minimization filter first adapts a model to the bundle of correlator values.

φ

Carrier phase offset [rad]

f

Doppler frequency [Hz]

α

Acceleration [Hz / s]

Alternatively, other cost functions and / or a higher number of parameters (eg acceleration change) can also be selected. An example of another cost function would be

A "roof" over a symbol denotes an estimated parameter. The allowable range for Doppler and acceleration may be limited based on the prior knowledge of visual line dynamics. The estimate itself is z. B. performed with a raster search algorithm for Doppler and acceleration. For fixed Doppler and acceleration values, the phase can be calculated analytically.

Based on the estimated parameters, the filtered correlator values are given by

The estimated wound carrier phase and Doppler frequency can be read directly from the above equation, or the filtered punctual correlator can be added over the burst period and standard methods such as the atan, atan2 discriminator can be used to obtain the wound carrier phase and Doppler frequency.

The cost minimization filter can be applied to early / late correlator values with model parameters taken from the prompt correlator estimate.

The early / late correlator values thus obtained can be added up over the bundle period and can be used for code phase determination similarly to the spectral filter (early-late tracking).

The invention will be described below with reference to an embodiment and with reference 7 explained.

The digitized and diminished signal is displayed in 702 tracked in the DLL (code tracking loop) or FLL (frequency tracking loop). Instead of the DLL / FLL circuits, a vector tracking control loop can also be used. For the determination of the code phase z. For example, at 50 Hz, the early and late correlator values, and for determining the carrier phase, the punctual correlator values to a ring buffer 706 transfer. Synchronized on z. For example, the 1 Hz Rinex epochs 710 . 712 The bundle extractor extracts the correlator values from the ring buffer 706 ,

As in 8th First, the correlator values are shown in bundles 802 . 804 grouped. The burst length is equal to the RINEX observation rate (eg 1 s or 30 s) plus a certain overlap period (eg 20 ms or 0.5 s). The bundle ends just before the RINEX observation epoch 806 respectively. 808 ,

Assuming, for example, a 1 Hz RINEX rate and a 50 Hz correlator rate, then a cluster consists of approximately 75 complex correlator values, with the first 25 correlator values overlapping the previous burst. For each correlator value, the corresponding time periods and NCO (Numerical Controlled Oscillator) values are also stored. Optionally one could also consider including some correlator values in the respective bundles (after the RINEX epoch, eg 100 ms), but this would increase the overall latency of the receiver. It is advantageous to minimize the overlap of two bundles to ensure that the estimates are statistically independent and the phase difference at the boundary is then a clear indication of whether continuous carrier phase tracking is taking place or oh so-called cycle slips occur.

Cycle slip refers to sudden changes in the integer phase ambiguity of the carrier phase measurement, which usually results from a brief loss of signal tracking.

After a bundle has been formed, a data stripping process multiplies 716 the correlator values with the corresponding navigation data bits or navigation data symbols. The bits or symbols are from the reference station 308 made available.

Using raw carrier phase observations 708 from the reference station 308 become the difference correlators 726 educated. This is in 7 in 720 shown.

In 720 The raw correlator values are converted into single difference values or double difference values. Optionally, it can be configured to operate in a bypass mode and then output the raw, ie non-differentiated correlator values.

If supporting information 718 is available, for. If, for example, this channel is part of an integrated GPS / INS (Global Positioning System / Inertial Navigation System), the inertial / geometry-based user dynamics are corrected in the correlator values.

The bundles with the correlator values become a filter 730 . 740 fed. This can be for example a frequency domain filter 730 or a cost minimization filter 740 be.

• – Transformation der Differenzen in den Frequenzbereich 728
• – Bestimmung der Dopplerverschiebung in Richtung der Quelle des Signals 732, d. h. der Sichtlinie, die durch die Antenne des Empfängers und z. B. der Satellitensendeantenne, die das hochfrequente Signal abstrahlt.
• – Adaptive Dopplerfilterung 734
• – Rücktransformation in den Zeitbereich 736
• – Ausgabe der gefilterten Differenzen 744

For the example of the frequency domain filter, ie an adaptive spectral filter, the following steps are now carried out:

• - Transformation of the differences in the frequency domain 728
• Determination of the Doppler shift towards the source of the signal 732 , ie the line of sight through the antenna of the receiver and z. B. the satellite broadcast antenna, which radiates the high-frequency signal.
• - Adaptive Doppler filtering 734
• - Back transformation into the time domain 736
• - Output of the filtered differences 744

The edition 744 . 746 the filter 730 . 740 is a bunch of filtered correlator values.

• – Für alle Korrelatorwerte in einem Bündel wird der Phasendiskriminator (z. B. atan2) gebildet (750). Der Phasendiskriminator 750 stellt die gewickelte Trägerphase dar.

To obtain the rover carrier phase observations, the following steps are performed:

• For all correlator values in a bundle, the phase discriminator (eg atan2) is formed ( 750 ). The phase discriminator 750 represents the wound carrier phase.

The wound carrier phase is in 750 handled, starting with 0 cycles at the beginning of a bundle. A polynomial adaptation 752 smooths the unwound phase during a burst interval. Thereafter, the polynomial at the previous RINEX epoch is evaluated and the result is compared with the previous carrier phase output ( 745 ). Ideally, the difference is very close to an integer value and this integer value can be used to resolve the carrier phase ambiguity for the current burst. If the fractional fraction of this difference is greater than z. B. 0.25 cycles, a cycle slip is explained. This process is called carrier phase continuation. Using the resolved ambiguity, the polynomial is evaluated at the current RINEX epoch ( 756 ). This represents the carrier phase estimate. If supporting information 718 is present (eg, if this channel is part of an integrated GPS / INS system) then the inertial or geometry based phase offset will increase. added to the carrier phase pseudorange. The procedure is repeated for all tracked GNSS signals independently and the carrier phase estimates are written to a RINEX file.

The filter option 1, ie the spectral adaptive filter 730 , also outputs filtered early / late correlator values, or in a more general sense, multiple correlator values. The correlator values are summed over the burst interval and the accumulated correlator values are used to construct a code discriminator 738 to calculate the code pseudoranges. Standard procedures can be used. The resulting code pseudorange is written to a RINEX file along with the carrier phase.

The invention can not only be used in the field of satellite navigation, but also z. B. in the mobile sector or other systems where a duration of signals for z. B. a synchronization or a position determination is determined.

Claims (11)

A method of increasing signal energy in a receiver - wherein the receiver receives a code spread signal modulated onto a carrier. Wherein the code and the carrier each have a phase, wherein the receiver tracks the phase of the code and the phase of the carrier, wherein the receiver for the phase tracking and the tracking of the code correlates the received signal with a predefined code, wherein in correlating internal correlator values are generated, characterized in that - the receiver additionally receives data from which the receiver forms reference correlator values, - the receiver forms differences between the internal correlator values and reference correlator values A method according to claim 1, characterized in that the receiver forms simple differences or double differences between the internal correlator values and reference correlator values Method according to claim 1, characterized in that - the receiver filters the differences and - The receiver uses the filtered differences for code phase determination and / or carrier phase determination. Method according to claim 3, characterized in that When filtering the differences, perform the following steps: - Transformation of the differences in the frequency domain Determination of the Doppler shift towards the source of the signal - Adaptive Doppler filtering - Back transformation into the time domain - Output of the filtered differences A method according to claim 3, characterized in that the filtering of the differences is performed with a cost minimization filter. Method according to Claim 1, characterized in that the carrier phase is obtained from the filtered differences and the internal correlator values. Method according to claim 3, characterized in that - the filtered differences include punctual, early and late differences - the filtered differences are processed in a code discriminator and - pseudoranges are calculated from the results of the code discriminator. Differenzkorrelator, - Having a first interface for receiving internal correlator values, wherein the internal correlator values are generated by a correlation of a digitized analog signal with a predefined reference signal, and Having a second interface, characterized in that The second interface comprises means for receiving reference correlator values, and wherein - The differential correlator comprises means for generating differences between the internal correlator values and the reference correlator values. Signal processing arrangement comprising A correlator for generating internal correlator values by correlation, wherein correlation correlates an internally available code with a received signal, wherein the received signal comprises a carrier and a carrier modulated code A receive interface for receiving carrier phase values characterized in that the signal processing arrangement A reference correlator value generation unit for generating reference correlator values from the carrier phases, and - has a difference correlator that forms the difference between the reference correlator values and the internal correlator values. A signal transit time measurement system comprising a reference receiver and a receiver, wherein - the reference receiver and the receiver are spatially separated The reference receiver has means to measure signal propagation times of an electromagnetic signal, - wherein the electromagnetic signal is a high-frequency signal - Wherein a code is modulated on the electromagnetic signal - wherein the reference receiver has means to measure carrier phases - wherein the reference receiver comprises means for transmitting the values of the measured carrier phases - wherein the receiver comprises means for receiving the carrier phase values transmitted by the reference receiver - wherein the receiver comprises means to receive the electromagnetic signal and to correlate with an internally provided code, so that internal correlator values are generated characterized in that - The receiver comprises means for generating reference correlator values from the carrier phase transmitted by the reference receiver - The receiver has a differential correlator, which forms the difference between the internal correlator values and the reference correlator values. A signal transit time measurement system according to claim 10, wherein the reference receiver comprises an antenna which receives the electromagnetic signal and to which the signal propagation time is related. characterized in that - The receiver has means for converting reference correlator values from the reference antenna position to another geographical position.

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