CN1333540C - Method for reducing multi-path interference and target type receiver - Google Patents

Method for reducing multi-path interference and target type receiver Download PDF


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CN1333540C CN 200410006129 CN200410006129A CN1333540C CN 1333540 C CN1333540 C CN 1333540C CN 200410006129 CN200410006129 CN 200410006129 CN 200410006129 A CN200410006129 A CN 200410006129A CN 1333540 C CN1333540 C CN 1333540C
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本发明提供一种降低多路径干扰的方法,用于CCK符号的解码。 The present invention provides a method for reducing multi-path interference, CCK symbols for decoding. 本方法首先依据目前CCK符号的ICI校正关联输出值,取得一组目前CCK符号的起始候选CCK字码,并用同一方法,取得一组下一CCK符号的起始候选CCK字码。 First, the method according to the current output value associated with CCK ICI corrected symbols to obtain a set of candidate starting current CCK codeword CCK symbol, and using the same method, to obtain a set of candidate starting the next CCK codeword CCK symbol. 接着,本方法对于目前CCK符号的每一个起始候选CCK字码,取得第一降低ISI关联输出值,以同时校正目前CCK符号产生的ICI,以及下一CCK符号产生的ISI。 Next, the method for each of the current CCK codeword CCK first candidate symbol, to obtain a first output value associated with reduced ISI, while correcting the ISI to the current symbol generated by the ICI CCK, CCK symbol and a next generated. 然后,对于目前CCK符号的每一个起始候选CCK字码,依据其第一降低ISI关联输出值,将其中因前一CCK符号产生的ISI校正后,取得第二降低ISI关联输出值。 Then, each of the first candidate for the current CCK codeword CCK symbols, associated with the output value according to ISI reducing its first, after which the ISI corrected a CCK symbol generated by the front acquires a second correlation output value reduce ISI. 最后,依据第二降低ISI关联输出值,对目前CCK符号进行解码。 Finally, according to the second output value associated with reduced ISI, the current CCK symbols are decoded.


降低多路径干扰遭遇的方法与应用其的耙式接收器 Encounter multipath interference reduction method and application thereof rake receiver

技术领域 FIELD

本发明是有关于无线通讯系统,例如但不受限于无线局域网络(wirelesslocal area network,WLAN),且特别是有关于可以减低由于多路径通道而产生的符号内(Intra-Symbol)和符号间(inter-symbol)干扰的一种802.11b互补码(Complementary Code Keying,以下简称互补码(或CCK)接收器(Receiver)。 The present invention relates to a wireless communication system, such as, but not limited to wireless local area networks (wirelesslocal area network, WLAN), and in particular relates to a symbol can be reduced due to multipath channel generated (Intra-Symbol) and inter-symbol 802.11b one kind of complementary code (inter-symbol) interference (complementary code Keying, hereinafter referred to as the complementary code (or CCK) receptor (receiver).

背景技术 Background technique

伟伯斯彻(Webster)等人的美国专利公开案号2001/0036223揭露了一种应用于室内多路径WLAN中的耙式(RAKE)接收器,其直接展频(directspread spectrum)的讯号采用长度比较短的字码(Codeword)。 Wei Bosi Che (Webster) et al., U.S. Patent Publication No. 2001/0036223 discloses a receiver is applied in the indoor multipath WLAN rake (RAKE), direct spread spectrum (directspread spectrum) signals using a length shorter word (codeword). 伟伯斯彻的图6、7、8以及10显示的RAKE接收器中,在接收器通道匹配滤波器(ChannelMatched Filter,简称CMF)与字码关联器间的讯号处理路径中具有一嵌入式以位为单位(Chip-Based)的决定回馈等化器(Decision Feedback Equalizer,以下简称DFE)结构。 7, 8 and a RAKE receiver 10 shown in Bosi Che Wei, the signal processing path between the vessel and the word associated with the receiver channel matched filter (CMF ChannelMatched Filter, referred to) embedded in a decision feedback equalizer of bit units (Chip-Based) a (decision feedback equalizer, hereinafter referred to as DFE) structure. 该决定回馈等化器用于抵消符号间干扰(inter-symbolinterference,简称ISI。在伟伯斯彻的802.11b CCK解码器中(亦即因多路径通道所造成的相邻互补码字码间的干扰)。 The decision feedback equalizer for canceling inter-symbol interference (inter-symbolinterference, referred to as ISI. Wei Bosi Che in the 802.11b CCK decoder (i.e., adjacent-channel caused by multipath interference between the complementary codeword ).

在伟伯斯彻的802.11b互补码解码器中图12-14显示一符号内位干扰(Intra-Symbol Chip Interference,简称ICI)抵消器,用于抵消因后游标(post-cursor)所产生的符号内位干扰ICI,使用繁冗的复数运算(乘法与加法),对每一个字码(最多达256个字码),须要一DFE旋积(convolution)方块与一字码关联方块。 Wei, Bo Siche in the 802.11b complementary code in FIG decoder 12-14 in a sign bit disturb (Intra-Symbol Chip Interference, abbreviated ICI) canceller for canceling because the cursor (post-cursor) generated the ICI bit symbol interference, the use of cumbersome complex arithmetic (multiplication and addition) for every word (up to 256 word), the product needs to spin a DFE (Convolution) block code word associated with the block. 对64或256个符号内位干扰ICI的输出,每一个字码均有各自处理路径独立计算。 Of 64 bits or 256 symbols output interference ICI, every word has its own independent path calculation processing.

在伟伯斯彻的图12中,对各字码(总计有高达256个字码)而言,需要三个基本建构方块:(i)一DFE旋积方块1220,利用各字码(含8个复数字元,标示为OW#Kchip)间的复数旋积与高达8个的复数DFE标签(tap)来计算符号内位干扰ICI在字码关联器1230前的逐位代表;(ii)一逐位减法器1210,从方块1220的输出减去各接收的8个复数字元(方块1203);以及(iii)一字码关联器1230,计算各字码(8个复数字元)与减法器1210的输出8个复数字元间的关联值。 Wei, Bo Siche in FIG 12, for each word (up to a total of 256 word), it requires three basic building blocks: (i) a product of spin DFE block 1220, by each word (containing 8 a plurality of characters, among a plurality of labeled OW # kchip) by spin ICI product before the representatives 1230 and up to eight complex DFE tag (TAP) to calculate symbol interference bit word correlator; (ii) a bitwise subtractor 1210 subtracts each 8 complex characters received (block 1203) from the output of block 1220; and (iii) the associated code word 1230, calculated for each word (8 complex characters) and subtraction output correlation value between 8 1210 complex characters. 以上所有运算都是复数的,且为每个字码所需要。 All the above operations are complex and require for each word. 简言之,在方块1220内独立计算256个符号内位干扰ICI偏差值输出1212,在方块1210处进行平行减法,接着从字码关联器1230获得256个关联输出。 Briefly, the independent calculation block 256 in the 1220 symbol interference offset value output ICI 1212, parallel subtraction block 1210, then the associated output 256 is obtained from the associated word 1230. 因为各字码关联器1230的输入1223在经符号内位干扰ICI偏差校正后变得不同,此架构使得原本可以用快速瓦许转换(Fast Walsh Transform,以下简称FWT)以共用的硬件来同时而有效地计算所有互补码字码的CCK关联值的硬件实现,变成不可能。 Since each word associated with the input 12231230 symbols are in the ICI bit disturb deviation correction become different, so this architecture could have many watts flash converter (Fast Walsh Transform, hereinafter referred to as FWT) simultaneously at a common hardware and All computing hardware effectively complementary code values ​​associated CCK codeword implemented, it becomes impossible.

在伟伯斯彻的图13的另一实施例中,对于各字码(共有64个字码),需要两个基本建构方块(具有上述的相同功能):(i)一DFE旋积方块1340与(ii)一字码关联器1330。 A product of spin DFE block 1340 Wei Bo Siche In another embodiment of the embodiment of FIG. 13, for each word (there are 64 word), two basic building blocks (having the same function as described above) :( i) and (ii) the associated code word 1330. 所需的DFE方块与关联方块的数量从256简化为64。 DFE block number associated with the block 256 from the desired reduced to 64. 经由复数运算,方块1330的64个符号内位干扰ICI输出展开成256个符号内位干扰输出。 Via a complex operation, the block 1330 is 64 symbols output bit disturb ICI developed into bit disturb the 256 symbols output. 相较于图12,此架构计算后互补码字码关联器校正1360的符号内位干扰偏差。 Compared to FIG. 12, the computing architecture is complementary codeword correlator 1360 symbol interference correction bit deviation. 因而,此互补码字码关联器可利用64元件快速瓦许转换器1320与1至4展开方块1350,来同时与有效计算所有的CCK关联值。 Thus, the complementary codeword correlator 64 may utilize flash W Xu converter element 1320 and expand block 1350 1-4, simultaneously with all the associated values ​​valid CCK calculation. 然而,此方法仍需利用复数旋积1340与复数关联1330来先计算64个后关联符号内位干扰偏差。 However, this method using a plurality of still associated with the complex spin product 1340 to 1330 to calculate the 64 bit disturb the associated symbol deviation. 接着利用1至4展开方块1350来产生全部的256个后关联符号内位干扰偏差。 Then using 1-4 expand block 1350 to generate interference within the full bit 256 after the associated symbol deviation. 接着从方块1350的对应的256个关联器输出,减去这些后关联符号内位干扰偏差。 Then the output from the correlator 256 corresponding to the block 1350, after subtracting the bit associated symbol interference within these deviations. 为实现此接收器架构,需要能独立操作的64个复数旋积1340与复数字码关联器1330。 To achieve this receiver architecture, it requires complex 64 can operate independently of the rotary product code 1340 associated with the digital complex 1330.

在伟伯斯彻的图14的又一实施例中,对于各字码(共有256个字码)需要两个基本建构方块(具有上述的相同功能):(i)一DFE旋积方块1440与(ii)一复数字码关联器1430。 Wei, Bo Siche In a further embodiment of the embodiment of FIG. 14, for each word (there are 256 word) requires two basic building blocks (having the same function as described above) :( i) a product of spin DFE block 1440 (ii) a complex digital code correlator 1430. 伟伯斯彻描述了,可预先计算与储存DFE标签。 Wei, Bo Siche described, can be pre-computed and stored DFE tag. 然而不变的是,需用方块1440独立计算256个符号内位干扰输出,接着用方块1430的输出找到256项关联值。 However, the same is required to calculate the bit block 1440 independently interference output 256 symbols, and then to find the value 256 associated with the output of block 1430.

总结来说,伟伯斯彻的架构需要一个位为单位的DFE来消除由前一互补码符号所产生的符号间干扰。 In summary, Wei Bosi Che's architecture requires a bit DFE unit to eliminate inter-symbol from the previous complementary code symbol interference generated. 另外,为消除由目前互补码符号所产生的符号内位干扰,伟伯斯彻的架构需要大量复杂硬件与大量的复数运算(复数旋积与复数关联所需的复数乘法与加法)。 Further, to eliminate the interference from the current symbol bit symbols generated by the complementary code, Wei, Bo Siche architecture requires a lot of hardware and a large number of complex arithmetic complex (complex product desired rotation and a plurality of associated complex multiplication and addition). 因此,为实现伟伯斯彻的第12-14图的实施例,需要大量功率消耗,复杂的硬件与冗长的处理时间。 Accordingly, to achieve great Bosi Che embodiment of FIG. 12-14, it requires a lot of power consumption, complicated hardware and lengthy processing times.

在2002年11月7号所申请的美国专利序案号第10/289,749号,全名为“以封包为单位的具有快速多路径干扰解码器(Fast Multipath InterferenceCipher,以下简称FMIC)的无乘法(Multiplication-Free)互补码解调器(Packet-Based)”(“Packet-based Multiplication-free CCK Demodulator with aFast Multipath Interference Cipher”)中,本发明的作者提出一个互补码的接收器,采用创新的FMIC,以减低符号内位干扰,如此专利申请的图2所示,所有256个CCK字码的符号内位干扰,能够由一个创新的FMIC方块在第一运算模式(Mode 1)时共同地且有效地计算出来。 In November 2002, No. 7 US patent application Order No. 10 / 289,749, the full name of the "fast packet units have multi-path interference decoder (Fast Multipath InterferenceCipher, hereinafter referred to as FMIC) no multiplication ( Multiplication-Free) complementary code demodulator (Packet-based) "(" Packet-based Multiplication-free CCK demodulator with aFast Multipath Interference Cipher "), the authors of the present invention proposes a complementary code of the receiver, innovative FMIC FIG sign bit to reduce the interference, so Patent application 2, the sign bit disturb all CCK codewords 256 can be made of an innovative FMIC block when a first operational mode (mode 1) in common and effective be calculated. 在下一个运算模式(Mode2)时,在CCK解码器方块内,由CCK关联器输出中减去相对应的符号内位干扰。 At the next calculation mode (Mode2), CCK decoder in the block by subtracting the CCK correlator output bit symbols corresponding to the interference. 一CCK关联器的架构,通常利用FWT以共同与有效地计算出CCK字码间的关联值。 A CCK correlator architecture, generally together with an effective use of FWT to calculate the correlation value between the CCK codeword. 上述FMIC方块的特色,在于采用类似于CCK关联器的架构,来实现快速多路径转换(Fast Multipath Transform,以下简称FMT),虽然在美国专利序号第10/289,749号所提出的接收器,能够成功地降低由于多路径传播(此多路径传播实质上并不在任何附加的硬件上传播)所产生的符号内位干扰,但是(尤其是FMIC)却不能降低由于多路径通道而产生的符号间干扰。 FMIC box above characteristics, similar to that using the CCK correlator architecture to implement a fast multi-path converter (Fast Multipath Transform, hereinafter referred to as the FMT), although the receiver U.S. Patent Serial No. 10 / 289,749 proposed to successfully reduced due to multipath propagation (multipath propagation, this does not propagate in substantially any additional hardware) bit symbol interference generated, but (especially FMIC) did not reduce inter-symbol channel due to multipath interference is generated.


本发明的目的,在提供一种CCK接收器,以有效地降低符号内位干扰和符号间干扰。 Object of the present invention, there is provided a CCK receiver to effectively reduce the inter-symbol interference and bit symbol interference.

本发明的另一目的,在提供一种CCK接收器,利用最少的复数运算来降低符号内位干扰和符号间干扰。 Another object of the present invention, there is provided a CCK receiver with minimal complex operation to reduce the inter-symbol interference and bit symbol interference.

本发明的又一目的是提供一种CCK接收器,可以使用简单的硬件结构,来降低符号内位干扰和符号间干扰。 A further object of the present invention is to provide a CCK receiver using a simple hardware structure to reduce the inter-symbol interference and bit symbol interference.

为达上述目的,本发明提供一种降低目前CCK符号遭遇到的多路径干扰的方法,本发明所提供的方法,首先依据用于目前CCK符号和下一CCK符号的内位干扰校正关联输出值,来分别获得对应于目前CCK符号和下一CCK符号的M1(复数)个目前起始候选互补码字码和M2(复数)个下一起始候选互补码字码。 To achieve the above object, the present invention provides a method currently encountered CCK symbol multipath interference reduction, the present invention provides the method, the first based on the current symbol bit disturb CCK and CCK symbol next correction value for the associated output to respectively correspond to the current symbol Ml CCK and CCK next symbol (s) complementary to a current first candidate codeword and M2 (a plurality of) a start of the next candidate complementary codeword. 接下来,本发明所提供的方法对于每一个目前起始候选互补码字码,计算出其所对应的第一降低符号间干扰关联输出值;经过此步骤所得的第一降低符号间干扰关联输出,除了已经将目前CCK符号产生的符号内位干扰加以校正,又进一步将因为下一CCK符号而产生的符号间干扰加以校正。 Next, the method of the present invention is provided for each current first candidate complementary codewords, the output value of the calculated correlation between the interference corresponds to a first symbol decrease; this step is obtained through a first inter-symbol interference reduction associated output in addition to the current CCK has been a symbol symbol interference generated bits to be corrected, corrected inter-symbol interference and further because the next CCK symbol generated. 接着,本发明对每一目前起始候选互补码字码,依据第一降低符号间干扰ISI关联输出,计算出其对应的第二降低符号间干扰关联输出,此步骤针对因为前一CCK符号而产生的符号间干扰,加以校正。 Next, the present invention is complementary to each of the current first candidate codeword, according to a first intersymbol interference ISI reducing an associated output, which corresponds to the calculated inter-symbol interference associated with a second reduced output, for this step because of the previous CCK symbols generating intersymbol interference is corrected. 然后依据各个目前起始候选互补码字码的第二降低符号间干扰关联输出值,以解码目前CCK符号。 Then based on each of the second current reducing intersymbol interference associated with the first candidate output value of the complementary codeword to decode the current CCK symbol.

本发明也提供一种使用于多路径环境的RAKE接收器,本发明的RAKE接收器用来接收连续的CCK符号,包括目前CCK符号、前一CCK符号和下一CCK符号。 The present invention also provides a multi-channel environment using RAKE receivers, RAKE receiver of the present invention for receiving successive symbols of CCK, CCK including the current symbol, the previous symbol and the next CCK CCK symbol. 本发明的RAKE接收器是包括用来估计通道脉冲响应的通道估测装置,以及FMIC偏差计算装置。 RAKE receiver of the present invention is a channel used to estimate the channel impulse response estimation means, and deviation calculating means FMIC. 此FMIC偏差计算装置,自通道估测装置获得接收回馈标签权值(Feedback Tap Weight)以及前馈标签权值(Feed-Forward Tap Weight),从而对每一个可能的CCK字码,计算出因多路径干扰对接收到的CCK符号(可以是目前或下一CCK符号),所造成的多路径干扰(Multipath Interference,简称MPI)偏差(bias)值;此多路径干扰偏差值,是符号内位干扰的后关联表示。 This FMIC deviation calculation means, means for obtaining channel estimation received from the reserved label weights (Feedback Tap Weight) tag and a feedforward weights (Feed-Forward Tap Weight), so that for each possible CCK codeword, calculated by plurality path interference received symbols CCK (CCK may be present or next symbol), caused by multipath interference (multipath interference, abbreviated MPI) bias (bIAS) value; this multipath interference offset value, the interference is the sign bit after the association represents. 本发明的RAKE接收器也包括通道匹配滤波器(Channel Matched Filter,简称CMF),用来接收由通道估测装置所产生的通道匹配滤波器标签权值(CMF Tap Weight),以及包括CCK关联器,用来接收通道匹配滤波器的输出,并且产生关联输出。 RAKE receiver of the present invention also includes a channel matched filter (Channel Matched Filter, referred to the CMF), for receiving the channel matched filter weights tag (CMF Tap Weight) generated by the channel estimation means, and including CCK correlator for receiving the output of the channel matched filter and generates an associated output. 本发明的RAKE接收器更包括解码器,其用来(i)从FMIC偏差计算装置中获得多路径干扰偏差值,用来降低目前CCK符号所产生的符号内位干扰。 RAKE receiver of the present invention further comprises a decoder, which is used to (i) means to obtain multipath interference offset value calculated from the FMIC deviation for reducing interference within the current symbol bit CCK generated symbols. (ii)从通道估测装置接收前馈标签权值和回馈标签权值,以计算由前一CCK符号和下一CCK符号所产生的符号间干扰偏差值。 (Ii) feeding the tag label weights and feedback weights, in order to calculate the inter-symbol from the previous symbol and the next CCK CCK symbol interference generated by the offset value from the channel estimation prior to receiving means. (iii)从CCK关联器接收其关联输出,以降低由目前CCK符号所产生的符号内位干扰,以及由下一CCK符号和前一CCK符号所产生的符号间干扰。 (Iii) receiving the output from its associated CCK correlator, in order to reduce the inter-symbol interference from the current CCK-bit symbols generated by the next CCK symbol and a symbol and a previous symbol interference generated by CCK.

为让本发明的上述和其他目的、特征和优点能更明显易懂,下文特举一较佳实施例,并配合所附图式,作详细说明如下。 To make the above and other objects, features and advantages of the present invention can be more fully understood by referring cite a preferred embodiment, and with the accompanying drawings, described in detail below.


图1是绘示一种802.11b封包格式。 FIG 1 is a schematic diagram showing an 802.11b packet format.

图2是绘示依照本发明的一实施例的接收器示意图。 FIG 2 is a schematic diagram of a receiver in accordance with an embodiment of the present invention.

图3是绘示依照一个多路径传播通道的现有习知的多路径强度示意图(也就是通道脉冲响应图)。 FIG 3 is a schematic diagram view of a conventional multi-path intensity of the conventional propagation path in accordance with a multi-path (i.e., channel impulse response graph).

图4是绘示在RAKE之后的综合通道脉冲响应(CIR)图。 FIG 4 is a schematic illustrating an integrated channel impulse response after the RAKE (CIR) FIG.

图5是绘示在RAKE之后的接收讯号中所有多路径成分示意图。 FIG 5 is a schematic diagram of a multi-path components of all the received signals after the RAKE.

图6和图7是绘示依照本发明的一实施例的以目前CCK符号来降低多路径干扰遭遇的方法流程图。 6 and FIG. 7 is a method to reduce the current CCK symbol multipath interference encountered shown according to an embodiment of the present invention. FIG.

22: 选择器 24: 通道脉冲响应估测装置25: 连接符号解码器 28: CMF(通道匹配滤波器)36a:FMIC偏差计算装置 36b:CCK关联器具体实施方式以下描述本发明的最佳实施模式。 22: selector 24: the channel impulse response estimation means 25: connection symbol decoder 28: CMF (channel matched filter) 36a: FMIC deviation calculating means 36b: CCK correlator DETAILED DESCRIPTION The following description of the preferred embodiment mode of the invention . 此描述非用于限制本发明,只为描述本发明实施例的一般原则。 This description of non-limiting for the present invention, only the general principles described embodiments of the present invention. 本发明的范围由申请专利范围定义。 Scope of the invention defined by the claims. 在某些例子中,省略现有习知装置,元件,机构与方法的细节以免模糊本发明的描述。 In certain instances, details of the existing conventional device is omitted, elements, methods and means to avoid obscuring the description of the present invention.

802.11b为一无线局域网络的国际标准。 802.11b global standard for the wireless local area network. 简化的802.11b封包格式显示于图1中,其包含两种操作模式。 Simplified 802.11b packet format shown in FIG. 1, which includes two modes of operation. 封包的序言(Preamble)(模式1)为巴克码(Barker Code),而资料部份(模式2)采以互补码作调变,将互补码编码后以位(chip)为单位表示。 Packet preamble (a Preamble) (mode 1) is a Barker code (Barker Code), and the information part (Mode 2) is complementary to the code adopted for modulation, encoding the complementary code after expressed in units of bits (chip). 因为不会同时操作此两模式,所以可利用共享模式1与2的硬件以降低硬件复杂度。 Because not simultaneously operate the two modes, it is possible to utilize the shared hardware modes 1 and 2 in order to reduce hardware complexity.

图2是依照本发明的接收器20的方块图。 FIG 2 is a block diagram in accordance with the present invention, the receiver 20. 以下,先对接收器20中各方块的功能加以描述。 Hereinafter, to be described to the receiver 20 of each function block.

802.11b传输器(Transmitter)一次传输一个资料封包至接收器20以进行处理。 802.11b transmission device (Transmitter) first transmission of a data packet to the receiver 20 for processing. 当传输CCK调变封包时,各8个资料位经CCK编码成8个复数字元(称为一个CCK符号或是一个CCK字码),此8个复数字元由该802.11b传输器逐位地依序传输。 CCK modulation when transmitting packets, each of the eight data bits, CCK encoded into eight complex characters (referred to as a CCK or a CCK codeword symbol), the eight characters of the plurality of transmission bit by bit 802.11b in sequence transmission. 此信号经过无线通道并到达802.11b接收器(比如为接收器20)。 This signal passes through the channel and reaches the 802.11b wireless receivers (such as the receiver 20). 一般无线通道的特征在于多路径传播,因此会让接收到的信号失真。 Characterized in that the general wireless channel multi-path propagation, the received signal will thus distortion. 此种多路径传播和其问题,将会在下文的相关的图5中再加以叙述。 Such multipath propagation and its problems, will be 5 coupled hereinafter be described relating to FIG. 因此,做出低成本与高性能的接收器以减轻多路径失真是非常重要的。 Therefore, to make low-cost and high-performance receiver to mitigate multipath distortion is very important.

接收器20包括选择器22,通道脉冲响应估测装置(CIR Estimation)24、FMIC偏差计算装置36a、通道匹配滤波器(CMF)28、CCK关联器模组36b和连接符号解码器25。 The receiver 20 includes a selector 22, a channel impulse response estimation unit (CIR Estimation) 24, FMIC deviation calculating means 36a, the channel matched filter (CMF) 28, CCK correlator module 36b and the symbol decoder 25 is connected. 选择器22是将接收讯号送至两条路径中其中一条。 The selector 22 is supplied to the signal received in one of two paths. 在序言处理期间,选择器22选择第一条路径(模式1),而当对接收CCK解码时,则连接第二条路径(模式2)。 During processing the preamble, the selector 22 selects a first path (mode 1), when the received CCK decoder, connected to the second path (mode 2).

通道脉冲响应估测装置24的输入耦接至选择器22,而其输出耦接至FMIC偏差计算装置(FMIC Bias Computation)36a、通道匹配滤波器(CMF)28和连接符号解码器(Joint Symbol Decoder)25。 Input channel impulse response estimation means 24 is coupled to selector 22, and an output coupled to the FMIC deviation calculating means (FMIC Bias Computation) 36a, the channel matched filter (CMF) 28 and the connector symbol decoder (Joint Symbol Decoder ) 25. 假设通道脉冲响应估测装置24在一个封包(Packet)周期内其通道脉冲响应维持不变,并且在处理每一封包的序言的期间于模式1工作。 Suppose the channel impulse response estimation means 24 in which a channel pulse response packet remain unchanged within the (Packet) period, and operating in a mode during processing of the preamble of each packet. 在模式1操作的期间,通道脉冲响应估测装置24利用巴克码关联来估计“通道脉冲响应”(CIR,又称作multipath intensity profile,即″多路径强度轮廓″)。 During mode 1 operation, the channel impulse response estimation means 24 uses the Barker code correlation to estimate the "channel impulse response" (the CIR, also known as multipath intensity profile, i.e., "multi-path intensity profile"). 通道脉冲响应估测装置24的其中一组输出为通道匹配滤波器的标签权值(tap weights)。 Wherein the channel impulse response estimation means 24 outputs a set of channel weights tag match filter (tap weights). 在第四版的数字通讯(Digital Communications,Fourth Edition,JGProakis,McGraw Hill,NewYork,1995,下文简称“Proakis”)一书的第14章中说到,最佳的通道匹配滤波器标签权值可以轻易第从估计的通道脉冲响应中取得。 In the fourth edition of Digital Communications (Digital Communications, Fourth Edition, JGProakis, McGraw Hill, NewYork, 1995, hereinafter referred to as "Proakis") Chapter 14 of the book said, the best channel matching filter weights can tag easily obtained from the first estimate of the channel impulse response. 而为了简化硬件的架构,使用者也可以采用临界标准(Threshold Criterion),将各多路径中能量较小的路径忽略不计。 In order to simplify the hardware architecture, the user may employ a standard threshold (Threshold Criterion), each of the smaller multi-path energy path is negligible. 通道脉冲响应估计装置24的另一组输出,是回馈FB标签权值(以下以B1、B2、...、B7表示)和前馈FF标签权值(以下以F1、F2、...、F7表示),其分别对应包含了通道匹配滤波器的综合通道频率响应的后游标和前游标部分。 Channel impulse response estimate another set of output device 24, a feedback FB tag weight (hereinafter to B1, B2, ..., B7 shown) and a feedforward FF tag weight (hereinafter to F1, F2, ..., F7 shown), which contains the corresponding cursor integrated channel matched filter frequency response and a front channel portion cursor. 这些标签权值用途如下:(i)FMIC偏差计算装置36a用来计算FMIC偏差,并输出至连接符号解码器25,(ii)供连接符号解码器25用来计算因为多路径通道,所产生的符号间干扰:包含前一CCK符号和下一CCK符号对目前CCK符号的符号间干扰。 These labels weights for the following purposes: (i) FMIC deviation calculating means 36a for calculating a deviation FMIC, and outputs to the symbol decoder 25 is connected, (II) for the symbol decoder 25 is connected for calculating the multipath channel because, generated ISI: included former CCK symbol and a sign next to the current CCK CCK symbol of inter-symbol interference.

在模式1中,FMIC偏差计算装置36a的功能,是与美国专利序号第10/289,749号的FMIC方块36a相同。 In mode. 1, FMIC deviation calculating means 36a function, with U.S. Patent No. FMIC block No. 10 / 289,749 36a identical. 而在本发明中的FMIC偏差计算装置36a的实际架构,也可以采用在美国专利序号第10/289,749号中所绘示的任一FMIC实际架构。 The actual architecture of the present invention, FMIC deviation calculation means 36a and depicted in U.S. Patent Serial No. 10 / 289,749 FMIC to any one of the actual architecture can also be employed. 在模式1期间,本发明内的FMIC偏差计算装置36a使用由通道脉冲响应估测装置24所提供的前馈FF和回馈FB标签权值,以共同地和有效地计算64(或32、或16、或8)个多路径干扰偏差值。 During mode 1, FMIC deviation within the present invention using the computing devices 36a estimation means 24 provides feedback and feedforward FF FB by the label weights channel impulse response, to collectively and efficiently computed 64 (or 32, or 16 or 8) multipath interference offset value. 连接符号解码器25是在模式2中使用这些多路径干扰偏差值,有效的抵消符号内位干扰,详情如下所述。 Symbol decoder 25 is connected to the use of these multi-path interference offset value in mode 2, the bit offset within an effective symbol interference, the following details. 这些由FMIC偏差计算装置36a所提供的多路径干扰偏差值,是接收到的CCK符号(8位长)所产生的符号内位干扰的后关联表示。 FMIC deviation calculated by these means 36a provided multipath interference offset value is received CCK symbols generated by the symbol (length 8 bits) indicates the associated bit interference. 假设多路径强度轮廓在一个封包的周期内不会改变。 Suppose the multi-path intensity profile will not change in the period of a packet. 则每一个封包只须在模式1时,计算一次所有可能的多路径干扰偏差值。 Each packet is only 1 when the mode is calculated once for all possible multipath interference offset value. 这些多路径干扰偏差值(以下记为α·ICIm=α·C‾mH(B^low+F^up)HC‾m(m=0,1,2,...,255)).]]>得到这些值的连接符号解码器25,在模式2时,从CCK关联器模组(CCK Correlator Bank)36b的输出,扣除相对应的多路径干扰值,便可补偿因符号内位干扰所造成的影响。 These multipath interference offset value (hereinafter referred to as & alpha; & CenterDot; ICIm = & alpha; & CenterDot; C & OverBar; mH (B ^ low + F ^ up) HC & OverBar; m (m = 0,1,2, ..., 255) ).]]> connection symbol decoder to obtain values ​​of 25, in mode 2, the output from the correlator module CCK (CCK Correlator Bank) 36b, the deduction value corresponding to multipath interference, and can compensate for the symbol bit interference effects caused. 如美国专利序号第10/289,749号的接收器所描述的消除内位干扰的功能,本发明不需要更多而复杂的硬件,而是利用相同的硬件结构用来实现模式1中的FMIC多路径干扰偏差计算装置36a,以及模式2中的CCK关联器模组36b。 FMIC multipath receivers as described in U.S. Serial No. No. 10 / 289,749 by eliminating the bit disturb functions described, the present invention does not require more complex hardware, but to use the same hardware configuration used to implement mode 1 interference deviation calculating means 36a, and a mode CCK correlator module 36b 2. 换句话说,同一种硬件结构可以被“分享(Shared)”,在两个不同的模式时被使用,且用以执行两种不同的功能。 In other words, the same hardware configuration may be "shared (the Shared)", when used in two different modes, and to perform two different functions.

在模式2的操作期间,通道匹配滤波器28会使用由通道脉冲估测装置24所提供的通道匹配滤波器标签权值,将通过多路径通道的接收讯号的能量进行有效合并。 During the operation mode 2, the channel matched filter 28 uses the channel matched filter weights tags by the channel estimation device 24 provides a pulse of energy will be effectively incorporated by multi-path reception of signals channels. 在图2中,通道匹配滤波器28的输出是标示为Rk。 In FIG. 2, the output of the channel matched filter 28 is marked as Rk. 对第k个CCK字码而言,每一个Rk包含了八个接收位[r8k,r8k+1,r8k+2,r8k+3,r8k+4,r8k+5,r8k+6,r8k+7]。 Of the k-th codeword in terms of CCK, each received bit Rk contains eight [r8k, r8k + 1, r8k + 2, r8k + 3, r8k + 4, r8k + 5, r8k + 6, r8k + 7] . 在多路径通道时Rk的组成,详示于图5,在以下会有详细的相关叙述。 Rk composition when the multi-path channel, shown in detail in FIG. 5, the following will be described in detail relevant.

在1999年IEEE所制定的802.11b的标准中,一个CCK字码包含8个CCK位。 In the IEEE 802.11b standard developed in 1999, a CCK CCK word contains 8 bits. 在下文中,以C={Cm,m=0,1,...,255}来表示CCK编码本(Codebook),Cm用于表示CCK编码本中第m个CCK字码。 Hereinafter, to C = {Cm, m = 0,1, ..., 255} to represent codebook CCK (Codebook), Cm represents a CCK for encoding m-th CCK codeword. 第k个传输CCK字码是记为Ck:Ck=[ck0,ck1,ck2,ck3,ck4,ck5,ck6,ck7,],在此,每一个CCK位cki为{1,ejπ/2,ejπ,ej3π/2}(或者是{ejπ/4,ej5π/4,ej7π/4})中的一个四相位键移(Quadra-Phase Shift Keying,以下简称QPSK)复数值,其中第一个指标k是表示为CCK字码传输的时间顺序,而第二个指标i是表示在CCK字码中的第i个CCK位。 K-th transfer CCK codeword is denoted as Ck: Ck = [ck0, ck1, ck2, ck3, ck4, ck5, ck6, ck7,], Here, each of CCK bit cki is {1, ejπ / 2, ejπ , ej3π / 2} (or {ejπ / 4, ej5π / 4, ej7π / 4}) in a four-phase shift key (Quadra-phase shift Keying, hereinafter referred to as QPSK) complex value, wherein the index k is a first represented as a time sequence of CCK codeword transmission, while the second index i is an i th bit in the CCK of CCK code word. 而Ck是CCK编码本中的其中一个Cm。 And Ck is the CCK encoding of one Cm. 当一个802.11b的传输器工作在11Mbps模式时,会将8个资料位组合一起以决定传送的CCK字码,而这8个位的二进位表示是用来决定传输的CCK字码的指标值。 When the transmission is operating at a 802.11b 11Mbps mode, 8 bits are data bits are grouped together to determine CCK codeword transmitted, which is 8-bit binary representation is used to determine an index value for transmission CCK codeword . 例如8个资料位为10000001(129的二进位表示),则会被编码为C129来传输。 For example, eight data bits is 10000001 (binary representation 129), it will be transmitted is coded as C129. 而802.11b接收器的目的,就是将此8个资料位正确地加以解码。 The purpose of 802.11b receiver, this is the 8-bit data to be decoded correctly.

CCK关联器模组36b是将接收位Rk(一次8个位)进行FWT,以计算8个接收位与256个CCK字码其中的64个CCK字码之间的关联值。 CCK correlator module 36b Rk is the received bit (a 8 bits) FWT, in order to calculate the correlation values ​​between the received bit and 8 256 CCK codewords 64 wherein CCK codeword. 这64个输出只要简单地延伸,就可以获得所有8个接收位与256个CCK字码之间的关联值,其在以下的第(2)方程式中被标示为RkHCm(m=0,1,...,255)(也如图2所示)。 This output 64 simply extends between the associated values ​​can be obtained for all eight bits of the received CCK codeword 256, which is indicated in the following section (2) for the equation RkHCm (m = 0,1, ..., 255) (also shown in FIG. 2). 本发明中的CCK关联器模组36b的架构,同样也可以与美国专利申请序号第10/289,749号所绘示的不同的CCK关联器模组36b相同。 In the present invention, the CCK correlator architecture module 36b, also may be the same U.S. Patent Application depicted different CCK correlator module 36b Serial No. 10 / 289,749.

本发明的图2所绘示的FMIC偏差计算装置36a和CCK关联器模组36b是分开的构件。 FIG 2 of the present invention depicted FMIC deviation calculating means 36a and the CCK correlator module 36b are separate members. 在一般的实施例中,FMIC偏差计算装置36a和CCK关联器模组36b这两个装置实际上可以分成两个构件,或者也可以共享同一个硬件结构(如美国专利序号第10/289,749所绘示)。 In a general embodiment, FMIC deviation calculation means 36a and 36b CCK correlator module the two devices may actually be divided into two members, or may share the same hardware configuration (e.g., U.S. Patent Serial No. 10 / 289,749 depicted shown).

最后,连接符号解码器25是撷取RkHCm(m=0,1,...,255)和预先计算的FMIC偏差值当作输入,并且使用以下所叙述的程序#1和#2,对CCK符号共同地解码。 Finally, the symbol decoder 25 is connected to retrieve RkHCm (m = 0,1, ..., 255), and the previously calculated deviation FMIC as input, and using the procedure described in the # 1 and # 2, CCK sign jointly decoded. 而解码资料是连接符号解码器25的输出,并且代表802.11b接收器的输出。 Decoded data output is connected to the output of the symbol decoder 25 and represents the 802.11b receiver.

在提供连接符号解码器25的完整描述之前,以下先要以一个数学模型来描述多路径通道。 Before providing a complete description of the connection of the symbol decoder 25, the following first to a mathematical model to describe the multi-path channel. 这将会看到此连接符号解码器是对多路径传播通道最佳的解码器。 This will see this symbol decoder is connected to a multi-path propagation path best decoder.

多路径传播时常伴随着讯号经由墙壁、家俱、人类身上、和其他物件的反射,而出现在无线局域网络传输器和接收器之间。 Often accompanied by signals between the walls, furniture, humans, and other reflective objects, now out of WLAN transmitters and receivers with multi-path propagation. 在一个经由802.11b的传输器发布的CCK讯号,经过多路径传播通道时,接收器会收到多个复本(Multiple Copies),每个复本到达接收器的时间延迟和强度不同。 In a publication via the 802.11b CCK signal transmitter, passes through multi-path propagation path, the receiver receives multiple copies (Multiple Copies), arrive at the receiver each time the replica delay and different intensities. 若没有适当地处理这些因复本或回音(Echoes)造成的影响,接收器的性能常会因而下降,甚至导致无法接受的收讯品质和/或使得传输范围的缩减。 If not properly treated, or those affected by the echo replica (Echoes) caused by the performance of the receiver often result in a lower, or even lead to unacceptable reception quality and / or reduced so that the transmission range.

为了加强在多路径传输环境中的讯杂比(Signal-to-Noise Ratio),通常会使用如装置28的通道匹配滤波器CMF,它也就是一般人所熟知的RAKE接收器。 To enhance to-noise environment in a multipath transmission ratio (Signal-to-Noise Ratio), channel means typically used as a matched filter 28 CMF, it is generally known of RAKE receiver. 通道匹配滤波器CMF 28采用来自于通道脉冲响应估测装置24提供的最佳的通道匹配滤波器标签权值。 Channel matched filter CMF 28 from the channel pulse response using the best channel estimation device 24 provides a matching tag value filter weights. 通道匹配滤波器CMF 28的输出,为综合通道脉冲响应,此是将通道脉冲响应估测装置24所估测的通道脉冲响应,以及通道匹配滤波器CMF 28的标签权值取其旋积。 CMF matched filter output channel 28, the channel impulse response is integrated, this is 24 the estimated channel impulse response to the channel impulse response estimation means, and a channel matched filter CMF tag weights 28 rotating whichever product. 在通道匹配滤波器CMF28的输出,可得到Proakis一书中所描述的多路径合并增益(Multipath Combining Gain)hcmf。 Channel matched filter output at the CMF28 obtained a book Proakis multipath combining gain (Multipath Combining Gain) described hcmf. 从图3中可以看到一个典型的无线局域网络多路径强度轮廓,图3是复制伟伯斯彻的图9。 It can be seen that a typical wireless local area network from the multi-path intensity profile in FIG. 3, FIG. 3 is a copy of FIG. 9 Wei Bosi Che. 这多路径强度轮廓有较短的前游标部分和较长的后游标部分。 This multi-path intensity profile with a short front portion of the cursor and cursor long rear portion. 在RAKE接收器执行通道匹配滤波器之后,一个典型的综合通道脉冲响应会具有大约同样长度的前游标部分和后游标部分。 After the RAKE receiver performs channel matched filter, a typical channel impulse response to be integrated with the cursor about the same front and rear portions of the length of the cursor. 如图4所显示,不同的前游标和后游标路径的复数增益值的分别对应标示为Fi和Bi,其中i是用来表示在CCK位中的相对路径延迟。 Shown in Figure 4, corresponding to different values ​​of the complex gain of the front and rear cursors cursor path denoted Fi and Bi, where i is used to indicate the relative path delays bits in the CCK. 目标讯号是存在于综合通道脉冲响应的中央,并且得到一个由RAKE产生的处理增益hcmf。 The target signal is present in the center of an integrated channel impulse responses, and to obtain a processing gain hcmf generated by the RAKE.

本发明的图5中显示多路径传播通道所造成的问题。 Display problems caused by multipath propagation path of FIG. 5 of the present invention. 对于一接收到的CCK符号,有三种多路径干扰形式:(i)由前一符号所产生的符号间干扰ISI,(ii)由目前符号所产生的符号内位干扰ICI,以及(iii)由下一符号所产生的符号间干扰ISI。 For a received CCK symbol, there are three multi-path interference in the form of: (i) between the symbol from the previous symbol interference generated by the ISI, (ii) bit disturb ICI the symbols from the current symbols generated, and (iii) a inter-symbol interference generated by the next symbol ISI. 在典型的802.11b无线局域网络中,一个传输器和一个接收器的距离,大约在一两百英尺(feet)之内。 In a typical 802.11b wireless local area network, a transmitter and a receiver distance, in a range of about two hundred feet (feet ') of. 一个典型的传输器是经由天线传送其讯号。 A typical transmitter transmits its signal via an antenna. 讯号在到达目标接收器之前,会在多路径传播的环境中行进。 Signal before reaching the target receiver, will travel in the multipath propagation environment. 在所有的接收路径中(不太可能只有一条路径),会具有许多的反射,这些反射讯号不太可能在超过700英尺的传输距离下,仍然强到能够被侦测出来。 In all receiving paths (one path may be less), will have a number of reflections, the reflected signal is less likely in excess of 700 feet transmission distance is still strong enough to be detected out. 一个CCK符号大约727ns长,我们可以安全地假设,多路径干扰只来自于目前CCK符号和相邻的两个CCK符号。 A CCK symbol about 727ns long, we can safely assume that multipath interference only from the current CCK CCK symbol and two adjacent symbols. 图5绘示了所有的多路径干扰成分。 5 illustrates all of the multipath interference component.

在图5中,假设有三个CCK符号,分别为#0(前一符号),#1(目前符号)和#2(下一符号),依顺序分别经由多路径传播通道传送出去,并且被CCK接收器20所接收。 In Figure 5, assume that three CCK symbols are # 0 (previous symbol), # 1 (the current symbol) and # 2 (the next symbol), sent out in sequence via a multipath propagation path, respectively, and are CCK The receiver 20 receives. 在图5中所显示的每一列(Row)表示一个特定的路径,而所显示的每一行(Column)是一个特定的时间(以chip为单位)。 Each row (Row) shown in FIG. 5 shows a particular path, and each row (Column) shown is a specific time (in chip units). C的第一个下标为CCK字码(符号)的数目,而第二个下标是CCK字码中位的指标。 The first number of the subscript C for CCK codeword (symbol), and the second subscript is the index of the CCK-bit word. 接收位rk是从通道匹配滤波器CMF 28的输出中取样(Sample),而rk等于具有来自于综合通道频率响应的前游标权值Fi和后游标权值Bi所加权的行总和。 Rk bits received from the channel matched filter 28 in CMF output samples (Sample), while having equal rk cursor right front channel frequency response from the integrated value Fi and the sum of the weights after a cursor line weighted by Bi. 例如r8的这一行,合并参照图4之后我们可以得到:r8=F7c17+F6c16+F5c15+F4c14+F3c13+F2c12+F1c11+hcmfc10+B1c07+B2c06+B3c05+B4c04+B5c03+B6c02+B7c01在通道匹配滤波器CMF 28的输出,假如我们从CCK符号#1来看每一个接收位rk,会看到有一个目标讯号(请看到图5有一条被命名为″目标CCK字码″的路径),以及其他的14个多路径干扰成分:其分别由前一CCK符号(符号#0)、目前CCK符号(符号#1)和下一CCK符号(符号#2)所产生的七个前游标多路径干扰(请看图5中的前游标多路径ISI(符号间干扰)和ICI(符号内位干扰)),以及七个后游标多路径干扰(请看图5中的后游标多路径ISI和ICI)。 E.g. r8 this line, refer to FIGS. 4 After we obtain: r8 = F7c17 + F6c16 + F5c15 + F4c14 + F3c13 + F2c12 + F1c11 + hcmfc10 + B1c07 + B2c06 + B3c05 + B4c04 + B5c03 + B6c02 + B7c01 channel matched filtering CMF output device 28, if we look at each receiver from a CCK symbol bit # RK, will see a target signal (please see FIG. 5 has been named a "target CCK codeword" path), and 14 other multipath interference component: CCK respectively from the previous symbol (symbol # 0), the current CCK symbol (symbol # 1) and the next CCK symbol (symbol # 2) produced seven pre cursor multipath interference (see pre cursor multipath ISI (intersymbol interference) and ICI in Figure 5 (the sign bit disturb)), and after seven cursor multipath interference (multipathing see the cursor ISI and ICI in FIG. 5) .

假设忽略外加的杂讯,则图5中所表示的所有讯号和多路径干扰成分,可以用一个矩阵方程式来描述:R1=BupC0+HBhFC1+FlowC2在此,分别将接收符号R1和三个传输符号(前一符号C0、目前符号C1和下一符号C2)表示为:R=[r8,r9,r10,r11,r12,r13,r14,r15]TC0=c00,c01,c02,c03,c04,c05,c06,c07]TC1=[c10,c11,c12,c13,c14,c15,c16,c17]TC2=[c20,c21,c22,c23,c24,c25,c26,c27]T而由前一符号和下一符号所产生的多路径干扰则用以下的Bup和Flow表示: Plus the noise is assumed to ignore all signals and multipath interference component is represented in FIG. 5, can be described by a matrix equation: R1 = BupC0 + HBhFC1 + FlowC2 Here, each received transmission symbol and three symbols R1 (previous symbol C0, the current symbol and the next symbol C1 C2) is expressed as: R = [r8, r9, r10, r11, r12, r13, r14, r15] TC0 = c00, c01, c02, c03, c04, c05 , c06, c07] TC1 = [c10, c11, c12, c13, c14, c15, c16, c17] TC2 = [c20, c21, c22, c23, c24, c25, c26, c27] T and the previous symbol and next symbol interference generated by multipath is represented by the following Bup and Flow:

Bup=0B7B6B5B4B3B2B100B7B6B5B4B3B2000B7B6B5B4B30000B7B6B5B400000B7B6B5000000B7B60000000B700000000;]]>Flow=00000000F70000000F6F7000000F5F6F700000F4F5F6F70000F3F4F5F6F7000F2F3F4F5F6F700F1F2F3F4F5F6F70]]>同时,接收讯号也包括符号内位干扰ICI(例如因为多路径传播所产生的目前CCK符号#1的多个复本)。 Bup = 0B7B6B5B4B3B2B100B7B6B5B4B3B2000B7B6B5B4B30000B7B6B5B400000B7B6B5000000B7B60000000B700000000;]]> Flow = 00000000F70000000F6F7000000F5F6F700000F4F5F6F70000F3F4F5F6F7000F2F3F4F5F6F700F1F2F3F4F5F6F70]]> At the same time, receive signals including the ICI bit symbol interference (e.g. current CCK symbol # 1 because of the plurality of copies of the generated multipath propagation). 因此以下以一个矩阵,来描述所有CCK符号#1的多路径影响:HBhF=hcmfF1F2F3F4F5F6F7B1hcmfF1F2F3F4F5F6B2B1hcmfF1F2F3F4F5B3B2B1hcmfF1F2F3F4B4B3B2B1hcmfF1F2F3B5B4B3B2B1hcmfF1F2B6B5B4B3B2B1hcmfF1B7B6B5B4B3B2B1hcmf=Blow+hcmfI+Fup]]>在此,由目前符号的后游标和前游标所产生的多路径干扰,是由以下的Blow和Fup矩阵来描述:Blow=00000000B10000000B2B1000000B3B2B100000B4B3B2B10000B5B4B3B2B1000B6B5B4B3B2B100B7B6B5B4B3B2B10]]> Therefore the following in a matrix, to describe the multipath effects of all CCK symbols # 1: HBhF = hcmfF1F2F3F4F5F6F7B1hcmfF1F2F3F4F5F6B2B1hcmfF1F2F3F4F5B3B2B1hcmfF1F2F3F4B4B3B2B1hcmfF1F2F3B5B4B3B2B1hcmfF1F2B6B5B4B3B2B1hcmfF1B7B6B5B4B3B2B1hcmf = Blow + hcmfI + Fup]]> Here, multipath interference from the current rear cursors and pre cursor symbol generated is Blow described by the following matrix and Fup: Blow = 00000000B10000000B2B1000000B3B2B100000B4B3B2B10000B5B4B3B2B1000B6B5B4B3B2B100B7B6B5B4B3B2B10]]>

Fup=0F1F2F3F4F5F6F700F1F2F3F4F5F6000F1F2F3F4F50000F1F2F3F400000F1F2F3000000F1F20000000F100000000]]>在R内的目标讯号是可以用一个8乘8的单位矩阵来描述,其具有一个实数值的通道匹配滤波器增益hcmf。 Fup = 0F1F2F3F4F5F6F700F1F2F3F4F5F6000F1F2F3F4F50000F1F2F3F400000F1F2F3000000F1F20000000F100000000]]> R target signal is in a can 8 by 8 matrix is ​​described, which has a real value of the channel matched filter gain hcmf.

一般来说,接收讯号Rk用于对目前符号Ck来解码,其可以被描述为:Rk=[r8k,r8k+1,r8k+2,r8k+3,r8k+4,r8k+5,r8k+6,r8k+7]=BupCk-1+HBhFCk+FlowCk+1=BupCk-1+BlowCk+hcmfCk+FupCk+FlowCk+1Eq.(1)在实际的接收器运作中,通道脉冲响应估测装置24提供了多路径强度轮廓的估测,其中多路径强度轮廓可以描述为(Bup,HBhF,Flow)或是(Bup,Blow,hcmf,Fup,Flow)。 In general, the received signal Rk is used to decode the current symbol Ck, which can be described as: Rk = [r8k, r8k + 1, r8k + 2, r8k + 3, r8k + 4, r8k + 5, r8k + 6 , r8k + 7] = BupCk-1 + HBhFCk + FlowCk + 1 = BupCk-1 + BlowCk + hcmfCk + FupCk + FlowCk + 1Eq. (1) in the actual operation of the receiver, the channel impulse response estimation means 24 provides estimate of a multipath intensity profile, wherein the multi-path intensity profile can be described as (Bup, HBhF, Flow) or (Bup, Blow, hcmf, Fup, Flow). 以下, the following, with 被用来分别表示Bup,Blow,hcmf,Fup以及Flow的估计值。 Is used to denote an estimated value Bup, Blow, hcmf, Fup and the Flow.

请参照图5中被命名为目前符号的数行,从其中可以辨识出四个多路径干扰成分。 Referring to FIG. 5 are designated as the number of rows of the current symbol, which can be identified from the path over four interference components. 每一个多路径干扰的成分,都形成一个三角形(用虚线围起来的区域)。 Each multipath interference component, a triangle is formed (a region surrounded by a broken line). 其中,第一个三角形(范围内所有的CCK符号C的第一个下标都是0)是用来表示由前一符号所产生的符号间干扰ISI,第二和第三个三角形(范围内所有的CCK符号C的第一个下标都是1)是用来表示由目前符号所产生的符号内位干扰ICI,以及第四个三角形(范围内所有的CCK符号C的第一个下标都是2)是用来表示由下一符号所产生的符号间干扰ISI。 Wherein the first triangle (the first subscript all CCK symbols C are in the range of 0) is used to represent a symbol by the first inter-symbol interference generated by the ISI, the (range of the second and third triangles the first subscript of all CCK symbols are C 1) bit is used to indicate interference ICI, and a fourth triangle (the first subscript of all CCK symbols within a symbol C in the range from the current symbol generated is 2) it is used to indicate the next symbol by the inter-symbol interference generated by ISI.

总结来说,上述完整的数学模型,是用来描述在典型的无线局域网络中多路径传播通道的情形。 In summary, the above-described complete mathematical model is used to describe the case of a typical wireless local area network of the multi-path propagation path. 因此我们可以观察到,当解码一个CCK符号时,一个最佳的CCK符号解码器,需要考虑到所有因为CCK符号本身以及两个相邻CCK符号所造成的多路径干扰。 Thus it can be observed, when decoding a CCK symbol, a symbol decoder optimal CCK, CCK needs to take into account all symbol itself as well as two multi-path interference caused by adjacent CCK symbols.

本发明提供一低复杂度且近似完美的连接符号解码器,是使用接收到相邻的符号,以共同地对目前CCK符号解码。 The present invention provides a low complexity and approximately perfect connection symbol decoder, using the received symbols next to collectively current CCK symbol decoding. 在本发明的图2中的接收器20,具有CCK关联器模组36b,是将一个RAKE接收器或是通道匹配滤波器CMF28的输出(在第(1)方程式和图2中的Rk)当作输入,来获得接收讯号Rk和所有可能的(Potential)CCK字码Cm(m=0,1,...,255)的关联值:RkHC‾m=(BupCk-1+BlowCk+hcmjCk+FupCk+FlowCk+1)HC‾m---Eq.(2)]]>在此,{}H代表hermitian(即所谓“复数转置(Complex Transpose)”)。 In the receiver 2 of FIG. 20 according to the present invention, having a CCK correlator module 36b, the receiver is a RAKE path or the output of the matched filter CMF28 (the first (1) Rk of Equation 2 and FIG.) When as input, and obtains a reception signal Rk all possible (potential) CCK codeword Cm (m = 0,1, ..., 255) associated value: RkHC & OverBar; m = (BupCk-1 + BlowCk + hcmjCk + FupCk + FlowCk + 1) HC & OverBar; m --- Eq (2)]]> here, {} H Representative Hermitian (i.e., so-called "complex transpose (complex transpose)").. 连接符号解码器25是将遭到多路径干扰所影响的关联输出RkHCm(m=0,1,...,255)、通道脉冲响应估测装置24所计算出的前馈FF和后馈FB标签权值、以及FMIC偏差计算装置36a预先计算的FMIC偏差值(在模式1期间取得)当作输入,并且进行连接符号多路径干扰抵消,以获得近似完美的解码效果。 Symbol decoder 25 is connected to an associated output RkHCm being affected by multipath interference (m = 0,1, ..., 255), the channel impulse response of the feedforward FF and FB 24 after the feeding of the calculated estimation means tag weight, and FMIC deviation calculation means 36a FMIC deviation (acquired during mode 1) is calculated in advance as input, and performs the connection symbol multipath interference cancellation, in order to obtain near perfect decoded results.

要更完全地来认识本发明,首先必须指出,实现一个连接符号解码器的最主要的挑战,是在于其本身的复杂度。 To more fully to understand the invention, first of all it must be noted, the main challenge to achieve a connection symbol decoder, is in its own complexity. 在对第k个传输CCK字码Ck解码时,在每一个256个关联输出中,目标讯号是以下式来表示:(hcmfCk)HCm对正确的CCK字元符号(当Ck=Cm)而言,目标讯号为一个实数。 When the transmission of the k-th codeword Ck CCK decoder, 256 associated with each output, the target signal is represented by the following formula: (hcmfCk) HCm CCK characters of the correct symbol (if Ck = Cm) concerned, target signal is a real number. 而我们可以观察到,第(2)方程式实际上是一个复数运算。 And we can observe, first (2) equation is actually a complex operation. 当接收器运作时,为了节省硬件的架构,在第(2)方程式中所有的目标讯号和多路径干扰的项目,我们可以忽略其虚数(Imaginary)的部分,而只要对其实数部分加以运算。 When the receiver operation, in order to save hardware architecture, the first (2) project in the equation all target signals and multipath interference, we can ignore some of its imaginary (Imaginary), and be operational as long as its real part. 在以下,当我们为了节省硬件的架构,而只计算目标讯号或是多路径干扰的项目的实数部分时,并不会特别地去强调。 In the following, when the real part of the hardware architecture in order to save us, but only to calculate the target signal or multipath interference items, not particularly to emphasize. 熟习此技艺者当可以理解,一个简化的实施例只需考虑实部(Real Part)的运算。 When the person skilled in this art may be appreciated, embodiments of a simplified embodiment only consider calculating the real part (Real Part) of. 在CCK关联器模组36b的输出,其多路径干扰失真是表示为:(a)由前一字码Ck-1所产生的符号间干扰ISI:(BupCk-1)HCm(b)由目前字码Ck所产生的符号内位干扰ICI:(BlowCk+FupCk)HCm(c)由下一字码Ck+1所产生的符号间干扰ISI:(FlowCk+1)HCm对上述(a)中起因于前一CCK符号的多路径失真而言,我们可以用 CCK correlator output module 36b, the multipath interference distortion is expressed as: (a) by the inter-symbol code word Ck-1 produced by the front interference ISI: (BupCk-1) HCm (b) from the current word the symbol bit code Ck generated interference ICI: (BlowCk + FupCk) HCm (c) by the next code word Ck + 1 is generated inter-symbol interference ISI: (FlowCk + 1) HCm above in (a) due to CCK symbol before a multipath distortion, we can use with 来估算对每一可能的目前字码Cm的多路径失真。 To estimate the multipath each possible code word currently Cm distortion. 若是 if 非常接近Bup,并且前一符号的解码是正确的(即C^k-1=Ck-1]]>),我们就可以自第m个CCK相关器的输出,扣除失真造成的偏差值 With Bup very close, and the decoding of the preceding symbol is correct (i.e., C ^ k-1 = Ck-1]]>), we can CCK from the m-th correlation output, the distortion caused by the offset value deducted 以有效地抵消此多路径失真。 In order to effectively counteract this multipath distortion.

关于上述(b)中起因于目前符号的多路径失真,我们可以用下列的式子来估计符号内位干扰ICI偏差值:ICIm=C‾mH(B^low+F^up)HC‾m]]>其中m=0,1,2,...,255,在此m用来表示第m个CCK候选字码(共256个)。 About (b) above due to the current multipath symbol distortion, we can use the following formula to estimate the bit disturb ICI deviation within a symbol: ICIm = C & OverBar; mH (B ^ low + F ^ up) HC & OverBar; m] ]> where m = 0,1,2, ..., 255, here m denotes the m-th CCK for word candidates (a total of 256). 我们可以观察到,(b)中所示的符号内位干扰ICI偏差具有一个Ck和一个Cm,而估计的符号内位干扰ICI偏差则具有两个Cm,故上述(b)中所示的符号内位干扰ICI偏差,并不等于上式中估计的符号内位干扰ICI偏差。 We can observe interference ICI deviations in the position shown in symbol (b) in having a Ck and Cm is a, and the offset within the bit symbol estimates interference ICI is Cm is two, so as shown in (b) above symbols ICI inner bit disturb deviation is not equal to the sign bit of formula estimated interference ICI deviation. 此时应注意到的是,Ck还没有被解码。 At this point it should be noted that, Ck has not been decoded. 上式估计的符号内位干扰ICI偏差,只能当成一个估计值来看,此因为目前CCK符号(Ck)只可以是256个CCK字码(Cm)中的一个。 The estimated bit symbol interference ICI formula bias, only as an estimate of view, since this is currently CCK symbol (Ck) can only be 256 CCK codewords (Cm is) one. 因此,用上式估计的符号内位干扰ICI偏差只有对正确的目前CCK符号才是正确的。 Therefore, spend within symbolic estimated bit interference ICI deviation only to correct the current CCK symbol is correct. 对每一其他不正确的255个CCK符号,当以上式估计的符号内位干扰ICI偏差,被用在相对应的CCK关联器模组36b的关联输出时,可能失真会更加的严重。 To each other incorrect CCK symbol 255, if the sign bit of the above formula estimated interference ICI deviation, is used in a corresponding association CCK correlator output module 36b, it may be more serious distortion. 当我们需要保持简单的接收器架构时,这种多路径失真,只能够降低而无法完全消除。 When we need to keep it simple receiver architecture, this multi-path distortion, can only be reduced but not completely eliminated. 为了提供最好的降低效果,在以下我们提供了一个软偏差(Soft Bias):α·ICIm=α·C‾mH(B^low+F^up)HC‾m]]>其中0≤α≤1。 In order to provide the best effect of reducing, in the following we provide a soft deviation (Soft Bias): & alpha; & CenterDot; ICIm = & alpha; & CenterDot; C & OverBar; mH (B ^ low + F ^ up) HC & OverBar; m]]> wherein 0≤α≤1. 而最佳的α值可以由电脑模拟来寻得。 The optimum value α can be obtained from either a computer simulation.

基于同样的原因,上述(c)中起因于下一CCK符号(Ck+1)的多路径失真也只能被降低而无法完全消除。 For the same reason, multipath distortion (c) above due to the next CCK symbol (Ck + 1) can only be lowered and not be completely eliminated. 以下我们可以提供一个软偏差来降低起因于下一CCK符号(Ck+1)的多路径干扰:β·C‾mHF^lowHC‾m]]>其中0≤β≤1。 Below we offer a soft bias can be reduced to the next due to multipath interference CCK symbol (Ck + 1) is: & beta; & CenterDot; C & OverBar; mHF ^ lowHC & OverBar; m]]> wherein 0≤β≤1. 而最佳的β值可以由电脑模拟来寻得。 And either the optimum β value may be obtained by a computer simulation.

总合来说,当对第k个传输CCK字码Ck解码时:1)多路径干扰失真(a)可以用回馈FB标签和前一字码Ck-1的估计值来移除。 Aggregate, when the k-th transmission of CCK codeword decoding Ck: 1) multipath interference distortion (a) may be the front and back labels FB Ck-1 code word estimates removed.

2)多路径干扰失真(b)可以用回馈FB和前馈FF标签的估计值来降低。 2) multipath interference distortion (b) can be reserved and a feedforward estimation value FB FF tag is reduced.

3)多路径干扰失真(c)可以用前馈FF标签和下一字码Ck+1的估计值来降低。 3) multipath interference distortion (c) can be fed and the next estimated value FF tag word Ck + 1 to be reduced before use.

为了让系统有最好的表现,理论上我们要降低在图5中四个三角形所表示的所有多路径干扰失真。 For the system to have the best performance, in theory, we all want to reduce multipath interference in Figure 5 in four triangles represented distortion. 这需要回馈FB和前馈FF标签的估计值,以及前一字码和下一字码的估计值。 This requires an estimate of feedback and feedforward FB FF label, and the estimated value of the previous code word and the next code word. 一般来说,当考虑一个字码Ck的解码时,我们可以假设前一字码Ck-1已经被解码,故我们只需要下一字码Ck+1的估计值。 Generally, when considering the decoding of a word Ck, we can assume that the previous word Ck-1 code has been decoded, so we only need to estimate the value of the next codeword Ck + 1. 因此最理想的情况,第k个字码是依据256个关联器输出(RkHCm)、多路径通道描述 Ideally therefore, the k-th word is based on the correlator 256 output (RkHCm), multipath channel described 解出的前一字码 Solved before the code word 和下一接收字码的估计值来进行解码。 And the estimated value of the next received codeword to be decoded. 在此需要注意的是,为了要取得下一字码的估计值,在解码目前CCK字码时,必需延迟一个CCK字码。 It should be noted here that, in order to obtain an estimate of the next code word, when decoding the current CCK codeword, a necessary delay CCK code word.

因为有256个CCK字码候选者,我们可以想象,在目前CCK字码Ck进行最后解码之前,一个连接符号解码器,需要计算目前字码和下一字码所有组合的可能性(共65536=2562个)。 Because CCK codewords have 256 candidate, we can imagine that, before the current CCK codeword Ck final decoding, connected to a symbol decoder, necessary to calculate the likelihood of the current code word and the next word of all combinations (a total of 65536 = 2562). 这个巨大数字,使得在设计此种连接解码器上变得不切实际。 This huge number, so that such designs impractical on a decoder is connected. 因此,目的在于设计一个低复杂度的连接解码器,其可以不用去计算所有65536个字码组合而能有效地降低所有的多路径干扰失真。 Accordingly, an object is to design low-complexity decoder is connected, which do not have to be calculated for all combinations of word 65536 and all can effectively reduce multipath interference distortion.

以下描述一个近似完美的低复杂度连接符号解码器25。 The following describes a low-complexity near perfect symbol decoder 25 is connected. 此连接符号解码器25的要点,在于进行连接解码之前,找出用于目前字码Ck的Mk个起始估计值(Mk是远小于256),以及用于下一字码Ck+1的Mk+1个起始估计值(Mk+1是远小于256)。 Before this point is connected to a symbol decoder 25, that is connected decode Mk currently used to identify a start estimation values ​​Ck of the codeword (Mk is much less than 256), and a Mk next word Ck + 1 is +1 start estimation value (Mk + 1 is much less than 256). 因此,当解码目前字码Ck时,只有MkMk+1个字码组合需要用连接符号解码器25来运算。 Therefore, when decoding the current codeword Ck, 25 computes only MkMk + 1 code word combinations need connector symbol decoder. 这么少组合数目,不仅使接收器20大为简化,并且在移除或是降低多路径干扰失真(即四个三角形)上,都有极佳的表现。 Such a small number of combinations, so that only the receiver 20 is greatly simplified, and removing or reducing multipath interference distortion (i.e. four triangles) on, have a very good performance.

连接符号解码器25包括了执行以下所叙述的程序#1和#2的硬件,依程序#1和#2的顺序,共同而有效地对受到多路径干扰影响的CCK符号解码。 Connection symbol decoder 25 comprises the following hardware procedure described in # 1 and # 2, according to the order of CCK symbol decoding program # 1 and # 2, together effectively the interference by multipath. 熟习此技艺者可以轻易地依据以下程序#l和#2的描述,来实现连接符号解码器。 Those skilled in this art can easily described based on the following procedures #l and # 2, symbol decoder connection is realized. 以下程序#l的描述,是指出对目前CCK字码Ck和下一CCK字码Ck+1来取得起始估计值,程序#1:对第k个传输CCK字码Ck和第k+1个传输CCK字码Ck+1来取得起始估计值:需要的输入:(a)来自CCK关联器模组36b的输入:RkHCm(m=0,1,2,…,255)和Rk+1HCm。 The following description #l procedure is noted that the current CCK codeword next CCK codeword Ck and Ck + 1 to obtain initial estimates, the program # 1: For k-th transfer CCK code word Ck and the k + 1 th CCK codeword transmission Ck + 1 to obtain initial estimates: input required: (a) input from the CCK correlator module 36b: RkHCm (m = 0,1,2, ..., 255) and Rk + 1HCm.

(b)来自FMIC偏差计算装置36a的输入:α·ICIm=α·C‾mH(B^low+F^up)HC‾m:]]>FMIC偏差计算装置36a(可以与美国专利序号第10/289,749号的方块36a相同)是用来对每一个CCK字码(用m来标示),有效地预先计算这些多路径干扰输出,通常α是一个介于0至1的数字,最佳的α(值可由电脑模拟决定。这些多路径干扰输出,是目前CCK字码的内位干扰ICI偏差估计值。在每一个封包的序言期间,多路径干扰输出只需对所有的256个字码,计算一次,并且将的储存。 (B) input from the FMIC deviation calculating means 36a of: & alpha; & CenterDot; ICIm = & alpha; & CenterDot; C & OverBar; mH (B ^ low + F ^ up) HC & OverBar; m:]]> FMIC deviation calculating means 36a (may be the United States 10 of the same block / Patent serial No. 289,749 36a) is used for each CCK codewords (with m to mark), these pre-computed efficiently multipath interference output, usually α is a number ranging from 0 to 1, the optimum [alpha] (the value determined by computer simulation. these multipath interference output, the bit is CCK codeword interference ICI offset estimation value. during each packet a preamble, only multi-path interference output for all 256 word, calculated once and stored.

步骤1:对第k个(例如目前的)传输字码(或是符号),依据CCK关联器模组36b的关联输出RkHCm,及FMIC偏差计算装置36a预先计算的符号内位干扰ICI偏差,以获得ICI校正关联输出(ICI-Corrected correlation Outputs)。 Step 1: k-th (e.g., current) transmission word (or symbol), based on an associated output RkHCm CCK correlator module 36b, the deviation calculating means and FMIC bit symbol interference ICI deviations in the pre-calculated 36a to ICI obtained calibration-related output (ICI-corrected correlation outputs). 更详细地说,对第k个(例如目前的)传输字码(或是符号),将每一个来自于CCK关联器36b的关联输出RkHCm(用m为指标,共256个),扣除其对应的多路径干扰α·ICIm,以获得ICI校正关联输出。 More specifically, for the k-th (e.g., current) transmission word (or symbol), each of CCK from the associated correlator output is RkHCm 36b (with m as an index of 256), the corresponding deduction the multipath interference α · ICIm, ICI corrected to obtain a correlation output. 为了执行此步骤,可使用CCK字码和多路径干扰偏差的对称特性以将偏差的数目降低,是由256减至32或是16,就如美国专利序号第10/289,749号所述,并且仍然可以达到同样的表现。 To perform this step, may be used and the symmetrical nature of CCK codeword multipath interference variation to reduce the number of deviations, are reduced by the 32 256 or 16, as No. 10 / 289,749 U.S. Serial No., and still you can achieve the same performance.

步骤2:从由步骤1所取得的256个ICI校正关联输出中,选择Mk个CCK字码,其ICI校正关联输出是最大的。 Step 2: 256 from ICI corrected by the correlation output acquired in step 1, select a Mk CCK codeword, which ICI calibration-related output is the largest. 这些Mk个CCK字码作为目前CCK字码Ck的起始估计值,标示为 These two Mk as the current CCK codeword starting CCK codeword estimation value Ck is labeled as (i=1,2,…,Mk)。 (I = 1,2, ..., Mk). 以下为求方便描述,用pk,来表示对应于 The following description for the sake of convenience, with pk, represented corresponding to (i=1,2,…,Mk)的ICI校正关联输出。 (I = 1,2, ..., Mk) associated ICI corrected output.

步骤3:在一个延迟之后,当关联器提供下一(第k+1个)CCK字码(或是符号)关联输出Rk+1HCm的时候,重复上述的步骤1,以获得用于第k+1个传输CCK符号的ICI校正关联输出,并且重复上述的步骤2,以获得用于下一CCK字码Ck+1的Mk+1个起始估计值,标示为 Step 3: After a delay, when the next-correlator (k + 1-th) of CCK word (or symbol) associated Rk 1HCm output when the above steps are repeated + 1, to obtain for the k + ICI correcting an associated output a CCK symbol transmission, and repeats the above step 2, to obtain CCK Ck + Mk 1 + 1 initial estimates for the next codeword, denoted (j=1,2,…,Mk+1)。 (J = 1,2, ..., Mk + 1). 同时由pk+1,j,来表示对应于 By both pk + 1, j, corresponding to the expressed (j=1,2,…,Mk+1)的ICI校正关联输出。 ICI associated correction (j = 1,2, ..., Mk + 1) output.

程序#1结束注意到在上述的步骤1,因目前CCK符号本身的多路径失真(即符号内位干扰ICI),已经由校正关联输出来降低,因而我们可以根据校正关联输出,有效地在步骤2中来决定Mk或是Mk+1个起始估计值。 Procedure # 1 In the above-noted end of step 1, due to the current CCK symbol multipath distortion itself (i.e., the sign bit interference ICI), has an associated output is reduced by the correction, so that we can correct an associated output, effectively at step Mk 2 determines a start Mk + 1 or an estimated value.

利用程序#1中所取得的目前和下一CCK符号的起始估计值,我们可使用程序#2来实现连接符号解码器。 Initial estimates of the current symbol and the next CCK using program # 1 was achieved, we may be achieved using a symbol decoder connected to the program # 2. use 来表示已解码的前一CCK字码Ck-1。 Represented decoded previous CCK codeword Ck-1. 对每一个目前CCK符号的Mk个起始估计值,有Mk+1个下一CCK符号的起始估计值。 For each Mk current CCK symbol estimate a start, there is an estimated value Mk + 1 start of the next CCK symbol. 我们需要去评估这些MkMk+1个候选者,并且决定目前CCK符号。 We need to evaluate these candidates who MkMk + 1, and decided the current CCK symbol.

程序#2:对第k个传输CCK字码Ck进行连接符号解码:需要的输入:(a)从程序#1产生的输入:(a1)目前CCK字码 Procedure # 2: transmission of the k-th connection CCK codeword symbol decoding Ck: Input required: (a) generated from the input program # 1: (a1) the current CCK codeword (i=1,2,…,Mk)的Mk个起始估计值,以及其对应的ICI校正关联输出pk,i。 (I = 1,2, ..., Mk) of a start estimation value Mk, and associated corresponding correction ICI output pk, i.

(a2)下一CCK字码 (A2) next CCK codeword (j=1,2,…,Mk+1)的Mk+1个起始估计值,以及其对应的ICI校正关联输出pk+1,j。 (J = 1,2, ..., Mk + 1) is a start Mk + 1 estimation values, and associated corresponding correction ICI output pk + 1, j.

(b)从通道脉冲响应估测装置24所产生的输入: (B) estimating channel impulse responses from the input device 24 produced by: with (或是相等地Fi和Bi,i=1,2,…,7)步骤1:为了降低起因于下一CCK符号的多路径干扰(符号间干扰ISI)的冲击(1a)对每一个Mk个起始估计值 (Or equally Fi and Bi, i = 1,2, ..., 7) Step 1: In order to reduce the multipath interference due to the next CCK symbols (intersymbol interference ISI) impact (1a) for each one Mk the initial estimate of (i=1,2…,Mk),及其对应的每一个(Mk+1个)起始估计值 (I = 1,2 ..., Mk), and each (Mk + 1 th) corresponding to the initial estimate (j=1,2,…,Mk+1),依据以下的公式来计算起因于下一符号的多路径干扰:β·C^k+1,jHF^lowHC^k,i]]>在此,β是一个介于1和0之间的数字,其最佳值是由电脑模拟而决定。 (J = 1,2, ..., Mk + 1), according to the following formula to calculate the multipath interference due to the next symbol: & beta; & CenterDot; C ^ k + 1, jHF ^ lowHC ^ k, i]]> here, beta] is a number between 0 and 1, the optimum value is determined by the computer simulation.

(1b)依据每一个从步骤(1a)获得的结果设置一个最小值。 (1b) based on each result obtained from step (1a) is provided a minimum. 更详细地说,依据i=1,2,…,Mk个起始估计值,固定指标i并且让指标j由1变化到Mk+1,然后找出一指标jmin以用于该个i,即β·C^k+1,jminHF^lowHC^k,i=minj=1,2,...,Mk+1{β·C^k+1,jHF^lowHC^k,i}]]>(1c)现在,对每一个具有固定的指标i的Mk个起始估计值,从对应的第i个ICI校正关联输出pk,i(从程序#1中取得)扣除β·C^k+1,jminHF^lowHC^k,i]]>以取得新的一组Mk个关联输出Qk,i(i=1,2,…,Mk):Qk,i=Pk,i-β·C^k+1,jminHF^lowHC^k,i]]>这些关联输出对字码候选者 More specifically, according to i = 1,2, ..., Mk estimate a start, a fixed index i and index j by one so that changes to Mk + 1, then find a jmin metrics for the i th, i.e., & beta; & CenterDot; C ^ k + 1, jminHF ^ lowHC ^ k, i = minj = 1,2, ..., Mk + 1 {& beta; & CenterDot; C ^ k + 1, jHF ^ lowHC ^ k, i} ]]> (1c) now, for each Mk a start estimation values ​​with a fixed index i, the output from the associated corresponding to the i-th ICI correction PK, i (obtained from the program # 1) deducting & beta; & CenterDot; C ^ k + 1, jminHF ^ lowHC ^ k, i]]> to get a new set of associations Mk output Qk, i (i = 1,2, ..., Mk): Qk, i = Pk, i- & beta ; & CenterDot; C ^ k + 1, jminHF ^ lowHC ^ k, i]]> associated with the output of word candidates (i=1,2,…,Mk)来说是″最佳″的关联输出,此时由目前和下一符号所产生的多路径干扰已经降低。 (I = 1,2, ..., Mk) is the "best" correlation output at this time from the present and the next symbol generated by multipath interference has been reduced.

步骤2:为了移除起因于前一CCK符号Ck-1的多路径干扰(符号间干扰ISI)(2a)对每一个Mk个起始估计值,依据以下的方程式:C^k-1HB^upHC^k,i]]> Step 2: To remove the front due to a multipath interference CCK symbol Ck-1 (inter-symbol interference ISI) (2a) for each of a start estimation value Mk, according to the following equation: C ^ k-1HB ^ upHC ^ k, i]]>

计算起因于在CCK关联器模组36b的关联输出中,已解码的前一符号 Calculating an associated output due to the CCK correlator module 36b, the decoded previous symbol 造成的多路径干扰偏差。 Multipath interference caused by deviation. 在此, here, 为对应于目前(第k个)传输CCK字码Ck的第i个起始估计值(i=1,2,…,Mk,共Mk个CCK候选符号)。 Corresponding to the current (k-th) initial estimate of the i-th transmission of CCK codeword Ck (i = 1,2, ..., Mk, Mk a total of candidate symbol CCK).

(2b)从上述步骤(1c)中所获得的第i个关联输出中,扣除 (2b) associated with the i-th output obtained from the above step (1C) in deduct . 而为了以下讨论的方便,今Wk,I来标示这些关联输出:Wk,i=Qk,iC^k-1HB^upHC^k,i]]>在此i=1,2,…,Mk这些关联输出对字码候选者 In order to facilitate the following discussion, this Wk, I associated output label these: these associations Wk, i = Qk, iC ^ k-1HB ^ upHC ^ k, i]]> Here i = 1,2, ..., Mk outputting word candidates (i=1,2,…,Mk)来说是“最佳”的关联输出,由上一,目前和下一符号所产生的多路径干扰,已经降低或消除。 (I = 1,2, ..., Mk) is the "best" correlation output, from the previous, current and next symbol interference produced by multi-path, it has been reduced or eliminated.

步骤3:连接符号解码对于第k个传输字码Ck,连接符号解码器25选择 Step 3: symbol decoder connected to the k-th transfer code word Ck, connected to the decoder 25 selects the symbol (即为最大值)当作解码的CCK字码 (I.e. maximum value) as the decoded codeword CCK ,其具有指标imax,对应于上述步骤(2b)中的最大关联输出Wk,i。 Having IMAX index, corresponding to the step (2b) of the highest correlation output Wk, i. 换句话说,C^k=C^k,imax]]>,在此imax为Wk,imax=maxi=1,2,...,Mk{Wk,i}]]>令CL表示解码的第k个CCK字码(C^k=C‾L)]]>解码的资料是L的二进位表示。 In other words, C ^ k = C ^ k, imax]]>, here imax is Wk, imax = maxi = 1,2, ..., Mk {Wk, i}]]> denotes the decoding order CL CCK codeword of k (C ^ k = C & OverBar; L)]]> L is decoded binary information bits.

程序#2结束当实现上述程序#1和#2的连接解码器25时,熟习此技艺者当不需要完全按照上述步骤来达到同等的解码结果。 # 2 ends when the program which the program # 1 and # 2 connected to the decoder 25, those skilled in this art when it is not exactly follow the steps above to achieve the same decoding result. 例如我们在进行程序#2中步骤3中的资料解码决定之前,可以对所有的指标i(i=1,2,…,Mk)和j(j=1,2,…,Mk+1)来计算所有Mk×Mk+1的多路径干扰校正关联输出。 Before we performed e.g. program information # 2 in the decoding decision step 3, all of the indicators may i (i = 1,2, ..., Mk), and j (j = 1,2, ..., Mk + 1) to All computing Mk × Mk + 1 multi-path interference calibration-related output.

对于第k个字码Ck来说,起始估计值的数目用Mk标示。 For the k-th word Ck, the initial estimate of the number designated by Mk. 一个连接判断规则(Joint Decision Rule),是利用前一字码 A connection determination rule (Joint Decision Rule), using the previous word code 的解码字码、目前字码 The decoded code word, current code word (i=1,2,…,Mk)的Mk个候选者,以及下一字码 (I = 1,2, ..., Mk) of a candidate Mk, and a next word (j=1,2,…,Mk+1)的Mk+1个候选者来对第k个目前字码Ck解码。 (J = 1,2, ..., Mk + 1) th of Mk + 1 to a k-th candidate codeword Ck currently decoded pairs. 换句话说,当对第k个字码Ck解码时,我们需要计算:(1)Mk个与已解码的前一字码 In other words, when the k-th codeword decoding Ck, we need to calculate: (1) Mk and a decoded code word before 的关联,以对由前一字码所产生的多路径干扰进行补偿:以及(2)Mk×Mk+1个关联,代表Mk+1个下一候选字码,所造成的不同的多路径干扰偏差值。 Correlation, to compensate for the multipath interference by the first code word generated: and (2) Mk × Mk 1 associations + Mk + 1 th representative of the next candidate word, different multipath interference caused by Deviation.

因此,复杂度是与Mk+Mk×Mk+1的值成正比。 Thus, the complexity is Mk + Mk × Mk + 1 is proportional to the value.

若是Mk和/或Mk+1为非常大的数字,则此种连接符号解码器不太可能实现。 And if Mk / Mk + 1 or to very large numbers, such a connection is unlikely symbol decoder. 因此,在降低成本的原则下,应该要取得较小的Mk和Mk+1值。 Therefore, under the principle of cost reduction, it should be made smaller Mk and Mk + 1 value. 在不失去普遍性下,我们可以假设在以下所讨论中,将候选字码Mk的数目固定为M(Mk=M,_k)。 Without the loss of generality, we can assume that in the following discussion, the number of candidates is fixed to Mk word M (Mk = M, _k). 回到802.11b采用256个CCK字码的例子,我们可以选择M=256,但是如此会导致组合的数目过大,而使得此连接符号解码器不切实际。 Back 256 using 802.11b CCK codeword example, we may choose M = 256, but so will cause the number of combinations is too large, so that this connector symbol decoder impractical.

因此,为了达到近似完美的解码表现,在实现连接符号解码器25时最需要关切的,是降低所需的候选字码数目M。 Accordingly, in order to achieve near perfect decoding performance, achieved in the symbol decoder 25 is connected when needed most concern is to reduce the required number of candidate codeword M. 以下的观察提供有用的建议:1)在大多数实际的环境下,多路径传播的影响在±1个CCK符号(8位或是727ns的长度)之内。 The following observations provide useful suggestions: 1) in most practical circumstances, the influence of multipath propagation within the ± 1 th CCK symbol (eight 727ns or length). 而一个连接解码演算法在解码目前符号时,只需依据三个接收符号(前一符号、目前符号和下一符号),即可提供最佳的系统性能。 And a connecting decoding algorithm in decoding the current symbol, based on only three received symbol (the previous symbol, the next symbol and the current symbol), to provide the best system performance.

2)依照美国专利序号第10/289,749号的发明的FMIC偏差计算装置36a,是提供一种低成本且可靠的方法来降低起始估计值M的数目。 2) In accordance with U.S. Patent No. FMIC deviation invention No. 10 / 289,749 computing device 36a, it is to provide a low cost and reliable method to reduce the number M of the initial estimate. 这使得一个近似完美低复杂度的连接符号解码器25合理的被实现。 This enables a low complexity near perfect connection symbol decoder 25 is implemented reasonable. 更详细地说,FMIC偏差计算装置36a是提供一种低成本和可靠的手段来计算ICI校正关联输出。 More specifically, FMIC deviation calculating means 36a to provide a low cost and reliable means to calculate correlation outputs corrected ICI. 依据这些ICI校正关联输出,我们可以得到一个很小的起始估计值(起始候选字码)数目,用于一个低复杂度连接符号解码器。 These calibration-related output based ICI, we can get a small starting value estimate (the first candidate word) number, is connected to a low-complexity decoder symbols. 依据电脑模拟的结果,在M=2或3时即可达到最佳的解码性能。 Based on the results of computer simulation, when M = 2 or 3 can achieve the best decoding performance. 这么小的数字,就目前的工艺技术而言,可以轻易地做成相当具有竞争力的产品。 Such a small number, the current technology, the products can easily be made quite competitive.

虽然上述将焦点放在用于运作在11 Mbps模式的802.11b CCK的连接符号解码器25,该模式使用256个CCK字码为其编码本,但熟习此技艺者可以轻易地用相同的原理和接收器架构,对运作在5.5Mbps模式的802.11bCCK来进行解码,其编码本包含16个CCK字码,为256个字码的一个子集合。 Although the above-described operation for the focus 25, which uses 256 CCK codewords in the codebook for 802.11b CCK 11 Mbps mode, the symbol decoder is connected, but those skilled in this art can easily use the same principle and receiver architecture, the operation in 802.11bCCK 5.5Mbps mode is decoded, which contains 16 CCK codebook word, a subset of 256 codewords.

以下的例子,进一步对程序#1和#2提供图示。 The following examples provide further illustration of the program # 1 and # 2. 图6是绘示程序#1的步骤,图7是绘示程序#2的步骤。 FIG 6 is a schematic diagram of the procedural steps # 1, FIG. 7 is a schematic diagram of procedures in step # 2. 以下例子所提供的数值仅仅用于图示的目的,并不是实际的资料。 The following numerical example is provided for purposes of illustration only, not actual data.

在图6中,有256个CCK字码候选者,图6中的每一列,是用来表示每一个(共256个)CCK字码候选者的结果。 In FIG. 6, there are 256 candidates CCK codeword, each column of FIG. 6, is used to indicate the results of each (256 total) CCK word candidates. 第一行是用来提供每一个CCK字码的指标m。 The first row index m is used to provide each of a CCK codeword. 第二行显示关于第k个CCK字码的CCK关联器模组36b的关联输出。 The second line shows the k-th correlation output on CCK CCK codeword correlator module 36b. 第三行显示从FMIC偏差计算装置36a所取得的预先计算FMIC偏差。 The third row shows a pre-calculation means 36a FMIC deviation calculated from the acquired offset FMIC. 第四行是绘示采用第二和第三行的资料,执行程序#1的步骤1。 The fourth row is a schematic illustrating use of the second and third rows of data, a step of executing program # 1. 第五行是绘示程序#1的步骤2。 The fifth row shows a procedure of step 1 # 2. 依据第四行显示的ICI校正关联输出,第五行显示用于第k个字码的起始的Mk(Mk=3)个候选者(在图6中被圈起者)为C1,C254和C3。 ICI corrected based on the associated output of the fourth line shows, the fifth line shows the starting Mk for the k-th codeword (Mk = 3) th candidate (by circled in FIG. 6) is C1, C254 and C3 . 图6中的最后三行(第六、七和八行)是绘示程序#1的步骤3,为了第k+1个字码,实际地重复步骤1和2,以决定第k+1个字码的Mk+1(Mk+1=3)个起始候选者。 The last three lines (the sixth, seventh and eighth lines) in FIG. 6 is a schematic diagram of the program # 1 in step 3, k + 1 for the first code word, the actual repeating steps 1 and 2, to determine the k + 1-th codeword Mk + 1 (Mk + 1 = 3) th initial candidates. 在此,用于步骤3的FMIC偏差,是与之前图6的第三行完全相同。 Here, the deviation for the FMIC step 3, the third row is the same as before in FIG. 6. 本例中步骤3的结果,得到第k+1个字码的起始的Mk+1(Mk+1=3)个候选者为C0,C253和C4(被圈起者)。 Results of this embodiment, step 3, to obtain the k + 1 initial codeword Mk + 1 (Mk + 1 = 3) are candidates for the C0, C253 and C4 (circled persons). 在整个例子中应该要注意的是,每一处显示在函数Re{}中的数值,在数学上是正确的。 Throughout the examples should be noted that, in every display the value of the function Re {}, mathematically correct.

图7中第一行是显示指标Mk。 The first line in FIG. 7 is a metrics Mk. 所有其他的各行不是被用于显示程序#2的步骤,就是显示程序#2的子步骤。 All other lines are not being used in each step of displaying the program # 2, program # 2 is to display sub-step. 详细地说,第二行显示程序#2的步骤(1a),对于第k个字码(以i作为指标,而i=1,2,3(=Mk))的三个起始字码候选者(C1,C254和C3)之一,我们对每一个第k+1个字码(以j作为指标,而j=1,2,3(=Mk+1))的3个起始的字码候选者(C0,C253和C4),来计算起因于下一符号的多路径干扰。 In detail, the program # step (1a) 2 The second line shows, for the k-th word (to i as an index, and i = 1,2,3 (= Mk)) of the three candidate start word by one (C1, C254 and C3), we have for each k + 1-word code (as an index to j, and j = 1,2,3 (= Mk + 1)) of the three initial words candidate code (C0, C253 and C4), to calculate the multipath interference due to the next symbol. 第三行是显示步骤(1b),此处jmin是在步骤(1a)中对应最小值的j指标,对于第k个字码的三个起始候选者,取得各自的jmin值。 The third step is a row (. IB), where jmin is corresponding to the minimum index j (1a) in step, the starting candidate for the three k-th codeword, to obtain the respective values ​​jmin. 以起首的字码候选者C1作为例子,将步骤(1a)中所取得的结果(1.1,0.5和-0.2)作比较,并且决定最小值(-0.2)(图7中被圈起者)。 In the First word candidate C1 as an example, the result of the step (1a) obtained in (1.1,0.5 and -0.2) for comparison, and determines (by circled in FIG. 7) the minimum value (-0.2) . 对每一个用于第k个字码的起始候选者而言,此最小值是起因于下一CCK字码的多路径干扰估计值。 The starting candidate for each of the k-th codeword in terms of minimum value, which is due to multipath next CCK codeword interference estimate. 第四行是显示步骤(1c)。 The fourth row is a step (1c). 在步骤(1c)中,对每一个用于第k个CCK字码的起始候选者,将下一字码(第k+1个CCK字码)造成的多路径干扰,由ICI校正关联输出Pki中扣除以得到Qk,i。 In step (1C) in each of the starting candidate for the k-th CCK codeword, the next word (k + 1-th CCK codeword) caused by multipath interference, the associated output corrected by ICI Pki deducted to give Qk, i. 在本例中,Qk,1=9.2,Qk,2=9.4和Qk,3=7.9。 In this embodiment, Qk, 1 = 9.2, Qk, 2 = 9.4 and Qk, 3 = 7.9. 第五行是显示程序#2中的步骤(2a)。 The fifth line is the step (2a) in the program # 2 is displayed. 随着第k-1个CCK字码已经被解码(第k-1个CCK字码是标示为 With the k-1 first CCK codeword has been decoded (the k-1 first CCK codeword is marked as 我们可以对每一个用于第k个字码的起始候选者,计算起因于前一CCK字码的多路径干扰,如下所示:C^k-1HB^upHC^k,1=C^k-1HB^k-1HC‾1]]>C^k-1HB^upHC^k,2=C^k-1HB^k-1HC‾254]]>C^k-1HB^upHC^k,3=C^k-1HB^k-1HC‾3]]>在本例中,它们的数值可以被找到分别为0.2,-1和0.6。 We can start every candidate for the k-th codeword calculated due to multipath interference previous CCK codeword, as follows: C ^ k-1HB ^ upHC ^ k, 1 = C ^ k -1HB ^ k-1HC & OverBar; 1]]> C ^ k-1HB ^ upHC ^ k, 2 = C ^ k-1HB ^ k-1HC & OverBar; 254]]> C ^ k-1HB ^ upHC ^ k, 3 = C ^ k-1HB ^ k-1HC & OverBar; 3]]> in the present embodiment, their values ​​can be found respectively 0.2, 1 and 0.6. 第六行是显示步骤(2b)和3。 The sixth step is a line (2b) and 3. 在步骤(2b)中,对每一第k个CCK字码的起始候选者,从Qk i减去起因于前一CCK字码的多路径干扰以获得Wk,i。 In step (2b), a starting candidate for each of the k-th CCK codeword, Qk i is subtracted from the former due to the multipath interference CCK codeword to obtain Wk, i. 在本例中,Wk,1=9,Wk,2=10.4和Wk,3=7.3。 In the present embodiment, Wk, 1 = 9, Wk, 2 = 10.4 and Wk, 3 = 7.3. Wk,i是表示具有对每一个用于第k个CCK字码的起始候选者,经过所有多路径干扰校正后的关联输出。 Wk, i is having a starting candidate for each of the k-th CCK codeword, after all multipath interference associated with the corrected output. 在步骤(3)中,第k个字码解码完成。 In step (3), the k-th codeword decoded. 在本例中,在经过所有的多路径干扰校正后,Wk,2(图7中被圈起者)为最大的关联输出。 In the present embodiment, after all of the multipath interference correction, Wk, 2 (circled in FIG. 7 persons) the maximum correlation output. 因此,解码的CCK字码为C^k=C^k,2=C‾254,]]>而解码的资料为11111110,为254的二进位表示。 Thus, CCK codeword is decoded C ^ k = C ^ k, 2 = C & OverBar; 254,]]> 11111110 decoded data is expressed as binary 254.

虽然本发明已以实例揭露如上,然其并非用以限定本发明的应用范围,任何熟习此技艺者,在不脱离本发明的精神和范围内,当可作些许的更动与润饰,因此本发明的范围当视所附的权利要求所界定者为准。 Although the invention has been described by way of example, they are not intended to limit the scope of application of the present invention, any person skilled in this art, without departing from the spirit and scope of the present invention, it is intended that modifications and variations, so this the scope of the invention is best defined of the appended claims and their equivalents.

Claims (8)

  1. 1.一种降低多路径干扰的方法,用于接收到的互补码符号的解码,该方法包括下列步骤:I、预先计算每一个可能的互补码字码对应的多路径干扰偏差值,上述多路径干扰偏差值是以字码内位干扰(ICI)的后关联表示;II、对于接收到的目前互补码符号,计算其所有可能的互补码字码的关联输出值;III、对于接收到的目前互补码符号,依据上述步骤II中得到所有可能的互补码字码的关联输出值,及在步骤I中得到的此互补码字码对应的多路径干扰偏差值,计算经过内位干扰校正后的校正关联输出值;IV、对于接收到的目前互补码符号,依据上述步骤III中得到的所有可能的互补码字码的校正关联输出值,选取M1个目前起始候选互补码字码;V、对于接收到的下一互补码符号,执行与上述步骤II,步骤III中相同的计算,亦即计算出所有可能的互补码字码的关联输 1. A method for reducing multipath interference, for decoding the received complementary code symbol, the method comprising the steps of: I, pre-computed for each possible multi-path complementary codeword interference offset value corresponding to the plurality path interference offset value is associated with the code word bits after the interference (ICI) is represented; II, for the current received code symbols complementary, calculating correlation values ​​of the output codes which is complementary to all possible codewords; III, for the received after the current complementary code symbols, obtained according to the above-described step II all possible complementary codes associated with the output value codeword, and obtained in step I multipath this complementary codeword corresponding to the interference offset value is calculated through the bit disturb correction correcting an associated output value; IV, for the current received code symbols complementary, correction based on correlation values ​​for all possible complementary output codeword is obtained in the above step III, a currently selected first candidate M1 complementary codeword; V , for the next received code symbols complementary, and perform the above-described step II, III, the same calculation step, i.e., the calculated input associated with all possible complementary codeword of 值及经过内位干扰校正后的校正关联输出值;VI、对于接收到的下一互补码符号,依据上述V中得到的所有可能的互补码字码的校正关联输出值,选取M2个下一起始候选互补码字码;VII、对于M1个目前起始候选互补码字码,根据其经过内位干扰校正后的校正关联输出值,计算第一降低符号间干扰的关联输出值,上述第一降低符号间干扰的关联输出值,同时校正了内位干扰,以及因下一互补码符号产生的符号间干扰;VIII、对于M1个目前起始候选互补码字码,依据上述步骤VII中得到的第一降低符号间干扰的关联输出值,进一步对前一互补码符号产生的符号间干扰加以校正,产生第二降低符号间干扰的关联输出值;以及IX、依据上述第二降低符号间干扰的关联输出值,以解码目前互补码符号。 And the correction values ​​associated with the output value after the correction bit disturb; Vl, for the next received code symbols complementary, correction based on correlation values ​​for all possible complementary output codeword V obtained above, select the one with the next M2 complementary starting candidate codeword; VII correlation output value, for a current first candidate M1 complementary code word, according to its output value after the correction associated with the correction of the position of interference, calculation of the first reducing intersymbol interference, the first reduce correlation output value intersymbol interference, while correcting the position of the interference, because the next complementary and intersymbol interference generated code symbol; VIII, for a current first candidate M1 the complementary codewords, obtained according to the above-described step VII a first output value associated with reduced intersymbol interference, inter-symbol prior to a further complementary code symbol interference generated is corrected, the output value associate a second reduced intersymbol interference; and IX, according to the above second reduced intersymbol interference correlation output value, complementary codes to decode the current symbol.
  2. 2.根据权利要求1所述的一种降低多路径干扰的方法,其特征在于其中所述的步骤VII更包括下列步骤:对每一个目前起始候选互补码字码,计算因下一起始候选互补码字码所产生的多路径干扰偏差,上述多路径干扰偏差是以后关联符号间干扰偏差值表示;对每一个目前起始候选互补码字码,在M2个已计算出的后关联符号间干扰偏差值中,选择一最小值,作为其后关联符号间干扰偏差值;以及将目前起始候选互补码字码的校正关联输出值,减去其对应的后关联符号间干扰偏差值,以获得其对应的第一降低符号间干扰的关联输出值。 The one of the 1 multipath interference reduction method as claimed in claim, wherein said step VII wherein further comprising the steps of: for each current candidate complementary start codeword, is calculated by the next candidate initial complementary codeword multipath interference generated by a deviation, the deviation is above multipath interference associated with future inter-symbol interference offset value represents; each complementary current first candidate codeword, between M2 has been calculated after the associated symbol interference offset values, select a minimum value, as inter-symbol interference offset value associated thereafter; and the current first candidate codeword complementary codes associated with the output value correction, subtracting between the associated symbol interference offset value corresponding to to obtain the corresponding reduced intersymbol interference associated with the first output value.
  3. 3.根据权利要求1所述的一种降低多路径干扰的方法,其特征在于其中所述的步骤VIII更包括下列步骤:对每一个目前起始候选互补码字码,计算前一互补码符号产生的多路径干扰偏差,上述多路径干扰偏差是以后关联符号间干扰偏差值表示;以及将每一个目前起始候选互补码字码的第一降低符号间干扰的关联输出值,减去因前一互补码符号所产生的后关联符号间干扰偏差值,以获得其对应的第二降低符号间干扰的关联输出值。 1, according to one of the multi-path interference reduction method as claimed in claim, wherein said step VIII wherein further comprising the steps of: for each current candidate complementary start codeword, before calculating a complementary code symbol generating multi-path interference offset the deviation between the multipath interference after interference offset value associated symbol represents; and associating an output value between a first lowered current first candidate symbols per codeword complementary interference, by subtracting the former intersymbol interference associated with the output value of the second symbol associated with a decrease between the complementary code symbol interference generated by the offset value, to obtain the corresponding.
  4. 4.根据权利要求2所述的以一种降低多路径干扰的方法,其特征在于其中所述的步骤VIII更包括下列步骤:对每一个目前起始候选互补码字码,计算前一互补码符号产生的后关联符号间干扰偏差值;以及将每一个目前起始候选互补码字码的第一降低符号间干扰的关联输出值,减去因前一互补码符号所产生的后关联符号间干扰偏差值,以获得其对应的第二降低符号间干扰的关联输出值。 4. A method for reducing multipath interference according to claim 2, wherein said step of VIII wherein further comprising the steps of: for each current candidate complementary start codeword, before calculating a complementary code after the inter-symbol interference generated by the associated symbol offset value; and associating an output value of a first intersymbol interference reduction for each current candidate complementary start codeword, and by subtracting the previous symbol associated complementary code generated inter-symbol interference offset value, to obtain a correlation between the output value corresponding to a second symbol interference reduction.
  5. 5.根据权利要求1所述的以一种降低多路径干扰的方法,其特征在于其中所述的步骤IX的解码的动作,是由M1个目前起始候选互补码字码中,选择具有最大的第二降低符号间干扰的关联输出值者来实现,而步骤IX所产生的解码资料,是将被选中的互补码字码的指标以二进位表示。 5. A method for reducing multipath interference according to claim 1, wherein the step of decoding IX wherein said operation is determined by a current first candidate M1 complementary codeword in, with the largest associated with the output value of the intersymbol interference by a second reduction to achieve, and decoding the information generated in step IX, is selected to be complementary indicator codeword is represented in binary.
  6. 6.根据权利要求4所述的以一种降低多路径干扰的方法,其特征在于其中所述的步骤IX的解码的动作,是由M1个目前起始候选互补码字码中,选择具有最大的第二降低符号间干扰的关联输出值者来实现,而步骤IX所产生的解码资料,是将被选中互补码字码的指标以二进位表示。 6. A method for reducing multipath interference according to claim 4, wherein the step of decoding IX wherein said operation is determined by a current first candidate M1 complementary codeword in, with the largest associated with the output value of the intersymbol interference by a second reduction to achieve, and decoding the information generated in step IX, is selected to be complementary to the codeword index represented in binary.
  7. 7.一种使用于多路径通道的耙式接收器,此接收器用于接收连续的互补码符号,包含一个目前互补码符号、一个前一互补码符号和一个下一互补码符号,其特征在于此接收器包括:一通道估测装置,依据巴克码关联来估算通道脉冲响应,并产生通道匹配滤波器的诸标签权值、以及诸回馈权值和诸前馈卷标权值;一快速多路径干扰偏差计算装置,是根据上述诸回馈标签权值及诸前馈标签权值,用以计算每一个可能的互补码字码对应的多路径干扰偏差值,上述多路径干扰偏差是以互补码字码内位干扰的后关联值表示;一通道匹配滤波器,耦接该通道判断装置,以取得通道匹配滤波器的上述诸标签权值,该通道匹配滤波器具有一输出端;一互补码关联器,其输入端耦接该通道匹配滤波器的输出端,取得经由通道匹配滤波器滤波后的接收讯号,即诸互补码 A rake receiver used in a multipath channel, the receiver for receiving a continuous complementary code symbol, comprising a current complementary code symbol, a previous symbol and a complementary code of the next complementary code symbol, wherein the receiver comprising: a channel estimation apparatus, according to Barker code correlation to estimate the channel impulse response, the channel matched filter and generates various label weights, weights and various feedback and feedforward label such weights; a fast multipole path interference deviation calculating means is based on the weights of various reserved tag label Jizhu feedforward weights for each possible multi-path calculating complementary codeword interference offset value corresponding to the deviation of multipath interference in a complementary code after the codeword bits associated value indicates interference; a channel matched filter coupled to the channel determination means, to acquire the various labels matched filter weights the channel, the channel matched filter having an output terminal; a complementary code associated having an input terminal coupled to the output of the channel matched filter, the reception signal acquired via the channel matched filter filters, i.e. all complementary code 号,用以产生所有互补码字码的关联输出值;以及一解码器,具有以下三输入端,其一输入端耦接至该快速多路径干扰解码偏差计算装置,用以取得诸上述多路径干扰偏差,以降低该目前互补码符号所产生的符号内位干扰;第二个输入端耦接至该通道估测装置的输出端,取得上述诸前馈标签权值及诸回馈标签权值后,用以计算由前一互补码符号和下一互补码符号对目前互补码符号所产生的多路径干扰偏差;第三个输入端耦接至该互补码关联器的输出端,取得相关的互补码关联值后,用以降低由目前互补码符号所产生的符号内位干扰,以及由下一互补码符号和前一互补码符号所产生的符号间干扰。 Number, to generate an associated output values ​​of all the complementary codeword; and a decoder, having the three-input, one input terminal is coupled to the fast decoding multipath interference deviation calculating means for obtaining the above-described various multipath after a second input terminal coupled to the output of the channel estimation apparatus, obtains the various feedforward weights Jizhu reserved tag label weights; interference variation to reduce interference within the current symbol bit symbols generated by the complementary code for multipath calculated by the previous and the next code symbol is complementary to the complementary code symbols complementary codes the current symbol interference generated by variations; a third input terminal coupled to the output of the complementary code correlator acquires related complementary after the code correlation value, to reduce the interference from the current symbol bit complementary code symbols generated by the next symbol and inter code symbols complementary front and a complementary code symbol interference generated.
  8. 8.根据权利要求7所述的使用于多路径通道的耙式接收器,其特征在于其更包括一选择器,其输入为接收讯号,此选择器具有两个输出端,可以将接收讯号耦接至该通道估测装置或该通道匹配滤波器,用以在两种模式中,择其一运作。 Use according to claim rake receiver in the multi-path channel 7, characterized in that it further comprises a selector, which receives the input signal, the selector has two output terminals, can be coupled to receive signals the channel estimation means connected to the channel or matched filter, for in both modes, select one of the operation.
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